THS770006
SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
Broadband, Fully-Differential, 14-/16-Bit
ADC DRIVER AMPLIFIER
Check for Samples: THS770006
FEATURES
DESCRIPTION
•
•
•
•
The THS770006 is a fixed-gain of +6dB, wideband,
fully-differential amplifier designed and optimized
specifically for driving 16-bit analog-to-digital
converters (ADCs) at input frequencies up to
130MHz, and 14-bit ADCs at input frequencies up to
200MHz. This device provides high bandwidth,
high-voltage output with low distortion and low noise,
critical in high-speed data acquisition systems that
require very high dynamic range, such as wireless
base stations and test and measurement
applications. This device also makes an excellent
differential amplifier for general-purpose, high-speed
differential signal chain and short line driver
applications.
1
2.4GHz Bandwidth
3100V/µs Slew Rate, VOUT =2V Step
Fixed Voltage Gain: +6dB
IMD3: –107dBc, VOUT = 2VPP, RL = 400Ω,
f = 100MHz
OIP3: 48dBm, f = 100MHz
Noise Figure: 11dB, f = 100MHz
23
•
•
APPLICATIONS
•
•
14-/16-bit ADC Driver
ADC Driver for Wireless Base Station Signal
Chains: GSM, WCDMA, MC-GSM
ADC Driver for High Dynamic Range Test and
Measurement Equipment
•
THS770006 Driving 16-Bit ADC
100W
RO
50W
VINVIN+
50W
100W
VOCM
AIN+
30MHz
Bandpass
Filter
VOCM
AIN-
16-Bit
ADC
RO
RELATED DEVICES
DEVICE
THS770006
FFT Plot with Two-Tone Input at 96MHz and
100MHz (see Application Information section).
dBFS
The THS770006 operates on a nominal +5V single
supply, offers very fast, 7.5ns maximum recovery
time from overdrive conditions, and has a
power-down mode for power saving. The THS770006
is offered in a Pb-free (RoHS compliant) and green,
QFN-24
thermally-enhanced
package.
It
is
characterized for operation over the industrial
temperature range of –40°C to +85°C.
THS4509
Wideband, low-noise, low-distortion,
fully-differential amplifier
PGA870
Wideband, low-noise, low-distortion,
fully-differential, digitally-programmable
gain amplifier
ADS5481 to
ADS5485
10
0
-10
-20
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
130
140
0
5
10 15
20
25 30 35 40
Frequency (MHz)
45
50
55
DESCRIPTION
16-bit, 80MSPS to 200MSPS ADCs
ADS6145
14-bit, 125MSPS ADC
ADS6149
14-bit, 250MSPS ADC
61
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2010–2012, Texas Instruments Incorporated
THS770006
SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
PACKAGE/ORDERING INFORMATION (1)
PRODUCT
PACKAGE
TYPE
PACKAGE
DESIGNATOR
SPECIFIED
TEMPERATURE
RANGE
THS770006
VQFN-24
RGE
–40°C to +85°C
(1)
PACKAGE
MARKING
ORDERING
NUMBER
TRANSPORT
MEDIA, QUANTITY
THS770006IGRE
THS770006IRGET
Tape and reel, 250
THS770006IGRE
THS770006IRGER
Tape and reel, 3000
For the most current package and ordering information see the Package Option Addendum at the end of this document, or visit the
device product folder on www.ti.com.
DEVICE MARKING INFORMATION
= Pin 1 designator
THS7700
06IRGE
THS770006IRGE = device name
TI = TI LETTERS
YM = YEAR MONTH DATE CODE
TI YMS
LLLL
S = ASSEMBLY SITE CODE
LLLL = ASSY LOT CODE
ABSOLUTE MAXIMUM RATINGS (1)
Over operating free-air temperature range, unless otherwise noted.
Power supply (VS+ to GND)
THS770006
UNIT
5.5
V
Input voltage range
Ground to VS+
V
Differential input voltage, VID
Ground to VS+
V
Continuous input current, II
10
mA
Continuous output current, IO
100
mA
–40°C to +125°C
°C
Maximum junction temperature, TJ
+150
°C
Maximum junction temperature, continuous operation, long term reliability
+125
°C
Human body model (HBM)
2500
V
Charged device model (CDM)
1000
V
Machine model (MM)
100
V
Storage temperature range, Tstg
ESD ratings
(1)
2
Stresses above these ratings may cause permanent damage. Exposure to absolute maximum conditions for extended periods may
degrade device reliability. These are stress ratings only, and functional operation of the device at these or any other conditions beyond
those specified is not implied.
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THS770006
SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
THERMAL INFORMATION
THS770006
THERMAL METRIC (1)
RGE
UNITS
24 PINS
θJA
Junction-to-ambient thermal resistance
θJC(top)
Junction-to-case(top) thermal resistance
θJB
Junction-to-board thermal resistance
19
ψJT
Junction-to-top characterization parameter
0.5
ψJB
Junction-to-board characterization parameter
18.8
θJC(bottom)
Junction-to-case(bottom) thermal resistance
8.9
(1)
44.1
35
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
ELECTRICAL CHARACTERISTICS
Test conditions are at TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input
and output, and input and output referenced to midsupply, unless otherwise noted. Measured using evaluation module as
discussed in Test Circuits section.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
AC PERFORMANCE
Small-signal bandwidth
VOUT = 200mVPP
2.4
GHz
C
VOUT = 2VPP
780
MHz
C
VOUT = 3VPP
485
MHz
C
VOUT = 2VPP
360
MHz
C
VOUT = 3VPP
325
MHz
C
VOUT = 2V step
3100
V/µs
C
VOUT = 4V step
3200
V/µs
C
Rise time
VOUT = 2V step
0.6
ns
C
Fall time
VOUT = 2V step
0.6
ns
C
Settling time to 0.1%
VOUT = 2V step
2.2
ns
C
Input return loss, s11
See s-Parameters section, f < 200MHz
–20
dB
C
Output return loss, s22
See s-Parameters section, f < 200MHz
–20
dB
C
Reverse isolation, s12
See s-Parameters section, f < 200MHz
–70
dB
C
f = 10MHz
–87
dBc
C
f = 50MHz
–81
dBc
C
f = 100MHz
–78
dBc
C
f = 200MHz
–74
dBc
C
f = 10MHz
–103
dBc
C
f = 50MHz
–91
dBc
C
f = 100MHz
–86
dBc
C
f = 200MHz
–77
dBc
C
f = 50MHz, 10MHz spacing
–80
dBc
C
f = 100MHz, 10MHz spacing
–79
dBc
C
f = 150MHz, 10MHz spacing
–77
dBc
C
f = 200MHz, 10MHz spacing
–76
dBc
C
f = 50MHz, 10MHz spacing
–107
dBc
C
f = 100MHz, 10MHz spacing
–107
dBc
C
f = 150MHz, 10MHz spacing
–97
dBc
C
f = 200MHz, 10MHz spacing
–82
dBc
C
RL = 20Ω
19.6
dBm
C
RL = 400Ω
8.7
dBm
C
Large-signal bandwidth
Bandwidth for 0.1dB flatness
Slew rate
Second-order harmonic
distortion
Third-order harmonic distortion
Second-order intermodulation
distortion
Third-order intermodulation
distortion
1dB compression point
(1)
f = 100MHz
Test levels: (A) 100% tested at +25°C. Over-temperature limits by characterization and simulation. (B) Limits set by characterization and
simulation. (C) Typical value only for information.
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3
THS770006
SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
Test conditions are at TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input
and output, and input and output referenced to midsupply, unless otherwise noted. Measured using evaluation module as
discussed in Test Circuits section.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
Output third-order intercept
point
At device outputs, RL = 400Ω, f = 100MHz
48
dBm
C
Input-referred voltage noise
f > 100kHz
1.7
nV/√Hz
C
Ouput-referred voltage noise
f > 100kHz
3.4
nV/√Hz
C
10.5
dB
C
f = 100 MHz
11
dB
C
f = 200 MHz
13
dB
C
f = 50 MHz
Noise figure
100Ω differential
source
Overdrive recovery
Overdrive = ±0.5V
Output balance error
f = 200MHz
Output impedance
f = 100MHz
5
7.5
ns
B
–60
dB
C
4.4
Ω
C
DC PERFORMANCE
Gain
Output offset
Common-mode rejection ratio
TA = +25°C, RL = 400Ω
5.75
6
6.25
dB
A
TA = +25°C, RL = 100Ω
5.5
5.7
5.9
dB
B
TA = –40°C to +85°C, RL = 400Ω
5.7
6.3
dB
B
TA = –40°C to +85°C, RL = 100Ω
5.45
5.95
dB
B
TA = +25°C
–10
10
mV
A
12.5
mV
B
dB
A
dB
B
115
Ω
A
2.25
2.75
V
A
1.5
3.1
V
C
V
A
V
B
1.4
V
A
1.45
V
B
V
A
V
B
1.5
V
A
1.55
V
B
VPP
B
VPP
B
TA = –40°C to +85°C
±1
–12.5
TA = +25°C
36
TA = –40°C to +85°C
35
60
INPUT
Differential input resistance
85
Inputs shorted together, VOCM = 2.5V
Input common-mode range
VOCM = 2.5V, VOUT = 3VPP, HD degradation < 3dB,
see Figure 13
100
OUTPUT
Most positive output voltage
Each output with
200Ω to midsupply
TA = +25°C
3.64
TA = –40°C to +85°C
3.59
Least positive output voltage
Each output with
200Ω to midsupply
TA = +25°C
Most positive output voltage
Each output with
50Ω to midsupply
TA = +25°C
3.59
TA = –40°C to +85°C
3.54
Least positive output voltage
Each output with
50Ω to midsupply
TA = +25°C
Differential output voltage
Differential output current drive
3.7
1.3
TA = –40°C to +85°C
3.6
1.3
TA = –40°C to +85°C
TA = +25°C, RL = 400Ω
4.4
TA = –40°C to +85°C, RL = 400Ω
4.2
4.85
TA = +25°C, RL = 10Ω
80
mA
B
TA = –40°C to +85°C, RL =10Ω
80
mA
B
OUTPUT COMMON-MODE VOLTAGE CONTROL
VOCM small-signal bandwidth
VOUT_CM = 200mVPP
525
MHz
C
VOCM slew rate
VOUT_CM = 500mVPP
180
V/µs
C
VOCM voltage range
Supplied by external source (2)
2.25
2.5
2.75
V
C
VOCM gain
VOCM = 2.5V
0.98
1
1.02
V/V
A
Output common-mode offset
from VOCM input
VOCM = 2.5V
–30
12
30
mV
A
VOCM input bias current
2.25V ≤ VOCM ≤ 2.75V
–400
±30
400
µA
A
(2)
4
Limits set by best harmonic distortion with VOUT = 3VPP. VOCM voltage range can be extended if lower output swing is used or distortion
degradation is allowed, and increased bias current into pin is acceptable. For more information, see Figure 12 and Figure 31.
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THS770006
SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
Test conditions are at TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input
and output, and input and output referenced to midsupply, unless otherwise noted. Measured using evaluation module as
discussed in Test Circuits section.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST
LEVEL (1)
POWER SUPPLY
Specified operating voltage
Quiescent current
Power-supply rejection ratio
4.75
5
5.25
V
C
TA = +25°C
85
100
115
mA
A
TA = –40°C to +85°C
80
125
mA
B
TA = +25°C, VCC ±0.25V
60
dB
A
TA = –40°C to +85°C, VCC ±0.5V
59
dB
B
V
A
V
A
A
90
POWER-DOWN
Enable voltage threshold
Device powers on below 0.5V
Disable voltage threshold
Device powers down above 2.0V
0.5
2
Power-down quiescent current
0.8
3
mA
Input bias current
80
100
µA
A
10
µs
C
0.15
µs
C
Turn-on time delay
Time to VOUT = 90% of final value
Turn-off time delay
Time to VOUT = 10% of original value
THERMAL CHARACTERISTICS
–40
Specified operating range
Thermal resistance, θJC (3)
Junction to case (bottom)
Thermal resistance, θJA (3)
Junction to ambient
(3)
°C
C
8.9
°C/W
C
44.1
°C/W
C
+85
Tested using JEDEC High-K test PCB. Thermal management of the final printed circuit board (PCB) should keep the junction
temperature below +125°C for long term reliability.
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THS770006
SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
PIN CONFIGURATION
NC
NC
VS+
VS+
VS+
VS+
NC
24
23
22
21
20
19
RGE PACKAGE
VQFN-24
(TOP VIEW)
1
18
NC
17
Unused
16
VOUT+
15
VOUT-
100W
PD
2
50W
VIN-
3
VOCM
50W
VIN+
4
100W
11
12
NC
NC
GND
NC
10
13
GND
6
9
NC
GND
Unused
8
14
GND
5
7
VOCM
PIN DESCRIPTIONS
PIN
NO.
NAME
1
NC
No internal connection
2
PD
Power down. High = low power (sleep) mode. Low = active.
3
VIN–
Inverting input pin
4
VIN+
Noninverting input pin
5
VOCM
Output common-mode voltage control input pin
6, 7
NC
8, 9, 10, 11
GND
12, 13
NC
14
No internal connection
Ground. Must be connected to thermal pad.
No internal connection
Unused Bonded to die, but not used. Tie to GND.
15
VOUT–
Inverting output pin
16
VOUT+
Noninverting output pin
17
Unused Bonded to die, but not used. Tie to GND.
18, 19
NC
No internal connection
20, 21, 22,
23
VS+
Power supply pins, +5V nominal
24
NC
No internal connection
Thermal pad
6
DESCRIPTION
Thermal pad on bottom of device is used for heat dissipation and must be tied to GND
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SBOS520B – JULY 2010 – REVISED JANUARY 2012
www.ti.com
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
TITLE
FIGURE
Frequency Response Magnitude (with Transformers)
Figure 1
Frequency Response Magnitude (no Transformers)
Figure 2
Frequency Response Phase (no Transformers)
Figure 3
Small- and Large-Signal Pulse Response
Figure 4
Slew Rate vs Output Voltage Step
Figure 5
Overdrive Recovery
Figure 6
Single-Ended Input Harmonic Distortion vs Frequency
Figure 7
Harmonic Distortion vs Frequency
Figure 8
Harmonic Distortion vs Frequency, VOUT = 0.9VPP
Figure 9
Harmonic Distortion vs VOUT
Figure 10
Harmonic Distortion vs RL
Figure 11
Harmonic Distortion vs VOCM
Figure 12
Intermodulation Distortion vs Frequency, VOUT = 2VPP, 3VPP Envelope
Figure 14
Intermodulation Distortion vs Frequency, VOUT = 0.9VPP Envelope
Figure 15
Output Intercept Point vs Frequency
Figure 16
Maximum Differential Output Voltage Swing Peak-to-Peak vs Differential Load Resistance
Figure 17
Maximum/Minimum Single-Ended Output Voltage vs Differential Load Resistance
Figure 18
Differential Output Impedance vs Frequency
Figure 19
s-Parameters (Magnitude)
Figure 20
Frequency Response vs Capacitive Load
Figure 21
Recommended RO vs Capacitive Load
Figure 22
Common-Mode Rejection Ratio vs Frequency
Figure 23
Power-Supply Rejection Ratio vs Frequency
Figure 24
VOCM Pulse Response
Figure 25
Turn-On Time
Figure 26
Turn-Off Time
Figure 27
Input and Output Voltage Noise vs Frequency
Figure 28
Output Balance Error vs Frequency
Figure 29
VOCM Small-Signal Frequency Response
Figure 30
VOCM Input Bias Current vs VOCM Input Voltage
Figure 31
Noise Figure vs Frequency
Figure 32
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TYPICAL CHARACTERISTICS
At TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input and output, and
input and output pins referenced to midsupply, unless otherwise noted. Measured using EVM as discussed in Test Circuits
section.
FREQUENCY RESPONSE MAGNITUDE
(WITH TRANSFORMERS)
3
0
-3
VO = 200mVPP
VO = 2VPP
VO = 3VPP
-6
-9
10M
VOUT = 200mVPP
9
6
Gain Magnitude (dB)
6
Gain Magnitude (dB)
12
Measured on EVM using transformers.
See Frequency Response: 200mVPP, 2VPP, 3VPP section.
3
0
-3
VOUT = 2VPP
Measured on EVM;
no transformers.
See Frequency Response:
200mVPP, 2VPP, 3VPP section.
-6
-9
100M
1G
Frequency (Hz)
-12
100k
10G
1M
10M
100M
Frequency (Hz)
Figure 1.
SMALL- AND LARGE-SIGNAL PULSE RESPONSE
45
3
2
0
Differential VOUT (V)
Gain Phase (°)
VOUT = 200mVPP
-45
-90
VOUT = 2VPP
Measured on EVM;
no transformers.
VOUT = 3VPP
See Frequency Response:
200mVPP, 2VPP, 3VPP section.
-180
100k
1M
10M
100M
Frequency (Hz)
1
0
-1
-2
1G
0.5V Step Input
2.5V Step Input
-3
10G
0
20
Figure 3.
40
60
TIme (ns)
100
120
OVERDRIVE RECOVERY
2.0
4000
4
VIN
Differential VOUT
1.5
3000
VIN (V)
Slew Rate (V/mS)
80
Figure 4.
SLEW RATE
vs OUTPUT VOLTAGE STEP
2000
1000
1
2
3
4
5
3
1.0
2
0.5
1
0
0
-0.5
-1
-1.0
-2
-1.5
-3
-2.0
0
-4
0
VOUT (VPP)
Figure 5.
8
10G
1G
Figure 2.
FREQUENCY RESPONSE PHASE
(NO TRANSFORMERS)
-135
VOUT = 3VPP
Differential VOUT (V)
9
FREQUENCY RESPONSE MAGNITUDE
(NO TRANSFORMERS)
20
40
60
Time (ns)
80
100
Figure 6.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input and output, and
input and output pins referenced to midsupply, unless otherwise noted. Measured using EVM as discussed in Test Circuits
section.
SINGLE-ENDED INPUT
HARMONIC DISTORTION vs FREQUENCY
HARMONIC DISTORTION vs FREQUENCY
-60
-60
-70
HD2, VOUT = 2VPP
-75
HD2, VOUT = 1VPP
-80
-85
-90
-95
-100
HD3, VOUT = 3VPP
-105
HD3, VOUT = 2VPP
-110
HD3, VOUT = 1VPP
-115
10M
100M
Frequency (Hz)
1G
-70
HD2, VOUT = 1VPP
-75
-80
-85
-90
-95
-100
HD3, VOUT = 3VPP
-105
HD3, VOUT = 2VPP
-110
HD3, VOUT = 1VPP
100M
Frequency (Hz)
1G
Figure 7.
Figure 8.
HARMONIC DISTORTION vs FREQUENCY
VOUT = 0.9VPP
HARMONIC DISTORTION
vs VOUT
-65
Harmonic Distortion (dBc)
RL = 400W
VOUT = 0.9VPP
-40
-50
-60
-70
HD2
-80
-90
HD3
f = 100MHz
RL = 400W
-70
-75
HD2
-80
HD3
-85
-90
-95
-100
-110
100M
-100
1G
0
1
2
3
VOUT Differential (V)
Frequency (Hz)
4
Figure 9.
Figure 10.
HARMONIC DISTORTION
vs RL
HARMONIC DISTORTION
vs VOCM
5
-20
-65
f = 100MHz
VOUT = 3VPP
f = 100MHz
RL = 400W
-30
-70
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
RL = 400W
HD2, VOUT = 2VPP
-115
10M
-30
Harmonic Distortion (dBc)
HD2, VOUT = 3VPP
-65
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
HD2, VOUT = 3VPP
RL = 400W
-65
-75
HD2
-80
HD3
-85
-40
HD2, VOUT = 3VPP
-50
HD3, VOUT = 3VPP
-60
-70
-80
-90
HD3, VOUT = 2VPP
-100
HD2, VOUT = 2VPP
-110
-90
0
200
400
600
800
1k
2
RL (W)
Figure 11.
2.25
2.5
VOCM (V)
2.75
3
Figure 12.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input and output, and
input and output pins referenced to midsupply, unless otherwise noted. Measured using EVM as discussed in Test Circuits
section.
HARMONIC DISTORTION
vs INPUT COMMON-MODE RANGE
INTERMODULATION DISTORTION
vs FREQUENCY, VOUT = 2VPP, 3VPP ENVELOPE
-20
-65
VOCM = 2.5V
f = 100MHz
RL = 400W
-40
-50
-60
-70
-80
-75
-80
-85
-90
-95
-100
1.0
1.5
2.0
2.5
3.0
Input Common-Mode Voltage at Device Pins (V)
-110
50M
3.5
100M
125M
150M
Frequency (Hz)
175M
Figure 14.
INTERMODULATION DISTORTION
vs FREQUENCY, VOUT = 0.9VPP ENVELOPE
OUTPUT INTERCEPT POINT
vs FREQUENCY
200M
90
RL = 400W
VOUT = 0.9VPP Envelope
OIP2
80
Output Intercept Point (dBm)
-40
-50
-60
IMD2
-70
-80
IMD3
-90
-100
70
60
OIP3
50
40
30
20
RL = 400W
VOUT = 3VPP Envelope
10
-110
100M
0
50M
1G
Frequency (Hz)
100M
150M
Frequency (Hz)
200M
Figure 15.
Figure 16.
MAXIMUM DIFFERENTIAL OUTPUT VOLTAGE SWING
PEAK-TO-PEAK vs DIFFERENTIAL LOAD RESISTANCE
MAXIMUM/MINIMUM SINGLE-ENDED OUTPUT VOLTAGE
vs DIFFERENTIAL LOAD RESISTANCE
4.0
5.5
5.0
3.5
4.0
Single-Ended VOUT (V)
Maximum
Differential
VOUT_PP
4.5
Differential VOUT (V)
75M
Figure 13.
-30
Harmonic Distortion (dBc)
IMD3, VOUT = 3VPP Envelope
IMD3, VOUT = 2VPP Envelope
-105
-90
RL = 400W
IMD2, VOUT = 3VPP Envelope
IMD2, VOUT = 2VPP Envelope
-70
Harmonic Distortion (dBc)
Harmonic Distortion (dBc)
-30
HD2 VOUT = 3VPP
HD3 VOUT = 3VPP
HD2 VOUT = 2VPP
HD3 VOUT = 2VPP
3.5
3.0
2.5
2.0
Maximum Single-Ended VOUT
3.0
2.5
2.0
Minimum Single-Ended VOUT
1.5
1.5
1.0
1.0
10
100
Load Resistance (W)
1k
10
Figure 17.
10
100
Load Resistance (W)
1k
Figure 18.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input and output, and
input and output pins referenced to midsupply, unless otherwise noted. Measured using EVM as discussed in Test Circuits
section.
DIFFERENTIAL OUTPUT IMPEDANCE
vs FREQUENCY
s-PARAMETERS
(MAGNITUDE)
1k
20
s22
Gain Magnitude (dB)
Differential ZOUT (W)
0
100
10
-20
s11
-40
-60
s12
-80
1
-100
1M
10M
100M
Frequency (Hz)
1G
1M
10G
10M
100M
Frequency (Hz)
Figure 19.
Figure 20.
FREQUENCY RESPONSE
vs CAPACITIVE LOAD
RECOMMENDED RO
vs CAPACITIVE LOAD
9
1G
10G
1k
6
RO (W)
Gain (dB)
100
3
0
-3
10
1
100M
1G
Frequency (Hz)
10G
1
10
100
Capacitive Load (pF)
1k
Figure 21.
Figure 22.
COMMON-MODE REJECTION RATIO
vs FREQUENCY
POWER-SUPPLY REJECTION RATIO
vs FREQUENCY
100
100
90
90
80
80
70
70
PSRR (dB)
CMRR (dB)
-6
10M
CL = 10pF, RO = 35W
CL = 24pF, RO = 18.7W
CL = 44pF, RO = 13W
CL = 94pF, RO = 8.2W
CL = 660pF, RO = 0.7W
60
50
40
60
50
40
30
30
20
20
10
10
0
0
1M
10M
100M
Frequency (Hz)
1G
10k
Figure 23.
100k
1M
10M
Frequency (Hz)
100M
1G
Figure 24.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input and output, and
input and output pins referenced to midsupply, unless otherwise noted. Measured using EVM as discussed in Test Circuits
section.
VOCM PULSE RESPONSE
TURN-ON TIME
3.0
4
Power-Down Input and VOUT (V)
2.9
VOUT Common-Mode (V)
2.8
2.7
2.6
2.5
2.4
2.3
2.2
2.1
2.0
Power-Down
2
1
0
VOUT
-1
0
50
100
150
200
250
Time (ns)
300
350
400
0
Input and Output Voltage Noise (nVÖHz)
1
0
Power-Down
-1
6
28
32
36
8
10
12
Time (ms)
14
16
18
20
40
10
Output Noise
Input Noise
1
10k
Figure 27.
100k
1M
Frequency (Hz)
10M
100M
Figure 28.
OUTPUT BALANCE ERROR
vs FREQUENCY
VOCM SMALL-SIGNAL FREQUENCY RESPONSE
0
3
-10
0
-20
-3
-30
Gain (dB)
Output Balance Error (dB)
16
20
24
Time (ms)
INPUT AND OUTPUT VOLTAGE NOISE
vs FREQUENCY
VOUT
4
12
TURN-OFF TIME
2
2
8
Figure 26.
3
0
4
Figure 25.
4
Power-Down Input and VOUT (V)
3
-40
-50
-6
-9
-60
-12
-70
-15
-80
-18
-90
1M
10M
100M
Frequency (Hz)
1G
10G
1M
Figure 29.
12
10M
100M
Frequency (Hz)
1G
Figure 30.
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TYPICAL CHARACTERISTICS (continued)
At TA = +25°C, VS+ = +5V, VOCM = +2.5V, VOUT = 2VPP, RL = 400Ω differential, G = +6dB, differential input and output, and
input and output pins referenced to midsupply, unless otherwise noted. Measured using EVM as discussed in Test Circuits
section.
VOCM INPUT BIAS CURRENT
vs VOCM INPUT VOLTAGE
NOISE FIGURE
vs FREQUENCY
20
VO = 200mVPP
150
18
100W Differential Source
16
100
Noise Figure (dB)
VOCM Input Bias Current (mA)
200
50
0
-50
14
12
10
8
6
-100
4
-150
2
-200
2.1
2.3
2.5
2.7
VOCM Input Voltage (V)
2.9
0
10M
Figure 31.
100M
200M
Frequency (Hz)
Figure 32.
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TEST CIRCUITS
OVERVIEW
The standard THS770006 evaluation module (EVM) is used for testing the typical performance shown in the
Typical Characteristics, with changes as noted below. The EVM schematic is shown in Figure 33. The signal
generators and analyzers used for most tests have single-ended 50Ω input and output impedance. The
THS770006 EVM is configured to convert to and from a differential 50Ω impedance by using RF transformers or
baluns (CX2156NL from Pulse, supplied as a standard configuration of the EVM). For line input termination, two
49.9Ω resistors (R5 and R6) are placed to ground on the input transformer output pins (terminals 1 and 3). In
combination with the 100Ω input impedance of the device, the total impedance seen by the line is 50Ω.
A resistor network is used on the amplifier output to present various loads (RL) and maintain line output
termination to 50Ω. Depending on the test conditions, component values are changed as shown in Table 1, or as
otherwise noted. As a result of the voltage divider on the output formed by the load component values, the
amplifier output is attenuated. The Loss column in Table 1 shows the attenuation expected from the resistor
divider. The output transformer causes slightly more loss, so these numbers are approximate.
Table 1. Load Component Values (1)
(1)
14
LOAD RL
R15 AND R17
R16
LOSS
100Ω
25Ω
Open
6dB
200Ω
86.6Ω
69.8Ω
16.8dB
400Ω
187Ω
57.6Ω
25.5dB
1kΩ
487Ω
52.3Ω
31.8dB
The total load includes 50Ω termination by the test equipment. Components are chosen to achieve load and 50Ω line termination
through a 1:1 transformer.
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Figure 33. THS770006IRGE EVM Schematic
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TEST DESCRIPTIONS
The following sections describe how the tests were performed, as well as the EVM circuit modifications that were
made (if any). Modifications made for test purposes include changing capacitors to resistors, resistors to
capacitors, the shorting/opening of components, etc., as noted. Unless otherwise noted, C1, C2, C9, and C13
are all changed to 0.1µF.
Frequency Response: 200mVPP, 2VPP, 3Vpp
This test is run with and without transformers in the signal path.
For tests with transformers, the standard EVM is used and only the gain magnitude is shown. A network analyzer
is connected to the input and output of the EVM with 50Ω coaxial cables and set to measure the forward transfer
function (s21). The input signal frequency is swept with the signal level set for the desired output amplitude. The
use of transformers gives better magnitude response that correlates best with detailed design simulation in terms
of peaking in the response due to better control of parasitic capacitance at the device output pins, but also
results in excess phase shift. So only magnitude is plotted.
For tests without transformers, the standard EVM is used, with the gain magnitude and phase shown. A network
analyzer is connected to the input of the EVM with 50Ω coaxial cable, the output is terminated with a 50Ω load,
and a high impedance differential probe is used for the measurement. The analyzer is set to measure the
forward transfer function (s21). The analyzer with a probe input is calibrated at the input pins of the device and
signal is measured at the output pin, thus effectively removing the transformers from the transfer function. The
input signal frequency is swept with signal level set for desired output amplitude. Not using transformers gives
better phase response that correlates best with detailed design simulations, but as a result of extra parasitic
capacitance at the device output pins gives significantly more peaking in the magnitude response. The –3dB
points of the magnitude response measured without transformers correlates better with measured slew rate, so
both magnitude and phase are plotted.
s-Parameters: s11, s22, and s12
The standard EVM is used with both R15 and R17 = 24.9Ω, and R16 = open, to test the input return loss, output
return loss, and reverse isolation. A network analyzer is connected to the input and output of the EVM with 50Ω
coaxial cables and set to measure the appropriate transfer function: s11, s22, or s12. Note the transformers are
included in the signal chain in order to retrieve proper measurements with single-ended test equipment. The
impact is minimal from 10MHz to 200MHz, but further analysis is required to fully de-embed the respective
effects.
Frequency Response with Capacitive Load
The standard EVM is used with R15 and R17 = RO, R16 = CLOAD, C9 and C13 = 953Ω, R21 = open, T2
removed, and jumpers placed across terminals 3 to 4 and 1 to 6. A network analyzer is connected to the input
and output of the EVM with 50Ω coaxial cables and set to measure the forward transfer function (s21). Different
values of load capacitance are placed on the output (at R16) and the output resistor values (R15 and R17)
changed until an optimally flat frequency response is achieved with maximum bandwidth.
Distortion
The standard EVM is used for measurement of single-tone harmonic distortion and two-tone intermodulation
distortion. For differential distortion measurements, the standard EVM is used with no modification. For
single-ended input distortion measurements, the standard EVM is used with with T1 removed and jumpers
placed across terminals 3 to 4 and 1 to 6, and R5 and R6 = 100Ω. A signal generator is connected to the J1
input of the EVM with 50Ω coaxial cables, with filters inserted inline to reduce distortion from the generator. The
J3 output of the EVM is connected with 50Ω coaxial cables to a spectrum analyzer to measure the
fundamental(s) and distortion products.
Noise Figure
The standard EVM is used with T1 changed to a 1:2 impedance ratio transformer (Mini-Circuits ADT2), R15 and
R17 = 24.9Ω, and R5, R6, and R16 = open. A noise figure analyzer is connected to the input and output of the
EVM with 50Ω coaxial cables. The noise figure analyzer provides a 50Ω (noise) source so that the data are
adjusted to refer to a 100Ω source.
16
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Transient Response, Slew Rate, Overdrive Recovery
The standard EVM is used with T1 and T2 removed and jumpers placed across terminals 3 to 4 and 1 to 6; R15,
R17, and R25 = 49.9Ω; C1, C2, C9, and C13 = 0Ω; and R5, R6, R16, and R21 = open. A differential waveform
generator is connected to the input of the EVM with 50Ω coaxial cables at J1 and J2. The differential output at J3
and J4 is connected with 50Ω coaxial cables to an oscilloscope to measure the outputs. Waveform math in the
oscilloscope is used to combine the differential output of the device.
Power-Down
The standard EVM is used with T1 and T2 removed, jumpers placed across terminals 3 to 4 and 1 to 6, R15 and
R17 = 49.9Ω, C9 and C13 = 0Ω, and R5, R6, R16, and R21 = open. A waveform generator is connected to the
power-down input of the EVM with a 50Ω coaxial cable at J8. The differential output at J3 and J4 is connected
with 50Ω coaxial cables to an oscilloscope to measure the outputs. J1 is left disconnected so that the output is
driven to the VOCM voltage when the device is active, and discharged through the resistive load on the output
when disabled. Both outputs are the same and only one is shown.
Differential Z-out
The standard EVM is used with R15 and R17 = 24.9Ω, and R16 = open. A network analyzer is connected to the
output of the EVM at J3 with 50Ω coaxial cable, both inputs are terminated with a 50Ω load, and a
high-impedance differential probe is used for the measurement. The analyzer is set to measure the forward
transfer function (s21). The analyzer with probe input is calibrated across the open resistor pads of R16 and the
signal is measured at the output pins of the device. The output impedance is calculated using the known resistor
values and the attenuation caused by R15 and R17.
Output Balance Error
The standard EVM is used with R15 and R17 = 100Ω, and R16 = 0Ω. A network analyzer is connected to the
input of the EVM with 50Ω coaxial cable, the output is left open, and a high-impedance differential probe is used
for the measurement. The analyzer is set to measure the forward transfer function (s21). The analyzer with probe
input is calibrated at the input pins of the device and the signal is measured from the shorted pads of R16 to
ground.
Common-Mode Rejection
The standard EVM is used with T1 removed and jumpers place across terminals 3 to 4, 1 to 6, and 1 to 3. A
network analyzer is connected to the input and output of the EVM with 50Ω coaxial cable and set to measure the
forward transfer function (s21).
VOCM Frequency Response
The standard EVM is used with T2 removed and jumpers across terminals 3 to 4 and 1 to 6; R10, R15, and
R17 = 49.9Ω; C3 and C4 = 0Ω; and R9, R16, and R21 = open. A network analyzer is connected to the VOCM
input of the EVM at J7 and output of the EVM with 50Ω coaxial cable, and set to measure the forward transfer
function (s21). The input signal frequency is swept with the signal level set for 200mV. Each output at J3 and J4
is measured as single-ended, and because both are the same, only one output is shown.
VOCM Slew Rate and Pulse Response
The standard EVM is used with T2 removed and jumpers across terminals 3 to 4 and 1 to 6; R10, R15, and
R17 = 49.9Ω; C9 and C13 = 0Ω; and C3, C4, R9, R16, and R21 = open. A waveform generator is connected to
the VOCM input of the EVM at J7 with 50Ω coaxial cable. The differential output at J3 and J4 is connected with
50Ω coaxial cable to an oscilloscope to measure the outputs. J1 is left disconnected so that the output is driven
to the VOCM voltage. Both outputs are the same, so only one is shown.
Input/Output Voltage Noise, Settling Time, and Power-Supply Rejection
These parameters are taken from simulation.
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THEORY OF OPERATION
GENERAL DESCRIPTION
The THS770006 is a fixed-gain of +6dB, wideband, fully-differential amplifier designed and optimized specifically
for driving 14-bit and 16-bit ADCs at input frequencies up to 200MHz. This device provides high bandwidth, low
distortion, and low noise, which are critical parameters in high-speed data acquisition systems that require very
high dynamic range, such as wireless base stations and test and measurement applications. It also makes an
excellent differential amplifier for general-purpose, high-speed differential signal chain and short line-driver
applications. The device has an operating power-supply range of 4.75V to 5.5V. The THS770006 has proprietary
circuitry to provide very fast recovery from overdrive conditions and has a power-down mode for power saving.
The THS770006 is offered in a Pb-free (RoHS compliant) and green, QFN-24 thermally-enhanced package. It is
characterized for operation over the industrial temperature range of –40°C to +85°C.
The amplifier uses two negative-feedback loops. One is for the primary differential amplifier and the other
controls the common-mode operation.
Primary Differential Amplifier
The primary amplifier of the THS770006 is a fully-differential op amp with on-chip gain setting resistors (RF =
100Ω and RG = 50Ω) that fix the differential gain at 2V/V, or 6dB, by use of negative feedback.
VOCM Control Loop
The output common-mode voltage is controlled through a second negative-feedback loop. The output
common-mode voltage is internally sensed and compared to the VOCM pin. The loop then works to drive the
difference, or error voltage, to zero in order to maintain the output common-mode voltage = VOCM (within the loop
gain and bandwidth of the loop). For more details on fully-differential amplifier theory and use, see application
report SLOA054, Fully-Differential Amplifiers, available for download from www.ti.com.
OPERATION
Differential to Differential
The THS770006 is a fixed gain of 6dB, fully-differential amplifier that can be used to amplify differential input
signals to differential output signals. A basic block diagram of the circuit is shown in Figure 34. The differential
input to differential output configuration gives the best performance; the signal source and load should be
balanced.
100W
Differential
Input
Differential
Output
50W
VIN-
VOUT+
VIN+
VOUT50W
100W
THS770006
Figure 34. Differential Input to Differential Output Amplifier
18
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Single-Ended to Differential
The THS770006 can be used to amplify and convert single-ended input signals to differential output signals. A
basic block diagram of the circuit is shown in Figure 35. The gain from the single-ended input to the differential
output is 6dB. In order to maintain proper balance in the amplifier and avoid offsets at the output, the alternate
input must be biased and the impedance matched to the signal input. For example, if a 50Ω source biased to
2.5V provides the input, the alternate input should be tied to 2.5V through 50Ω. If a 50Ω source is ac-coupled to
the input, the alternate input should be ac-coupled to ground through 50Ω. Note that the ac coupling should
provide a similar frequency response to balance the gain over frequency.
VREF
100W
Bias and
Impedance
Match
Differential
Output
50W
VOUT+
Single-Ended
Input
VOUT50W
100W
THS770006
Figure 35. Single-Ended Input to Differential Output Amplifier
Setting the Output Common-Mode Voltage
The VOCM input controls the output common-mode voltage. VOCM has no internal biasing network and must be
driven by an external source or resistor divider network to the positive power supply. In ac-coupled applications,
the VOCM input impedance and bias current are not critical, but in dc-coupled applications where more accuracy
is desired, the input bias current of the pin should be considered. For best harmonic distortion with VOUT = 3VPP,
the VOCM input should be maintained within the operating range of 2.25V to 2.75V. The VOCM input voltage can
be operated outside this range if lower output swing is used or distortion degradation is allowed, and increased
bias current into the pin is acceptable. For more information, see Figure 12 and Figure 31. It is recommended to
use a 0.1µF decoupling capacitor from the VOCM pin to ground to prevent noise and other spurious signals from
coupling into the common-mode loop of the amplifier.
Input Common-Mode Voltage Range
The THS770006 is designed primarily for ac-coupled operation. With input dc blocking, the input common-mode
voltage of the device is driven to the same voltage as VOCM by the outputs. Therefore, as long as the VOCM input
is maintained within the operating range of 2.25V to 2.75V, the input common-mode of the main amplifier is also
maintained within its linear operating range of 2.25V to 2.75V. If the device is used with dc coupled input, the
driving source needs to bias the input to its linear operating range of 2.25V to 2.75V for proper operation.
Operation with Split Supply ±2.5V
The THS770006 can be operated using a split ±2.5V supply. In this case, VS+ is connected to +2.5V, and GND
(and any other pin noted to be connected to GND) is connected to –2.5V. As with any device, the THS770006 is
impervious to what the user decides to name the levels in the system. In essence, it is simply a level shift of the
power pins by –2.5V. If everything else is level-shifted by the same amount, the device sees no difference. With
a ±2.5V power supply, the VOCM range is 0V ±0.25V; therefore, power-down levels are –2.5V = on and +2.5V =
off, and input and output voltage ranges are symmetrical about 0V. This design has certain advantages in
systems where signals are referenced to ground, and as noted in the following section, for driving ADCs with low
input common-mode voltage requirements in dc-coupled applications.
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Driving Capcitive Loads
The THS770006 is tested as described previously, with the data shown in the typical graphs. As a result of the
fixed gain architecture of the device, the only practical means to avoid stability problems such as
overshoot/ringing, gain peaking, and oscillation when driving capacitive loads is to place small resistors in series
with the outputs (RO) to isolate the phase shift caused by the capacitive load from the feedback loop of the
amplifier. The Typical Characteristics graphs show recommended values for an optimally flat frequency response
with maximum bandwidth. Smaller values of RO can be used if more peaking is allowed, and larger values can
be used to reduce the bandwidth.
Driving ADCs
The THS770006 is designed and optimized for the highest performance to drive differential input ADCs.
Figure 36 shows a generic block diagram of the THS770006 driving an ADC. The primary interface circuit
between the amplifier and the ADC is usually a filter of some type for antialias purposes, and provides a means
to bias the signal to the input common-mode voltage required by the ADC. Filters range from single-order real
RC poles to higher-order LC filters, depending on the requirements of the application. Output resistors (RO) are
shown on the amplifier outputs to isolate the amplifier from any capacitive loading presented by the filter.
100W
50W
VOUT+
RO
VIN-
Filter
and
Bias
VOCM
VIN+
VOUT-
50W
100W
RO
AIN+
ADC
AIN-
CM
VOCM
THS770006
Figure 36. Generic ADC Driver Block Diagram
The key points to consider for implementation are described in the following three subsections.
SNR Considerations
The signal-to-noise ratio (SNR) of the amplifier and filter can be calculated from the amplitude of the signal and
the bandwidth of the filter. The noise from the amplifier is band-limited by the filter with the equivalent brick-wall
filter bandwidth. The amplifier and filter noise can be calculated using the following equations:
SNRAMP+FILTER = 10 × log
V2O
e
= 20 × log
2
FILTEROUT
VO
eFILTEROUT
Where:
eFILTEROUT = eNAMPOUT • √ENB
eNAMPOUT = the output noise density of the THS770006 (3.4nV/√Hz)
ENB = the brick-wall equivalent noise bandwidth of the filter
VO is the amplifier output signal.
(1)
For example, with a first-order (N = 1) band-pass or low-pass filter with 30MHz cutoff, the ENB is 1.57 • f–3dB =
1.57 • 30MHz = 47.1MHz. For second-order (N = 2) filters, the ENB is 1.22 • f–3dB. As the filter order increases,
the ENB approaches f–3dB (N = 3 → ENB = 1.15 • f–3dB; N = 4 → ENB = 1.13 • f–3dB). Both VO and eFILTEROUT are
in RMS voltages. For example, with a 2VPP (0.707VRMS) output signal and 30MHz first-order filter, the SNR of the
amplifier and filter is 70.7dB with eFILTEROUT = 3.4nV/√Hz • √47.1MHz = 23μVRMS.
The SNR of the amplifier, filter, and ADC sum in RMS fashion, as shown in Equation 2 (SNR values in dB):
SNRSYSTEM = -20 × log
-SNRAMP+FILTER
10
10
-SNRADC
+ 10
10
(2)
20
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This formula shows that if the SNR of the amplifier and filter equals the SNR of the ADC, the combined SNR is
3dB lower (worse). Thus, for minimal impact (< 1dB) on the ADC SNR, the SNR of the amplifier and filter should
be ≥ 10dB greater than the ADC SNR. The combined SNR calculated in this manner is usually accurate to within
±1dB of actual implementation.
SFDR Considerations
The SFDR of the amplifier is usually set by second-order or third-order harmonic distortion for single-tone inputs,
and by second-order or third-order intermodulation distortion for two-tone inputs. Harmonics and second-order
intermodulation distortion can be filtered to some degree, but third-order intermodulation spurs cannot be filtered.
The ADC generates the same distortion products as the amplifier, but as a result of the sampling and clock
feedthrough, additional spurs (not linearly related to the input signal) are included.
When the spurs from the amplifier and filter are known, each individual spur can be directly added to the same
spur from the ADC, as shown in Equation 3, to estimate the combined spur (spur amplitudes in dBc):
HDxSYSTEM = -20 × log 10
-HDxADC
-HDxAMP+FILTER
20
+ 10
20
(3)
This calculation assumes the spurs are in phase, but usually provides a good estimate of the final combined
distortion.
For example, if the spur of the amplifier and filter equals the spur of the ADC, then the combined spur is 6dB
higher. To minimize the amplifier contribution (< 1dB) to the overall system distortion, it is important that the spur
from the amplifier and filter be ~15dB better than the converter. The combined spur calculated in this manner is
usually accurate to within ±6dB of actual implementation; however, higher variations have been observed as a
result of phase shift in the filter, especially in second-order harmonic performance.
This worst-case spur calculation assumes that the amplifier/filter spur of interest is in phase with the
corresponding spur in the ADC, such that the two spur amplitudes can be added linearly. There are two
phase-shift mechanisms that cause the measured distortion performance of the amplifier-ADC chain to deviate
from the expected performance calculated using Equation 3: common-mode phase shift and differential phase
shift.
Common-mode phase shift is the phase shift seen equally in both branches of the differential signal path
including the filter. Common-mode phase shift nullifies the basic assumption that the amplifier/filter and ADC spur
sources are in phase. This phase shift can lead to better performance than predicted as the spurs become phase
shifted, and there is the potential for cancellation as the phase shift reaches 180°. However, there is a significant
challenge in designing an amplifier-ADC interface circuit to take advantage of common-mode phase shift for
cancellation: the phase characteristic of the ADC spur sources are unknown, thus the necessary phase shift in
the filter and signal path for cancellation is also unknown.
Differential phase shift is the difference in the phase response between the two branches of the differential filter
signal path. Differential phase shift in the filter as a result of mismatched components caused by nominal
tolerance can severely degrade the even-order distortion of the amplifier-ADC chain. This effect has the same
result as mismatched path lengths for the two differential traces, and causes more phase shift in one path than
the other. Ideally, the phase response over frequency through the two sides of a differential signal path are
identical, such that even-order harmonics remain optimally out of phase and cancel when the signal is taken
differentially. However, if one side has more phase shift than the other, then the even-order harmonic
cancellation is not as effective.
Single-order RC filters cause very little differential phase shift with nominal tolerances of 5% or less, but
higher-order LC filters are very sensitive to component mismatch. For instance, a third-order Butterworth
bandpass filter with 100MHz center frequency and 20MHz bandwidth shows up to a 20° differential phase
imbalance in a Spice Monte Carlo analysis with 2% component tolerances. Therefore, while a prototype may
work, production variance is unacceptable. In ac-coupled applications that require second- and higher-order
filters between the THS770006 and ADC, a transformer or balun is recommended at the ADC input to restore the
phase balance. For dc-coupled applications where a transformer or balun at the ADC input cannot be used, it is
recommended to use first- or second-order filters to minimize the effect of differential phase shift because of
component tolerance.
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ADC Input Common-Mode Voltage Considerations—AC-Coupled Input
The input common-mode voltage range of the ADC must be respected for proper operation. In an ac-coupled
application between the amplifier and the ADC, the input common-mode voltage bias of the ADC is
accomplished in different ways depending on the ADC. Some ADCs use internal bias networks such that the
analog inputs are automatically biased to the required input common-mode voltage if the inputs are ac-coupled
with capacitors (or if the filter between the amplifier and ADC is a bandpass filter). Other ADCs supply their
required input common-mode voltage from a reference voltage output pin (often called CM or VCM). With these
ADCs, the ac-coupled input signal can be re-biased to the input common-mode voltage by connecting resistors
from each input to the CM output of the ADC, as Figure 37 illustrates. However, the signal is attenuated because
of the voltage divider created by RCM and RO.
RO
RCM
AIN+
Amp
ADC
AIN-
RCM
CM
RO
Figure 37. Biasing AC-Coupled ADC Inputs Using the ADC CM Output
The signal can be re-biased when ac coupling; thus, the output common-mode voltage of the amplifier is a don’t
care for the ADC.
ADC Input Common-Mode Voltage Considerations—DC-Coupled Input
DC-coupled applications vary in complexity and requirements, depending on the ADC. One typical requirement is
resolving the mismatch between the common-mode voltage of the driving amplifier and the ADC. Devices such
as the ADS5424 require a nominal 2.4V input common-mode, while others such as the ADS5485 require a
nominal 3.1V input common-mode; still others such as the ADS6149and the ADS4149 require 1.5V and 0.95V,
respectively. As shown in Figure 38, a resistor network can be used to perform a common-mode level shift. This
resistor network consists of the amplifier series output resistors and pull-up or pull-down resistors to a reference
voltage. This resistor network introduces signal attenuation that may prevent the use of the full-scale input range
of the ADC. ADCs with an input common-mode closer to the typical 2.5V THS770006 output common-mode are
easier to dc-couple, and require little or no level shifting.
VREF
VAMP+
RO
RP
ADC
VADC+
Amp
RIN
VAMP-
RO
RP
CIN
VADC-
VREF
Figure 38. Resistor Network to DC Level-Shift Common-Mode Voltage
For common-mode analysis of the circuit in Figure 38, assume that VAMP± = VOCM and VADC± = VCM (the
specification for the ADC input common-mode voltage). VREF is chosen to be a voltage within the system higher
than VCM (such as the ADC or amplifier analog supply) or ground, depending on whether the voltage must be
pulled up or down, respectively; RO is chosen to be a reasonable value, such as 24.9Ω. With these known
values, RP can be found by using Equation 4:
RP = RO
22
VADC - VREF
VAMP - VADC
(4)
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Shifting the common-mode voltage with the resistor network comes at the expense of signal attenuation.
Modeling the ADC input as the parallel combination of a resistance (RIN) and capacitance (CIN) using values
taken from the ADC data sheet, the approximate differential input impedance (ZIN) for the ADC can be calculated
at the signal frequency. The effect of CIN on the overall calculation of gain is typically minimal and can be ignored
for simplicity (that is, ZIN = RIN). The ADC input impedance creates a divider with the resistor network; the gain
(attenuation) for this divider can be calculated by Equation 5:
GAIN =
2RP || ZIN
2RO + 2RP || ZIN
(5)
With ADCs that have internal resistors that bias the ADC input to the ADC input common-mode voltage, the
effective RIN is equal to twice the value of the bias resistor. For example, the ADS5485 has a 1kΩ resistor tying
each input to the ADC VCM; therefore, the effective differential RIN is 2kΩ.
The introduction of the RP resistors also modifies the effective load seen by the amplifier. Equation 6 shows the
effective load seen by the amplifier:
RL = 2RO + 2RP || ZIN
(6)
The RP resistors act in parallel to the ADC input such that the effective load (output current) seen by the amplifier
is increased. Higher current loads limit the THS770006 differential output swing.
Using the gain and knowing the full-scale input of the ADC (VADC
with the network can be calculated using Equation 7:
V
VAMP PP = ADC FS
GAIN
FS),
the required amplitude to drive the ADC
(7)
Using the ADC examples given previously, Table 2 shows sample calculations of the value of RP and VAMP FS for
full-scale drive, and then for –1dB (often times, the ADC drive is backed off from full-scale in applications, so
lower amplitudes may be acceptable). All voltage values are in volts, resistor values in ohms (the nearest
standard value should be used), and gain values are as noted. Table 2 does not include the ADS5424 because
no level shift is required with this device.
Table 2. Example RP for Various ADCs
ADC RIN
(Ω)
RO
(Ω)
RP
(Ω)
GAIN
(V/V)
GAIN
(dB)
VADC FS
(VPP)
VAMP PP
FOR
0dBFS(V
PP)
5
2k
50
158.3
0.73
–2.72
3
4.10
3.66
0
6k (1)
50
75.0
0.59
–4.56
2
3.38
3.01
0
5k (1)
50
30.6
0.38
–8.49
2
5.31
4.74
(1)
50
81.6
0.61
–4.31
2
3.28
2.93
5k (1)
50
213.2
0.79
–2.05
2
2.53
2.26
ADC
VAMP
(VDC)
VADC
(VDC)
VREF
(VDC)
ADS5485
2.5
3.1
ADS6149
2.5
1.5
2.5
0.95
ADS4149
(1)
(2)
(2)
0.95
2.5
0 (2)
0.95
5
0
5k
VAMP PP
FOR –1d
BFS
(VPP)
At 70MHz.
THS770006 with ±2.5V supply.
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The calculated values for the ADS5485 give the lowest attenuation. As a result of the high VFS of 3VPP, 3.66VPP
is required from the amplifier to drive to –1dBFS. Performance of the THS770006 is still very good up to 130MHz
at this level, but for best performance, back off further from full-scale and consider trading reduced SNR
performance for better SFDR performance.
The values calculated for the ADS6149 show reasonable design targets and should work with good performance.
Note the ADS6149 does not have buffered inputs, and the inputs have equivalent resistive impedance that varies
with sampling frequency. The calculation shown in Table 2 uses a value of 70MHz for RIN, taken from the
ADS6149 data sheet.
The values calculated for the low input common-mode voltage of the ADS4149 result in a large attenuation of the
amplifier signal leading to 5.31VPP being required for full-scale ADC drive. This amplitude is greater than the
maximum capability of the device. With a single +5V supply, the THS770006 is not suitable to drive this ADC in
dc-coupled applications unless the ADC input is backed off towards –6dBFS.
Another option is to operate the THS770006 with a split ±2.5V supply. The RP and gains are shown in the last
two rows of Table 2 for pull-up voltages of 2.5V and 5V. For this situation, if the +2.5V is used as the pull-up
reference voltage, only 2.93VPP is required for the –1dBFS input to the ADS4149. If a 5V reference is used, only
2.26VPP is required to reach the –1dBFS input to the ADS4149. See the Operation with Split Supply ±2.5V
section for more detail. Note that, similar to the ADS6149, the ADS4149 does not have buffered inputs and the
inputs have equivalent resistive impedance that varies with sampling frequency. The RIN value at 70MHz taken
from the ADS4149 data sheet was used in the calculation.
As with any design, testing is recommended to validate whether it meets the specific design goals.
24
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APPLICATION INFORMATION
THS770006 DRIVING 16-BIT ADC
To illustrate the performance of the THS770006 as an ADC driver, the device is tested with a 16-bit ADC.
TESTING WITH AN AC-COUPLED BANDPASS FILTER
For testing purposes, a 30MHz, third-order Butterworth bandpass filter with center frequency at 100MHz is
designed. The design target for the source impedance is 40Ω differential, and for load impedance is 400Ω
differential. Therefore, approximately 1dB insertion loss is expected in the pass-band, requiring the amplifier
output amplitude to be 2.5VPP to drive the ADC to –1dBFS.
The output noise voltage specification for the THS770006 is 3.4 nV/√Hz. With 2.5VPP amplifier output voltage
swing and 30MHz bandwidth, the expected SNR from the amplifier + antialias filter is 93.5dB. When added in
combination with the 16-bit ADC, the expected total SNR is 75.1dBFS for the typical case.
Figure 39 shows the resulting FFT plot when driving the ADC to –1dBFS with a single-tone 100MHz sine wave,
and sampling at 125MSPS. Test results show 98dBc SFDR from the second-order harmonic and 75.6 dBFS
SNR; analysis of the plot is shown in Table 3 versus typical ADC specifications. The test results from circuit
board to circuit board shows over 10dB of variation in the second order harmonic and a balun is inserted
between the filter and ADC inputs to get repeatable performance. With balun, the minimum expected results
should be better than 90dBc SFDR and 75dBFS SNR.
Figure 39 shows the same circuit with a two-tone input at 96MHz and 100MHz. The near-in 3rd order
intermodulation terms are about -100dBc.
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0
-10
-20
dBFS
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
0
5
10
15
20 25 30 35 40 45
Frequency (MHz)
50
55
62.5
Figure 39. FFT Plot of THS770006 + 30MHz BPF + 16-Bit ADC with Single-Tone at 100MHz
Table 3. Analysis of FFT for THS770006 + BPF + 16-Bit ADC at 100MHz vs Typical ADC Specifications
ADC INPUT
SNR
THS770006 + BPF + 16-Bit ADC
–1dBFS
16-Bit ADC Only (typ)
–1dBFS
dBFS
CONFIGURATION
HD2
HD3
75.6dBFS
–98dBc
–107dBc
75.2dBFS
–100dBc
–100dBc
10
0
-10
-20
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
130
140
0
5
10 15
20
25 30 35 40
Frequency (MHz)
45
50
55
61
Figure 40. FFT Plot of THS770006 + 30MHz BPF + 16-Bit ADC with Two-Tone Input at 96MHz and 100MHz
26
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TESTING WITH AN AC-COUPLED LOW-PASS FILTER
For testing purposes, a 150MHz, first-order, low-pass filter is built. The design gives approximately 1.6dB
insertion loss at low frequency, requiring the amplifier signal be 2.7VPP in order to drive the ADC to –1dBFS.
With 2.7VPP amplifier output voltage swing and 180MHz (–3dB) bandwidth, the expected SNR from the amplifier
+ antialias filter is 84.4dB. When added in combination with the 16-bit, 130MSPS ADC, the total expected SNR is
74.7dBFS for the typical case. Note the frequency response is approximately –1dB at 100MHz, which requires
even higher amplitude for the following test.
dBFS
Figure 41 shows the resulting FFT plot when driving the ADC to –1dBFS with a 100MHz sine wave, and
sampling at 125MSPS. Test results showed 91dBc SFDR from second- and third-order harmonic and 73.1dBFS
SNR; analysis of the plot is shown in Table 4 versus typical ADC specifications. As a result of harmonic
attenuation and phase shift between the amplifier and ADC, harmonic performance is better than predicted from
the worst-case scenario described previously. Typical expected results should be approximately 90dBc SFDR
and 73dBFS SNR.
0
-10
-20
-30
-40
-50
-60
-70
-80
-90
-100
-110
-120
-130
0
5
10
15
20 25 30 35 40 45
Frequency (MHz)
50
55
62.5
Figure 41. FFT Plot of THS770006 + 180MHz LPF + 16-Bit ADC with Single-Tone at 100MHz
Table 4. Analysis of FFT for THS770006 + 180MHz LPF + 16-Bit ADC at 100MHz vs Typical ADC
Specifications
CONFIGURATION
ADC INPUT
SNR
HD2
HD3
THS770006 + BPF + 16-Bit ADC
–1dBFS
73.1dBFS
–91dBc
–91dBc
16-Bit ADC Only (typ)
–1dBFS
75.2dBFS
–100dBc
–100dBc
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EVM AND LAYOUT RECOMMENDATIONS
Figure 33 is the THS770006RGE EVM schematic, and Figure 42 through Figure 45 show the layout details of the
EVM PCB. Table 5 is the bill of materials for the EVM as supplied from TI. It is recommended to follow the layout
of the external components as close as possible to the amplifier, ground plane construction, and power routing.
General layout guidelines are:
1. Place a 2.2µF to 10µF capacitor on each supply pin within 2 inches from the device. It can be shared among
other op amps.
2. Place a 0.01µF to 0.1µF capacitor on each supply pin to ground as close as possible to the device.
Placement within 1mm of the device supply pins ensures best performance.
3. Keep input and output traces as short as possible to minimize parasitic capacitance and inductance. Doing
so reduces unwanted characteristics such as reduced bandwidth and peaking in the frequency response,
overshoot, and ringing in the pulse response, and results in a more stable design.
4. To reduce parasitic capacitance, ground plane and power-supply planes should be removed from device
input pins and output pins.
5. The VOCM pin must be biased to a voltage between 2.25V to 2.75V for proper operation. Place a 0.1µF to
0.22µF capacitor to ground as close as possible to the device to prevent noise coupling into the
common-mode.
6. For best performance, drive circuits and loads should be balanced and biased to keep the input and output
common-mode voltage between 2.25V to 2.75V. AC-coupling is a simple way to achieve this performance.
7. The THS770006 is provided in a thermally enhanced PowerPAD™ package. The package is constructed
using a downset leadframe on which the die is mounted. This arrangement results in low thermal resistance
to the thermal pad on the underside of the package. Excellent thermal performance can be achieved by
following the guidelines in TI application reports SLMA002, PowerPAD™ Thermally-Enhanced Package and
SLMA004, PowerPAD™ Made Easy. For proper operation, the thermal pad on the bottom of the device must
be tied to the same voltage potential as the GND pin on the device.
Figure 42. EVM Layout: Top Layer
28
Figure 43. EVM Layout: Bottom Layer
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Figure 44. EVM Layout: Layer 2
Figure 45. EVM Layout: Layer 3
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Table 5. THS770006RGE EVM Bill of Materials
ITEM
30
DESCRIPTION
SMD
SIZE
REFERENCE
DESIGNATOR
QTY
MANUFACTURER
PART NUMBER
DISTRIBUTOR
PART NUMBER
1
CAP, 10.0uF, CERAMIC, X7R, 10V
1206
C4, C5, C6
3
(TDK) C3216X7R1A106K
(DIGI-KEY) 445-4043-1-ND
2
CAP, 0.1uF, CERAMIC, X7R, 16V
0603
C7, C8
2
(AVX) 0603YC104KAT2A
(DIGI-KEY) 478-1239-1-ND
3
CAP, 0.01uF, CERAMIC, X7R, 16V
0402
C10, C11
2
(AVX) 0402YC103KAT2A
(DIGI-KEY) 478-1114-1-ND
4
CAP, 100pF, CERAMIC, NPO, 50V
0402
C12
1
(AVX) 04025A101KAT2A
(DIGI-KEY) 478-4979-1-ND
5
(AVX) 04025C102KAT2A
(DIGI-KEY) 478-1101-1-ND
5
CAP, 1000pF, CERAMIC, X7R, 50V
0402
C1, C2, C3, C9,
C13
6
OPEN
0402
R11, R12, R13,
R14
4
7
RESISTOR, 0 OHM
0402
R4, R21
2
(PANASONIC) ERJ-2GE0R00X
(DIGI-KEY) P0.0JCT-ND
8
RESISTOR, 49.9 OHM, 1/10W, 1%
0402
R5, R6
2
(PANASONIC) ERJ-2RKF49R9X
(DIGI-KEY) P49.9LCT-ND
9
RESISTOR, 57.6 OHM, 1/10W, 1%
0402
R16
1
(PANASONIC) ERJ-2RKF57R6X
(DIGI-KEY) P57.6LCT-ND
10
RESISTOR, 187 OHM, 1/10W, 1%
0402
R15, R17
2
(PANASONIC) ERJ-2RKF1870X
(DIGI-KEY) P187LCT-ND
11
RESISTOR, 1K OHM, 1/10W, 1%
0402
R9, R10
2
(PANASONIC) ERJ-2RKF1001X
(DIGI-KEY) P1.00KLCT-ND
12
RESISTOR, 10K OHM, 1/10W, 1%
0603
R25, R26
2
(PANASONIC) ERJ-3EKF1002V
(DIGI-KEY) P10.0KHCT-ND
13
TRANSFORMER, BALUN
T1, T2
2
(PULSE) CX2156NL
(DIGI-KEY) 553-1499-ND
14
JACK, BANANA RECEPTANCE, 0.25"
DIA. HOLE
J5, J6
3
(SPC) 15459
(NEWARK) 79K5034
15
CONNECTOR, SMA PCB JACK
(NEWARK) 34C8151
16
CONNECTOR, EDGE, SMA PCB JACK
17
HEADER, 0.1" CTRS, 0.025" SQ. PINS
18
SHUNTS
19
TEST POINT, RED
20
TEST POINT, BLACK
21
IC, THS770006
22
STANDOFF, 4-40 HEX, 0.625" LENGTH
4
(KEYSTONE) 1808
(DIGI-KEY) 1808K-ND
23
SCREW, PHILLIPS, 4-40, .250"
4
PMSSS 440 0025 PH
(DIGI-KEY) H703-ND
24
BOARD, PRINTED CIRCUIT
3 POS.
J7, J8
2
(AMPHENOL) 901-144-8RFX
J1, J2, J3, J4
4
(JOHNSON) 142-0701-801
(NEWARK) 90F2624
JP1, JP2
2
(SULLINS) PBC36SAAN
(DIGI-KEY) S1011E-36-ND
JP1, JP2
2
(SULLINS) SSC02SYAN
(DIGI-KEY) S9002-ND
TP3
1
(KEYSTONE) 5000
(DIGI-KEY) 5000K-ND
TP1, TP2
2
(KEYSTONE) 5001
(DIGI-KEY) 5001K-ND
U1
1
(TI) THS770006RGE
(TI) EDGE# 6515711 REV.A
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REVISION HISTORY
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (May 2011) to Revision B
Page
•
Replaced "ADS5493" with "16-Bit Driver" in figure and figure title ....................................................................................... 1
•
Deleted ADS5493 from Related Devices table ..................................................................................................................... 1
•
Deleted reference to ADS5493 from text ............................................................................................................................ 24
•
Replaced references to "ADS5493" with "16-bit ADC" in Application Information section ................................................. 25
Changes from Original (July 2010) to Revision A
Page
•
Changed large-signal bandwidth from 675 to 780 in Electrical Characteristics ................................................................... 3
•
Added new row for Input Common-Mode Range parameter in Electrical Characteristics ................................................... 4
•
Added Figure 13, Harmonic Distortion vs Input Common-Mode Range ............................................................................ 10
•
Changed SNR Considerations section. .............................................................................................................................. 20
•
Changed SFDR Considerations section ............................................................................................................................. 21
•
Changed ADC Input Common-Mode Voltage Considerations section to show ac-coupled input. ..................................... 22
•
Added new subsection titled ADC Input Common-Mode Voltage Considerations—DC-Coupled Input ............................. 22
•
Deleted figure and last two paragraphs from the THS770006 Driving ADS5493 section .................................................. 25
•
Deleted text from first sentence in the Testing the ADS5493 with an AC-Coupled Bandpass Filter section ..................... 25
•
Deleted text from first paragraph in the Testing the ADS5493 with an AC-Coupled Low-Pass Filter section ................... 27
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Evaluation Board/Kit Important Notice
Texas Instruments (TI) provides the enclosed product(s) under the following conditions:
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR EVALUATION PURPOSES
ONLY and is not considered by TI to be a finished end-product fit for general consumer use. Persons handling the product(s) must have
electronics training and observe good engineering practice standards. As such, the goods being provided are not intended to be complete
in terms of required design-, marketing-, and/or manufacturing-related protective considerations, including product safety and environmental
measures typically found in end products that incorporate such semiconductor components or circuit boards. This evaluation board/kit does
not fall within the scope of the European Union directives regarding electromagnetic compatibility, restricted substances (RoHS), recycling
(WEEE), FCC, CE or UL, and therefore may not meet the technical requirements of these directives or other related directives.
Should this evaluation board/kit not meet the specifications indicated in the User’s Guide, the board/kit may be returned within 30 days from
the date of delivery for a full refund. THE FOREGOING WARRANTY IS THE EXCLUSIVE WARRANTY MADE BY SELLER TO BUYER
AND IS IN LIEU OF ALL OTHER WARRANTIES, EXPRESSED, IMPLIED, OR STATUTORY, INCLUDING ANY WARRANTY OF
MERCHANTABILITY OR FITNESS FOR ANY PARTICULAR PURPOSE.
The user assumes all responsibility and liability for proper and safe handling of the goods. Further, the user indemnifies TI from all claims
arising from the handling or use of the goods. Due to the open construction of the product, it is the user’s responsibility to take any and all
appropriate precautions with regard to electrostatic discharge.
EXCEPT TO THE EXTENT OF THE INDEMNITY SET FORTH ABOVE, NEITHER PARTY SHALL BE LIABLE TO THE OTHER FOR ANY
INDIRECT, SPECIAL, INCIDENTAL, OR CONSEQUENTIAL DAMAGES.
TI currently deals with a variety of customers for products, and therefore our arrangement with the user is not exclusive.
TI assumes no liability for applications assistance, customer product design, software performance, or infringement of patents or
services described herein.
Please read the User’s Guide and, specifically, the Warnings and Restrictions notice in the User’s Guide prior to handling the product. This
notice contains important safety information about temperatures and voltages. For additional information on TI’s environmental and/or
safety programs, please contact the TI application engineer or visit www.ti.com/esh.
No license is granted under any patent right or other intellectual property right of TI covering or relating to any machine, process, or
combination in which such TI products or services might be or are used.
FCC Warning
This evaluation board/kit is intended for use for ENGINEERING DEVELOPMENT, DEMONSTRATION, OR EVALUATION PURPOSES
ONLY and is not considered by TI to be a finished end-product fit for general consumer use. It generates, uses, and can radiate radio
frequency energy and has not been tested for compliance with the limits of computing devices pursuant to part 15 of FCC rules, which are
designed to provide reasonable protection against radio frequency interference. Operation of this equipment in other environments may
cause interference with radio communications, in which case the user at his own expense will be required to take whatever measures may
be required to correct this interference.
EVM Warnings and Restrictions
It is important to operate this EVM within the input voltage range of 0V to +5.5V and the output voltage range of 0V to +5.5V.
Exceeding the specified input range may cause unexpected operation and/or irreversible damage to the EVM. If there are questions
concerning the input range, please contact a TI field representative prior to connecting the input power.
Applying loads outside of the specified output range may result in unintended operation and/or possible permanent damage to the EVM.
Please consult the EVM User's Guide prior to connecting any load to the EVM output. If there is uncertainty as to the load specification,
please contact a TI field representative.
During normal operation, some circuit components may have case temperatures greater than +85°C. The EVM is designed to operate
properly with certain components above +85°C as long as the input and output ranges are maintained. These components include but are
not limited to linear regulators, switching transistors, pass transistors, and current sense resistors. These types of devices can be identified
using the EVM schematic located in the EVM User's Guide. When placing measurement probes near these devices during operation,
please be aware that these devices may be very warm to the touch.
Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265
Copyright © 2012, Texas Instruments Incorporated
32
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Copyright © 2010–2012, Texas Instruments Incorporated
Product Folder Link(s): THS770006
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
THS770006IRGER
ACTIVE
VQFN
RGE
24
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
THS7700
06IRGE
THS770006IRGET
ACTIVE
VQFN
RGE
24
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
THS7700
06IRGE
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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