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TL594
SLVS052I – APRIL 1988 – REVISED SEPTEMBER 2016
TL594 Pulse-Width-Modulation Control Circuit
1 Features
3 Description
•
•
The TL594 device incorporates all the functions
required in the construction of a pulse-widthmodulation (PWM) control circuit on a single chip.
Designed primarily for power-supply control, this
device offers the systems engineer the flexibility to
tailor the power-supply control circuitry to a specific
application.
1
•
•
•
•
•
•
Complete PWM Power-Control Circuitry
Uncommitted Outputs for 200-mA Sink or Source
Current
Output Control Selects Single-Ended or Push-Pull
Operation
Internal Circuitry Prohibits Double Pulse at Either
Output
Variable Dead Time Provides Control Over Total
Range
Internal Regulator Provides a Stable 5-V
Reference Supply Trimmed to 1%
Circuit Architecture Allows Easy Synchronization
Undervoltage Lockout (UVLO) for Low-VCC
Conditions
2 Applications
•
•
•
•
•
•
•
White Goods
Power Supplies: AC/DC, Isolated,
With PFC, > 90 W
Server PSUs
Solar Micro-Inverters
Power Supplies: AC/DC, Isolated,
No PFC, < 90 W
Power: Telecom/Server AC/DC Supplies
Solar Power Inverters
The TL594 device contains two error amplifiers, an
on-chip adjustable oscillator, a dead-time control
(DTC) comparator, a pulse-steering control flip-flop, a
5-V regulator with a precision of 1%, an undervoltage
lockout control circuit, and output control circuitry.
The uncommitted output transistors provide either
common-emitter or emitter-follower output capability.
Each device provides for push-pull or single-ended
output operation, with selection by means of the
output-control function. The architecture of these
devices prohibits the possibility of either output being
pulsed twice during push-pull operation. The
undervoltage lockout control circuit locks the outputs
off until the internal circuitry is operational.
Device Information
PART NUMBER
PACKAGE
BODY SIZE (NOM)
TL594D
SOIC (16)
9.90 mm × 3.91 mm
TL594N
PDIP (16)
19.30 mm × 6.35 mm
TL594NS
SO (16)
10.30 mm × 5.30 mm
TL594PW
TSSOP (16)
5.00 mm × 4.40 mm
Block Diagram
OUTPUT CTRL
(see Function Table)
13
6
RT
5
CT
DTC
Oscillator
0.1 V
4
9
PWM
Comparator
11
10
+
1
í
IN+
INí
15
C2
E2
12
+
2
í
Reference
Regulator
3
VCC
Undervoltage
Lockout
Control
14
FEEDBACK
E1
Pulse-Steering
Flip-Flop
Error Amplifier 2
16
C1
C1
Error Amplifier 1
IN+ 1
2
INí
8
1D
DTC
Comparator
7
REF
GND
0.7 mA
Copyright © 2016, Texas Instruments Incorporated
For OUTPUT CTRL function, see Table 1.
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TL594
SLVS052I – APRIL 1988 – REVISED SEPTEMBER 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
8
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
4
5
6
6
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information .................................................
Electrical Characteristics...........................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Parameter Measurement Information .................. 7
Detailed Description .............................................. 9
8.1 Overview .................................................................. 9
8.2 Functional Block Diagram ......................................... 9
8.3 Feature Description ................................................ 10
8.4 Device Functional Modes ....................................... 12
9
Application and Implementation ........................ 13
9.1 Application Information............................................ 13
9.2 Typical Application .................................................. 13
10 Power Supply Recommendations ..................... 19
11 Layout................................................................... 19
11.1 Layout Guidelines ................................................. 19
11.2 Layout Example .................................................... 20
12 Device and Documentation Support ................. 21
12.1
12.2
12.3
12.4
12.5
12.6
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
21
21
21
21
21
21
13 Mechanical, Packaging, and Orderable
Information ........................................................... 21
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (January 2014) to Revision I
Page
•
Added Applications section, ESD Ratings table, Feature Description section, Device Functional Modes, Application
and Implementation section, Power Supply Recommendations section, Layout section, Device and Documentation
Support section, and Mechanical, Packaging, and Orderable Information section................................................................ 1
•
Changed values in the Thermal Information table from 73 to 73.5 (D), from 67 to 43.5 (N), from 64 to 73.6 (NS), and
from 108 to 101.5 (PW) .......................................................................................................................................................... 4
Changes from Revision G (January 2007) to Revision H
Page
•
Deleted Ordering Information table; see POA at the end of the data sheet........................................................................... 1
•
Updated document to new TI data sheet format - no specific changes ................................................................................. 1
•
Added ESD warning ............................................................................................................................................................... 1
2
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SLVS052I – APRIL 1988 – REVISED SEPTEMBER 2016
5 Pin Configuration and Functions
D, N, NS, or PW Package
16-Pin SOIC, PDIP, SO, or TSSOP
Top View
1IN+
1
16
2IN+
1IN±
2
15
2IN±
FEEDBACK
3
14
REF
DTC
4
13
OUTPUT CTRL
CT
5
12
VCC
RT
6
11
C2
GND
7
10
E2
C1
8
9
E1
Not to scale
Pin Functions
PIN
NO.
NAME
I/O
DESCRIPTION
1
1IN+
I
Noninverting input to error amplifier 1
2
1IN–
I
Inverting input to error amplifier 1
3
FEEDBACK
I
Input pin for feedback
4
DTC
I
Dead-time control comparator input
5
CT
—
Capacitor terminal used to set oscillator frequency
6
RT
—
Resistor terminal used to set oscillator frequency
7
GND
—
Ground
8
C1
O
Collector terminal of BJT output 1
9
E1
O
Emitter terminal of BJT output 1
10
E2
O
Emitter terminal of BJT output 2
11
C2
O
Collector terminal of BJT output 2
12
VCC
—
Positive supply
13
OUTPUT CTRL
I
Selects single-ended, parallel output, or push-pull operation
14
REF
O
5-V reference regulator output
15
2IN–
I
Inverting input to error amplifier 2
16
2IN+
I
Noninverting input to error amplifier 2
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SLVS052I – APRIL 1988 – REVISED SEPTEMBER 2016
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MAX
UNIT
Supply voltage, VCC (2)
MIN
41
V
Amplifier input voltage
VCC + 0.3
V
Collector output voltage
41
V
Collector output current
250
mA
150
°C
150
°C
Operating junction temperature, TJ
Storage temperature, Tstg
(1)
(2)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltage values, except differential voltages, are with respect to the network ground terminal.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
1000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
VCC
Supply voltage
VI
Amplifier input voltage
VO
Collector output voltage
MIN
MAX
7
40
UNIT
V
–0.3
VCC – 2
V
40
V
Collector output current (each transistor)
200
mA
Current into FEEDBACK terminal
0.3
mA
CT
Timing capacitor
RT
Timing resistor
fosc
Oscillator frequency
TA
Operating free-air temperature
0.47
10000
nF
1.8
500
kΩ
1
300
kHz
0
70
–40
85
TL594C
TL594I
°C
6.4 Thermal Information
TL594
THERMAL METRIC (1)
RθJA
N (PDIP)
NS (SO)
PW (TSSOP)
16 PINS
16 PINS
16 PINS
16 PINS
UNIT
73.5
43.5
73.6
101.5
°C/W
RθJC(top) Junction-to-case (top) thermal resistance
32.8
30.6
30.3
29.4
°C/W
RθJB
Junction-to-board thermal resistance
30.8
23.5
34.4
47.3
°C/W
ψJT
Junction-to-top characterization parameter
6.1
15.3
3.4
1.4
°C/W
ψJB
Junction-to-board characterization parameter
30.6
23.4
34.1
46.6
°C/W
(1)
4
Junction-to-ambient thermal resistance
D (SOIC)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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6.5 Electrical Characteristics
VCC = 15 V, over recommended operating free-air temperature range (unless otherwise noted)
TEST CONDITIONS (1)
PARAMETER
MIN
TYP (2)
MAX
4.95
5
5.05
2
25
mV
14
35
mV
2
10
mV/V
35
50
mA
mV
UNIT
REFERENCE
Output voltage (REF)
IO = 1 mA, TA = 25°C
Input regulation
VCC = 7 V to 40 V, TA = 25°C
Output regulation
IO = 1 mA to 10 mA, TA = 25°C
Output-voltage change with temperature
ΔTA = MIN to MAX
Short-circuit output current (3)
Vref = 0
10
V
AMPLIFIER (SEE Figure 3)
Input offset voltage, error amplifier
FEEDBACK = 2.5 V
2
10
Input offset current
FEEDBACK = 2.5 V
25
250
nA
Input bias current
FEEDBACK = 2.5 V
0.2
1
µA
Common mode input voltage,
error amplifier
VCC = 7 V to 40 V
Open-loop voltage amplification,
error amplifier
ΔVO = 3 V, RL = 2 kΩ, VO = 0.5 V to 3.5 V
Unity-gain bandwidth
VO = 0.5 V to 3.5 V, RL = 2 kΩ
Common mode rejection ratio,
error amplifier
VCC = 40 V, TA = 25°C
Output sink current, FEEDBACK
Output source current, FEEDBACK
0.3 to
VCC – 2
70
V
95
dB
800
kHz
65
80
dB
VID = –15 mV to –5 V, FEEDBACK = 0.5 V
0.3
0.7
mA
VID = 15 mV to 5 V, FEEDBACK = 3.5 V
–2
mA
OSCILLATOR, CT = 0.01 µF, RT = 12 kΩ (SEE Figure 4)
Frequency
Standard deviation of frequency
(4)
All values of VCC, CT, RT, and TA constant
Frequency change with voltage
VCC = 7 V to 40 V, TA = 25°C
Frequency change with temperature (5)
ΔTA = MIN to MAX
10
kHz
100
Hz/kHz
1
Hz/kHz
50
Hz/kHz
–2
–10
µA
3
3.3
DEAD-TIME CONTROL (SEE Figure 4)
Input bias current
VI = 0 to 5.25 V
Maximum duty cycle, each output
DTC = 0 V
Input threshold voltage
0.45
Zero duty cycle
Maximum duty cycle
0
V
OUTPUT
VC = 40 V, VE = 0 V, VCC = 40 V
2
100
Collector off-state current
DTC and OUTPUT CTRL = 0 V, VC = 15 V,
VE = 0 V, VCC = 1 V to 3 V
4
200
Emitter off-state current
VCC = VC = 40 V, VE = 0
Collector-emitter saturation voltage
Output control input current
(1)
(2)
(3)
(4)
–100
Common emitter, VE = 0, IC = 200 mA
1.1
1.3
Emitter follower, VC = 15 V, IE = –200 mA
1.5
2.5
VI = Vref
3.5
µA
µA
V
mA
For conditions shown as MIN or MAX, use the appropriate value specified under recommended operating conditions.
All typical values, except for parameter changes with temperature, are at TA = 25°C.
Duration of the short circuit must not exceed one second.
Standard deviation is a measure of the statistical distribution about the mean, as derived from the formula:
N
å (Xn - X)2
s=
(5)
n =1
N -1
Temperature coefficient of timing capacitor and timing resistor is not taken into account.
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Electrical Characteristics (continued)
VCC = 15 V, over recommended operating free-air temperature range (unless otherwise noted)
TEST CONDITIONS (1)
PARAMETER
MIN
TYP (2)
MAX
4
4.5
0.3
0.7
UNIT
PWM COMPARATOR (SEE Figure 4)
Input threshold voltage, FEEDBACK
Zero duty cycle
Input sink current, FEEDBACK
FEEDBACK = 0.5 V
V
mA
UNDERVOLTAGE LOCKOUT (SEE Figure 4)
TA = 25°C
Threshold voltage
6
ΔTA = MIN to MAX
3.5
Hysteresis (6)
6.9
100
V
mV
OVERALL DEVICE
Standby supply current
VCC = 15 V
RT at Vref,
All other inputs and outputs open VCC = 40 V
Average supply current
DTC = 2 V, see Figure 4
(6)
9
15
11
18
12.4
mA
mA
Hysteresis is the difference between the positive-going input threshold voltage and the negative-going input threshold voltage.
6.6 Switching Characteristics
VCC = 15 V, TA = 25°C, over recommended operating conditions (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Output-voltage rise time
Common-emitter configuration (see Figure 5)
100
200
ns
Output-voltage fall time
Common-emitter configuration (see Figure 5)
30
100
ns
Output-voltage rise time
Emitter-follower configuration (see Figure 6)
200
400
ns
Output-voltage fall time
Emitter-follower configuration (see Figure 6)
45
100
ns
6.7 Typical Characteristics
100 k
100
VCC = 15 V
TA = 25°C
40 k
VCC = 15 V
∆VO = 3 V
TA = 25°C
90
4k
0.01 µF
0%
0.1 µF
1k
400
100
80
0.001 µF
-1%
Voltage Amplification - dB
Oscillator Frequency - Hz
-2%
10 k
Df = 1%
(see Note A)
CT = 1 µF
70
60
50
40
30
20
40
10
10
1k
0
4k
10 k
40 k 100 k
RT - Timing Resistance - W
400 k
1M
10
100
1k
10 k
100 k
1M
f - Frequenc y - Hz
Frequency variation (Δf) is the change in oscillator frequency that
occurs over the full temperature range.
Figure 1. Oscillator Frequency and Frequency Variation
vs Timing Resistance
6
1
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Figure 2. Amplifier Voltage Amplification
vs Frequency
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7 Parameter Measurement Information
Amplifier Under Test
+
VI
FEEDBACK
-
+
Vref
Other Amplifier
Figure 3. Amplifier-Characteristics Test Circuit
VCC = 15 V
150 W
2W
12
VCC
4
Test
Inputs
3
12 kW
6
5
8
C1
DTC
TL594
RT
11
C2
Output 2
10
E2
CT
Output 1
9
E1
FEEDBACK
150 W
2W
0.01 µF
1
IN+
IN16 IN+
15 IN-
2
Error
Amplifiers
13 OUTPUT
CTRL
14
REF
GND
7
50 kW
TEST CIRCUIT
VCC
Voltage
at C1
0V
VCC
Voltage
at C2
0V
Voltage
at CT
Threshold Voltage
DTC Input
0V
Threshold Voltage
Feedback
Input
0.7 V
Duty Cycle
0%
MAX
0%
VOLTAGE WAVEFORMS
Figure 4. Operational Test Circuit and Waveforms
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Parameter Measurement Information (continued)
15 V
tf
68 W
2W
Each Output
Circuit
Output
tr
90%
90%
CL = 15 pF
(includes probe and
jig capacitance)
10%
10%
TEST CIRCUIT
OUTPUT-VOLTAGE WAVEFORM
Figure 5. Common-Emitter Configuration
15 V
Each Output
Circuit
90%
90%
Output
10%
10%
68 W
2W
CL = 15 pF
(includes probe and
jig capacitance)
TEST CIRCUIT
tr
tf
OUTPUT-VOLTAGE WAVEFORM
Figure 6. Emitter-Follower Configuration
8
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8 Detailed Description
8.1 Overview
The design of the TL594 not only incorporates the primary building blocks required to control a switching power
supply, but also addresses many basic problems and reduces the amount of additional circuitry required in the
total design. The TL594 is a fixed-frequency pulse-width-modulation (PWM) control circuit. Modulation of output
pulses is accomplished by comparing the sawtooth waveform created by the internal oscillator on the timing
capacitor (CT) to either of two control signals. The output stage is enabled during the time when the sawtooth
voltage is greater than the voltage control signals. As the control signal increases, the time during which the
sawtooth input is greater decreases; therefore, the output pulse duration decreases. A pulse-steering flip-flop
alternately directs the modulated pulse to each of the two output transistors.
The error amplifiers have a common-mode voltage range of –0.3 V to VCC – 2 V. The DTC comparator has a
fixed offset that provides approximately 5% dead time. The on-chip oscillator can be bypassed by terminating RT
to the reference output and providing a sawtooth input to CT, or it can be used to drive the common circuitry in
synchronous multiple-rail power supplies. For more information on the operation of the TL594, see Designing
Switching Voltage Regulators With the TL494 (SLVA001).
8.2 Functional Block Diagram
OUTPUT CTRL
(see Function Table)
13
6
RT
5
CT
DTC
Oscillator
DTC
Comparator
0.1 V
4
9
PWM
Comparator
11
10
+
1
í
IN+
INí
15
C2
E2
12
+
2
í
Reference
Regulator
3
VCC
Undervoltage
Lockout
Control
14
FEEDBACK
E1
Pulse-Steering
Flip-Flop
Error Amplifier 2
16
C1
C1
Error Amplifier 1
IN+ 1
2
INí
8
1D
7
REF
GND
0.7 mA
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For OUTPUT CTRL function, see Table 1.
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8.3 Feature Description
8.3.1 5-V Reference Regulator
The TL594 internal 5-V reference regulator output is the REF pin. In addition to providing a stable reference, it
acts as a preregulator and establishes a stable supply from which the output-control logic, pulse-steering flip-flop,
oscillator, dead-time control comparator, and PWM comparator are powered. The regulator employs a band-gap
circuit as its primary reference to maintain thermal stability of less than 100-mV variation over the operating freeair temperature range of 0°C to 70°C. Short-circuit protection is provided to protect the internal reference and
preregulator; 10 mA of load current is available for additional bias circuits. The reference is internally
programmed to an initial accuracy of ±1% and maintains a stability of less than 25-mV variation over an input
voltage range of 7 V to 40 V. For input voltages less than 7 V, the regulator saturates within 1 V of the input and
tracks it.
8.3.2 Undervoltage Lockout
The TL594 has circuitry to provide an undervoltage-lockout functionality. A minimum recommended VCC voltage
of 7 V is recommended for operation, but if the VCC voltage drops below 6 V during operation, then the device
shuts off. See Electrical Characteristics for additional information regarding the undervoltage lockout circuitry.
8.3.3 Oscillator
The oscillator provides a positive sawtooth waveform to the dead-time and PWM comparators for comparison to
the various control signals.
The frequency of the oscillator is programmed by selecting timing components RT and CT. The oscillator charges
the external timing capacitor, CT, with a constant current, the value of which is determined by the external timing
resistor, RT. This produces a linear-ramp voltage waveform. When the voltage across CT reaches 3 V, the
oscillator circuit discharges it, and the charging cycle is reinitiated. The charging current is determined by
Equation 1.
3V
ICHARGE =
RT
(1)
The period of the sawtooth waveform is Equation 2.
3 V ´ CT
T=
ICHARGE
(2)
The frequency of the oscillator becomes Equation 3.
1
fOSC =
R T ´ CT
(3)
However, the oscillator frequency is equal to the output frequency only for single-ended applications. For pushpull applications, the output frequency is one-half the oscillator frequency.
Single-ended applications are calculated with Equation 4.
1
f=
R T ´ CT
(4)
Push-pull applications are calculated with Equation 5.
1
f=
2RT ´ CT
(5)
10
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Feature Description (continued)
8.3.4 Dead-Time Control
The dead-time control input provides control of the minimum dead time (off time). The output of the comparator
inhibits switching transistors Q1 and Q2 when the voltage at the input is greater than the ramp voltage of the
oscillator. An internal offset of 110 mV ensures a minimum dead time of approximately 3% with the dead-time
control input grounded. Applying a voltage to the dead-time control input can impose additional dead time. This
provides a linear control of the dead time from its minimum of 3% to 100% as the input voltage is varied from 0 V
to 3.3 V, respectively. With full-range control, the output can be controlled from external sources without
disrupting the error amplifiers. The dead-time control input is a relatively high-impedance input (II < 10 µA) and
must be used where additional control of the output duty cycle is required. However, for proper control, the input
must be terminated. An open circuit is an undefined condition.
8.3.5 Comparator
The comparator is biased from the 5-V reference regulator. This provides isolation from the input supply for
improved stability. The input of the comparator does not exhibit hysteresis, so protection against false triggering
near the threshold must be provided. The comparator has a response time of 400 ns from either of the controlsignal inputs to the output transistors, with only 100 mV of overdrive. This ensures positive control of the output
within one-half cycle for operation within the recommended 300-kHz range.
8.3.6 Pulse-Width Modulation (PWM)
The comparator also provides modulation control of the output pulse width. For this, the ramp voltage across
timing capacitor CT is compared to the control signal present at the output of the error amplifiers. The timing
capacitor input incorporates a series diode that is omitted from the control signal input. This requires the control
signal (error amplifier output) to be approximately 0.7 V greater than the voltage across CT to inhibit the output
logic, and ensures maximum duty cycle operation without requiring the control voltage to sink to a true ground
potential. The output pulse width varies from 97% of the period to 0 as the voltage present at the error amplifier
output varies from 0.5 V to 3.5 V, respectively.
8.3.7 Error Amplifiers
Both high-gain error amplifiers receive their bias from the VI supply rail. This permits a common-mode input
voltage range from –0.3 V to 2 V less than VI. Both amplifiers behave characteristically of a single-ended singlesupply amplifier, in that each output is active high only. This allows each amplifier to pull up independently for a
decreasing output pulse-width demand. With both outputs ORed together at the inverting input node of the PWM
comparator, the amplifier demanding the minimum pulse out dominates. The amplifier outputs are biased low by
a current sink to provide maximum pulse width out when both amplifiers are biased off.
8.3.8 Output-Control Input
The output-control input determines whether the output transistors operate in parallel or push-pull. This input is
the supply source for the pulse-steering flip-flop. The output-control input is asynchronous and has direct control
over the output, independent of the oscillator or pulse-steering flip-flop. The input condition is intended to be a
fixed condition that is defined by the application. For parallel operation, the output-control input must be
grounded. This disables the pulse-steering flip-flop and inhibits its outputs. In this mode, the pulses seen at the
output of the dead-time control or PWM comparator are transmitted by both output transistors in parallel. For
push-pull operation, the output-control input must be connected to the internal 5-V reference regulator. Under this
condition, each of the output transistors is enabled, alternately, by the pulse-steering flip-flop.
8.3.9 Output Transistors
Two output transistors are available on the TL594. Both transistors are configured as open collector/open
emitter, and each is capable of sinking or sourcing up to 200 mA. The transistors have a saturation voltage of
less than 1.3 V in the common-emitter configuration and less than 2.5 V in the emitter-follower configuration. The
outputs are protected against excessive power dissipation to prevent damage, but do not employ sufficient
current limiting to allow them to be operated as current-source outputs.
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8.4 Device Functional Modes
When the OUTPUT CTRL pin is tied to ground, the TL594 is operating in single-ended or parallel mode. When
the OUTPUT CTRL pin is tied to VREF, the TL594 is operating in normal push-pull operation (see Table 1).
Table 1. Function Table
INPUT
OUTPUT CTRL
12
OUTPUT FUNCTION
VI = 0
Single-ended or parallel output
VI = Vref
Normal push-pull operation
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The TL594 device contains an adjustable oscillator, a dead-time control comparator, a pulse-steering flip flop,
two error amplifiers, and a 5-V regulator. The TL594 device can be used for a wide variety of switching converter
applications over a frequency range of 1 Hz to 300 kHz, where the oscillation frequency is set by the RT and CT
values. For additional information regarding designing switching voltage regulators with the TL594, see
Designing Switching Voltage Regulators With the TL494.
9.2 Typical Application
This design example uses the TL594 to create a 5-V, 10-A power supply. This application is from Designing
Switching Voltage Regulators With the TL494.
140 H
30
100
5.1 k
1k
4k
270
5.1 k
TL594
1 nF
50 k
51 k
9.1 k
510
5.1 k
5.1 k
1k
2.5 F
0.1
Copyright © 2016, Texas Instruments Incorporated
Figure 7. 32-V to 5-V, 10-A Power Supply Application
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Typical Application (continued)
9.2.1 Design Requirements
• VI = 32 V
• VO = 5 V
• IO = 10 A
• fOSC = 20-kHz switching frequency
• VR = 20-mV peak-to-peak (VRIPPLE)
• ΔIL = 1.5-A inductor current change
9.2.2 Detailed Design Procedure
9.2.2.1 Input Power Source
The 32-V dc power source for this supply uses a 120-V input, 24-V output transformer rated at 75 VA. The 24-V
secondary winding feeds a full-wave bridge rectifier, followed by a current-limiting resistor (0.3 Ω) and two filter
capacitors (see Figure 8).
Bridge Rectifier
0.3
3 A/50 V
24V
3A
120 VAC
+32 V
+ 20 mF + 20 mF
Figure 8. Input Power Source
The output voltage and current are determined by Equation 6 and Equation 7.
VRECTIFIER = VSECONDARY ´ 2 = 24 V ´ 2 = 34 V
IRECTIFIER(AVG) »
(6)
VO
5V
´ IO »
´ 10 A = 1.6 A
VI
32 V
(7)
The 3-A, 50-V full-wave bridge rectifier meets these calculated conditions. Figure 7 shows the switching and
control sections.
9.2.2.2 Control Circuits
9.2.2.2.1 Oscillator
Connecting an external capacitor and resistor to pins 5 and 6 controls the TL594 oscillator frequency. The
oscillator is set to operate at 20 kHz, using the component values calculated by Equation 8 and Equation 9.
1
fOSC =
R T ´ CT
(8)
Choose CT = 0.001 µF and calculate RT with Equation 9.
RT +
1
f OSC
CT
+
(20
10 3)
1
(0.001
10 *6)
+ 50 kW
(9)
9.2.2.2.2 Error Amplifier
The error amplifier compares a sample of the 5-V output to the reference and adjusts the PWM to maintain a
constant output current (see Figure 9).
14
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Typical Application (continued)
k
k
TL594
k
k
TL594
k
Figure 9. Error-Amplifier Section
The TL594 internal 5-V reference is divided to 2.5 V by R3 and R4. The output-voltage error signal also is
divided to 2.5 V by R8 and R9. If the output must be regulated to exactly 5 V, a 10-kΩ potentiometer can be used
in place of R8 to provide an adjustment.
To increase the stability of the error-amplifier circuit, the output of the error amplifier is fed back to the inverting
input through RT, reducing the gain to 101.
9.2.2.2.3 Current-Limiting Amplifier
The power supply was designed for a 10-A load current and an IL swing of 1.5 A; therefore, the short-circuit
current is calculated as Equation 10.
I
ISC = IO + L = 10.75 A
(10)
2
Figure 10 shows the current-limiting circuit.
k
k
TL594
TL594
k
Figure 10. Current-Limiting Circuit
Resistors R1 and R2 set the reference of about 1 V on the inverting input of the current-limiting amplifier.
Resistor R13, in series with the load, applies 1 V to the noninverting terminal of the current-limiting amplifier
when the load current reaches 10 A. The output-pulse width is reduced accordingly. The value of R13 is
calculated as Equation 11.
1V
R13 =
= 0.1W
10 A
(11)
9.2.2.2.4 Soft Start
To reduce stress on the switching transistors at start-up, the start-up surge that occurs as the output filter
capacitor charges must be reduced. The availability of the dead-time control makes implementation of a soft-start
circuit relatively simple (see Figure 11).
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Typical Application (continued)
Figure 11. Soft-Start Circuit
The soft-start circuit allows the pulse width at the output to increase slowly (see Figure 11) by applying a
negative slope waveform to the dead-time control input (pin 4).
Initially, capacitor C2 forces the dead-time control input to follow the 5-V regulator, which disables the outputs
(100% dead time). As the capacitor charges through R6, the output pulse width slowly increases until the control
loop takes command. With a resistor ratio of 1:10 for R6 and R7, the voltage at pin 4 after start-up is 0.1 × 5 V,
or 0.5 V.
The soft-start time generally is in the range of 25 to 100 clock cycles. If 50 clock cycles at a 20-kHz switching
rate is selected, the soft-start time is calculated as Equation 12.
1
1
t= =
= 50 msper clock cycle
f 20kHz
(12)
The value of the capacitor then is determined with Equation 13.
soft - start time 50 ms ´ 50 cycles
C2 =
=
= 2.5 mF
R6
1 kW
(13)
This helps eliminate any false signals that might be created by the control circuit as power is applied.
9.2.2.2.5 Setting the Dead Time
The primary function of the dead-time control is to control the minimum off time of the output of the TL594
device. The dead-time control input provides control from 5% to 100% dead time. The TL594 device can be
tailored to the specific power transistor switches that are used, to ensure that the output transistors never
experience a common on-time. Figure 12 shows the bias circuit for the basic function.
16
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Typical Application (continued)
VREF
R1
TD = RTCT(0.05 + 0.35R2)
R2 in kW
R1 + R2 = 5 kW
Dead-Time Control In
R2
Figure 12. Setting Dead Time
9.2.2.3 Inductor Calculations
Figure 13 shows the switching circuit used.
L
S1
VI
D1
C1
R1
VO
Figure 13. Switching Circuit
The size of the inductor (L) required is:
d
=
duty cycle = VO/VI = 5 V/32 V = 0.156
f
=
20 kHz (design objective)
ton
=
time on (S1 closed) = (1/f) × d = 7.8 µs
toff
=
time off (S1 open) = (1/f) – ton = 42.2 µs
L
≉
(VI – VO ) × ton/ΔIL
≉
[(32 V – 5 V) × 7.8 µs]/1.5 A
≉
140.4 µH
9.2.2.4 Output Capacitance Calculations
Once the filter inductor has been calculated, the value of the output filter capacitor is calculated to meet the
output ripple requirements. An electrolytic capacitor can be modeled as a series connection of an inductance, a
resistance, and a capacitance. To provide good filtering, the ripple frequency must be far below the frequencies
at which the series inductance becomes important. So, the two components of interest are the capacitance and
the effective series resistance (ESR). The maximum ESR is calculated with Equation 14 according to the relation
between the specified peak-to-peak ripple voltage and the peak-to-peak ripple current.
DVO(ripple)
V
=
» 0.067 W
ESR(max) =
DIL
1.5 A
(14)
The minimum capacitance of C3 necessary to maintain the VO ripple voltage at less than the 100-mV design
objective is calculated according to Equation 15.
DIL
1.5 A
C3 =
=
= 94 mF
8f DVO 8 ´ 20 ´ 103 ´ 0.1 V
(15)
A 220-mF, 60-V capacitor is selected because it has a maximum ESR of 0.074 Ω and a maximum ripple current
of 2.8 A.
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9.2.2.5 Transistor Power-Switch Calculations
The transistor power switch was constructed with an NTE153 pnp drive transistor and an NTE331 npn output
transistor. These two power devices were connected in a pnp hybrid Darlington circuit configuration (see
Figure 14).
TL594
Figure 14. Power-Switch Section
The hybrid Darlington circuit must be saturated at a maximum output current of IO + ΔIL/2 or 10.8 A. The
Darlington hFE at 10.8 A must be high enough not to exceed the 250-mA maximum output collector current of the
TL594. Based on published NTE153 and NTE331 specifications, the required power-switch minimum drive was
calculated by Equation 16 through Equation 18 to be 144 mA.
hFE (Q1) at IC of 3 A = 15
(16)
hFE (Q2) at IC of 10.0 A = 5
(17)
I
IO + L
2
³ 144mA
iB ³
hFE (Q2) ´ hFE (Q1)
(18)
The value of R10 was calculated by Equation 19.
V - [VBE (Q1) + VCE (TL494)] 32 - (1.5 + 0.7)
R10 £ I
=
iB
0.144
R10 £ 207 W
(19)
Based on these calculations, the nearest standard resistor value of 220 Ω was selected for R10. Resistors R11
and R12 permit the discharge of carriers in switching transistors when they are turned off.
The power supply described demonstrates the flexibility of the TL594 PWM control circuit. This power-supply
design demonstrates many of the power-supply control methods provided by the TL594, as well as the versatility
of the control circuit.
18
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9.2.3 Application Curve
VREF − Reference Voltage − (V)
6
5
4
3
2
1
0
0
1
2
3
4
5
6
7
VI − Input Voltage − (V)
Figure 15. Reference Voltage vs Input Voltage
10 Power Supply Recommendations
The TL594 is designed to operate from an input voltage supply range between 7 V and 40 V. This input supply
must be well regulated. If the input supply is placed more than a few inches from the device, additional bulk
capacitance may be required in addition to the ceramic bypass capacitors. A tantalum capacitor with a value of
47 µF is a typical choice; however this may vary depending upon the output power being delivered.
11 Layout
11.1 Layout Guidelines
Always try to use a low EMI inductor with a ferrite type closed core. Some examples would be toroid and
encased E core inductors. Open core can be used if they have low EMI characteristics and are placed a bit more
away from the low power traces and components. Make the poles perpendicular to the PCB as well if using an
open core. Stick cores usually emit the most unwanted noise.
11.1.1 Feedback Traces
Try to run the feedback trace as far from the inductor and noisy power traces as possible. The feedback trace
must be as direct as possible and wider to decrease impedance. These two sometimes involve a trade-off, but
keeping it away from inductor EMI and other noise sources is the more critical of the two. Run the feedback trace
on the side of the PCB opposite of the inductor, ideally with a ground plane separating the two.
11.1.2 Input or Output Capacitors
When using a low value ceramic input filter capacitor, it must be placed as close to the VCC pin of the IC as
possible. This eliminates as much trace inductance effects as possible and give the internal IC rail a cleaner
voltage supply. Some designs require the use of a feed-forward capacitor connected from the output to the
feedback pin as well, usually for stability reasons. In this case it must also be positioned as close to the IC as
possible. Using surface mount capacitors also reduces lead length and reduces the chance of noise coupling into
the effective antenna created by through-hole components.
11.1.3 Compensation Components
External compensation components for stability must also be placed close to the IC. Surface mount components
are recommended here as well for the same reasons discussed for the filter capacitors. These must not be
placed very close to the inductor either.
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Layout Guidelines (continued)
11.1.4 Traces and Ground Planes
Make all of the power (high current) traces as short and direct as possible, while trying to maximize trace width
for the appropriate current carrying capability. It is good practice on a standard PCB board to make the traces an
absolute minimum of 15 mils (0.381 mm) per Ampere. The inductor, output capacitors, and output diode must be
as close to each other possible. This helps reduce the EMI radiated by the power traces due to the high
switching currents through them. This also reduces lead inductance and resistance as well, which in turn reduces
noise spikes, ringing, and resistive losses that produce voltage errors.
The grounds of the IC, input capacitors, output capacitors, and output diode (if applicable) must be connected
close together directly to a ground plane. It would also be a good idea to have a ground plane on both sides of
the PCB. This reduces noise as well by reducing ground loop errors as well as by absorbing more of the EMI
radiated by the inductor. For multi-layer boards with more than two layers, a ground plane can be used to
separate the power plane (where the power traces and components are) and the signal plane (where the
feedback and compensation and components are) for improved performance.
On multi-layer boards the use of vias are required to connect traces and different planes. It is good practice to
use one standard via per 200 mA of current if the trace requires conduct to a significant amount of current from
one plane to the other. Arrange the components so that the switching current loops curl in the same direction.
Due to the way switching regulators operate, there are two power states. One state when the switch is on and
one when the switch is off. During each state there is a current loop made by the power components that are
currently conducting. Place the power components so that during each of the two states the current loop is
conducting in the same direction. This prevents magnetic field reversal caused by the traces between the two
half-cycles and reduces radiated EMI.
11.2 Layout Example
-
-
TL594
Copyright © 2016, Texas Instruments Incorporated
Figure 16. TL594 Layout Example
20
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12 Device and Documentation Support
12.1 Documentation Support
12.1.1 Related Documentation
For related documentation see the following:
Designing Switching Voltage Regulators With the TL494 (SLVA001)
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TL594CD
ACTIVE
SOIC
D
16
40
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
TL594C
Samples
TL594CDR
ACTIVE
SOIC
D
16
2500
RoHS & Green
NIPDAU | SN
Level-1-260C-UNLIM
0 to 70
TL594C
Samples
TL594CDRG4
ACTIVE
SOIC
D
16
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
TL594C
Samples
TL594CN
ACTIVE
PDIP
N
16
25
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
TL594CN
Samples
TL594CNE4
ACTIVE
PDIP
N
16
25
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
TL594CN
Samples
TL594CNSR
ACTIVE
SO
NS
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
TL594
Samples
TL594CPW
ACTIVE
TSSOP
PW
16
90
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
T594
Samples
TL594CPWR
ACTIVE
TSSOP
PW
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
T594
Samples
TL594CPWRG4
ACTIVE
TSSOP
PW
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
T594
Samples
TL594ID
ACTIVE
SOIC
D
16
40
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TL594I
Samples
TL594IDG4
ACTIVE
SOIC
D
16
40
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TL594I
Samples
TL594IDR
ACTIVE
SOIC
D
16
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TL594I
Samples
TL594IDRG4
ACTIVE
SOIC
D
16
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TL594I
Samples
TL594IN
ACTIVE
PDIP
N
16
25
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
TL594IN
Samples
TL594INSR
ACTIVE
SO
NS
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TL594I
Samples
TL594INSRG4
ACTIVE
SO
NS
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
TL594I
Samples
TL594IPWR
ACTIVE
TSSOP
PW
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
Z594
Samples
TL594IPWRG4
ACTIVE
TSSOP
PW
16
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
Z594
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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