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TLV2186
SBOS947A – JULY 2019 – REVISED JUNE 2020
TLV2186 Precision, Rail-to-Rail Input and Output, 24-V, Zero-Drift Operational Amplifier
1 Features
3 Description
•
The TLV2186 is a low-power, 24-V, rail-to-rail input
and output zero-drift operational amplifier (op amp).
The TLV2186 features only 10 µV of offset voltage
(typical) and 0.1 µV/°C of offset voltage drift over
temperature (typical). This device is a great choice for
precision instrumentation, signal measurement, and
active filtering applications.
1
•
•
•
•
•
High precision:
– Offset drift: 0.1 μV/°C
– Low offset voltage: 10 μV
Low quiescent current: 90 µA
Excellent dynamic performance:
– Gain bandwidth: 750 kHz
– Slew rate: 0.35 V/µs
Robust design:
– RFI/EMI filtered inputs
Rail-to-rail input/output
Supply range: 4.5 V to 24 V
Low quiescent current consumption (90 μA) makes
the TLV2186 an excellent option for power-sensitive
applications, such as battery-powered instrumentation
and portable systems.
Moreover, the high common-mode architecture along
with low offset voltage allows for high-side current
shunt monitoring at the positive rail. This device also
provides robust ESD protection during shipment,
handling, and assembly.
2 Applications
•
•
•
•
•
•
PC PSU and game console unit
Merchant DC/DC
Flow transmitter
Pressure transmitter
Merchant battery charger
Electricity meter
The device is specified for operation from –40°C to
+125°C. The TLV2186 is offered in WSON-8 and SO8 packages.
Device Information(1)
PART NUMBER
TLV2186
PACKAGE
BODY SIZE (NOM)
SOIC (8)
4.90 mm × 3.90 mm
WSON (8)
2.00 mm x 2.00 mm
(1) For all available packages, see the package option addendum
at the end of the data sheet.
High-Side Current Shunt Monitor Application
VOS vs Input Common Mode Voltage
4
6 V to 24 V
Microcontroller
Battery /
Power
Supply
TLV2186
+
0 V to 5 V
ADC
Input-referred Offset Voltage (µV)
RS
3
2
1
0
-1
-2
-3
-4
-5
-6
-12.5 -10
-7.5 -5 -2.5
0
2.5
5
7.5
Input Common-mode Voltage (V)
10
12.5
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TLV2186
SBOS947A – JULY 2019 – REVISED JUNE 2020
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 14
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
14
14
15
19
8
Application and Implementation ........................ 20
8.1 Application Information............................................ 20
8.2 Typical Applications ................................................ 22
9 Power Supply Recommendations...................... 28
10 Layout................................................................... 28
10.1 Layout Guidelines ................................................. 28
10.2 Layout Example .................................................... 29
11 Device and Documentation Support ................. 30
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Device Support......................................................
Documentation Support ........................................
Receiving Notification of Documentation Updates
Support Resources ...............................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
30
30
31
31
31
31
31
12 Mechanical, Packaging, and Orderable
Information ........................................................... 31
4 Revision History
Changes from Original (July 2019) to Revision A
•
2
Page
Added DSG (WSON-8) package to and associated content to data sheet............................................................................ 1
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SBOS947A – JULY 2019 – REVISED JUNE 2020
5 Pin Configuration and Functions
D Package
8-Pin SOIC
Top View
OUT A
1
8
V+
±IN A
2
7
OUT B
+IN A
3
6
±IN B
V±
4
5
+IN B
Not to scale
DSG Package
8-Pin WSON With Exposed Thermal Pad
Top View
OUT A
1
±IN A
2
+IN A
3
V±
4
Thermal
Pad
8
V+
7
OUT B
6
±IN B
5
+IN B
Not to scale
Pin Functions
PIN
I/O
DESCRIPTION
NAME
NO.
–IN A
2
I
Inverting input channel A
+IN A
3
I
Noninverting input channel A
–IN B
6
I
Inverting input channel B
+IN B
5
I
Noninverting input channel B
OUT A
1
O
Output channel A
OUT B
7
O
Output channel B
V–
4
—
Negative supply
V+
8
—
Positive supply
Thermal Pad
—
—
Exposed thermal die pad on underside; connect thermal die pad to V–.
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SBOS947A – JULY 2019 – REVISED JUNE 2020
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
MIN
VS
MAX
Supply voltage, VS = (V+) – (V–)
26
Common-mode
Input
voltage
(V–) –0.5
Differential
Output short-circuit
(V+) + 0.5
(V+) – (V–) + 0.2
(2)
UNIT
V
V
Continuous
TJ
Operating junction temperature
-40
150
°C
Tstg
Storage temperature
-65
150
°C
(1)
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Theseare stress ratings
only, which do not imply functional operation of the device at these or anyother conditions beyond those indicated under Recommended
OperatingConditions. Exposure to absolute-maximum-rated conditions for extended periods mayaffect device reliability.
Short-circuit to ground, one amplifier per package.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
4000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
1500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safemanufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safemanufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
Single supply
VS
Supply
Voltage
TA
Specified temperature
NOM
4.5
Dual supply
MAX
UNIT
24
±2.25
±12
V
–40
125
°C
6.4 Thermal Information
TLV2186
THERMAL METRIC (1)
D (SOIC)
DSG (WSON)
8 PINS
8 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
129.4
70.2
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
69.6
86.2
°C/W
RθJB
Junction-to-board thermal resistance
72.8
36.8
°C/W
ΨJT
Junction-to-top characterization parameter
20.8
2.9
°C/W
ΨJB
Junction-to-board characterization parameter
72.0
36.8
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
13.3
°C/W
(1)
4
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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6.5 Electrical Characteristics
at TA = 25°C, VS = ±2.25V to ±12V, RL = 10 kΩ connected to VS / 2, VCM = VS / 2, and VOUT = VS / 2 (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OFFSET VOLTAGE
VOS
Input offset voltage
±10
±250
μV
dVOS/dT
Input offset voltage drift
TA = –40°C to +125°C
±0.1
±1.0
μV/°C
PSRR
Power-supply rejection
ratio
TA = –40°C to +125°C
±0.05
±1
μV/V
0.1
0.6
INPUT BIAS CURRENT
IB
Input bias current
TA = –40℃ to +85℃
0.6
TA = –40℃ to +125℃
0.1
IOS
Input offset current
nA
5
TA = –40℃ to +85℃
1.2
1.2
TA = –40℃ to +125℃
nA
2
NOISE
Input voltage noise
f = 0.1 Hz to 10 Hz
110
nVRMS
eN
Input voltage noise density
f = 1 kHz
38
nV/√Hz
iN
Input current noise
f = 1 kHz
100
fA/√Hz
INPUT VOLTAGE
VCM
CMRR
Common-mode voltage
Common-mode rejection
ratio
(V–) – 0.2
(V+) + 0.2
(V–) – 0.1 < VCM < (V+) + 0.1 V,
TA = –40℃ to +125℃
VS = ±2.25 V
108
126
VS = ±12 V
110
134
(V–) – 0.1 < VCM < (V+) + 0.1 V,
TA = –40℃ to +125℃
VS = ±2.25 V
106
114
VS = ±12 V
106
120
V
dB
FREQUENCY RESPONSE
GBW
Gain-bandwidth product
SR
Slew rate
1-V step, G = 1
750
kHz
0.35
V/μs
tS
Settling time
Overload recovery time
To 0.1%, 1-V step , G = 1
7.5
μs
VIN × gain > VS
10
μs
INPUT CAPACITANCE
ZID
Differential
100 || 5
MΩ || pF
ZICM
Common-mode
50 || 2.5
GΩ || pF
OPEN-LOOP GAIN
AOL
Open-loop voltage gain
VS = ±12 V
(V–) + 0.3 V < VO < (V+) –
0.3 V, RL = 10 kΩ
120
140
(V–) + 0.3 V < VO < (V+) –
0.3 V, RL = 10 kΩ, TA =
–40°C to +125°C
120
134
(V–) + 0.65 V < VO < (V+) –
0.65 V, RL = 2 kΩ
120
140
(V–) + 0.65 V < VO < (V+) –
0.65 V, RL = 2 kΩ, TA =
–40°C to +125°C
120
134
dB
OUTPUT
No load
5
20
RL = 10 kΩ
60
100
RL = 2 kΩ
340
500
90
115
VO
Voltage output swing from
both rails
ISC
Short-circuit current
CLOAD
Capacitive load drive
See typical curves
RO
Open-loop output
impedance
See typical curves
RL = 10 kΩ, TA = –40℃ to +125℃
±20
mV
mA
POWER SUPPLY
IQ
Quiescent current per
amplifier
VS = ±2.25 to ±12 V
90
TA = –40°C to +125°C
130
150
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5
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6.6 Typical Characteristics
Table 1. Typical Characteristic Graphs
DESCRIPTION
FIGURE
Offset Voltage Distribution
Figure 1
Offset Voltage Drift (-40°C to +125C°C)
Figure 2
Input Bias Current Distribution
Figure 3
Input Offset Current Distribution
Figure 4
Offset Voltage vs Common-Mode Voltage
Figure 5
Offset Voltage vs Supply Voltage
Figure 6
Open-Loop Gain and Phase vs Frequency
Figure 7
Closed-Loop Gain vs Frequency
Figure 8
Input Bias Current and Offset Current vs Temperature
Figure 9
Output Voltage Swing vs Output Current (Sourcing)
Figure 10
Output Voltage Swing vs Output Current (Sinking)
Figure 11
CMRR and PSRR vs Frequency
Figure 12
CMRR vs Temperature
Figure 13
PSRR vs Temperature
Figure 14
0.1-Hz to 10-Hz Voltage Noise
Figure 15
Input Voltage Noise Spectral Density vs Frequency
Figure 16
THD+N vs Frequency
Figure 17
THD+N vs Output Amplitude
Figure 18
Quiescent Current vs Supply Voltage
Figure 19
Quiescent Current vs Temperature
Figure 20
Open-Loop Gain vs Temperature (10 kΩ)
Figure 21
Open-Loop Gain vs Temperature (2 kΩ)
Figure 22
Open-Loop Output Impedance vs Frequency
Figure 23
Small-Signal Overshoot vs Capacitive Load (Gain = –1, 10-mV step)
Figure 24
Small-Signal Overshoot vs Capacitive Load (Gain = 1, 10-mV step)
Figure 25
No Phase Reversal
Figure 26
Positive Overload Recovery
Figure 27
Negative Overload Recovery
Figure 28
Small-Signal Step Response (Gain = 1, 10-mV step)
Figure 29
Small-Signal Step Response (Gain = –1, 10-mV step)
Figure 30
Large-Signal Step Response (Gain = 1, 10-V step)
Figure 31
Large-Signal Step Response (Gain = –1, 10-V step)
Figure 32
Phase Margin vs Capacitive Load
Figure 33
Settling Time (1-V Step, 0.1% Settling)
Figure 34
Short Circuit Current vs Temperature
Figure 35
Maximum Output Voltage vs Frequency
Figure 36
EMIRR vs Frequency
Figure 37
Channel Separation
Figure 38
6
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50
30
45
27
40
24
35
21
Amplifiers (%)
Amplifiers (%)
at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
30
25
20
18
15
12
15
9
10
6
5
3
0
-50
0
-1
-40
-30
-20 -10
0
10
20
Input Offset Voltage (uV)
30
40
50
45
72
40
64
35
56
Amplifiers (%)
80
30
25
20
0.6
0.5
0.4
0.3
0.2
0.1
0
-1200
0
8
0
-0.1
16
5
-0.2
1
32
10
-0.3
0.8
40
24
-0.4
-0.4 -0.2
0
0.2 0.4 0.6
Input Offset Voltage Drift (µV/qC)
48
15
-0.5
-0.6
Figure 2. Offset Voltage Drift (-40°C to 125C°C)
50
-0.6
Amplifiers (%)
Figure 1. Offset Voltage Distribution
-0.8
-900
-600
-300
0
300
600
Input Offset Current (pA)
900
1200
Input Bias Current (nA)
Figure 3. Input Bias Current Distribution
Figure 4. Input Offset Current Distribution
5
3
4
2
1
Offset Voltage (µV)
Input-referred Offset Voltage (µV)
4
0
-1
-2
-3
3
2
1
0
-4
-1
-5
-6
-12.5 -10
-2
-7.5 -5 -2.5
0
2.5
5
7.5
Input Common-mode Voltage (V)
10
12.5
4
Figure 5. Offset Voltage vs Common-Mode Voltage
6
8
10
12
14
16
18
Supply Voltage (V)
20
22
Figure 6. Offset Voltage vs Supply Voltage
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at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
180
150
180
Gain
Phase 150
120
120
90
90
60
60
30
30
0
0
30
G= 1
G= 1
G= 10
Gain (dB)
Phase (q)
Gain (dB)
20
10
0
-10
-30
10m 100m
1
10
100
1k
10k
Frequency (Hz)
100k
1M
-30
10M
-20
100
Figure 7. Open-Loop Gain and Phase vs Frequency
1M
10M
12
Ibn
Ibp
Ios
2.4
2.1
10
8
1.8
Output Voltage (V)
Input Current (nA)
10k
100k
Frequency (Hz)
Figure 8. Closed-Loop Gain vs Frequency
2.7
1.5
1.2
0.9
0.6
0.3
6
4
2
0
-2
-4
-40qC
25qC
85qC
125qC
-6
0
-8
-0.3
-40
-10
-20
0
20
40
60
80
Temperature (qC)
100
120
140
0
Figure 9. Input Bias Current and Offset Current vs
Temperature
2.5
5
7.5
10 12.5 15 17.5
Output Current (mA)
20
22.5
25
Figure 10. Output Voltage Swing vs Output Current
(Sourcing)
12.5
160
-40qC
25qC
85qC
125qC
10
PSRR
PSRR
CMRR
140
Rejection Ratio (dB)
7.5
Output Voltage (V)
1k
5
2.5
0
-2.5
-5
120
100
80
60
40
-7.5
20
-10
-12.5
0
3
6
9
12
15
18
Output Current (mA)
21
24
27
0
100m
Figure 11. Output Voltage Swing vs Output Current
(Sinking)
8
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10
100
1k
10k
Frequency (Hz)
100k
1M
10M
Figure 12. CMRR and PSRR vs Frequency
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at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
170
145
Power Supply Rejection Ratio (dB)
140
135
130
125
120
115
110
105
-40
-25
-10
5
20 35 50 65
Temperature (qC)
80
95
160
150
140
130
120
-40
110 125
-25
Figure 13. CMRR vs Temperature
20 35 50 65
Temperature (qC)
80
95
110 125
Figure 14. PSRR vs Temperature
Voltage Noise Density (nV/—Hz)
Input Referred Voltage Noise (200 nV/div)
100
10
100m
0.01
-80
0.001
-100
-120
0.0001
100
1k
Frequency (Hz)
10k
100
1k
Frequency (Hz)
10k
100k
-40
Noise (dB)
G=
G=
G=
G=
Noise (%)
-60
Total Harmonic Distortion
1, RL = 10 k:
1, RL = 2 k:
1, RL = 10 k:
1, RL = 2 k:
Total Harmonic Distortion
0.1
G=
G=
G=
G=
10
1
-40
1
1
Figure 16. Input Voltage Noise Spectral Density vs
Frequency
Figure 15. 0.1-Hz to 10-Hz Voltage Noise
Noise ( )
5
1000
Time (1 s/div)
Total Harmonic Distortion
-10
1, RL = 10 k:
1, RL = 2 k:
1, RL = 10 k:
1, RL = 2 k:
0.1
-60
0.01
-80
0.001
10m
Figure 17. THD+N vs Frequency
100m
1
Output Amplitude (VRMS)
-100
10
Figure 18. THD+N vs Output Amplitude
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Total Harmonic Distortion + Noise (dB)
Common-Mode Rejection Ratio (dB)
150
9
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at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
110
100
VS = 4.5 V
VS = 24 V
90
100
Quiescent Current (µA)
Quiescent Current (µA)
80
70
60
VS Min = 4.5 V
50
40
30
90
80
70
20
10
60
-40
0
0
2
4
6
8
10 12 14 16
Supply Voltage (V)
18
20
22
24
Figure 19. Quiescent Current vs Supply Voltage
5
20 35 50 65
Temperature (qC)
80
95
110 125
180
Open-loop Gain (dB)
Open-loop Gain (dB)
-10
Figure 20. Quiescent Current vs Temperature
180
160
140
120
-40
-25
-25
-10
5
20 35 50 65
Temperature (qC)
80
95
160
140
120
-40
110 125
-25
-10
5
20 35 50 65
Temperature (qC)
80
95
110 125
RL = 2 kΩ
Figure 21. Open-Loop Gain vs Temperature
Figure 22. Open-Loop Gain vs Temperature
40
35
100
RISO = 0 :
RISO = 25 :
RISO = 50 :
30
Overshoot ( )
Open-Loop Output Impedance, ZO (:)
1000
10
1
25
20
15
0.1
10
0.01
10
100
1k
10k
Frequency (Hz)
100k
1M
5
10
100
Capactiance (pF)
1000
Gain = –1, 10-mV step
Figure 23. Open-Loop Output Impedance vs Frequency
10
Figure 24. Small-Signal Overshoot vs Capacitive Load
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at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
100
Voltage (5 V/div)
Overshoot ( )
80
VIN (V)
VOUT (V)
RISO = 0 :
RISO = 25 :
RISO = 50 :
60
40
20
0
10
100
Capactiance (pF)
Time (100 Ps/div)
1000
Gain = 1, 10-mV step
Figure 25. Small-Signal Overshoot vs Capacitive Load
Figure 26. No Phase Reversal
Voltage (5 V/div)
Voltage (5 V/div)
VIN
VOUT
VIN
VOUT
Time (10 Ps/div)
Time (10 Ps/div)
Figure 27. Positive Overload Recovery
Figure 28. Negative Overload Recovery
VIN
VOUT
Voltage (5 mV/div)
Voltage (5 mV/div)
VIN
VOUT
Time (10 Ps/div)
Time (10 Ps/div)
Gain = 1, 10-mV step
Gain = –1, 10-mV step
Figure 29. Small-Signal Step Response
Figure 30. Small-Signal Step Response
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at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
VIN (V)
VOUT (V)
Voltage (2 V/div)
Voltage (2 V/div)
VIN (V)
VOUT (V)
Time (10 Ps/div)
Time (10 Ps/div)
Gain = 1, 10-V step
Gain = –1, 10-V step
Figure 31. Large-Signal Step Response
Figure 32. Large-Signal Step Response
65
Falling
Rising
60
Output (1 mV/div)
Phase Margin (q)
55
50
45
40
35
30
25
20
15
10
100
CLOAD (pF)
Time (5 Ps/div)
1000
1-V step, 0.1% settling
Figure 33. Phase Margin vs Capacitive Load
Figure 34. Settling Time
32
30
Sinking
Sourcing
31
29
28
27
26
25
24
23
22
VS = r12 V
VS = r2.25 V
25
Output Voltage (VPP)
Short Circuit Current (mA)
30
20
15
10
5
21
20
-40
0
-20
0
20
40
60
Temperature (qC)
80
100
120
Figure 35. Short Circuit Current vs Temperature
12
1
10
100
1k
10k
Frequency (Hz)
100k
1M
Figure 36. Maximum Output Voltage vs Frequency
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175
-60
150
-80
Channel Seperation (dB)
EMIRR IN+ (dB)
at TA = 25°C, VS = ±12 V, VCM = VS / 2, RL = 10 kΩ (unless otherwise noted)
125
100
75
50
25
10M
-100
-120
-140
-160
100M
1G
Frequency (Hz)
10G
-180
1k
10k
100k
1M
Frequency (Hz)
Figure 37. EMIRR vs Frequency
Figure 38. Channel Separation
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7 Detailed Description
7.1 Overview
The TLV2186 operational amplifier combines precision offset and drift with excellent overall performance, making
the device a great choice for a wide variety of precision applications. The precision offset drift of only 0.1 µV/°C
provides stability over the entire operating temperature range of –40°C to +125°C. In addition, this device offers
excellent linear performance with high CMRR, PSRR, and AOL. As with all amplifiers, applications with noisy or
high-impedance power supplies require decoupling capacitors close to the device pins. In most cases, 0.1-µF
capacitors are adequate. See the Layout Guidelines section for details and a layout example.
The TLV2186 is part of a family of zero-drift, MUX-friendly, rail-to-rail output operational amplifiers. This device
operates from 4.5 V to 24 V, is unity-gain stable, and is designed for a wide range of general-purpose and
precision applications. The zero-drift architecture provides ultra-low input offset voltage and near-zero input offset
voltage drift over temperature and time. This choice of architecture also offers outstanding ac performance, such
as ultra-low broadband noise, zero flicker noise, and outstanding distortion performance when operating below
the chopper frequency.
7.2 Functional Block Diagram
The Functional Block Diagram shows a representation of the proprietary TLV2186 architecture.
C2
CHOP1
+IN
±IN
GM1
CHOP2
Notch
Filter
GM2
GM3
OUT
24-V
Differential
Front End
GM_FF
C1
14
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7.3 Feature Description
The TLV2186 operational amplifier has several integrated features to help maintain a high level of precision
through a variety of applications. These include a rail-to-rail inputs, phase-reversal protection, input bias current
clock feedthrough, EMI rejection, electrical overstress protection and MUX-friendly Inputs.
7.3.1 Rail-to-Rail Inputs
Unlike many chopper amplifiers, the TLV2186 has rail-to-rail inputs that allow the input common-mode voltage to
not only reach, but exceed the supply voltages by 200 mV. This configuration simplifies power-supply
requirements by not requiring headroom over the input signal range.
The TLV2186 is specified for operation from 4.5 V to 24 V (±2.25 V to ±12 V) with rail-to-rail inputs. Many
specifications apply from –40°C to +125°C. Parameters that can exhibit significant variance with regard to
operating voltage or temperature are presented in the Typical Characteristics section.
7.3.2 Phase-Reversal Protection
The TLV2186 has internal phase-reversal protection. Some op amps exhibit a phase reversal when the input is
driven beyond the linear common-mode range. This condition is most often encountered in noninverting circuits
when the input is driven beyond the specified common-mode voltage range, causing the output to reverse into
the opposite rail. The TLV2186 input prevents phase reversal with excessive common-mode voltage. Instead, the
output limits into the appropriate rail. This performance is shown in Figure 39.
Voltage (5 V/div)
VIN (V)
VOUT (V)
Time (100 Ps/div)
Figure 39. No Phase Reversal
7.3.3 Input Bias Current Clock Feedthrough
Zero-drift amplifiers such as the TLV2186 use a switching architecture on the inputs to correct for the intrinsic
offset and drift of the amplifier. Charge injection from the integrated switches on the inputs can introduce short
transients in the input bias current of the amplifier. The extremely short duration of these pulses prevents the
pulses from amplifying, however the pulses may be coupled to the output of the amplifier through the feedback
network. The most effective method to prevent transients in the input bias current from producing additional noise
at the amplifier output is to use a low-pass filter, such as an RC network.
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Feature Description (continued)
7.3.4 EMI Rejection
The TLV2186 uses integrated electromagnetic interference (EMI) filtering to reduce the effects of EMI
interference from sources such as wireless communications and densely-populated boards with a mix of analog
signal chain and digital components. EMI immunity can be improved with circuit design techniques; the TLV2186
benefits from these design improvements. Texas Instruments has developed the ability to accurately measure
and quantify the immunity of an operational amplifier over a broad frequency spectrum extending from 10 MHz to
6 GHz. Figure 40 shows the results of this testing on the TLV2186. Table 2 lists the EMIRR +IN values for the
TLV2186 at particular frequencies commonly encountered in real-world applications. Applications listed in
Table 2 may be centered on or operated near the particular frequency shown. Detailed information can also be
found in the EMI Rejection Ratio of Operational Amplifiers (SBOA128), available for download from www.ti.com.
175
EMIRR IN+ (dB)
150
125
100
75
50
25
10M
100M
1G
Frequency (Hz)
10G
Figure 40. EMIRR Testing
Table 2. TLV2186 EMIRR IN+ for Frequencies of Interest
FREQUENCY
APPLICATION AND ALLOCATION
EMIRR IN+
400 MHz
Mobile radio, mobile satellite, space operation, weather, radar, ultra-high frequency
(UHF) applications
48.4 dB
900 MHz
Global system for mobile communications (GSM) applications, radio
communication, navigation, GPS (to 1.6 GHz), GSM, aeronautical mobile, UHF
applications
52.8 dB
1.8 GHz
GSM applications, mobile personal communications, broadband, satellite, L-band
(1 GHz to 2 GHz)
69.1 dB
2.4 GHz
802.11b, 802.11g, 802.11n, Bluetooth®, mobile personal communications,
industrial, scientific and medical (ISM) radio band, amateur radio and satellite, Sband (2 GHz to 4 GHz)
88.9 dB
3.6 GHz
Radiolocation, aero communication and navigation, satellite, mobile, S-band
82.5 dB
802.11a, 802.11n, aero communication and navigation, mobile communication,
space and satellite operation, C-band (4 GHz to 8 GHz)
95.5 dB
5 GHz
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The electromagnetic interference (EMI) rejection ratio, or EMIRR, describes the EMI immunity of operational
amplifiers. An adverse effect that is common to many op amps is a change in the offset voltage as a result of RF
signal rectification. An op amp that is more efficient at rejecting this change in offset as a result of EMI has a
higher EMIRR and is quantified by a decibel value. Measuring EMIRR can be performed in many ways, but this
section provides the EMIRR +IN, which specifically describes the EMIRR performance when the RF signal is
applied to the noninverting input pin of the op amp. In general, only the noninverting input is tested for EMIRR for
the following three reasons:
• Op amp input pins are known to be the most sensitive to EMI, and typically rectify RF signals better than the
supply or output pins.
• The noninverting and inverting op amp inputs have symmetrical physical layouts and exhibit nearly matching
EMIRR performance
• EMIRR is more simple to measure on noninverting pins than on other pins because the noninverting input
terminal can be isolated on a PCB. This isolation allows the RF signal to be applied directly to the
noninverting input terminal with no complex interactions from other components or connecting PCB traces.
High-frequency signals conducted or radiated to any pin of the operational amplifier may result in adverse
effects, as the amplifier would not have sufficient loop gain to correct for signals with spectral content outside the
bandwidth. Conducted or radiated EMI on inputs, power supply, or output may result in unexpected dc offsets,
transient voltages, or other unknown behavior. Take care to properly shield and isolate sensitive analog nodes
from noisy radio signals and digital clocks and interfaces.
The EMIRR +IN of the TLV2186 is plotted versus frequency as shown in Figure 40. The TLV2186 unity-gain
bandwidth is 750 kHz. EMIRR performance below this frequency denotes interfering signals that fall within the op
amp bandwidth.
7.3.4.1 EMIRR +IN Test Configuration
Figure 41 shows the circuit configuration for testing the EMIRR +IN. An RF source is connected to the op amp
noninverting input terminal using a transmission line. The op amp is configured in a unity-gain buffer topology
with the output connected to a low-pass filter (LPF) and a digital multimeter (DMM). A large impedance mismatch
at the op amp input causes a voltage reflection; however, this effect is characterized and accounted for when
determining the EMIRR IN+. The multimeter samples and measures the resulting DC offset voltage. The LPF
isolates the multimeter from residual RF signals that may interfere with multimeter accuracy.
Ambient temperature: 25Û&
V+
±
Low-Pass
Filter
50
+
RF Source
DC Bias: 0 V
Modulation: None (CW)
Frequency Sweep: 201 pt. Log
V±
Sample /
Averaging
Digital
Multimeter
Not shown: 0.1 µF and 10 µF supply decoupling
Figure 41. EMIRR +IN Test Configuration
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7.3.5 Electrical Overstress
Designers often ask questions about the capability of an operational amplifier to withstand electrical overstress.
These questions tend to focus on the device inputs, but may involve the supply voltage pins or even the output
pin. Each of these different pin functions have electrical stress limits determined by the voltage breakdown
characteristics of the particular semiconductor fabrication process and specific circuits connected to the pin.
Additionally, internal electrostatic discharge (ESD) protection is built into these circuits to protect from accidental
ESD events both before and during product assembly.
Having a good understanding of this basic ESD circuitry and the relevance to an electrical overstress event is
helpful. See Figure 42 for an illustration of the ESD circuits contained in the TLV2186 (indicated by the dashed
line area). The ESD protection circuitry involves several current-steering diodes connected from the input and
output pins and routed back to the internal power-supply lines, where the diodes meet at an absorption device
internal to the operational amplifier. This protection circuitry is intended to remain inactive during normal circuit
operation.
An ESD event produces a short-duration, high-voltage pulse that is transformed into a short-duration, highcurrent pulse while discharging through a semiconductor device. The ESD protection circuits are designed to
provide a current path around the operational amplifier core to prevent damage. The energy absorbed by the
protection circuitry is then dissipated as heat.
When an ESD voltage develops across two or more amplifier device pins, current flows through one or more
steering diodes. Depending on the path that the current takes, the absorption device may activate. The
absorption device has a trigger or threshold voltage that is greater than the normal operating voltage of the
TLV2186, but less than the device breakdown voltage level. When this threshold is exceeded, the absorption
device quickly activates and clamps the voltage across the supply rails to a safe level.
When the operational amplifier connects into a circuit, as shown in Figure 42, the ESD protection components
are intended to remain inactive, and do not become involved in the application circuit operation. However,
circumstances may arise where an applied voltage exceeds the operating voltage range of a given pin. If this
condition occurs, there is a risk that some internal ESD protection circuits may be biased on, and conduct
current. Any such current flow occurs through steering-diode paths and rarely involves the absorption device.
(2)
TVS
RF
V+
RI
ESD CurrentSteering Diodes
IN
(3)
RS
Op Amp
Core
+IN
Edge-Triggered ESD
Absorption Circuit
ID
VIN
OUT
RL
(1)
V±
(2)
TVS
(1)
VIN = (V+) + 500 mV
(2)
TVS: 26 V > VTVSBR (min) > V+ ; where VTVSBR
breakdown voltage.
(3)
Suggested value is approximately 5 kΩ in example overvoltage condition.
(min)
is the minimum specified value for the transient voltage suppressor
Figure 42. Equivalent Internal ESD Circuitry Relative to a Typical Circuit Application
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Figure 42 shows a specific example where the input voltage (VIN) exceeds the positive supply voltage (V+) by
500 mV or more. Much of what happens in the circuit depends on the supply characteristics. If V+ can sink the
current, one of the upper input steering diodes conducts and directs current to +VS. Excessively high current
levels can flow with increasingly higher VIN. As a result, the data sheet specifications recommend that
applications limit the input current to 10 mA.
If the supply is not capable of sinking the current, VIN may begin sourcing current to the operational amplifier, and
then take over as the source of positive supply voltage. The danger in this case is that the voltage can rise to
levels that exceed the operational amplifier absolute maximum ratings.
Another common question involves what happens to the amplifier if an input signal is applied to the input while
the power supplies V+ or V– are at 0 V. Again, this question depends on the supply characteristic while at 0 V, or
at a level below the input signal amplitude. If the supplies appear as high impedance, then the operational
amplifier supply current may be supplied by the input source through the current-steering diodes. This state is not
a normal bias condition; the amplifier most likely does not operate normally. If the supplies are low impedance,
then the current through the steering diodes can become quite high. The current level depends on the ability of
the input source to deliver current, and any resistance in the input path.
If there is any uncertainty about the ability of the supply to absorb this current, external zener diodes must be
added to the supply pins, as shown in Figure 42. The zener voltage must be selected such that the diode does
not turn on during normal operation. However, the zener voltage must be low enough so that the zener diode
conducts if the supply pin begins to rise above the safe operating supply voltage level.
7.3.6 MUX-Friendly Inputs
The TLV2186 features a proprietary input stage design that allows an input differential voltage to be applied while
maintaining high input impedance. Typically, high-voltage CMOS or bipolar-junction input amplifiers feature
antiparallel diodes that protect input transistors from large VGS voltages that may exceed the semiconductor
process maximum and permanently damage the device. Large VGS voltages can be forced when applying a large
input step, switching between channels, or attempting to use the amplifier as a comparator.
The TLV2186 solves these problems with a switched-input technique that prevents large input bias currents
when large differential voltages are applied. This solves many issues seen in switched or multiplexed
applications, where large disruptions to RC filtering networks are caused by fast switching between large
potentials. The TLV2186 offers outstanding settling performance as a result of these design innovations and
built-in slew rate boost and wide bandwidth. The TLV2186 can also be used as a comparator. Differential and
common-mode Absolute Maximum Ratings still apply relative to the power supplies.
7.4 Device Functional Modes
The TLV2186 has a single functional mode, and is operational when the power-supply voltage is greater than 4.5
V (±2.25 V). The maximum power supply voltage for the TLV2186 is 24 V (±12 V).
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TLV2186 operational amplifier combines precision offset and drift with excellent overall performance, making
the device an excellent choice for many precision applications. The precision offset drift of only 0.1 µV/°C
provides stability over the entire temperature range. In addition, the device pairs excellent CMRR, PSRR, and
AOL dc performance with outstanding low-noise operation. As with all amplifiers, applications with noisy or highimpedance power supplies require decoupling capacitors close to the device pins. In most cases, 0.1-µF
capacitors are adequate.
The following application examples highlight only a few of the circuits where the TLV2186 can be used.
8.1.1 Basic Noise Calculations
Low-noise circuit design requires careful analysis of all noise sources. External noise sources can dominate in
many cases; consider the effect of source resistance on overall op amp noise performance. Total noise of the
circuit is the root-sum-square combination of all noise components.
The resistive portion of the source impedance produces thermal noise proportional to the square root of the
resistance. The source impedance is usually fixed; consequently, select the op amp and the feedback resistors
to minimize the respective contributions to the total noise.
Figure 43 illustrates both noninverting (A) and inverting (B) op amp circuit configurations with gain. In circuit
configurations with gain, the feedback network resistors also contribute noise. In general, the current noise of the
op amp reacts with the feedback resistors to create additional noise components. However, the extremely low
current noise of the TLV2186 means that the current noise contribution can be neglected.
The feedback resistor values can generally be chosen to make these noise sources negligible. Low impedance
feedback resistors load the output of the amplifier. The equations for total noise are shown for both
configurations.
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Application Information (continued)
(A) Noise in Noninverting Gain Configuration
R1
Noise at the output is given as EO, where
R2
GND
±
EO
+
RS
+
±
VS
Source
GND
'1 = l1 +
:2;
A5 = ¥4 „ G$ „ 6(-) „ 45
d
:3;
A41 æ42 = ¨4 „ G$ „ 6(-) „ d
8
41 „ 42
h d
h
41 + 42
¾*V
Thermal noise of R1 || R2
:4;
G$ = 1.38065 „ 10F23
Boltzmann Constant
:5;
,
h
-
6(-) = 237.15 + 6(°%)
(B) Noise in Inverting Gain Configuration
R1
RS
R2
h
>-?
Thermal noise of RS
Temperature in kelvins
:45 + 41 ; „ 42
42
2
p „ ¨:A0 ;2 + kA41 +45 æ42 o + FE0 „ H
IG
45 + 41
45 + 41 + 42
:6;
'1 = l1 +
+
:7;
:45 + 41 ; „ 42
8
I d
A41 +45 æ42 = ¨4 „ G$ „ 6(-) „ H
h
45 + 41 + 42
¾*V Thermal noise of (R1 + RS) || R2
GND
:8;
G$ = 1.38065 „ 10F23
:9;
6(-) = 237.15 + 6(°%)
±
+
±
d
8
¾*V
> 84/5 ?
Noise at the output is given as EO, where
EO
VS
42
41 „ 42 2
2
p „ ¨:A5 ;2 + :A0 ;2 + kA41 æ42 o + :E0 „ 45 ;2 + lE0 „ d
hp
41
41 + 42
:1;
Source
GND
d
,
h
-
2
> 84/5 ?
Boltzmann Constant
>-?
Temperature in kelvins
Copyright © 2017, Texas Instruments Incorporated
(1)
en is the voltage noise spectral density of the amplifier. For the TLV2186 series of operational amplifiers, en = 38 nV/
√Hz at 1 kHz.
(2)
For additional resources on noise calculations visit TI Precision Labs.
Figure 43. Noise Calculation in Gain Configurations
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8.2 Typical Applications
8.2.1 High-Side Current Sensing
RL
+
24 V
±
IL
Load Current
R1
R2
+ 24 V ±
+
R1 = R3, R2 = R4
VO
±
R3
TLV2186
R4
Figure 44. High-Side Current Monitor
8.2.1.1 Design Requirements
A common systems requirement is to monitor the current being delivered to a load. Monitoring makes sure that
normal current levels are being maintained, and also provides an alert if an overcurrent condition occurs.
Fortunately, a relatively simple current monitor solution can be achieved using a precision rail-to-rail input/output
op amp such as the TLV2186. This device has an input common-mode voltage (VCM) range that extends 200 mV
beyond each power supply rail allowing for operation at the supply rail.
The TLV2186 is configured as a difference amplifier with a predetermined gain. The difference amplifier inputs
are connected across a sense resistor through which the load current flows. The sense resistor may be
connected to the high side or low side of the circuit through which the load current flows. Commonly, high-side
current sensing is applied and an applicable TLV2186 configuration is in Figure 44. Low-side current sensing
may be applied as well if the sense resistor can be placed between the load and ground.
Use the following parameters for this design example:
• Single supply: 24 V
• Linear output voltage range: 0.3 V to 3.3 V
• Iload: 1 A to 11 A
The design details and equations below can be used to reconfigure this design for different output voltage ranges
and current loads.
8.2.1.2 Detailed Design Procedure
Designing a high-side current monitor circuit is straightforward providing the amplifier electrical characteristics are
carefully consideration so that linear operation is maintained. Other additional considerations, such as the input
voltage range of the analog to digital converter (ADC) that follows the current monitor stage, must be kept in
mind while configuring the system.
Consider the design of a TLV2186 high-side current monitor with an output voltage range set to be compatible
with the input of ADC with an input range of 3.3 V, such as one integrated in a microcontroller. The full-scale
input range of such a converter is 0 V to 3.3 V. The TLV2186 can be operated from a single 24-V supply,
referenced to ground. Although the TLV2186 is specified as a rail-to-rail input/output (RRIO) amplifier, the linear
output operating range (like all amplifiers) does not quite extend all the way to the supply rails. This linear
operating range must be taken into consideration.
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Typical Applications (continued)
The TLV2186 is powered by 24 V; therefore, the device is easily capable of providing the 3.3-V positive level, or
even more if the ADC has a wider input range. However, because the TLV2186 output does not swing
completely to 0 V, the specified lower swing limit must be observed in the design.
The best measure of an op amp linear output voltage range comes from the open-loop voltage gain (AOL)
specification listed in the Electrical Characteristics table. The AOL test conditions specify a linear swing range 300
mV from each supply rail (RL = 10 kΩ). Therefore, the linear swing limit on the low end (VoMIN) is 300 mV, and
3.3 V is the VoMAX limit, thus yielding an 11:1 VoMAX to VoMIN ratio. This ratio proves important in determining the
difference amplifier operating parameters.
An optimum load current, ILOAD of 10 A is used as an example. In most applications, however, the ability to
monitor current levels well below 10 A is useful. This situation is where the 11:1 VoMAX to VoMIN ratio is crucial. If
11 A is set as the maximum current, this current must correspond to a 3.3-V output. Using the 11:1 ratio, the
minimum current of 1 A corresponds to 300 mV.
Selection of the current sense resistor RS comes down to how much voltage drop can be tolerated at maximum
current and the permissible power loss, or dissipation. A good compromise for a 10-A sense application is an RS
of 10 mΩ. That value results in a power dissipation of 1 W, and a 0.1-V drop at 10 amps.
Next, determine the gain of the TLV2186 difference amplifier circuit. The maximum current of 11 A flowing
through a 10-mΩ sense resistor results in 110 mV across the resistor. That voltage appears as a differential
voltage, VR, that is applied across the TLV2186 difference amplifier circuit inputs:
VS
IL * RS
VS
11 A * 10 m :
110 mV
(1)
The TLV2186 required voltage gain is determined from:
VOMAX
GA
VS
GA
3.3 V
0.11 V
30
V
V
(2)
Now, checking the VoMIN using IL = 1 A:
VOMIN GA * ISMIN * RS
VOMIN
30
V
* 1 A * 10 m:
V
300 mV
(3)
The complete TLV2186 high-side current monitor is shown in Figure 45. The circuit is capable of monitoring a
current range of < 1 A to 11 A, with a VCM very close to the 24-V supply voltage.
RL = 10 m
+
24 V
±
IL
Load Current
1 A to 11 A
R1 = 1 k
R2 = 30 k
+ 3.3 V ±
+ 24 V ±
±
R3 = 1 k
+
R4 = 30 k
TLV2186
µController
ADC
VO = 300 mV to 3.3 V
Figure 45. TLV2186 Configured as a High-Side Current Monitor
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Typical Applications (continued)
In this example, the TLV2186 output voltage is intentionally limited to 3.3 V. However, because of the 24-V
supply, the output voltage could be much higher to allow for a higher-voltage data converter with a higher
dynamic range.
The circuit in Figure 45 was checked using the TINA Spice circuit simulation tool to verify the correct operation of
the TLV2186 high-side current monitor. The simulation results are seen in Figure 46. The performance is exactly
as expected. Upon careful inspection of the plots, one possible surprise is that VO continues towards zero as the
sense current drops below 1 A, where VO is 300 mV and less.
VLOAD
RL = 10 m
+
24 V
±
IL
Load Current
0 A to 11 A
R1 = 1 k
R2 = 30 k
+ 24 V ±
100 nF
R3 = 1 k
VO = 0 V to 3.3 V
±
+
TLV2186
R4 = 30 k
RL = 10 k
Figure 46. TLV2186 High-Side Current-Monitor Simulation Schematic
The TLV2186 output, as well as other CMOS output amplifiers, often swing closer to 0 V than the linear output
parameters suggest. The Electrical Characteristics table lists under the OUTPUT subsection VO, which is an
output slam to the rail measure. It is not an indication of the linear output range, but instead how close the output
can move towards the supply rail. In that region, the amplifier output approaches saturation, and the amplifier
ceases to operate linearly. Thus, in the current-monitor application, the current-measurement capability may
continue well below the 300 mV output level. However, keep in mind that the linearity errors are becoming large.
Lastly, some notes about maximizing the high-side current monitor performance:
• All resistor values are critical for accurate gain results. The resistor pairs of [R1 and R3] and [R2 and R4]
must be matched as closely as possible to minimize common-mode mismatch error. Use a 0.1% tolerance, or
better. Often, selecting two adjacent resistors on a reel provides close matching compared to random
selection.
• Keep the closed-loop gain, GA, to which the TLV2186 difference amplifier is set, to a reasonable value. Doing
so reduces gain error and can be used to maximize bandwidth. A GA of 30 V/V is used in the example.
• Although current monitoring is often used for monitoring dc supply currents, ac current can also be monitored.
The –3-dB bandwidth, or upper cutoff frequency, of the circuit of is:
GBW
fH
Noise Gain
where
•
•
GNG
24
GBW is the amplifier unity gain bandwidth; 750 kHz for the TLV2186.
Noise gain is equal to the gain as seen looking into the op amp noninverting input, as shown in Equation 5. (4)
1
R2
R1
(5)
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Typical Applications (continued)
For the TLV2186 circuit in Figure 45:
GNG
fH
30 k:
V
31
1 k:
V
750 kHz
24.2 kHz
31
1
Make sure that the amplifier slew rate is sufficient to support the expected output voltage swing range and
waveform. Also, if a single power supply such as 24 V is used, the ac power source applied to the sense input
must have a positive dc component to keep the VCM above 0 V. The input voltage cannot drop below 0 V if
normal operation is to be maintained.
The TLV2186 output can attain a 0 V output level if a small negative voltage is used to power the V– pin instead
of ground. The LM7705 is a switched capacitor voltage inverter with a regulated, low-noise, –0.23-V fixed voltage
output. Powering the TLV2186 V– pin at this level approximately matches the 300-mV linear output voltage swing
lower limit, thus extending the output swing to 0 V, or very near 0 V. Doing so greatly improves the resolution at
low sense current levels.
The LM7705 requires only about 78 μA of quiescent current, but be aware that the specified supply range is 3 V
to 5.25 V. The 3.3-V or 5-V supply used by the ADC could be tapped as a power source.
For more information about amplifier-based, high-side current monitors, see the TI Analog Engineer’s Circuit
Cookbook: Amplifiers.
8.2.1.3 Application Curve
4
24.2
VOUT
VLOAD
24.16
3.2
24.12
2.8
24.08
2.4
24.04
2
24
1.6
23.96
1.2
23.92
0.8
23.88
0.4
23.84
0
0
1
2
3
4
5
6
7
Input Current (A)
8
9
10
Load Voltage (V)
Output Voltage (V)
3.6
23.8
11
Figure 47. High-Side Results
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Typical Applications (continued)
8.2.2 Bridge Amplifier
Figure 48 shows the basic configuration for a bridge amplifier. Click the following link to download the TINA-TI
file: Bridge Amplifier Circuit.
VEX
R1
R
R
R
R
+5V
VOUT
VREF
Copyright © 2017, Texas Instruments Incorporated
Figure 48. Bridge Amplifier
8.2.3 Low-Side Current Monitor
Figure 49 shows the TLV2186 configured in a low-side current-sensing application. The load current (ILOAD)
creates a voltage drop across the shunt resistor (RSHUNT). This voltage is amplified by the TLV2186, with a gain
of 201. In this example, the load current is set from 0 A to 500 mA, and corresponds to an output voltage range
from 0 V to 10 V. The output range can be adjusted by changing the shunt resistor or gain of the configuration.
Click the following link to download the TINA-TI file: Current-Sensing Circuit.
VSYSTEM
Load
15 V
+
VOUT = ILOAD * RSHUNT(1 + RF / RIN)
TLV2186
ILOAD
RSHUNT
100 m
VOUT / ILOAD= 1 V / 49.75 mA
±
RIN
100
VOUT
RF
20 k
CF
150 pF
Figure 49. Low-Side Current Monitor
26
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Typical Applications (continued)
8.2.4 RTD Amplifier With Linearization
See the Analog Linearization of Resistance Temperature Detectors technical brief for an in-depth analysis of
Figure 50. Click the following link to download the TINA-TI file: RTD Amplifier with Linearization.
15 V
(5 V)
Out
REF5050
In
1 …F
1 …F
R2
49.1 kŸ
R3
60.4 kŸ
R1
4.99 kŸ
TLV2186
0°C = 0 V
200°C = 5 V
R5
(1)
105.8 kŸ
RTD
Pt100
(1)
V OUT
R4
1 kŸ
R5 provides positive-varying excitation to linearize output.
Figure 50. RTD Amplifier With Linearization
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9 Power Supply Recommendations
The TLV2186 is specified for operation from 4.5 V to 24 V (±2.25 V to ±12 V); many specifications apply from
–40°C to +125°C. The Typical Characteristics presents parameters that can exhibit significant variance with
regard to operating voltage or temperature.
CAUTION
Supply voltages larger than 40 V can permanently damage the device (see the
Absolute Maximum Ratings).
Place 0.1-μF bypass capacitors close to the power-supply pins to reduce errors coupling in from noisy or
high-impedance power supplies. For more detailed information on bypass capacitor placement, see the Layout
section.
10 Layout
10.1 Layout Guidelines
For best operational performance of the device, use good PCB layout practices, including:
• For the lowest offset voltage, avoid temperature gradients that create thermoelectric (Seebeck) effects in the
thermocouple junctions formed from connecting dissimilar conductors. Also:
– Use low thermoelectric-coefficient conditions (avoid dissimilar metals).
– Thermally isolate components from power supplies or other heat sources.
– Shield operational amplifier and input circuitry from air currents, such as cooling fans.
• Noise can propagate into analog circuitry through the power pins of the circuit as a whole and the op amp
itself. Bypass capacitors reduce the coupled noise by providing low-impedance power sources local to the
analog circuitry.
– Connect low-ESR, 0.1-µF ceramic bypass capacitors between each supply pin and ground, placed as
close as possible to the device. A single bypass capacitor from V+ to ground is applicable for singlesupply applications.
• Separate grounding for analog and digital portions of circuitry is one of the simplest and most effective
methods of noise suppression. One or more layers on multilayer PCBs are usually devoted to ground planes.
A ground plane helps distribute heat and reduces EMI noise pickup. Make sure to physically separate digital
and analog grounds paying attention to the flow of the ground current. For more detailed information, seeThe
PCB is a component of op amp design.
• To reduce parasitic coupling, run the input traces as far away as possible from the supply or output traces. If
these traces cannot be kept separate, crossing the sensitive trace perpendicular is much better as opposed to
in parallel with the noisy trace.
• Place the external components as close as possible to the device. As illustrated in Figure 51, keep the
feedback resistor (R3) and gain resistor (R4) close to the inverting input to minimize parasitic capacitance.
• Keep the length of input traces as short as possible. Always remember that the input traces are the most
sensitive part of the circuit.
• Consider a driven, low-impedance guard ring around the critical traces. A guard ring can significantly reduce
leakage currents from nearby traces that are at different potentials.
• For best performance, clean the PCB following board assembly.
• Any precision integrated circuit may experience performance shifts due to moisture ingress into the plastic
package. Following any aqueous PCB cleaning process, bake the PCB assembly to remove moisture
introduced into the device packaging during the cleaning process. A low-temperature, post-cleaning bake at
85°C for 30 minutes is sufficient for most circumstances.
28
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10.2 Layout Example
GND
+V
R3
Use ground pours for
shielding the input
signal pairs
Place bypass
capacitors as close to
device as possible
(avoid use of vias)
C3
C4
C3
R3
IN±
1
NC
NC
C4
8
IN±
IN+
1
NC
NC
8
2
±IN
V+
7
3
+IN
OUT
6
4
V±
NC
5
+V
R1
R1
2
±IN
±
V+
7
3
+IN
+
OUT
6
R2
4
V±
NC
5
OUT
OUT
R2
-V
C1
IN+
R4
GND
R4
C2
Place components
close to device and to
each other to reduce
parasitic errors
C1
-V
Use a lowESR,ceramic bypass
capacitor
C2
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Figure 51. Operational Amplifier Board Layout for Difference Amplifier Configuration
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 TINA-TI™ (Free Software Download)
TINA-TI™ is a simple, powerful, and easy-to-use circuit simulation program based on a SPICE engine. TINATI™ is a free, fully-functional version of the TINA™ software, preloaded with a library of macromodels in addition
to a range of both passive and active models. TINA-TI™ provides all the conventional dc, transient, and
frequency domain analysis of SPICE, as well as additional design capabilities.
Available as a free download from the Analog eLab Design Center, TINA-TI™ offers extensive post-processing
capability that allows users to format results in a variety of ways. Virtual instruments offer the ability to select
input waveforms and probe circuit nodes, voltages, and waveforms, creating a dynamic quick-start tool.
NOTE
These files require that either the TINA software (from DesignSoft™) or TINA-TI™
software be installed. Download the free TINA-TI™ software from the TINA-TI™ folder.
11.1.1.2 TI Precision Designs
TI Precision Designs are available online at http://www.ti.com/ww/en/analog/precision-designs/. TI Precision
Designs are analog solutions created by TI’s precision analog applications experts and offer the theory of
operation, component selection, simulation, complete PCB schematic and layout, bill of materials, and measured
performance of many useful circuits.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
• Texas Instruments, Zero-drift Amplifiers: Features and Benefits
• Texas Instruments, The PCB is a component of op amp design
• Texas Instruments, Op Amps for Everyone
• Texas Instruments, Operational amplifier gain stability, Part 3: AC gain-error analysis
• Texas Instruments, Operational amplifier gain stability, Part 2: DC gain-error analysis
• Texas Instruments, Using infinite-gain, MFB filter topology in fully differential active filters
• Texas Instruments, Op Amp Performance Analysis
• Texas Instruments, Single-Supply Operation of Operational Amplifiers
• Texas Instruments, Tuning in Amplifiers
• Texas Instruments, Shelf-Life Evaluation of Lead-Free Component Finishes
• Texas Instruments, Feedback Plots Define Op Amp AC Performance
• Texas Instruments, EMI Rejection Ratio of Operational Amplifiers
• Texas Instruments, Analog Linearization of Resistance Temperature Detectors
• Texas Instruments, TI Precision Design TIPD102 High-Side Voltage-to-Current (V-I) Converter
30
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11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
11.5 Trademarks
TINA-TI, E2E are trademarks of Texas Instruments.
Bluetooth is a registered trademark of Bluetooth SIG, Inc.
TINA, DesignSoft are trademarks of DesignSoft, Inc.
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical packaging and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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28-Sep-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TLV2186IDR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
T2186
TLV2186IDSGR
ACTIVE
WSON
DSG
8
3000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
PVDY
TLV2186IDSGT
ACTIVE
WSON
DSG
8
250
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
PVDY
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
