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TLV521
SNOSD26 – MAY 2016
TLV521 NanoPower, 350nA, RRIO, CMOS Input, Operational Amplifier
1 Features
2 Applications
•
•
•
•
•
•
•
1
Unmatched Price Performance
For VS = 3.3 V, Typical Unless Otherwise Noted
– Ultra-low Supply Current
– 350 nA Typical, 500 nA Maximum
– Wide Operating Voltage Range 1.7 V to 5.5 V
– Low TCVOS 1.5 µV/°C
– VOS 3 mV (Max)
– Input Bias Current 1 pA
– PSRR 100 dB
– CMRR 90 dB
– Open-Loop Gain 110 dB
– Gain Bandwidth Product 6 kHz
– Slew Rate 2.5 V/ms
– Input Voltage Noise at f = 100 Hz 300 nV/√Hz
– Temperature Range –40ºC to 125°C
– Rail to Rail Input and Output (RRIO)
Wireless Remote Sensors
Powerline Monitoring
Power Meters
Battery Powered Industrial Sensors
Micropower Oxygen Sensor and Toxic Gas
Sensor
Active RFID Readers
Zigbee Based Sensors for HVAC Control
Sensor Network Powered by Energy Scavenging
Current Sensing
Glucose Monitoring
•
•
•
•
•
3 Description
The TLV521 350 nA nanopower op amp offers
optimum price performance in TI's nanopower family
of operational amplifiers. The TLV521 has a carefully
designed CMOS input stage enabling very low Ibias
of 1 pA, thereby reducing IBIAS and IOS errors that
would otherwise impact sensitive applications like
Megaohm resistance, high-impedance photodiode
and charge sense situations. Additionally, built-in EMI
protection reduces sensitivity to unwanted RF signals
from sources like mobile phones and RFID readers.
The TLV521 is offered in the 5-pin SC70 package,
and operates from –40°C to 125°C.
Device Information(1)
PART NUMBER
TLV521
PACKAGE
BODY SIZE (NOM)
SC70 (5)
2.00 mm x 1.25 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Nanopower Supply Current
800
SUPPLY CURRENT (nA)
700
600
125°C
500
85°C
400
300
25°C
200
VCM = 0.3 V
-40°C
100
0
1
2
3
4
5
6
SUPPLY VOLTAGE (V)
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TLV521
SNOSD26 – MAY 2016
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Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
2
3
6.1
6.2
6.3
6.4
6.5
6.6
6.7
3
3
3
3
4
5
6
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information .................................................
Electrical Characteristics...........................................
AC Electrical Characteristics....................................
Typical Characteristics ..............................................
7.4 Device Functional Modes........................................ 11
8
Applications and Implementation ...................... 12
8.1 Application Information............................................ 12
8.2 Typical Applications ................................................ 13
9 Power Supply Recommendations...................... 17
10 Layout................................................................... 18
10.1 Layout Guidelines ................................................. 18
10.2 Layout Example .................................................... 18
11 Device and Documentation Support ................. 19
11.1
11.2
11.3
11.4
11.5
11.6
Detailed Description ............................................ 11
7.1 Overview ................................................................. 11
7.2 Functional Block Diagram ....................................... 11
7.3 Feature Description................................................. 11
Device Support ....................................................
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
19
19
19
19
19
19
12 Mechanical, Packaging, and Orderable
Information ........................................................... 20
4 Revision History
DATE
REVISION
NOTES
May 2016
*
Initial release.
5 Pin Configuration and Functions
DCK Package
5-Pin (SC70) SC70-5
Top View
1
5
OUT
V
-
+
V
2
+
-
3
4
IN-
IN+
Pin Functions
PIN
2
TYPE
DESCRIPTION
NO.
NAME
1
OUT
O
Output
2
V–
P
Negative Power Supply
3
IN+
I
Noninverting Input
4
IN–
I
Inverting Input
5
V+
P
Positive Power Supply
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6 Specifications
6.1 Absolute Maximum Ratings (1)
over operating free-air temperature range (unless otherwise noted) (1)
Any pin relative to V–
MIN
MAX
UNIT
−0.3
6
V
–
IN+, IN–, OUT Pins
+
V – 0.3 V
V+, V–, OUT Pins
V + 0.3 V
V
40
mA
Differential Input Voltage (VIN+ - VIN–)
–300
300
mV
Junction Temperature
–40
Mounting Temperature
150
°C
Infrared or Convection (30 sec.)
260
°C
Wave Soldering Lead Temp. (4 sec.)
260
°C
150
°C
−65
Storage temperature, Tstg
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
V(ESD)
(1)
(2)
Electrostatic discharge
(1)
UNIT
±2000
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
±1000
Machine Model
±200
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions (1)
over operating free-air temperature range (unless otherwise noted)
MIN
MAX
UNIT
Temperature Range
−40
125
°C
Supply Voltage (VS = V+ - V−)
1.7
5.5
V
(1)
Absolute Maximum Ratings indicate limits beyond which damage may occur. Recommended Operating Conditions indicate conditions
for which the device is intended to be functional, but specific performance is not ensured. For ensured specifications and test conditions,
see Electrical Characteristics.
6.4 Thermal Information
TLV521
THERMAL METRIC (1)
DCK (SC70)
UNIT
5 PINS
RθJA
Junction-to-ambient thermal resistance
269.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
93.7
°C/W
RθJB
Junction-to-board thermal resistance
48.8
°C/W
ψJT
Junction-to-top characterization parameter
2
°C/W
ψJB
Junction-to-board characterization parameter
47.9
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
N/A
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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6.5 Electrical Characteristics
Unless otherwise specified, all limits for TA = 25°C, V+ = 3.3 V, V− = 0 V, VCM = VO = V+/2, and RL > 1 MΩ. (1)
PARAMETER
VOS
Input Offset Voltage
TCVOS
Input Offset Voltage Drift
IBIAS
Input Bias Current
IOS
Input Offset Current
CMRR
Common Mode Rejection Ratio
TEST CONDITIONS
MIN
TYP
MAX
VCM = 0.3 V
–3
0.1
3
VCM = 3 V
–3
0.1
3
70
0 V ≤ VCM ≤ 2.2 V
1
pA
50
fA
90
dB
100
PSRR
Power Supply Rejection Ratio
V+ = 1.8 V to 3.3 V; VCM = 0.3 V
CMVR
Common Mode Voltage Range
CMRR ≥ 70 dB
AVOL
Large Signal Voltage Gain
VO = 0.5 V to 2.8 V
RL = 100 kΩ to V+/2
VO
Output Swing High
RL = 100 kΩ to V+/2
VIN(diff) = 100 mV
3
50
Output Swing Low
RL = 100 kΩ to V+/2
VIN (diff) = −100 mV
2
50
Output Current
Sourcing, VO to V−
VIN(diff) = 100 mV
11
Sinking, VO to V+
VIN(diff) = −100 mV
12
IO
IS
(1)
4
Supply Current
VCM = 0.3 V
80
100
0
80
mV
μV/°C
±1.5
0 V ≤ VCM ≤ 3.3 V
UNIT
dB
3.3
110
350
V
dB
mV from
either rail
mA
500
nA
Electrical Characteristics values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in
very limited self-heating of the device such that TJ = TA. Parametric performance, as indicated in the electrical tables, is not ensured
under conditions of self heating where TJ > TA.Absolute Maximum Ratings indicate junction temperature limits beyond which the device
may be permanently degraded, either mechanically or electrically.
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AC Electrical Characteristics (1)
6.6
Unless otherwise specified, all limits for TA = 25°C, V+ = 3.3 V, V− = 0 V, VCM = VO = V+/2, and RL > 1 MΩ.
PARAMETER
TEST CONDITIONS
GBW
Gain-Bandwidth Product
CL = 20 pF, RL = 100 kΩ
SR
Slew Rate
AV = +1,
VIN = 0 V to 3.3 V
MIN
(2)
TYP
(3)
6
Falling Edge
2.9
Rising Edge
2.5
MAX
(2)
UNIT
kHz
V/ms
θm
Phase Margin
CL = 20 pF, RL = 100 kΩ
73
Gm
Gain Margin
CL = 20 pF, RL = 100 kΩ
19
dB
en
Input-Referred Voltage Noise Density
f = 100 Hz
300
nV/√Hz
deg
Input-Referred Voltage Noise
0.1 Hz to 10 Hz
22
μVPP
In
Input-Referred Current Noise
f = 100 Hz
100
fA/√Hz
EMIRR
EMI Rejection Ratio, IN+ and IN− (4)
VRF_PEAK = 100 mVP (−20 dBP),
f = 400 MHz
121
VRF_PEAK = 100 mVP (−20 dBP),
f = 900 MHz
121
VRF_PEAK = 100 mVP (−20 dBP),
f = 1800 MHz
124
VRF_PEAK = 100 mVP (−20 dBP),
f = 2400 MHz
142
(1)
(2)
(3)
(4)
dB
Electrical Characteristics values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in
very limited self-heating of the device such that TJ = TA. Parametric performance, as indicated in the electrical tables, is not ensured
under conditions of self heating where TJ > TA. Absolute Maximum Ratings indicate junction temperature limits beyond which the device
may be permanently degraded, either mechanically or electrically.
All limits are ensured by testing, statistical analysis or design.
Typical values represent the most likely parametric norm at the time of characterization. Actual typical values may vary over time and
will also depend on the application and configuration. The typical values are not tested and are not ensured on shipped production
material.
The EMI Rejection Ratio is defined as EMIRR = 20log (VRF_PEAK/ΔVOS).
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6.7 Typical Characteristics
At TJ = 25°C, unless otherwise specified.
800
800
700
700
600
SUPPLY CURRENT (nA)
SUPPLY CURRENT (nA)
125°C
125°C
500
85°C
400
300
25°C
200
VCM = 0.3 V
-40°C
600
85°C
500
400
25°C
300
-40°C
200
VCM = VS ± 0.3 V
100
100
0
1
2
3
4
5
0
1
6
2
Figure 1. Supply Current vs. Supply Voltage
5
6
150
18
VS = 3.3V
100
TA = 25oC
16
VS = 3.3 V
-40°C
VCM = VS/2
14
VOS (μV)
50
12
10
8
25°C
0
-50
6
85°C
4
-100
2
125°C
-150
-0.1
0
-1.0 -0.8 -0.6 -0.4 -0.2 0.0 0.2 0.4 0.6 0.8 1.0
0.4
0.9
1.9
2.4
2.9
3.4
Figure 4. Input Offset Voltage vs. Input Common Mode
Figure 3. Offset Voltage Distribution
150
150
VCM = 0.3 V
-40°C
VCM = VS - 0.3 V
100
100
25°C
-40°C
25°C
50
VOS (μV)
50
1.4
VCM (V)
VOS (mV)
VOS (μV)
4
Figure 2. Supply Current vs. Supply Voltage
20
PERCENTAGE (%)
3
SUPPLY VOLTAGE (V)
SUPPLY VOLTAGE (V)
0
0
-50
-50
85°C
-100
85°C
125°C
-100
125°C
-150
1
2
3
4
5
6
-150
1
VS (V)
3
4
5
6
VS (V)
Figure 5. Input Offset Voltage vs. Supply Voltage
6
2
Figure 6. Input Offset Voltage vs. Supply Voltage
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Typical Characteristics (continued)
At TJ = 25°C, unless otherwise specified.
150
150
VS = 3.3 V
-40°C
100
50
25°C
50
VOS (μV)
VOS (μV)
VS = 3.3 V
-40°C
100
0
-50
25°C
0
-50
85°C
-100
85°C
-100
125°C
-150
0.0
0.5
1.0
1.5
125°C
2.0
2.5
3.0
-150
0.0
3.5
0.5
1.0
VOUT (V)
Figure 7. Input Offset Voltage vs. Output Voltage
2.0
Figure 8. Input Offset Voltage vs. Sourcing Current
150
100
1.5
ISOURCE (mA)
16
VS = 3.3 V
-40°C
VS = 3.3V
-40°C
12
25°C
ISOURCE (mA)
VOS (μV)
50
0
-50
85°C
25°C
8
85°C
4
-100
125°C
125°C
-150
0.0
0.5
1.0
1.5
0
0.0
2.0
ISINK (mA)
0.5
1.0
1.5
2.0
2.5
3.0
3.5
+
OUTPUT VOLTAGE REFERENCED TO V (V)
Figure 9. Input Offset Voltage vs. Sinking Current
Figure 10. Sourcing Current vs. Output Voltage
16
40
-40°C
VS = 3.3V
-40°C
30
25°C
ISOURCE (mA)
ISINK (mA)
12
VCM = VS/2
8
85°C
4
25°C
20
10
85°C
125°C
125°C
0
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
0
1
OUTPUT VOLTAGE REFERENCED TO V (V)
Figure 11. Sinking Current vs. Output Voltage
2
3
4
5
6
VS (V)
-
Figure 12. Sourcing Current vs. Supply Voltage
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Typical Characteristics (continued)
At TJ = 25°C, unless otherwise specified.
5
40
RL = 100 kΩ
VCM = VS/2
VOUT FROM RAIL (mV)
4
30
ISINK (mA)
-40°C
25°C
20
10
125°C
85°C
3
2
25°C
1
85°C
-40°C
125°C
0
1
2
3
4
5
0
1
6
2
3
4
5
6
VS (V)
VS (V)
Figure 14. Output Swing High vs. Supply Voltage
Figure 13. Sinking Current vs. Supply Voltage
5
50
VOUT FROM RAIL (mV)
RL = 100 kΩ
VS = 3.3V
40
30
125°C
4
25°C
IBIAS (fA)
20
85°C
3
10
0
-10
-40°C
-20
-40°C
-30
25°C
2
1
2
3
4
5
-40
0.0
6
0.5
1.0
1.5
VS (V)
2.0
2.5
3.0
3.5
VCM (V)
Figure 15. Output Swing Low vs. Supply Voltage
Figure 16. Input Bias Current vs. Common Mode Voltage
100
15
VS = 5 V
VS = 3.3V
10
VS = 1.8 V, 3.3 V, 5 V
80
VS = 3.3 V
125°C
PSRR (dB)
IBIAS (pA)
5
0
-5
85°C
0.5
1.0
1.5
2.0
2.5
3.0
3.5
VS = 1.8 V
+PSRR
40
20
-10
-15
0.0
60
0
10
-PSRR
100
1k
10k
100k
FREQUENCY (Hz)
VCM (V)
Figure 18. PSRR vs. Frequency
Figure 17. Input Bias Current vs. Common Mode Voltage
8
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Typical Characteristics (continued)
At TJ = 25°C, unless otherwise specified.
VS = 5V
VS = 3.3 V
VS = 1.8V, 3.3V, 5V
CL = 20 pF
90
PHASE
RL = 1 MΩ
40
GAIN (dB)
CMRR (dB)
80
VS = 1.8V
70
25°C
GAIN
90
50
125°C
30
60
-40°C
0
10
50
-10
100
1e2
1k
1e3
10k
1e4
-20
100
100k
1e5
Figure 20. Frequency Response vs. Temperature
Figure 19. CMRR vs. Frequency
VS = 3.3 V
RL = 100 kΩ
CL = 20 pF
130
90
70
GAIN
50
30
0
90
40
10
RL = 10 kΩ
GAIN
20
70
50
CL = 200 pF
30
0
10
-10
-30
100k
10k
-20
100
FREQUENCY (Hz)
1k
10k
-30
100k
FREQUENCY (Hz)
Figure 21. Frequency Response vs. RL
Figure 22. Frequency Response vs. CL
3.3
15
10
3.0
FALLING EDGE
5
2.7
5 PV/DIV
SLEW RATE (V/ms)
CL = 20 pF
CL = 100 pF
-10
1k
130
RL = 10 MΩ 110
CL = 50 pF
PHASE
GAIN (dB)
RL = 10 MΩ
-20
100
VS = 3.3 V
110
PHASE (°)
40
20
60
RL = 1 MΩ
PHASE
-30
100k
10k
FREQUENCY (Hz)
FREQUENCY (Hz)
60
1k
PHASE (°)
40
10
1e1
GAIN (dB)
110
70
85°C
20
130
PHASE (°)
60
100
2.4
RISING EDGE
2.1
-5
AV = +1
-10
VOUT = VS
1.8
1.5
2.3
3.1
3.9
4.7
0
5.5
VS = 3.3V
VCM = VS/2
-15
-2.5 -1.5 -0.5 0.5 1.5 2.5 3.5 4.5 5.5 6.5 7.5
SUPPLY VOLTAGE (V)
1s/DIV
Figure 23. Slew Rate vs. Supply Voltage
Figure 24. 0.1 to 10 Hz Time Domain Voltage Noise
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Typical Characteristics (continued)
At TJ = 25°C, unless otherwise specified.
INPUT
500 mV/DIV
500 mV/DIV
INPUT
OUTPUT
OUTPUT
VS = 5V
VS = 1.8V
RL = 100 k:
RL = 100 k:
200 Ps/DIV
200 Ps/DIV
Figure 25. Large Signal Pulse Response
Figure 26. Large Signal Pulse Response
170
4
INPUT
3
OUTPUT
EMIRRV_PEAK (dB)
2
1
1V/DIV
150
0
-1
130
110
70
50
-2
+
-3
90
V = +2.5V
VS = 5 V
30
VPEAK = -20 dBVp
-
V = -2.5V
10
0.1
1.0e-1
-4
2 ms/DIV
1
1.0
10
1.0e1
100
1.0e2
1000
1.0e3
10000
1.0e4
FREQUENCY (MHz)
Figure 28. EMIRR vs. Frequency
Figure 27. Overload Recovery Waveform
10
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7 Detailed Description
7.1 Overview
The TLV521 is fabricated with Texas Instruments' state-of-the-art VIP50 process. This proprietary process
dramatically improves the performance of Texas Instruments' low-power and low-voltage operational amplifiers.
The following sections showcase the advantages of the VIP50 process and highlight circuits which enable ultralow power consumption.
7.2 Functional Block Diagram
Figure 29. Block Diagram
7.3 Feature Description
The amplifier's differential inputs consist of a noninverting input (IN+) and an inverting input (IN–). The amplifier
amplifies only the difference in voltage between the two inputs, which is called the differential input voltage. The
output voltage of the op-amp Vout is given by Equation 1:
VOUT = AOL (IN+ - IN-)
(1)
where AOL is the open-loop gain of the amplifier, typically around 100 dB.
7.4 Device Functional Modes
7.4.1 Input Stage
The TLV521 has a rail-to-rail input which provides more flexibility for the system designer. Rail-to-rail input is
achieved by using in parallel, one PMOS differential pair and one NMOS differential pair. When the common
mode input voltage (VCM) is near V+, the NMOS pair is on and the PMOS pair is off. When VCM is near V−, the
NMOS pair is off and the PMOS pair is on. When VCM is between V+ and V−, internal logic decides how much
current each differential pair will get. This special logic ensures stable and low distortion amplifier operation
within the entire common mode voltage range.
Because both input stages have their own offset voltage (VOS) characteristic, the offset voltage of the TLV521
becomes a function of VCM. VOS has a crossover point at 1.0 V below V+. Refer to the ’VOS vs. VCM’ curve in the
Typical Performance Characteristics section. Caution should be taken in situations where the input signal
amplitude is comparable to the VOS value and/or the design requires high accuracy. In these situations, it is
necessary for the input signal to avoid the crossover point. In addition, parameters such as PSRR and CMRR
which involve the input offset voltage will also be affected by changes in VCM across the differential pair transition
region.
7.4.2 Output Stage
The TLV521 output voltage swings 3 mV from rails at 3.3-V supply, which provides the maximum possible
dynamic range at the output. This is particularly important when operating on low supply voltages.
The TLV521 Maximum Output Voltage Swing defines the maximum swing possible under a particular output
load. The TLV521 output swings 50 mV from the rail at 5-V supply with an output load of 100 kΩ.
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8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TLV521 is specified for operation from 1.7 V to 5.5 V (±0.85 V to ±2.275 V). The TLV521 features rail to rail
input and rail-to-rail output swings while consuming only nanowatts of power. Parameters that can exhibit
significant variance with regard to operating voltage or temperature are presented in the Typical Characteristics
section.
8.1.1 Driving Capacitive Load
The TLV521 is internally compensated for stable unity gain operation, with a 6-kHz, typical gain bandwidth.
However, the unity gain follower is the most sensitive configuration to capacitive load. The combination of a
capacitive load placed at the output of an amplifier along with the amplifier’s output impedance creates a phase
lag, which reduces the phase margin of the amplifier. If the phase margin is significantly reduced, the response
will be under damped which causes peaking in the transfer and, when there is too much peaking, the op amp
might start oscillating.
-
RISO
VOUT
VIN
+
CL
Figure 30. Resistive Isolation of Capacitive Load
In order to drive heavy capacitive loads, an isolation resistor, RISO, should be used, as shown in Figure 30. By
using this isolation resistor, the capacitive load is isolated from the amplifier’s output. The larger the value of
RISO, the more stable the amplifier will be. If the value of RISO is sufficiently large, the feedback loop will be
stable, independent of the value of CL. However, larger values of RISO result in reduced output swing and
reduced output current drive.
Recommended minimum values for RISO are given in the following table, for 5-V supply. Figure 31 shows the
typical response obtained with the CL = 50 pF and RISO = 154 kΩ. The other values of RISO in the table were
chosen to achieve similar dampening at their respective capacitive loads. Notice that for the TLV521 with larger
CL a smaller RISO can be used for stability. However, for a given CL a larger RISO will provide a more damped
response. For capacitive loads of 20 pF and below no isolation resistor is needed.
Table 1. Capacitive Loads vs. Needed Isolation Resistors
12
CL
RISO
0 – 20 pF
not needed
50 pF
154 kΩ
100 pF
118 kΩ
500 pF
52.3 kΩ
1 nF
33.2 kΩ
5 nF
17.4 kΩ
10 nF
13.3 kΩ
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VIN
20 mV/DIV
VOUT
VS = 5V
200 Ps/DIV
Figure 31. Step Response
8.1.2 EMI Suppression
The near-ubiquity of cellular, Bluetooth, and Wi-Fi signals and the rapid rise of sensing systems incorporating
wireless radios make electromagnetic interference (EMI) an evermore important design consideration for
precision signal paths. Though RF signals lie outside the op amp band, RF carrier switching can modulate the
DC offset of the op amp. Also some common RF modulation schemes can induce down-converted components.
The added DC offset and the induced signals are amplified with the signal of interest and thus corrupt the
measurement. The TLV521 uses on chip filters to reject these unwanted RF signals at the inputs and power
supply pins; thereby preserving the integrity of the precision signal path.
Twisted pair cabling and the active front-end’s common-mode rejection provide immunity against low-frequency
noise (i.e. 60-Hz or 50-Hz mains) but are ineffective against RF interference. Even a few centimeters of PCB
trace and wiring for sensors located close to the amplifier can pick up significant 1 GHz RF. The integrated EMI
filters of the TLV521 reduce or eliminate external shielding and filtering requirements, thereby increasing system
robustness. A larger EMIRR means more rejection of the RF interference. For more information on EMIRR,
please refer to AN-1698.
8.2 Typical Applications
8.2.1 60-Hz Twin T-Notch Filter
VBATT = 3V o 2V @ end of life
CR2032 Coin Cell
225 mAh = 5 circuits @ 9.5 yrs.
10 M:
10 M:
VBATT
-
Remote Sensor
10 M:
+
VIN
Signal
+
60 Hz
To ADC
VOUT
10 M:
270 pF
270 pF
10 M:
10 M:
Signal × 2
(No 60 Hz)
60 Hz Twin T Notch Filter
270 pF
AV = 2 V/V
270 pF
Figure 32. 60-Hz Notch Filter
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Typical Applications (continued)
8.2.1.1 Design Requirements
Small signals from transducers in remote and distributed sensing applications commonly suffer strong 60-Hz
interference from AC power lines. The circuit of Figure 32 notches out the 60 Hz and provides a gain AV = 2 for
the sensor signal represented by a 1-kHz sine wave. Similar stages may be cascaded to remove 2nd and 3rd
harmonics of 60 Hz. Thanks to the nA power consumption of the TLV521, even 5 such circuits can run for 9.5
years from a small CR2032 lithium cell. These batteries have a nominal voltage of 3 V and an end of life voltage
of 2 V. With an operating voltage from 1.7 V to 5.5 V the TLV521 can function over this voltage range.
8.2.1.2 Detailed Design Procedure
The notch frequency is set by F0 = 1 / 2πRC. To achieve a 60-Hz notch use R = 10 MΩ and C = 270 pF. If
eliminating 50-Hz noise, which is common in European systems, use R = 11.8 MΩ and C = 270 pF.
The Twin T Notch Filter works by having two separate paths from VIN to the amplifier’s input. A low frequency
path through the resistors R - R and another separate high frequency path through the capacitors C - C.
However, at frequencies around the notch frequency, the two paths have opposing phase angles and the two
signals will tend to cancel at the amplifier’s input.
To ensure that the target center frequency is achieved and to maximize the notch depth (Q factor) the filter
needs to be as balanced as possible. To obtain circuit balance, while overcoming limitations of available
standard resistor and capacitor values, use passives in parallel to achieve the 2C and R/2 circuit requirements
for the filter components that connect to ground.
To make sure passive component values stay as expected clean board with alcohol, rinse with deionized water,
and air dry. Make sure board remains in a relatively low humidity environment to minimize moisture which may
increase the conductivity of board components. Also large resistors come with considerable parasitic stray
capacitance which effects can be reduced by cutting out the ground plane below components of concern.
Large resistors are used in the feedback network to minimize battery drain. When designing with large resistors,
resistor thermal noise, op amp current noise, as well as op amp voltage noise, must be considered in the noise
analysis of the circuit. The noise analysis for the circuit in Figure 32 can be done over a bandwidth of 5 kHz,
which takes the conservative approach of overestimating the bandwidth (TLV521 typical GBW/AV is lower). The
total noise at the output is approximately 800 µVpp, which is excellent considering the total consumption of the
circuit is only 540 nA. The dominant noise terms are op amp voltage noise (550 µVpp), current noise through the
feedback network (430 µVpp), and current noise through the notch filter network (280 µVpp). Thus the total
circuit's noise is below ½ LSB of a 10 bit system with a 2-V reference, which is 1 mV.
8.2.1.3 Application Curve
Figure 33. 60-Hz Notch Filter Waveform
14
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Typical Applications (continued)
8.2.2 Portable Gas Detection Sensor
100 M:
V
1 M:
+
VOUT
+
-
V
RL
OXYGEN SENSOR
Figure 34. Precision Oxygen Sensor
8.2.2.1 Design Requirements
Gas sensors are used in many different industrial and medical applications. They generate a current which is
proportional to the percentage of a particular gas sensed in an air sample. This current goes through a load
resistor and the resulting voltage drop is measured. The TLV521 makes an excellent choice for this application
as it only draws 350 nA of current and operates on supply voltages down to 1.7 V. Depending on the sensed gas
and sensitivity of the sensor, the output current can be in the order of tens of microamperes to a few
milliamperes. Gas sensor data sheets often specify a recommended load resistor value or they suggest a range
of load resistors to choose from.
Oxygen sensors are used when air quality or oxygen delivered to a patient needs to be monitored. Fresh air
contains 20.9% oxygen. Air samples containing less than 18% oxygen are considered dangerous. This
application detects oxygen in air. Oxygen sensors are also used in industrial applications where the environment
must lack oxygen. An example is when food is vacuum packed. There are two main categories of oxygen
sensors, those which sense oxygen when it is abundantly present (i.e. in air or near an oxygen tank) and those
which detect traces of oxygen in ppm.
8.2.2.2 Detailed Design Procedure
Figure 34 shows a typical circuit used to amplify the output of an oxygen detector. The oxygen sensor outputs a
known current through the load resistor. This value changes with the amount of oxygen present in the air sample.
Oxygen sensors usually recommend a particular load resistor value or specify a range of acceptable values for
the load resistor. The use of the nanopower TLV521 means minimal power usage by the op amp and it
enhances the battery life. With the components shown in Figure 34 the circuit can consume less than 0.5 µA of
current ensuring that even batteries used in compact portable electronics, with low mAh charge ratings, could
last beyond the life of the oxygen sensor. The precision specifications of the TLV521, such as its very low offset
voltage, low TCVOS , low input bias current, high CMRR, and high PSRR are other factors which make the
TLV521 a great choice for this application.
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Typical Applications (continued)
8.2.2.3 Application Curve
5.0
4.5
4.0
VOUT (V)
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
0
10
20
30
40
50
VSENSOR (mV)
C001
Figure 35. Calculated Oxygen Sensor Circuit Output (Single 5V Supply)
8.2.3 High-Side Battery Current Sensing
ICHARGE
RSENSE
+
V
-
+
+
V
R1
24.9 k:
Q1
2N2907
RSENSE X R3
VOUT =
R1
-
R2
24.9 k:
+
10:
LOAD
X ICHARGE
VOUT
R3
10 M:
Figure 36. High-Side Current Sensing
8.2.3.1 Design Requirements
The rail-to-rail common mode input range and the very low quiescent current make the TLV521 ideal to use in
high-side and low-side battery current sensing applications. The high-side current sensing circuit in Figure 36 is
commonly used in a battery charger to monitor the charging current in order to prevent over charging. A sense
resistor RSENSE is connected in series with the battery.
8.2.3.2 Detailed Design Procedure
The theoretical output voltage of the circuit is VOUT = [ RSENSE × R3) / R1 ] × ICHARGE. In reality, however, due to
the finite Current Gain, β, of the transistor the current that travels through R3 will not be ICHARGE, but instead, will
be α × ICHARGE or β/( β+1) × ICHARGE. A Darlington pair can be used to increase the β and performance of the
measuring circuit.
Using the components shown in Figure 36 will result in VOUT ≈ 4000 Ω × ICHARGE. This is ideal to amplify a 1 mA
ICHARGE to near full scale of an ADC with VREF at 4.1 V. A resistor, R2 is used at the noninverting input of the
amplifier, with the same value as R1 to minimize offset voltage.
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Typical Applications (continued)
Selecting values per Figure 36 will limit the current traveling through the R1 – Q1 – R3 leg of the circuit to under 1
µA which is on the same order as the TLV521 supply current. Increasing resistors R1 , R2 , and R3 will decrease
the measuring circuit supply current and extend battery life.
Decreasing RSENSE will minimize error due to resistor tolerance, however, this will also decrease VSENSE =
ICHARGE × RSENSE, and in turn the amplifier offset voltage will have a more significant contribution to the total error
of the circuit. With the components shown in Figure 36 the measurement circuit supply current can be kept below
1.5 µA and measure 100 µA to 1 mA.
8.2.3.3 Application Curve
5.0
4.5
4.0
VOUT (V)
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
0
0.25
0.5
0.75
1
ICHARGE (mA)
1.25
1.5
C001
Figure 37. Calculated High-Side Current Sense Circuit Output
9 Power Supply Recommendations
The TLV521 is specified for operation from 1.7 V to 5.5 V (±0.85 V to ±2.275 V) over a –40°C to 125°C
temperature range. Parameters that can exhibit significant variance with regard to operating voltage or
temperature are presented in the Typical Characteristics.
CAUTION
Supply voltages larger than 6 V can permanently damage the device.
Low bandwidth nanopower devices do not have good high frequency (>1kHz) AC PSRR rejection against highfrequency switching supplies and other kHz and above noise sources, so extra supply filtering is recommended if
kHz range noise is expected on the power supply lines.
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10 Layout
10.1 Layout Guidelines
For best operational performance of the device, use good printed circuit board (PCB) layout practices, including:
• Noise can propagate into analog circuitry through the power pins of the circuit as a whole and op amp itself.
Bypass capacitors are used to reduce the coupled noise by providing low-impedance power sources local to
the analog circuitry.
• Connect low-ESR, 0.1-μF ceramic bypass capacitors between each supply pin and ground, placed as close
to the device as possible. A single bypass capacitor from V+ to ground is applicable for single supply
applications.
• Separate grounding for analog and digital portions of circuitry is one of the simplest and most-effective
methods of noise suppression. One or more layers on multilayer PCBs are usually devoted to ground planes.
A ground plane helps distribute heat and reduces EMI noise pickup. Make sure to physically separate digital
and analog grounds paying attention to the flow of the ground current. For more detailed information refer to
Circuit Board Layout Techniques, SLOA089.
• In order to reduce parasitic coupling, run the input traces as far away from the supply or output traces as
possible. If it is not possible to keep them separate, it is much better to cross the sensitive trace perpendicular
as opposed to in parallel with the noisy trace.
• Place the external components as close to the device as possible. As shown in Layout Example, keeping RF
and RG close to the inverting input minimizes parasitic capacitance.
• Keep the length of input traces as short as possible. Always remember that the input traces are the most
sensitive part of the circuit.
• Consider a driven, low-impedance guard ring around the critical traces. A guard ring can significantly reduce
leakage currents from nearby traces that are at different potentials.
10.2 Layout Example
Figure 38. Noninverting Layout Example
18
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
TLV521 PSPICE Model, SNOM024
TINA-TI SPICE-Based Analog Simulation Program, http://www.ti.com/tool/tina-ti
TI Filterpro Software, http://www.ti.com/tool/filterpro
DIP Adapter Evaluation Module, http://www.ti.com/tool/dip-adapter-evm
TI Universal Operational Amplifier Evaluation Module, http://www.ti.com/tool/opampevm
Evaluation board for 5-pin, north-facing amplifiers in the SC70 package, SNOA487.
Manual for LMH730268 Evaluation board 551012922-001
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation, see the following:
• Feedback Plots Define Op Amp AC Performance, SBOA015 (AB-028)
• Circuit Board Layout Techniques, SLOA089
• Op Amps for Everyone, SLOD006
• AN-1698 A Specification for EMI Hardened Operational Amplifiers, SNOA497
• EMI Rejection Ratio of Operational Amplifiers, SBOA128
• Capacitive Load Drive Solution using an Isolation Resistor, TIPD128
• Handbook of Operational Amplifier Applications, SBOA092
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
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12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
20
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TLV521DCKR
ACTIVE
SC70
DCK
5
3000
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
14F
TLV521DCKT
ACTIVE
SC70
DCK
5
250
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
14F
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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