TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
D
D
D
D
D
D
D
D
Integrated Depop Circuitry
High Power with PC Power Supply
– 2 W/Ch at 5 V into a 3-Ω Load
– 800 mW/Ch at 3 V
Fully Specified for Use With 3-Ω Loads
Ultra-Low Distortion
– 0.05% THD+N at 2 W and 3-Ω Load
Bridge-Tied Load (BTL) or Single-Ended
(SE) Modes
Stereo Input MUX
Surface-Mount Power Package
24-Pin TSSOP PowerPAD
Shutdown Control . . . IDD = 5 µA
CFR
PWP PACKAGE
(TOP VIEW)
GND/HS
TJ
LOUT+
LLINEIN
LHPIN
LBYPASS
LVDD
SHUTDOWN
MUTE OUT
LOUT–
MUTE IN
GND/HS
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
GND/HS
NC
ROUT+
RLINEIN
RHPIN
RBYPASS
RVDD
NC
HP/LINE
ROUT–
SE/BTL
GND/HS
RFR
RIR
CIR
NC
21
RLINEIN
20
RHPIN
19
RBYPASS
Right
MUX
ROUT+ 22
–
+
ROUT – 15
COUTR
RVDD 18
VDD
CB
CS
System
Control
11
9
8
MUTE IN
MUTE OUT
SHUTDOWN
Bias, Mute,
Shutdown,
and SE/BTL
MUX Control
NC
RIL
LBYPASS
5
LHPIN
4
LLINEIN
1 kΩ
SE/BTL 14
100 kΩ
HP/LINE 16
LVDD 7
6
100 kΩ
VDD
1 kΩ
COUTL
Left
MUX
LOUT+ 3
+
–
LOUT – 10
CIL
CFL
RFL
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments Incorporated.
Copyright 2000, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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1
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
description
The TPA0202 is a stereo audio power amplifier in a 24-pin TSSOP thermal package capable of delivering
greater than 2 W of continuous RMS power per channel into 3-Ω loads. The TPA0202 simplifies design and frees
up board space for other features. Full power distortion levels of less than 0.1% THD+N from a 5-V supply are
typical. Low-voltage applications are also well served by the TPA0202 providing 800-mW per channel into 3-Ω
loads with a 3.3-V supply voltage.
The TPA0202 has integrated depop circuitry that virtually eliminates transients that cause noise in the speakers
during power up and when using the mute and shutdown modes.
Amplifier gain is externally configured by means of two resistors per input channel and does not require external
compensation for settings of 2 to 20 in BTL mode (1 to 10 in SE mode). An internal input MUX allows two sets
of stereo inputs to the amplifier. In notebook applications, where internal speakers are driven as BTL and the
line (often headphone drive) outputs are required to be SE, the TPA0202 automatically switches into SE mode
when the SE/BTL input is activated. Using the TPA0202 to drive line outputs up to 700 mW/channel into external
3-Ω loads is ideal for small non-powered external speakers in portable multimedia systems. The TPA0202 also
features a shutdown function for power sensitive applications, holding the supply current at 5 µA.
The PowerPAD package† (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA0202 to operate at full power into 3-Ω loads at ambient temperature of up to
85°C with 300 CFM of forced-air cooling. Into 8-Ω loads, the operating ambient temperature increases to 100°C.
AVAILABLE OPTIONS
PACKAGE
TA
TSSOP‡
(PWP)
– 40°C to 85°C
TPA0202PWP
‡ The PWP packages are available taped and reeled. To order a taped
and reeled part, add the suffix R (e.g., TPA0202PWPR).
† See Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (Literature Number SLMA002) for more
information on the PowerPAD package.
2
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TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
GND/HS
1, 12,
13, 24
HP/LINE
16
LBYPASS
6
LHP IN
5
I
Left channel headphone input, selected when HP/LINE terminal (16) is held high
LLINE IN
4
I
Left channel line input, selected when HP/LINE terminal (16) is held low
LOUT+
3
O
Left channel + output in BTL mode, + output in SE mode
LOUT–
10
O
Left channel – output in BTL mode, high-impedance state in SE mode
LVDD
MUTE IN
7
I
Supply voltage input for left channel and for primary bias circuits
11
I
Mute all amplifiers, hold low for normal operation, hold high to mute
MUTE OUT
9
O
Follows MUTE IN terminal (11), provides buffered output
NC
Ground connection for circuitry, directly connected to thermal pad
I
Tap to voltage divider for left channel internal mid-supply bias
17, 23
RBYPASS
19
RHPIN
20
RLINEIN
ROUT+
Input MUX control input, hold high to select LHP IN or RHP IN (5, 20), hold low to select LLINE IN or
RLINE IN (4, 21)
No internal connection
Tap to voltage divider for right channel internal mid–supply bias
I
Right channel headphone input, selected when HP/LINE terminal (16) is held high
21
I
Right channel line input, selected when HP/LINE terminal (16) is held low
22
O
Right channel + output in BTL mode, + output in SE mode
ROUT–
15
O
Right channel – output in BTL mode, high impedance state in SE mode
RVDD
SE/BTL
18
I
Supply voltage input for right channel
14
I
Hold low for BTL mode, hold high for SE mode
SHUTDOWN
8
I
Places entire IC in shutdown mode when held high, IDD = 5 µA
TJ
2
O
Sources a current proportional to the junction temperature. This terminal should be left unconnected
during normal operation. For more information, see the junction temperature measurement section of
this document.
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3
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
absolute maximum ratings over operating free-air temperature range (unless otherwise noted)†
Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V
Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VDD +0.3 V
Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 85°C
Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to 150°C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
PWP‡
TA ≤ 25°C
2.7 W
DERATING FACTOR
21.8 mW/°C
TA = 70°C
1.7 W
TA = 85°C
1.4 W
‡ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.
recommended operating conditions
MIN
NOM
MAX
3
5
5.5
Supply Voltage, VDD
Operating free-air temperature, TA
Common mode input voltage,
voltage VICM
VDD = 5 V,
250 mW/ch average power,
4-Ω stereo BTL drive,
with proper PCB design
–40
85
VDD = 5 V,
2 W/ch average power,
3-Ω stereo BTL drive,
with proper PCB design
and 300 CFM forced-air
cooling
–40
85
1.25
4.5
1.25
2.7
VDD = 5 V
VDD = 3.3 V
UNIT
V
°C
V
dc electrical characteristics, TA = 25°C
PARAMETER
VDD = 5 V
IDD
TYP†
MAX
Stereo BTL
19
30
mA
Stereo SE
9
18
mA
Mono BTL
9
18
mA
Mono SE
3
10
mA
Stereo BTL
13
20
mA
Stereo SE
5
10
mA
Mono BTL
5
10
mA
3
6
mA
5
25
mV
TEST CONDITIONS
Supply current
VDD = 3
3.3
3V
Mono SE
VOO
IDD(MUTE)
Output offset voltage (measured differentially)
Supply current in mute mode
VDD = 5 V,
VDD = 5 V
IDD(SD)
IDD in shutdown
VDD = 5 V
Gain = 2,
NOTE 1: At 3 V < VDD < 5 V the dc output voltage is approximately VDD/2.
4
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See Note 1
1.5
5
UNIT
mA
20
µA
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
ac operating characteristics, VDD = 5 V, TA = 25°C, RL = 3 Ω (unless otherwise noted)
PARAMETER
PO
Output power ((each channel)) see
Note 2
THD+N
Total harmonic distortion plus noise
BOM
Maximum output power bandwidth
Phase margin
Supply ripple rejection ratio
TEST CONDITIONS
TYP
ZI
See Figure 3
2
THD = 1%,
BTL,
See Figure 3
2.2
Po = 2W,
VI = 1 V,
f = 20 – 20 kHz,
See Figure 5
200
RL = 10 kΩ,
m%
THD < 1 %,
AV = 1 V/V
See Figure 5
100
AV = 10 V/V
RL = 4 Ω,
>20
kHz
Open Loop,
See Figure 43
85°
f = 1 kHz,
See Figure 37
80
f = 20 – 20 kHz,
See Figure 37
60
f = 1 kHz,
See Figure 39
W
m%
dB
85
dB
85
dB
Line/HP input separation
100
dB
BTL attenuation in SE mode
100
dB
2
MΩ
95
dB
21
µV(rms)
Input impedance
Signal-to-noise ratio
Vn
UNIT
BTL,
Mute attenuation
Channel-to-channel output separation
MAX
THD = 0.2%,
Output noise voltage
Po = 500 mW,
See Figure 35
BTL
NOTE 2: Output power is measured at the output terminals of the IC at 1 kHz.
ac operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 3 Ω
PARAMETER
PO
Output power ((each channel)) see
Note 2
THD+N
Total harmonic distortion plus noise
BOM
Maximum output power bandwidth
Phase margin
Supply ripple rejection ratio
TEST CONDITIONS
TYP
See Figure 10
800
THD = 1%,
BTL,
See Figure 10
900
Po = 800 mW,
VI = 1 V,
f = 20 – 20 kHz,
See Figure 11
350
RL = 10 kΩ,
200
m%
AV = 10 V/V
RL = 4 Ω,
THD < 1 %,
AV = 1 V/V
See Figure 11
>20
kHz
Open Loop,
See Figure 44
85°
f = 1 kHz,
See Figure 37
70
f = 20 – 20 kHz,
See Figure 37
55
f = 1 kHz,
See Figure 40
mW
m%
dB
85
dB
85
dB
Line/HP input separation
100
dB
BTL attenuation in SE mode
100
dB
2
MΩ
Channel-to-channel output separation
Input impedance
Signal-to-noise ratio
Vn
UNIT
BTL,
Mute attenuation
ZI
MAX
THD = 0.2%,
Output noise voltage
Po = 500 mW,
See Figure 37
BTL
95
dB
21
µV(rms)
NOTE 2: Output power is measured at the output terminals of the IC at 1 kHz.
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5
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
PARAMETER MEASUREMENT INFORMATION
RF
CI
MUX
RI
4.7 µF
CB
RL = 3 Ω or 8 Ω
SE/BTL
HP/LINE
Figure 1. BTL Test Circuit
RF
CI
CO
MUX
RI
RL = 3 Ω, 8 Ω, or 32 Ω
VDD
4.7 µF
CB
SE/BTL
HP/LINE
Figure 2. SE Test Circuit
6
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TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
4, 5, 7, 8, 11, 12, 14, 15, 17, 18, 20,
21, 23, 24, 26, 27, 29, 30 32, 33
vs Frequency
THD + N
Total harmonic distortion plus noise
3, 6, 9, 10, 13, 16, 19, 22, 25, 28, 31,
34
vs Output power
Vn
Output noise voltage
vs Frequency
35,36
Supply ripple rejection ratio
vs Frequency
37,38
Crosstalk
vs Frequency
39 – 42
Open loop response
vs Frequency
43,44
45, 48
Closed loop response
vs Frequency
IDD
Supply current
vs Supply voltage
49
PO
Output power
vs Supply voltage
vs Load resistance
50, 51
52, 53
PD
Power dissipation
vs Output power
54 – 57
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
10
VDD = 5 V
f = 1 kHz
BTL
RL = 3 Ω
1
RL = 8 Ω
0.1
0.01
0
0.25 0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5
VDD = 5 V
PO = 1.5 W
RL = 4 Ω
BTL
1
AV = –10 V/V (RL = 3 Ω, PO = 2 W)
AV = –10 V/V
AV = –20 V/V
0.1
AV = –2 V/V
0.01
20
PO – Output Power – W
100
1k
10 k 20 k
f – Frequency – Hz
Figure 3
Figure 4
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7
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
10
VDD = 5 V
RL = 4 Ω
AV = –2 V/V
BTL
1
PO = 1.5 W
PO = 2 W, RL = 3 Ω
PO = 0.75 W
0.1
PO = 0.25 W
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
0.01
20
100
1k
f – Frequency – Hz
10
VDD = 5 V
RL = 3 Ω
BTL
1
f = 20 kHz
f = 20 Hz
0.1
f = 1 kHz
0.01
0.01
10 k 20 k
1
0.1
PO – Output Power – W
Figure 5
Figure 6
10
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
BTL
1
PO = 0.5 W
0.1
PO = 1 W
PO = 0.25 W
10
VDD = 5 V
PO = 1 W
RL = 8 Ω
BTL
1
AV =– 20 V/V
AV = –10 V/V
0.1
AV = –2 V/V
0.01
0.01
20
100
1k
f – Frequency – Hz
10 k
20 k
20
100
1k
f – Frequency – Hz
Figure 7
8
10
Figure 8
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10 k
20 k
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
10
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
BTL
1
f = 20 kHz
0.1
f = 1 kHz
f = 20 Hz
10
VDD = 3.3 V
f = 1 kHz
BTL
1
RL = 8 Ω
0.1
0.01
0.01
0.01
0.1
1
PO – Output Power – W
0
10
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
PO – Output Power – W
Figure 9
1
Figure 10
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
RL = 3 Ω
VDD = 3.3 V
PO = 0.75 W
RL = 4 Ω
BTL
1
AV = –10 V/V
AV = –20 V/V
0.1
AV = –2 V/V
AV = –10 V/V (RL = 3 Ω, PO = 800 mW)
0.01
10
VDD = 3.3 V
RL = 4 Ω
AV = –2 V/V
BTL
1
PO = 0.35 W
PO = 800 mW
(RL = 3 Ω)
PO = 0.75 W
0.1
PO = 0.1 W
0.01
20
100
1k
10 k
20 k
20
100
1k
f – Frequency – Hz
f – Frequency – Hz
Figure 11
Figure 12
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10 k 20 k
9
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
10
VDD = 3.3 V
RL = 3 Ω
AV = –2 V/V
BTL
f = 20 kHz
1
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 20 Hz
0.1
f = 1 kHz
0.01
0.1
10
VDD = 3.3 V
PO = 0.4 W
RL = 8 Ω
BTL
1
AV = –20 V/V
0.1
AV = –10 V/V
AV = –2 V/V
0.01
1
2
20
PO – Output Power – W
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
VDD = 3.3 V
RL = 8 Ω
AV = –2 V/V
BTL
1
PO = 0.25 W
PO = 0.4 W
PO = 0.1 W
0.01
1k
10 k 20 k
10
VDD = 3.3 V
RL = 8 Ω
AV = –2 V/V
BTL
1
0.1
f = 20 kHz
f = 1 kHz
f = 20 Hz
0.01
0.01
f – Frequency – Hz
Figure 15
10
20 k
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
100
10 k
Figure 14
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
20
1k
f – Frequency – Hz
Figure 13
0.1
100
1
0.1
PO – Output Power – W
Figure 16
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10
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
10
VDD = 5 V
PO = 0.5 W
RL = 4 Ω
SE
1
AV = –10 V/V
0.1
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
AV = –5 V/V
AV = –1 V/V
10
VDD = 5 V
RL = 4 Ω
AV = –2 V/V
SE
1
PO = 0.5 W
PO = 0.25 W
0.1
PO = 0.1 W
0.01
0.01
20
100
1k
f – Frequency – Hz
20
10 k 20 k
100
Figure 17
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
VDD = 5 V
RL = 4 Ω
AV = –2 V/V
SE
1
f = 20 kHz
0.1
f =100 Hz
f = 1 kHz
0.01
0.001
10 k 20 k
Figure 18
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
1k
f – Frequency – Hz
10
VDD = 5 V
PO = 0.25 W
RL = 8 Ω
SE
1
AV = –10 V/V
0.1
AV = –5 V/V
AV = –1 V/V
0.01
0.01
0.1
PO – Output Power – W
1
20
100
1k
10 k 20 k
f – Frequency – Hz
Figure 19
Figure 20
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11
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
VDD = 5 V
RL = 8 Ω
SE
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
1
0.1
PO = 0.25 W
PO = 0.1 W
PO = 0.05 W
0.01
20
100
1k
10 k 20 k
10
VDD = 5 V
RL = 8 Ω
AV = –2 V/V
SE
1
f = 20 kHz
0.1
f = 1 kHz
f = 100 Hz
0.01
0.001
f – Frequency – Hz
0.01
Figure 21
1
AV = –10 V/V
AV = –5 V/V
AV = –1 V/V
0.01
10
VDD = 5 V
RL = 32 Ω
SE
1
0.1
PO = 50 mW
PO = 75 mW
PO = 25 mW
0.01
20
12
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
VDD = 5 V
PO = 0.075 W
RL = 32 Ω
SE
0.1
100
1
Figure 22
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
0.1
PO – Output Power – W
1k
10 k 20 k
f – Frequency – Hz
1k
f – Frequency – Hz
Figure 23
Figure 24
POST OFFICE BOX 655303
20
• DALLAS, TEXAS 75265
100
10 k 20 k
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
VDD = 5 V
RL = 32 Ω
SE
1
f = 20 kHz
0.1
f = 20 Hz
f = 1 kHz
0.01
0.001
10
VDD = 3.3 V
PO = 0.2 W
RL = 4 Ω
SE
1
AV = –10 V/V
0.1
AV = –5 V/V
AV = –1 V/V
0.01
0.01
0.1
PO – Output Power – W
1
20
Figure 25
VDD = 3.3 V
RL = 4 Ω
SE
PO = 0.2 W
PO = 0.1 W
0.1
PO = 0.05 W
0.01
100
10 k 20 k
1k
10 k 20 k
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
10
20
1k
f – Frequency – Hz
Figure 26
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
1
100
10
VDD = 3.3 V
RL = 4 Ω
AV = –2 V/V
SE
1
f = 20 kHz
0.1
f = 1 kHz
f = 100 Hz
0.01
0.001
f – Frequency – Hz
0.01
0.1
1
PO – Output Power – W
Figure 27
Figure 28
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• DALLAS, TEXAS 75265
13
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
VDD = 3.3 V
PO = 100 mW
RL = 8 Ω
SE
1
AV = –10 V/V
0.1
AV = –5 V/V
AV = –1 V/V
10
VDD = 3.3 V
RL = 8 Ω
SE
1
PO = 100 mW
PO = 50 mW
0.1
PO = 25 mW
0.01
0.01
20
100
20
10 k 20 k
1k
f – Frequency – Hz
100
Figure 29
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
0.1
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
1
VDD = 3.3 V
RL = 8 Ω
SE
f = 20 kHz
f = 1 kHz
f = 100 Hz
0.01
0.001
10
VDD = 3.3 V
PO = 30 mW
RL = 32 Ω
SE
1
AV = –10 V/V
0.1
AV = –5 V/V
AV = –1 V/V
0.01
0.01
0.1
PO – Output Power – W
1
20
100
1k
f – Frequency – Hz
Figure 31
14
10 k 20 k
Figure 30
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
1k
f – Frequency – Hz
Figure 32
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
10 k 20 k
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
10
VDD = 3.3 V
RL = 32 Ω
SE
1
0.1
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
THD+N –Total Harmonic Distortion + Noise – %
THD+N –Total Harmonic Distortion + Noise – %
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
PO = 20 mW
PO = 30 mW
0.01
PO = 10 mW
0.001
20
100
VDD = 3.3 V
RL = 32 Ω
SE
1
f = 20 kHz
0.1
f = 1 kHz
f = 20 Hz
0.01
0.001
0.001
10 k 20 k
1k
10
f – Frequency – Hz
0.01
0.1
PO – Output Power – W
Figure 33
Figure 34
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100
VDD = 5 V
BW = 22 Hz to 22 kHz
RL = 4Ω
VO BTL
VO+
10
VO–
V n – Output Noise Voltage – µ V (rms)
100
V n – Output Noise Voltage – µ V (rms)
1
VDD = 3.3 V
BW = 22 Hz to 22 kHz
RL = 4Ω
VO BTL
VO+
10
VO–
1
1
20
100
1k
10 k 20 k
20
100
1k
f – Frequency – Hz
f – Frequency – Hz
Figure 35
Figure 36
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10 k 20 k
15
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
RL = 4 Ω
CB = 4.7 µF
BTL
–10
–20
Supply Ripple Rejection Ratio – dB
Supply Ripple Rejection Ratio – dB
0
–30
–40
–50
–60
VDD = 3.3 V
–70
–80
VDD = 5 V
–90
RL = 4 Ω
CB = 4.7 µF
SE
–10
–20
–30
–40
–50
VDD = 5 V
–60
–70
VDD = 3.3 V
–80
–90
–100
–100
20
100
1k
10 k 20 k
20
100
1k
f – Frequency – Hz
Figure 37
Figure 38
CROSSTALK
vs
FREQUENCY
–40
CROSSTALK
vs
FREQUENCY
–40
VDD = 5 V
PO = 1.5 W
RL = 4 Ω
BTL
–50
–60
Crosstalk – dB
Crosstalk – dB
VDD = 3.3 V
PO = 0.75 W
RL = 4 Ω
BTL
–50
–60
Left to Right
–70
–80
–90
–70
Left to Right
–80
–90
Right to Left
Right to Left
–100
–100
–110
–110
–120
–120
20
16
10 k 20 k
f – Frequency – Hz
100
10 k 20 k
1k
20
100
1k
f – Frequency – Hz
f – Frequency – Hz
Figure 39
Figure 40
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
10 k 20 k
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
CROSSTALK
vs
FREQUENCY
–40
CROSSTALK
vs
FREQUENCY
–40
VDD = 5 V
PO = 75 mW
RL = 32 Ω
SE
–50
–50
–60
Crosstalk – dB
–60
–70
–80
Left to Right
–90
–100
–70
Left to Right
–80
–90
–100
Right to Left
Right to Left
–110
–110
–120
–120
100
10 k 20 k
1k
20
100
1k
f – Frequency – Hz
f – Frequency – Hz
Figure 41
Figure 42
10 k 20 k
OPEN LOOP RESPONSE
100
VDD = 5 V
RL = 4 Ω
BTL
80
60
Phase
180°
90°
40
Phase
20
Gain – dB
Crosstalk – dB
VDD = 3.3 V
PO = 35 mW
RL = 32 Ω
SE
Gain
20
0°
0
–90°
–20
–40
0.01
0.1
1
10
100
1000
–180°
10000
f – Frequency – kHz
Figure 43
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• DALLAS, TEXAS 75265
17
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
OPEN LOOP RESPONSE
80
180°
VDD = 3.3 V
RL = 4 Ω
BTL
60
90°
Gain – dB
0°
20
Phase
Phase
40
Gain
0
–90°
–20
–40
0.01
0.1
1
10
100
1000
–180°
10000
f – Frequency – kHz
Figure 44
CLOSED LOOP RESPONSE
0°
10
VDD = 5 V
AV = –2 V/V
PO = 1.5 W
BTL
9
8
– 45°
7
– 90°
5
– 135°
4
Phase
– 180°
3
2
– 225°
1
0
20
100
1k
10 k
– 270°
100 k 200 k
f – Frequency – Hz
Figure 45
18
POST OFFICE BOX 655303
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Phase
Gain – dB
Gain
6
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
0°
10
VDD = 3.3 V
AV = –2 V/V
PO = 0.75 W
BTL
9
8
– 45°
7
– 90°
5
– 135°
4
Phase
Gain – dB
Gain
6
Phase
– 180°
3
2
– 225°
1
0
20
100
1k
10 k
– 270°
100 k 200 k
f – Frequency – Hz
Figure 46
CLOSED LOOP RESPONSE
0°
0
Gain
–1
– 45°
–2
– 90°
–4
–5
– 135°
–6
Phase
Gain – dB
–3
Phase
– 180°
–7
VDD = 5 V
AV = –1 V/V
PO = 0.5 W
SE
–8
–9
–10
20
100
1k
10 k
– 225°
– 270°
100 k 200 k
f – Frequency – Hz
Figure 47
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• DALLAS, TEXAS 75265
19
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
0°
0
Gain
–1
– 45°
–2
– 90°
–4
–5
– 135°
–6
Phase
Gain – dB
–3
Phase
– 180°
–7
VDD = 3.3V
AV = –1 V/V
PO = 0.25 W
SE
–8
–9
–10
20
100
1k
10 k
– 225°
– 270°
100 k 200 k
f – Frequency – Hz
Figure 48
SUPPLY CURRENT
vs
SUPPLY VOLTAGE
OUTPUT POWER
vs
SUPPLY VOLTAGE
3
30
2.5
ÁÁ
ÁÁ
PO – Output Power – W
I DD – Supply Current – mA
25
20
Stereo BTL
15
10
THD+N = 1%
BTL
Each Channel
2
RL = 4Ω
1.5
RL = 8 Ω
1
Stereo SE
0.5
5
0
3
4
5
VDD – Supply Voltage – V
6
0
2.5
3
3.5
4
4.5
Figure 50
POST OFFICE BOX 655303
5
VDD – Supply Voltage – V
Figure 49
20
RL = 3 Ω
• DALLAS, TEXAS 75265
5.5
6
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
SUPPLY VOLTAGE
1
OUTPUT POWER
vs
LOAD RESISTANCE
3
THD+N = 1%
SE
Each Channel
2.5
PO – Output Power – W
PO – Output Power – W
0.8
RL = 4 Ω
0.6
RL = 8 Ω
0.4
THD+N = 1%
BTL
Each Channel
0.2
2
1.5
VDD = 5 V
1
0.5
RL = 32 Ω
VDD = 3.3 V
0
0
2.5
3
3.5
4
4.5
5
VDD – Supply Voltage – V
5.5
0
6
4
24
8
12
16
20
RL – Load Resistance – Ω
Figure 51
32
Figure 52
OUTPUT POWER
vs
LOAD RESISTANCE
POWER DISSIPATION
vs
OUTPUT POWER
1.8
1
THD+N = 1%
SE
Each Channel
PD – Power Dissipation – W
0.6
0.4
RL = 3 Ω
1.6
0.8
PO – Output Power – W
28
VDD = 5 V
1.4
RL = 4 Ω
1.2
1
0.8
0.6
RL = 8 Ω
0.4
0.2
VDD = 5 V
BTL
Each Channel
0.2
VDD = 3.3 V
0
0
4
24
8
12
16
20
RL – Load Resistance – Ω
28
32
0
0
Figure 53
0.5
1
1.5
PO – Output Power – W
2
2.5
Figure 54
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21
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
TYPICAL CHARACTERISTICS
POWER DISSIPATION
vs
OUTPUT POWER
POWER DISSIPATION
vs
OUTPUT POWER
0.8
0.8
PD – Power Dissipation – W
PD – Power Dissipation – W
RL = 3 Ω
0.7
0.6
RL = 4 Ω
0.5
0.4
0.3
RL = 8 Ω
0.2
0
0
0.75
0.25
0.5
PO – Output Power – W
RL = 4 Ω
0.4
RL = 8 Ω
0.2
RL = 32Ω
VDD = 3.3 V
BTL
Each Channel
0.1
0.6
0
0
1
0.1
0.4
0.2
0.3
PO – Output Power – W
Figure 55
Figure 56
POWER DISSIPATION
vs
OUTPUT POWER
0.6
PD – Power Dissipation – W
VDD = 3.3V
SE
Each Channel
RL = 4 Ω
0.4
RL = 8 Ω
0.2
RL = 32Ω
0
0
0.05
0.1
0.15
0.2
PO – Output Power – W
Figure 57
22
VDD = 5 V
SE
Each Channel
POST OFFICE BOX 655303
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0.25
0.5
0.6
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 58)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface-mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements (< 2 mm) of many of
today’s advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
Figure 58. Views of Thermally Enhanced PWP Package
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23
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
bridged-tied load versus single-ended mode
Figure 59 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA0202 BTL amplifier
consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power equation, where
voltage is squared, yields 4× the output power from the same supply rail and load impedance (see equation 1).
V
+
(rms)
V
+
V
Power
O(PP)
Ǹ
2 2
2
(1)
(rms)
R
L
VDD
VO(PP)
RL
2x VO(PP)
VDD
–VO(PP)
Figure 59. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement —
which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 60. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 2.
fc
24
+ 2 p R1 C
(2)
L C
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
bridged-tied load versus single-ended mode (continued)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD
–3 dB
VO(PP)
CC
RL
VO(PP)
fc
Figure 60. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the thermal considerations
section.
BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD.
The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 61).
IDD
VO
IDD(RMS)
V(LRMS)
Figure 61. Voltage and Current Waveforms for BTL Amplifiers
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TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency
+ P PL
(3)
SUP
Where:
+
V rms 2
L
P
L
R
L
V
P
V rms
L
2
+ 2R
Vp
2
L
+Ǹ
V
2V
P
P
+
V
I rms + DD
DD DD
SUP
pR
L
I
rms
DD
+ p2VRP
L
Efficiency of a BTL Configuration
p VP
+ 2V
DD
+
p
ǒ Ǔ
P R
L L
2
2V
ń
1 2
(4)
DD
Table 1 employs equation 4 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 1. Efficiency Vs Output Power in 5-V 8-Ω BTL Systems
OUTPUT POWER
(W)
EFFICIENCY
(%)
PEAK-TO-PEAK
VOLTAGE
(V)
INTERNAL
DISSIPATION
(W)
0.25
31.4
2.00
0.55
0.50
44.4
2.83
0.62
1.00
62.8
4.00
4.47†
0.59
1.25
70.2
† High peak voltages cause the THD to increase.
0.53
A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 4, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
26
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TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
For example, if the 5-V supply is replaced with a 3.3-V supply (TPA0202 has a maximum recommended VDD
of 5.5 V) in the calculations of Table 1, then efficiency at 0.5 W would rise from 44% to 67% and internal power
dissipation would fall from 0.62 W to 0.25 W at 5 V. Then for a stereo 0.5-W system from a 3.3-V supply, the
maximum draw would only be 1.5 W as compared to 2.24 W from 5 V. In other words, use the efficiency analysis
to choose the correct supply voltage and speaker impedance for the application.
selection of components
Figure 62 and Figure 63 are a schematic diagrams of a typical notebook computer application circuits.
CFR
5 pF
RFR
50 kΩ
RIR
10 kΩ
CIR
1 µF
NC
21
RLINEIN
20
RHPIN
19
RBYPASS
Right
MUX
ROUT+ 22
–
+
ROUT – 15
RVDD 18
CB
1 µF
System
Control
MUTE IN
11
MUTE OUT
9
SHUTDOWN
8
CS
0.1 µF
(see Note A)
Bias, Mute,
Shutdown,
and SE/BTL
MUX Control
RIL NC
10 kΩ
LBYPASS
5
LHPIN
4
LLINEIN
Left
MUX
100 kΩ
1 kΩ
SE/BTL 14
HP/LINE 16
LVDD 7
6
VDD
COUTR
330 µF
100 kΩ
0.1 µF
VDD
LOUT+ 3
+
–
1 kΩ
COUTL
330 µF
LOUT – 10
CIL
1 µF
CFL
5 pF
RFL
50 kΩ
NOTE A: A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger aluminum
electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 62. TPA0202 Minimum Configuration Application Circuit
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27
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
CFRLINE
5 pF
RFRLINE
50 kΩ
RFRHP
10 kΩ
CIRLINE
1 µF RIRLINE
10 kΩ
21
RLINEIN
20
RHPIN
RIRHP
CIRHP 10 kΩ
1 µF
CBR
0.1 µF
19
RBYPASS
System
Control
11
ROUT+ 22
–
+
ROUT – 15
RVDD 18
9
See Note A
Right
MUX
8
MUTE IN
MUTE OUT
SHUTDOWN
CSR
0.1 µF
(see Note B)
Bias, Mute,
Shutdown,
and SE/BTL
MUX Control
COUTR
330 µF
VDD
100 kΩ
1 kΩ
SE/BTL 14
100 kΩ
HP/LINE 16
0.1 µF
LVDD 7
6
LBYPASS
VDD
CSR
0.1 µF
(see Note B)
CBL
1 µF
CILHP
1 µF
RILHP
10 kΩ
1 kΩ
COUTL
330 µF
5
4
LHPIN
LLINEIN
Left
MUX
LOUT+ 3
+
–
LOUT – 10
RILLINE
CILLINE 10 kΩ
1 µF
RFLHP
10 kΩ
CFLLINE
5 pF
RFLLINE
50 kΩ
NOTES: A. This connection is for ultra-low current in shutdown mode.
B. A 0.1 µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
aluminum electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 63. TPA0202 Full Configuration Application Circuit
28
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
gain setting resistors, RF and RI
ǒǓ
The gain for each audio input of the TPA0202 is set by resistors RF and RI according to equation 5 for BTL mode.
BTL Gain
+ *2
R
F
R
I
(5)
BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA0202 is a MOS amplifier, the input impedance is very high,
consequently input leakage currents are not generally a concern although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in equation 6.
Effective Impedance
+ RRF)RRI
F
(6)
I
As an example consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of the
amplifier would be –10 and the effective impedance at the inverting terminal would be 8.3 kΩ, which is well within
the recommended range.
For high performance applications metal film resistors are recommended because they tend to have lower noise
levels than carbon resistors. For values of RF above 50 kΩ the amplifier tends to become unstable due to a pole
formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kΩ. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.
–3 dB
f c(lowpass)
+ 2 p R1 C
(7)
F F
fc
For example, if RF is 100 kΩ and Cf is 5 pF then fc is 318 kHz, which is well outside of the audio range.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
29
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.
–3 dB
f c(highpass)
+ 2 p R1 C
(8)
I I
fc
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
CI
+ 2 p 1R fc
(9)
I
In this example, CI is 0.40 µF so one would likely choose a value in the range of 0.47 µF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher
that the source dc level. Please note that it is important to confirm the capacitor polarity in the application.
power supply decoupling, CS
The TPA0202 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VDD lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near
the audio power amplifier is recommended.
30
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
midrail bypass capacitor, CB
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N. The capacitor is fed from a 100-kΩ source inside the amplifier. To keep the start-up pop as low as
possible, the relationship shown in equation 10 should be maintained.
ǒ
CB
1
100 kΩ
Ǔv ǒ
CI RI
) R FǓ
1
(10)
As an example, consider a circuit where CB is 1 µF, CI is 0.22 µF, RF is 50 kΩ, and RI is 10 kΩ. Inserting these
values into the equation 10 we get 10 ≤ 75, which satisfies the rule. Bypass capacitor, CB, values of 0.1 µF to
1 µF ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance.
In Figure 63, the full feature configuration, two bypass capacitors are used. This provides the maximum
separation between right and left drive circuits. When absolute minimum cost and/or component space is
required, one bypass capacitor can be used as shown in Figure 62. It is critical that terminals 6 and 19 be tied
together in this configuration.
load considerations
Extremely low impedance loads (below 4 Ω) coupled with certain external component selections, board layouts,
and cabling can cause oscillations in the system. Using a single air-cored inductor in series with the load
eliminates any spurious oscillations that might occur. An inductance of approximately 1 µH has been shown to
eliminate such oscillations. There are no special considerations when using 4 Ω and above loads with this
amplifier.
optimizing depop operation
Circuitry has been included in the TPA0202 to minimize the amount of popping heard at power-up and when
coming out of shutdown mode. Popping occurs whenever a voltage step is applied to the speaker. If high
impedances are used for the feedback and input resistors, it is possible for the input capacitor to drift downwards
from mid-rail during mute and shutdown. A high gain amplifier intensifies the problem as the small delta in
voltage is multiplied by the gain. So it is advantageous to use low-gain configurations, and to limit the size of
the gain-setting resistors. The time constant of the input coupling capacitor (CI ) and the gain-setting resistors
(RI and RF ) needs to be shorter than the time constant formed by the bypass capacitor (CB ) and the output
impedance of the mid-rail generator, which is nominally 100 kΩ (see equation 10).
The effective output impedance of the mid-rail generator is actually greater than 100 kΩ due to a PNP transistor
clamping the input node (see Figure 64).
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
31
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
VDD
100 kΩ
50 kΩ
BYPASS
100 kΩ
Figure 64. PNP Transistor Clamping of BYPASS Terminal
The PNP transistor limits the voltage drop across the 50 kΩ resistor by slewing the internal node slowly when
power is applied. At start-up, the xBYPASS capacitor is at 0. The PNP is pulling the mid-point of the bias circuit
down, so the capacitor sees a lower effective voltage, and thus charges slower. This appears as a linear ramp
(while the PNP transistor is conducting), followed by the expected exponential ramp of an R-C circuit.
If the expression in equation 10 cannot be fulfilled or the small amount of pop is still unacceptable for the
application, then external circuitry must be added that can eliminate the pop heard during power up and while
transitioning out of mute or shutdown modes.
By holding the device in SE mode when the pop normally occurs, no pop can be heard through the
BTL-connected speakers (as the negative output is in a high impedance state when the amplifier is in SE mode).
From a hardware point of view, the easiest way to implement this is to drive the SE/BTL terminal from the
general-purpose input-output (GPIO) in the system. If the SE/BTL terminal is normally connected to a
headphone socket (as shown in Figure 65), then the GPIO signal must either be taken through an OR gate (see
Figure 65) or isolated with a diode (any signal diode) (see Figure 66).
VDD
Right
Channel
Rm1
100 kΩ
SE/BTL
0.1 µF
Rm2
100 kΩ
From GPIO
Left
Channel
Figure 65. Implementation with an OR Gate
32
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
VDD
Right
Channel
Rm1
100 kΩ
SE/BTL
0.1 µF
From GPIO
Rm2
100 kΩ
Left
Channel
Figure 66. Implementation with a Diode
The OR gate and diode isolate the GPIO terminal from the headphone switch. In these implementations, the
headphone switch has priority.
When the amplifier is in mute mode, the output stage continues to be biased. This causes the transition out of
mute mode to be very fast with only a short delay (from 100 ms to 500 ms). During power up or the transition
out of shutdown mode, a longer delay ( from 1 s to 2 s) is required. The exact delay time required is dependent
on the values of the external components used with the amplifier (see Figure 67).
System Control:
MUTE or SHUTDOWN
Delay
Output of Delay Circuit
(Input to SE/BTL)
Figure 67. Transition Delay Timing
single-ended operation
In SE mode (see Figure 59 and Figure 60), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 22 and 3).
ǒǓ
In SE mode the gain is set by the RF and RI resistors and is shown in equation 11. Since the inverting amplifier
is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included.
SE Gain
+*
RF
(11)
RI
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship (see equation 12):
ǒ
CB
1
25 kΩ
1 Ơ 1
v
Ǔ ǒC R Ǔ R C
I I
(12)
L C
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
33
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
output coupling capacitor, CC
In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 14.
–3 dB
f c(high)
+ 2 p R1 C
(14)
L C
fc
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of CC are required to pass low
frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω,
4 Ω, 8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 2 summarizes the frequency response characteristics of each
configuration.
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
RL
CC
330 µF
LOWEST FREQUENCY
3Ω
4Ω
330 µF
120 Hz
8Ω
330 µF
60 Hz
161 Hz
32 Ω
330 µF
15 Hz
10,000 Ω
330 µF
0.05 Hz
47,000 Ω
330 µF
0.01 Hz
As Table 2 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
34
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
SE/BTL operation
The ability of the TPA0202 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0202, two separate amplifiers drive OUT+ and OUT–. The SE/BTL input (terminal 14)
controls the operation of the follower amplifier that drives LOUT– and ROUT– (terminals 10 and 15). When
SE/BTL is held low, the amplifier is on and the TPA0202 is in the BTL mode. When SE/BTL is held high, the OUT–
amplifiers are in a high output impedance state, which configures the TPA0202 as an SE driver from LOUT+
and ROUT+ (terminals 3 and 22). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 68.
21
RLINE IN
20
RHP IN
MUX
–
+
ROUT+ 22
–
+
ROUT – 15
COUTR
Rm3
1 kΩ
Bypass
VDD
Rm1
100 kΩ
SE/BTL 14
HP/LINE 16
Rm2
100 kΩ
0.1 µF
Left
Channel
Figure 68. TPA0202 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT– amplifier is
shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.
As shown in the full feature application (Figure 63), the input MUX control can be tied to the SE/BTL input. The
benefits of doing this are described in the following input MUX operation section.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
35
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
Input MUX operation
Working in concert with the SE/BTL feature, the HP/LINE MUX feature gives the audio designer the flexibility
of a multichip design in a single IC (see Figure 69). The primary function of the MUX is to allow different gain
settings for BTL versus SE mode. Speakers typically require approximately a factor of 10 more gain for similar
volume listening levels as compared to headphones. To achieve headphone and speaker listening parity, the
resistor values would need to be set as follows:
SE Gain (HP)
+*
ǒ Ǔ
ǒ Ǔ
R F(HP)
(15)
R I(HP)
If, for example RI(HP) = 10 kΩ and RF(HP) = 10 kΩ then SE Gain(HP) = –1
BTL Gain (LINE)
+ *2
R F(LINE)
(16)
R I(LINE)
If, for example RI(LINE) = 10 kΩ and RF(LINE) = 50 kΩ then BTL Gain(LINE) = –10
RFRHP
CIRLINE R
IRLINE
RFRLINE
21
RLINE IN
–
+
MUX
20
CIRHP
RHP IN
ROUT+ 22
ROUT – 15
RIRHP
Right Channel
MID
VDD
SE/BTL 14
HP/LINE 16
0.1 µF
Left Channel
Figure 69. TPA0202 Example Input MUX Circuit
Another advantage of using the MUX feature is setting the gain of the headphone channel to –1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SE/BTL operation section for a description of the headphone jack control circuit.
36
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
mute and shutdown modes
The TPA0202 employs both a mute and a shutdown mode of operation designed to reduce supply current, IDD,
to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input
terminal should be held low during normal operation when the amplifier is in use. Pulling SHUTDOWN high
causes the outputs to mute and the amplifier to enter a low-current state, IDD = 5 µA. SHUTDOWN or MUTE
IN should never be left unconnected because amplifier operation would be unpredictable. Mute mode alone
reduces IDD to 1.5 mA.
Table 3. Shutdown and Mute Mode Functions
INPUTS†
OUTPUT
SE/BTL
HP/LINE
MUTE IN
Low
Low
X
X
X
X
Low
High
High
High
AMPLIFIER STATE
SHUTDOWN
MUTE OUT
INPUT
Low
Low
Low
L/R Line
BTL
—
High
—
X
Mute
High
—
High
X
Mute
Low
Low
Low
L/R HP
BTL
Low
Low
Low
Low
L/R Line
SE
High
Low
Low
Low
L/R HP
SE
OUTPUT
† Inputs should never be left unconnected.
X = do not care
using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.
5-V versus 3.3-V operation
The TPA0202 operates over a supply range of 3 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability goes.
For 3.3-V operation, supply current is reduced from 19 mA (typical) to 13 mA (typical). The most important
consideration is that of output power. Each amplifier in TPA0202 can produce a maximum voltage swing of
VDD – 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed to
VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into an 8-Ω load
before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power for a given output-power level than operation from 5-V supplies.
When the application demands less than 500 mW, 3.3-V operation should be strongly considered, especially
in battery-powered applications to improve the efficiency.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
37
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA0202 data sheet, one can see that when the
TPA0202 is operating from a 5-V supply into a 3-Ω speaker that 2 W peaks are available. Converting watts to
dB:
P dB
+ 10 Log
ǒǓ
PW
ǒǓ
+ 10 Log 21
+ 3.0 dB
(17)
P ref
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
* 15 dB + * 12 dB (15 dB headroom)
3.0 dB * 12 dB + * 9 dB (12 dB headroom)
3.0 dB * 9 dB + * 6 dB (9 dB headroom)
3.0 dB * 6 dB + * 3 dB (6 dB headroom)
3.0 dB * 3 dB + 0 dB (3 dB headroom)
3.0 dB
Converting dB back into watts:
PW
+ 10PdBń10 Pref
+ 63 mW (15 dB headroom)
+ 120 mW (12 dB headroom)
+ 250 mW (9 dB headroom)
+ 500 mW (6 dB headroom)
+ 1000 mW (3 dB headroom)
(18)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with 0 dB of
headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0202
and maximum ambient temperatures is shown in Table 4.
38
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
headroom and thermal considerations (continued)
Table 4. TPA0202 Power Rating, 5-V, 3-Ω, Stereo
PEAK OUTPUT POWER
(W)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE
2
2W
1.7
– 3°C
2
1000 mW (3 dB)
1.6
6°C
2
500 mW (6 dB)
1.4
24°C
2
250 mW (9 dB)
1.1
51°C
2
120 mW (12 dB)
0.8
78°C
2
63 mW (15 dB)
0.6
96°C
ÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
ÁÁÁÁÁÁÁ
DISSIPATION RATING TABLE
PACKAGE
PWP†
TA ≤ 25°C
2.7 W
DERATING FACTOR
21.8 mW/°C
TA = 70°C
1.7 W
TA = 85°C
1.4 W
2.8 W
22.1 mW/°C
1.8 W
1.4 W
PWP‡
† This parameter is measured with the recommended copper heat sink pattern on a 1-layer PCB, 4 in2 5-in × 5-in PCB,
1 oz. copper, 2-in × 2-in coverage.
‡ This parameter is measured with the recommended copper heat sink pattern on an 8-layer PCB, 6.9 in2 1.5-in × 2-in PCB,
1 oz. copper with layers 1, 2, 4, 5, 7, and 8 at 5% coverage (0.9 in2) and layers 3 and 6 at 100% coverage (6 in2).
The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
and 300 CFM data from the dissipation rating table, the derating factor for the PWP package with 6.9 in2 of
copper area on a multilayer PCB is 22 mW/°C and 54 mW/°C respectively. Converting this to ΘJA:
Θ JA
1
+ Derating
(19)
For 0 CFM :
1
+ 0.022
+ 45°CńW
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given ΘJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0202 is
150 °C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
39
TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205B – FEBRUARY 1998 – REVISED DECEMBER 2000
APPLICATION INFORMATION
headroom and thermal considerations (continued)
T A Max
+ TJ Max * ΘJA PD
+ 150 * 45 (0.6 2) + 96°C (15 dB headroom,
(20)
0 CFM)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB headroom per channel.
Table 4 shows that for some applications no airflow is required to keep junction temperatures in the specified
range. The TPA0202 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 4 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-Ω
speakers dramatically increases the thermal performance by increasing amplifier efficiency.
junction temperature measurement
Characterizing a PCB layout with respect to thermal impedance is very difficult, as it is usually impossible to
know the junction temperature of the IC in question. The TPA0202 terminal 2 (TJ) sources a current proportional
to the junction temperature. The circuit internal to TJ is shown in Figure 70.
VDD
R
R
5R
TJ
Figure 70. TJ Terminal Internal Circuit
Connect an ammeter between TJ and ground to measure the current. As the resistors have a tolerance of ± 20%,
this measurement must be calibrated on each device. The intent of this function is in characterization of the PCB
and end equipment and not a real-time measurement of temperature. Typically a 25°C reading is –120 µA for
a 3.3-V supply and –135 µA for a 5-V supply. The slope is approximately 0.25 µA/°C for both VDD = 3.3 V and
VDD = 5 V. To reduce quiescent current, do not ground TJ in normal operation. It can be connected to VDD or
left floating as it has a resistor connected across the base-emitter junction.
40
POST OFFICE BOX 655303
• DALLAS, TEXAS 75265
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPA0202PWP
ACTIVE
HTSSOP
PWP
24
60
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
TPA0202
TPA0202PWPR
ACTIVE
HTSSOP
PWP
24
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
TPA0202
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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