TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
2.6-W STEREO AUDIO POWER AMPLIFIER WITH FOUR
SELECTABLE GAIN SETTINGS AND MUX CONTROL
FEATURES
•
•
•
•
•
•
•
•
•
•
Compatible With PC 99 Desktop Line-Out Into
10-kΩ Load
Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
2.6-W/Ch Output Power Into 3-Ω Load
Input MUX Select Terminal
PC-Beep Input
Depop Circuitry
Stereo Input MUX
Fully Differential Input
Low Supply Current and Shutdown Current
Surface-Mount Power Packaging 24-Pin
TSSOP PowerPAD™
PWP PACKAGE
(TOP VIEW)
1
2
3
4
5
6
7
8
9
10
11
12
GND
GAIN0
GAIN1
LOUT+
LLINEIN
LHPIN
PVDD
RIN
LOUT–
LIN
BYPASS
GND
24
23
22
21
20
19
18
17
16
15
14
13
GND
RLINEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
HP/LINE
ROUT–
SE/BTL
PC-BEEP
GND
DESCRIPTION
The TPA0312 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2.6 W of continuous RMS power per channel into 3-Ω loads. This device minimizes the number of
external components needed, simplifying the design, and freeing up board space for other features. When driving
1 W into 8-Ω speakers, the TPA0312 has less than 0.65% THD+N across its specified frequency range. Included
within this device is integrated depop circuitry that virtually eliminates transients that cause noise in the speakers.
Amplifier gain is internally configured and controlled by way of two terminals (GAIN0 and GAIN1). BTL gain
settings of 6 dB, 10 dB, 15.6 dB, and 21.6 dB (inverting) are provided, whereas SE gain is always configured as
4.1 dB for headphone drive. An internal input MUX allows two sets of stereo inputs to the amplifier. The HP/LINE
terminal allows the user to select which MUX input is active, regardless of whether the amplifier is in SE or BTL
mode. In notebook applications, where internal speakers are driven as BTL and the line outputs (often
headphone drive) are required to be SE, the TPA0312 automatically switches into SE mode when the SE/BTL
input is activated, and this reduces the gain to 4.1 dB.
The TPA0312 consumes only 6 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to 150 µA.
The PowerPAD™ package (PWP) delivers a level of thermal performance that was previously achievable only in
TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA0312 to operate at full power into 8-Ω loads at an ambient temperature of 85°C.
AVAILABLE OPTIONS
TA
–40°C to 85°C
(1)
PACKAGED DEVICE
TSSOP (1) (PWP)
TPA0312PWP
The PWP package is available taped and reeled. To order a taped
and reeled part, add the suffix R to the part number (e.g.,
TPA0312PWPR).
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2000–2004, Texas Instruments Incorporated
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
These devices have limited built-in ESD protection. The leads should be shorted together or the device
placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates.
FUNCTIONAL BLOCK DIAGRAM
RHPIN
RLINEIN
R
MUX
Volume
Control
−
GAIN0
ROUT+
+
GAIN1
Volume
Control
RIN
−
ROUT−
PC-BEEP
SE/BTL
HP/LINE
LHPIN
LLINEIN
+
PC
Beep
MUX
Control
L
MUX
Depop
Circuitry
Volume
Control
Power
Management
PVDD
VDD
BYPASS
SHUTDOWN
GND
−
LOUT+
+
LIN
Volume
Control
−
LOUT−
+
2
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
TERMINAL FUNCTIONS
TERMINAL
NAME
NO.
I/O
DESCRIPTION
BYPASS
11
GAIN0
2
I
Bit 0 of gain control
GAIN1
3
I
Bit 1 of gain control
GND
Tap to voltage divider for internal mid-supply bias generator
1, 12, 13,
24
Ground connection for circuitry. Connected to the thermal pad.
LHPIN
6
I
Left-channel headphone input, selected when SE/BTL is held high
LIN
10
I
Common left input for fully differential input. AC ground for single-ended inputs.
LLINEIN
5
I
Left-channel line input, selected when SE/BTL is held low
LOUT+
4
O
Left-channel positive output in BTL mode and positive output in SE mode
LOUT-
9
O
Left-channel negative output in BTL mode and high-impedance in SE mode
PC-BEEP
14
I
The input for PC Beep mode. PC-BEEP is enabled when a > 1.5-V (peak-to-peak) square wave is input
to PC-BEEP
HP/LINE
17
I
HP/LINE is the input MUX control input. When the HP/LINE terminal is held high, the headphone inputs
(LHPIN or RHPIN [6, 20]) are active. When the HP/LINE terminal is held low, the line inputs (LLINEIN or
RLINEIN [5, 23]) are active.
PVDD
7, 18
I
Power supply for output stage
RHPIN
20
I
Right-channel headphone input, selected when SE/BTL is held high
RIN
8
I
Common right input for fully differential input. AC ground for single-ended inputs.
RLINEIN
23
I
Right-channel line input, selected when SE/BTL is held low
ROUT+
21
O
Right-channel positive output in BTL mode and positive output in SE mode
ROUT-
16
O
Right-channel negative output in BTL mode and high-impedance in SE mode
SHUTDOWN
22
I
Places entire IC in shutdown mode when held low, except PC-BEEP remains active
SE/BTL
15
I
Hold SE/BTL low for BTL mode and hold high for SE mode.
VDD
19
I
Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.
Thermal Pad
Connect to ground. Must be soldered down in all applications to properly secure the device on the PC
board.
3
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
VDD
Supply voltage
VI
Input voltage
6V
–0.3 V to VDD +0.3 V
Continuous total power dissipation
Internally limited (see Dissipation Rating Table)
TA
Operating free-air temperature range
–40°C to 85°C
TJ
Operating junction temperature range
–40°C to 150°C
Tstg
Storage temperature range
–65°C to 85°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
260°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
TA≤ 25°C
PACKAGE
PWP
(1)
2.7
DERATING FACTOR
TA = 70°C
TA = 85°C
21.8 mW/°C
1.7 W
1.4 W
W (1)
See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD of the before-mentioned document.
RECOMMENDED OPERATING CONDITIONS
VDD
Supply voltage
VIH
High-level input voltage
VIL
Low-level input voltage
SE/BTL, HP/LINE, GAIN0, GAIN1
SHUTDOWN
MIN
MAX
4.5
5.5
V
0.8 x VDD
V
2
SE/BTL, HP/LINE
0.6 x VDD
GAIN0, GAIN1
0.4 x VDD
SHUTDOWN
TA
UNIT
V
0.8
Operating free-air temperature
–40
°C
85
ELECTRICAL CHARACTERISTICS
at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
|VOO|
Output offset voltage (measured differentially)
VI = 0, Av = 6 dB
PSRR
Power supply rejection ratio
VDD = 4.5 V to 5.5 V
|IIH|
High-level input current
|IIL|
Low-level input current
IDD
Supply current
IDD(SD)
Supply current, shutdown mode
4
MIN
TYP
MAX
UNIT
25
mV
VDD = 5.5 V, VI = VDD
1
µA
VDD = 5.5 V, VI = 0 V
1
µA
77
dB
BTL mode
6
10
SE mode
3
5
150
300
mA
µA
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
OPERATING CHARACTERISTICS
VDD = 5 V, TA = 25°C, RL = 8 Ω , Gain = 6 dB, BTL mode
PARAMETER
TEST CONDITIONS
PO
Output power
RL = 3 Ω
THD + N
Total harmonic distortion plus noise
PO = 1 W,
BOM
Maximum output power bandwidth
THD = 5%
Supply ripple rejection ratio
f = 1 kHz, CB = 0.47 µF
SNR
Noise output voltage
2.6
THD + N = 1%,
2.05
f = 20 Hz to 15 kHz
ZI
Input impedance
CB = 0.47 µF, f = 20 Hz
to 20 kHz
TYP
THD + N= 10%
BTL mode
Signal-to-noise ratio
Vn
MIN
MAX
UNIT
W
0.65%
>15
kHz
72
dB
105
dB
BTL mode
20
SE mode
18
µVRMS
See Table 1
TYPICAL CHARACTERISTICS
TABLE OF GRAPHS
FIGURE
vs Output power
1, 4-6, 9-11,
14-16, 18
vs Frequency
2, 3, 7, 8, 12,
13, 17, 19
THD+N
Total harmonic distortion plus noise
Vn
Output noise voltage
vs Bandwidth
21
Supply ripple rejection ratio
vs Frequency
22, 23
Crosstalk
vs Frequency
24, 25
Shutdown attenuation
vs Frequency
26
Signal-to-noise ratio
vs Frequency
vs Output voltage
SNR
Closed-loop response
PO
PD
Output power
Power dissipation
20
27
28-30
vs Load resistance
31, 32
vs Output power
33, 34
vs Ambient temperature
35
5
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
AV = 6 dB
f = 1 kHz
BTL
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
10%
RL = 4 Ω
1%
RL = 8 Ω
RL = 3 Ω
0.1%
0.01%
0.5 0.75
1
1.25 1.5 1.75
2
2.25 2.5 2.75
PO = 1.75 W
RL = 3 Ω
BTL
1%
AV = 21.6 dB
0.1%
AV = 15.6 dB
0.01%
20
3
PO − Output Power − W
1k
10k 20k
f − Frequency − Hz
Figure 2.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 3 Ω
AV = 6 dB
BTL
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
100
Figure 1.
10%
1%
PO = 1.0 W
PO = 0.5 W
0.1%
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
RL = 3 Ω
AV = 6 dB
BTL
PO = 1.75 W
0.01%
20
100
1k
f − Frequency − Hz
Figure 3.
6
AV = 6 dB
10k 20k
0.01%
0.01
0.1
1
PO − Output Power − W
Figure 4.
10
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
10%
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 3 Ω
AV = 15.6 dB
BTL
0.01%
0.01
0.1
1
PO − Output Power − W
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 3 Ω
AV = 21.6 dB
BTL
0.01%
0.01
10
0.1
1
PO − Output Power − W
10
Figure 5.
Figure 6.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
PO = 1.75 W
RL = 3 Ω
BTL
1%
AV = 21.6 dB
AV = 6 dB
0.1%
AV = 15.6 dB
0.01%
20
100
1k
10k 20k
THD+N −Total Harmonic Distortion + Noise
10%
THD+N −Total Harmonic Distortion + Noise
f = 15 kHz
RL = 4 Ω
AV = 6 dB
BTL
1%
PO = 1.5 W
0.1%
PO = 0.25 W
PO = 1.0 W
0.01%
20
100
1k
f − Frequency − Hz
f − Frequency − Hz
Figure 7.
Figure 8.
10k 20k
7
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 4 Ω
AV = 6 dB
BTL
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
10%
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
0.1
1
PO − Output Power − W
f = 1 kHz
0.1%
f = 20 Hz
RL = 4 Ω
AV = 15.6 dB
BTL
0.1
1
PO − Output Power − W
10
Figure 9.
Figure 10.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
1%
0.01%
0.01
10
10%
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 4 Ω
AV = 21.6 dB
BTL
0.01%
0.01
0.1
1
PO − Output Power − W
Figure 11.
8
f = 15 kHz
10
RL = 8 Ω
AV = 6 dB
BTL
1%
0.1%
PO = 0.25 W
PO = 1.0 W
0.01%
20
PO = 0.5 W
100
1k
f − Frequency − Hz
Figure 12.
10k 20k
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
PO = 1 W
RL = 8 Ω
BTL
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
10%
1%
AV = 21.6 dB
AV = 6 dB
0.1%
AV = 15.6 dB
0.01%
20
100
1k
RL = 8 Ω
AV = 6 dB
BTL
f = 15 kHz
1%
f = 1 kHz
0.1%
f = 20 Hz
0.01%
0.01
10k 20k
f − Frequency − Hz
Figure 13.
Figure 14.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
f = 15 kHz
1%
f = 1 kHz
0.1%
0.01%
0.01
10
10%
RL = 8 Ω
AV = 15.6 dB
BTL
THD+N −Total Harmonic Distortion + Noise
10%
THD+N −Total Harmonic Distortion + Noise
0.1
1
PO − Output Power − W
f = 20 Hz
0.1
1
PO − Output Power − W
Figure 15.
10
f = 15 kHz
1%
f = 1 kHz
f = 20 Hz
0.1%
RL = 8 Ω
AV = 21.6 dB
BTL
0.01%
0.01
0.1
1
PO − Output Power − W
10
Figure 16.
9
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL = 32 Ω
AV = 4.1 dB
SE
1%
PO = 25 mW
0.1%
PO = 50 mW
PO = 75 mW
0.01%
20
100
1k
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
10%
RL = 32 Ω
AV = 4.1 dB
SE
1%
f = 15 kHz
0.1%
f = 1 kHz
f = 20 Hz
0.01%
0.01
10k 20k
f − Frequency − Hz
Figure 17.
Figure 18.
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE
THD+N −Total Harmonic Distortion + Noise
THD+N −Total Harmonic Distortion + Noise
RL = 10 kΩ
AV = 4.1 dB
SE
1%
0.1%
VO = 1 VRMS
0.01%
0.001%
20
1
10%
10%
100
1k
f − Frequency − Hz
Figure 19.
10
0.1
PO − Output Power − W
10k 20k
RL = 10 kΩ
AV = 4.1 dB
SE
1%
0.1%
f = 20 Hz
f = 15 kHz
0.01%
f = 1 kHz
0.001%
0.1
1
VO − Output Voltage − VRMS
Figure 20.
3
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
OUTPUT NOISE VOLTAGE
vs
BANDWIDTH
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
0
100
V n − Output Noise Voltage − µ V
90
Supply Ripple Rejection Ratio − dB
VDD = 5 V
RL = 4Ω
80
70
60
50
40
AV = 21.6 dB
30
AV = 15.6 dB
20
−20
RL = 8 Ω
CB = 0.47 µF,
AV = 6 dB
BTL
−40
−60
−80
−100
10
AV = 6 dB
0
10
100
1k
−120
20
10k
BW − Bandwidth − Hz
Figure 21.
Figure 22.
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY
CROSSTALK
vs
FREQUENCY
10k 20k
0
RL = 32 Ω
CB = 0.47 µF,
AV =4.1 dB
SE
−20
−40
Crosstalk − dB
Supply Ripple Rejection Ratio − dB
1k
f − Frequency − Hz
0
−20
100
−60
PO = 1 W
RL = 8 Ω
Av = 6 dB
BTL
−40
−60
−80
−80
−100
−100
−120
20
−120
20
LEFT TO RIGHT
RIGHT TO LEFT
100
1k
10k 20k
100
1k
f − Frequency − Hz
f − Frequency − Hz
Figure 23.
Figure 24.
10k 20k
11
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
CROSSTALK
vs
FREQUENCY
SHUTDOWN ATTENUATION
vs
FREQUENCY
0
VI = 1 VRMS
−20
Shutdown Attenuation − db
Crosstalk − dB
−20
0
VO = 1 VRMS
RL = 10 kΩ
Av = 4.1 dB
SE
−40
−60
LEFT TO RIGHT
−80
−100
RL = 10 kΩ, SE
−40
−60
RL = 32 Ω, SE
−80
−100
RL = 8 Ω, BTL
RIGHT TO LEFT
−120
20
100
1k
−120
20
10k 20k
100
1k
f − Frequency − Hz
f − Frequency − Hz
Figure 25.
Figure 26.
SIGNAL-TO-NOISE RATIO
vs
FREQUENCY
140
SNR − Signal-To-Noise Ratio − dB
130
PO = 1 W
RL = 8 Ω
BTL
120
AV = 6 dB
AV = 15.6 dB
110
100
90
AV = 21.6 dB
80
70
60
20
100
1k
f − Frequency − Hz
Figure 27.
12
10k 20k
10k 20k
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
CLOSED-LOOP RESPONSE
180°
10
7.5
Gain
90°
5
Phase
0°
0
Phase
Gain − dB
2.5
−2.5
−5
RL = 8 Ω
AV = 6 dB
BTL
−90°
−7.5
−10
10
−180°
100
1k
10k
100k
1M
f − Frequency − Hz
Figure 28.
CLOSED-LOOP RESPONSE
180°
30
25
90°
20
Gain
Phase
0°
10
Phase
Gain − dB
15
5
0
RL = 8 Ω
AV = 15.6 dB
BTL
−90°
−5
−10
10
−180°
100
1k
10k
100k
1M
f − Frequency − Hz
Figure 29.
13
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
CLOSED-LOOP RESPONSE
180°
30
25
Gain
90°
20
Phase
0°
10
Phase
Gain − dB
15
5
0
RL = 8 Ω
AV = 21.6 dB
BTL
−90°
−5
−10
10
−180°
100
1k
10k
1M
100k
f − Frequency − Hz
Figure 30.
OUTPUT POWER
vs
LOAD RESISTANCE
OUTPUT POWER
vs
LOAD RESISTANCE
3.5
1500
AV = 6 dB
BTL
1250
PO− Output Power − mW
PO − Output Power − W
3
AV = 4.1 dB
SE
2.5
2
10% THD+N
1.5
1
1% THD+N
0.5
1000
750
10% THD+N
500
250
1% THD+N
0
0
0
8
16
24
32
40
48
RL − Load Resistance − Ω
Figure 31.
14
56
64
0
8
16
24
32
40
48
RL − Load Resistance − Ω
Figure 32.
56
64
TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
POWER DISSIPATION
vs
OUTPUT POWER
POWER DISSIPATION
vs
OUTPUT POWER
1.8
0.4
3Ω
PD − Power Dissipation − W
0.35
1.4
1.2
4Ω
1
0.8
0.6
8Ω
0.3
f = 1 kHz
BTL
Each Channel
0.2
0
0
0.5
1
1.5
PO − Output Power − W
2
4Ω
0.25
0.2
0.15
8Ω
0.1
0.4
0.05
0
0
2.5
f = 1 kHz
SE
Each Channel
32 Ω
0.1
0.2
0.3
0.4
0.5
0.6
PO − Output Power − W
Figure 33.
0.7
0.8
Figure 34.
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7
ΘJA4
6
PD − Power Dissipation − W
PD − Power Dissipation − W
1.6
ΘJA1 = 45.9°C/W
ΘJA2 = 45.2°C/W
ΘJA3 = 31.2°C/W
ΘJA4 = 18.6°C/W
5
4
ΘJA3
3
ΘJA1,2
2
1
0
−40 −20
0
20 40 60 80 100 120 140 160
TA − Ambient Temperature − °C
Figure 35.
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TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure
36) to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down
TO-220-type packages have leads formed as gull wings to make them applicable for surface-mount applications.
These packages, however, have only two shortcomings: they do not address the low profile (< 2 mm)
requirements of many of today's advanced systems, and they do not offer a terminal-count high enough to
accommodate increasing integration. On the other hand, traditional low-power, surface-mount packages require
power-dissipation derating that severely limits the usable range of many high-performance analog circuits.
The PowerPAD™ package (thermally enhanced TSSOP) combines fine-pitch, surface-mount technology with
thermal performance comparable to much larger power packages.
The PowerPAD™ package is designed to optimize the heat transfer to the PWB. Because of the small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths
that remove heat from the component. The thermal pad is formed using a patented lead-frame design and
manufacturing technique to provide a direct connection to the heat-generating IC. When this pad is soldered or
otherwise thermally coupled to an external heat dissipator, high power dissipation in the ultrathin, fine-pitch,
surface-mount package can be reliably achieved.
DIE
Side View (a)
Thermal
Pad
DIE
End View (b)
Bottom View (c)
Figure 36. Views of Thermally Enhanced PWP Package
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TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
APPLICATION INFORMATION
SELECTION OF COMPONENTS
Figure 37 and Figure 38 are schematic diagrams of typical notebook computer application circuits.
Right CIRHP
Head− 0.47 µF
phone
Input
20
Signal
CIRLINE
Right 0.47 µF
Line
Input
Signal
23
RHPIN
RLINEIN
R
MUX
Volume
Control
−
+
8
RIN
CRIN
0.47 µF
PC-BEEP
14
Input
Signal
CPCB
0.47 µF
ROUT+
21
Volume
Control
COUTR
330 µF
PC-BEEP
−
+
PCBeep
ROUT−
16
VDD
1 kΩ
100 kΩ
2, 3
17
15
GAIN0
GAIN1
HP/LINE
SE/BTL
Gain/
MUX
Control
Depop
Circuitry
Power
Management
Left CILHP
Head− 0.47 µF
phone
Input
Signal
CILLINE
Left 0.47 µF
Line
Input
Signal
6
LHPIN
5
LLINEIN
10
CLIN
0.47 µF
LIN
PVDD
18
VDD
19
BYPASS
SHUT−
DOWN
11
GND
L
MUX
Volume
Control
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
CBYP
0.47 µF
To
System
Control
−
+
LOUT+
4
−
+
LOUT−
9
1 kΩ
1, 12,
13, 24
COUTL
330 µF
Volume
Control
100 kΩ
A.
A 0.1-µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower frequency noise
signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 37. Typical TPA0312 Application Circuit Using Single-Ended Inputs and Input MUX
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TPA0312
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APPLICATION INFORMATION (continued)
CIRHP−
0.47 µF
Right
Negative
Differential
Input Signal
Right
Positive
Differential
Input Signal
PC-BEEP
Input
Signal
20
CIRIN−
0.47 µF
23
RHPIN
RLINEIN
R
MUX
Volume
Control
−
ROUT+
21
+
CIRIN+
0.47 µF
8
RIN
14
PC-BEEP
Volume
Control
COUTR
330 µF
−
PCBeep
CPCB
0.47 µF
ROUT−
16
VDD
+
1 kΩ
100 kΩ
2, 3
GAIN0
GAIN1
17
HP/LINE
15
SE/BTL
Gain/
MUX
Control
Depop
Circuitry
Power
Management
CILHP
0.47 µF
Left
Negative
Differential
Input Signal
Left
Positive
Differential
Input Signal
6
LHPIN
5
LLINEIN
PVDD
18
VDD
19
BYPASS
SHUT−
DOWN
11
GND
L
MUX
CILIN−
0.47 µF
Volume
Control
−
LOUT+
See Note A
VDD
CSR
0.1 µF
VDD
CSR
0.1 µF
22
CBYP
0.47 µF
To
System
Control
4
1 kΩ
1, 12,
13, 24
+
10
LIN
COUTL
330 µF
Volume
Control
CILIN
0.47 µF
−
LOUT−
9
+
100 kΩ
A.
A 0.1-µF ceramic capacitor should be placed as close as possible to the IC. For filtering lower frequency noise
signals, a larger electrolytic capacitor of 10 µF or greater should be placed near the audio power amplifier.
Figure 38. Typical TPA0312 Application Circuit Using Differential Inputs
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TPA0312
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GAIN SETTING VIA GAIN0 AND GAIN1 INPUTS
The gain of the TPA0312 is set by two input terminals, GAIN0 and GAIN1.
Table 1. GAIN SETTINGS
GAIN0
GAIN1
SE/BTL
AV
0
0
0
6 dB
0
1
0
10 dB
1
0
0
15.6 dB
1
1
0
21.6 dB
X
X
1
4.1 dB
The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, ZI, to be dependant on the gain setting. The actual gain settings are controlled by
ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input impedance
will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kΩ, which is the absolute minimum input impedance of the TPA0312. At the lower gain settings,
the input impedance could increase as high as 115 kΩ.
INPUT RESISTANCE
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the –3-dB or
cutoff frequency also changes by over 6 times. If an additional resistor is connected from the input pin of the
amplifier to ground, as shown in the following figure, the variation of the cutoff frequency is much reduced.
ZF
C
IN
Input
Signal
ZI
R
The typical input impedance at each gain setting is given in the table below:
Av
ZI
21.6 dB
25 kΩ
15.6 dB
45 kΩ
10 dB
70 kΩ
6 dB
90 kΩ
The –3-dB frequency can be calculated using Equation 1:
1
ƒ –3 dB
2 C R R
I
(1)
If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.
INPUT CAPACITOR, CI
In the typical application, an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a
high-pass filter with the corner frequency determined in Equation 2.
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−3 dB
fc(highpass)
1
2 ZI C I
fc
(2)
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 26 kΩ and the specification calls for a flat bass response down to 65 Hz.
Equation 2 is reconfigured as Equation 3.
1
C
I
2 Z f c
I
(3)
In this example, CI is 94 nF; so, one would likely choose a value in the range of 0.1 nF to 1 µF. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high-gain applications. For this reason, a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at VDD/2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.
POWER SUPPLY DECOUPLING, CS
The TPA0312 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 µF, placed as close as possible to the device VDD lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio
power amplifier is recommended.
MIDRAIL BYPASS CAPACITOR, CBYP
The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor values of 0.47-µF to 1-µF ceramic or tantalum low-ESR capacitors are recommended for the
best THD and noise performance.
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OUTPUT COUPLING CAPACITOR, CC
In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias at
the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the output
coupling capacitor and impedance of the load form a high-pass filter governed by Equation 4.
−3 dB
fc(high)
1
2 RL C C
fc
(4)
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of CC are required to pass low
frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 3 Ω, 4 Ω,
8 Ω, 32 Ω, 10 kΩ, to 47 kΩ. Table 2 summarizes the frequency response characteristics of each configuration.
Table 2. COMMON LOAD IMPEDANCES VS LOW FREQUENCY OUTPUT CHARACTERISTICS IN SE MODE
RL (Ω)
CC (µF)
LOWEST FREQUENCY( Hz)
3
330
161
4
330
120
8
330
60
32
330
15
10,000
330
0.05
47,000
330
0.01
As Table 2 indicates, most of the bass response is attenuated into a 4-Ω load, an 8-Ω load is adequate,
headphone response is good, and drive into line level inputs (a home stereo, for example) is exceptional.
USING LOW-ESR CAPACITORS
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance, the more the real capacitor behaves like an ideal capacitor.
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TPA0312
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BRIDGE-TIED LOAD VERSUS SINGLE-ENDED MODE
Figure 39 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0312 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground-referenced load. Plugging 2 × VO(PP) into the power equation, where
voltage is squared, yields 4× the output power from the same supply rail and load impedance (see Equation 5).
V
O(PP)
V (rms)
2 2
2
V
(rms)
Power
R
L
(5)
VDD
VO(PP)
RL
2x VO(PP)
VDD
–VO(PP)
Figure 39. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-Ω speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement—which is loudness that can be heard. In addition to increased power, there are frequency response
concerns. Consider the single-supply SE configuration shown in Figure 40. A coupling capacitor is required to
block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to
1000 µF); so, they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of
limiting low-frequency performance of the system. This frequency limiting effect is due to the high-pass filter
network created with the speaker impedance and the coupling capacitance and is calculated with Equation 6.
1
fc
2 R C
L C
(6)
For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
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TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
VDD
–3 dB
VO(PP)
CC
RL
VO(PP)
fc
Figure 40. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the Crest Factor and Thermal
Considerations section.
SINGLE-ENDED OPERATION
In SE mode the load is driven from the primary amplifier output for each channel (OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 4.1 dB.
BTL AMPLIFIER EFFICIENCY
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sine-wave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD.
The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 41).
VO
IDD
IDD(avg)
V(LRMS)
Figure 41. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are
different between SE and BTL configurations. In an SE application, the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
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TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
P
Efficiency of a BTL amplifier
P
L
SUP
Where:
2
V rms 2
V
V
P L
, and V
P , therefore, P P
L
LRMS
L
2
R
2R
L
L
1
and P SUP VDD I DDavg and I DDavg
2V
V
P
P sin(t) dt 1 P [cos(t)]
0
R
R
R
L
L
0
L
V
Therefore,
2V
V
DD P
R
L
substituting PL and PSUP into Equation 7,
P
SUP
PL = Power delivered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
VP = Peak voltage on BTL load
IDDavg = Average current drawn from the
power supply
VDD = Power supply voltage
ηBTL = Efficiency of a BTL amplifier
2
Efficiency of a BTL amplifier
Where:
V
P
VP
2 RL
2 V DD V P
RL
VP
4 VDD
2 PL RL
(7)
Therefore,
BTL
2 PL RL
4V
DD
(8)
Table 3 employs Equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in
a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at full
output power is less than in the half-power range. Calculating the efficiency for a specific system is the key to
proper power supply design. For a stereo 1-W audio system with 8-Ω loads and a 5-V supply, the maximum draw
on the power supply is almost 3.25 W.
Table 3. EFFICIENCY VS OUTPUT POWER IN 5-V, 8-Ω, BTL SYSTEMS
(1)
24
OUTPUT POWER
(W)
EFFICIENCY
(%)
PEAK VOLTAGE
(V)
INTERNAL DISSIPATION
(W)
0.25
31.4
2.00
0.55
0.50
44.4
2.83
0.62
1.00
62.8
4.00
0.59
1.25
70.2
4.47 (1)
0.53
High peak voltages cause the THD to increase.
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in Equation 8, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.
CREST FACTOR AND THERMAL CONSIDERATIONS
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0312 data sheet, one can see
that when the TPA0312 is operating from a 5-V supply into a 3-Ω speaker, 4-W peaks are available. Converting
watts to dB:
P
P
10Log W 10Log 4 W 6 dB
dB
1W
P
ref
(9)
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
• 6 dB - 15 dB = -9 dB (15-dB crest factor)
• 6 dB - 12 dB = -6 dB (12-dB crest factor)
• 6 dB - 9 dB = -3 dB (9-dB crest factor)
• 6 dB - 6 dB = 0 dB (6-dB crest factor)
• 6 dB - 3 dB = 3 dB (3-dB crest factor)
Converting dB back into watts:
P
W
10PdB10 P
ref
63 mW (18−dB crest factor)
125 mW (15−dB crest factor)
250 mW (9−dB crest factor)
500 mW (6−dB crest factor)
1000 mW (3−dB crest factor)
2000 mW (15−dB crest factor)
(10)
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TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3-dB crest
factor, against 12-dB and 15-dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-Ω system, the internal dissipation in the TPA0312 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0312 POWER RATING, 5-V, 3-Ω, STEREO
PEAK OUTPUT POWER
(W)
(1)
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE (1)
4
2 W (3 dB)
1.7
-3°C
4
1000 mW (6 dB)
1.6
6°C
4
500 mW (9 dB)
1.4
24°C
4
250 mW (12 dB)
1.1
51°C
4
125 mW (15 dB)
0.8
78°C
4
63 mW (18 dB)
0.6
85°C
Package limited to 85°C ambient
Table 5. TPA0312 POWER RATING, 5-V, 8-Ω, STEREO
(1)
PEAK OUTPUT POWER
AVERAGE OUTPUT POWER
POWER DISSIPATION
(W/Channel)
MAXIMUM AMBIENT
TEMPERATURE (1)
2.5 W
1250 mW (3-dB crest factor)
0.55
85°C
2.5 W
1000 mW (4-dB crest factor)
0.62
85°C
2.5 W
500 mW (7-dB crest factor)
0.59
85°C
2.5 W
250 mW (10-dB crest factor)
0.53
85°C
Package limited to 85°C ambient
The maximum dissipated power, PDmax, is reached at a much lower output power level for a 3-Ω load than for an
8-Ω load. As a result, this simple formula for calculating PDmax may be used for a 3-Ω application:
2V2
DD
P Dmax
2R L
(11)
However, in the case of an 8-Ω load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula for
an 8-Ω load, but do not exceed the maximum ambient temperature of 85°C.
The maximum ambient temperature depends on the heat-sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to θJA:
1
1 45°CW
Θ
JA
0.022
Derating Factor
(12)
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are per
channel so the dissipated power needs to be doubled for two-channel operation. Given θJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0312 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T A Max T J Max ΘJA P D
150 45(0.6 2) 96°C (15−dB crest factor)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2.6-W system with 15-dB crest factor
per channel. Package limited to 85°C
26
(13)
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
Table 4 and Table 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0312 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Table 4 and Table 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-Ω speakers dramatically increases the thermal performance by increasing amplifier
efficiency.
SE/BTL OPERATION
The ability of the TPA0312 to easily switch between BTL and SE modes is one of its most important cost-saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0312, two separate amplifiers drive OUT+ and OUT-. The SE/BTL input (terminal 15) controls
the operation of the follower amplifier that drives LOUT- and ROUT- (terminals 9 and 16). When SE/BTL is held
low, the amplifier is on and the TPA0312 is in the BTL mode. When SE/BTL is held high, the OUT- amplifiers are
in a high-output impedance state, which configures the TPA0312 as an SE driver from LOUT+ and ROUT+
(terminals 4 and 21). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL input can be
from a logic-level CMOS source or, more typically, from a resistor divider network as shown in Figure 42.
20
23
RHPIN
RLINEIN
R
MUX
Volume
Control
−
+
8
RIN
ROUT+
21
Volume
Control
VDD
−
+
ROUT−
16
100 kΩ
SE/BTL
COUTR
330 µF
15
1 kΩ
100 kΩ
Figure 42. TPA0312 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3,5-mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed, the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT- amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.
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TPA0312
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SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
INPUT MUX OPERATION
CIRHP
0.47 µF
Right
Headphone
Input Signal
CIRLINE
0.47 µF
20
RHPIN
23
RLINEIN
R
MUX
Volume
Control
Right Line
Input Signal
8
CRIN
0.47 µF
RIN
−
+
ROUT+
−
+
ROUT− 16
21
Volume
Control
SE/BTL
15
HP/LINE
2
Figure 43. TPA0312 Example Input MUX Circuit
The TPA0312 offers the capability for the designer to use separate headphone inputs (RHPIN, LHPIN) and line
inputs (RLINEIN, LLINEIN). The inputs can be different if the input signal is single-ended. If using a differential
input signal, the inputs must be the same because the inputs share a common RIN, LIN. Although the typical
application in Figure 37 shows the input mux control signal HP/LINE tied to SE/BTL, that configuration is not
required. The input mux can be used to select between two inputs that are used in both SE and BTL modes.
If using the TPA0312 with a single-ended input, the RIN and LIN terminals must be tied through a capacitor to
ground, as shown in Figure 43. RIN and LIN must not be tied to bypass or an offset occurs on the output causing
the device to pop when turning on and off.
Input coupling capacitors can be eliminated when using differential inputs, but are used to obtain maximum
output power. If the input capacitors are eliminated, the dc offset must match the voltage on BYPASS or the
output power is limited.
28
TPA0312
www.ti.com
SLOS335A – DECEMBER 2000 – REVISED OCTOBER 2004
PC-BEEP OPERATION
The PC-BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC-BEEP input is active,
both LINEIN and HPIN inputs are deselected, and both the left and right channels are driven in BTL mode with
the signal from PC-BEEP. The gain from the PC-BEEP input to the speakers is fixed at 0.3 V/V and is
independent of the volume setting. When the PC-BEEP input is deselected, the amplifier returns to the previous
operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC-BEEP takes
the device out of shutdown, outputs the PC-BEEP signal, then returns the amplifier to shutdown mode.
The preferred input signal is a square wave or pulse train. To be accurately detected, the signal must have a
minimum of 1.5-Vpp amplitude, rise and fall times of less than 0.1 µs and a minimum of eight rising edges. When
the signal is no longer detected, the amplifier returns to its previous operating mode and volume setting.
To ac-couple the PC-BEEP input, choose a coupling-capacitor value to satisfy Equation 14:
C PCB
1
2 ƒ PCB (100 k)
(14)
The PC-BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally rests at
midrail when no signal is present.
SHUTDOWN MODES
The TPA0312 employs a shutdown mode of operation designed to reduce supply current, IDD, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal should
be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the outputs to
mute and the amplifier to enter a low-current state, IDD = 150 µA. SHUTDOWN should never be left unconnected
because amplifier operation would be unpredictable.
Table 6. HP/LINE, SE/BTL, AND SHUTDOWN FUNCTIONS
INPUTS (1)
(1)
(2)
HP/LINE
SE/BTL
X (2)
AMPLIFIER STATE
SHUTDOWN
INPUT
OUTPUT
X (2)
Low
X (2)
Mute
Low
Low
High
Line
BTL
Low
High
High
Line
SE
High
Low
High
HP
BTL
High
High
High
HP
SE
Inputs should never be left unconnected.
X = do not care
29
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPA0312PWP
ACTIVE
HTSSOP
PWP
24
60
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
TPA0312
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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