TPA122DGNG4

TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 150-mW STEREO AUDIO POWER AMPLIFIER FEATURES • • • • • • • DESCRIPTION 150-mW Stereo Output PC Power Supply Compatible – Fully Specified for 3.3-V and 5-V Operation – Operation to 2.5 V Pop Reduction Circuitry Internal Midrail Generation Thermal and Short-Circuit Protection Surface-Mount Packaging – PowerPAD™ MSOP – SOIC Pin Compatible With LM4880 and LM4881 (SOIC) The TPA122 is a stereo audio power amplifier packaged in either an 8-pin SOIC, or an 8-pin PowerPAD™ MSOP package capable of delivering 150 mW of continuous RMS power per channel into 8-Ω loads. Amplifier gain is externally configured by means of two resistors per input channel and does not require external compensation for settings of 1 to 10. THD+N when driving an 8-Ω load from 5 V is 0.1% at 1 kHz, and less than 2% across the audio band of 20 Hz to 20 kHz. For 32-Ω loads, the THD+N is reduced to less than 0.06% at 1 kHz, and is less than 1% across the audio band of 20 Hz to 20 kHz. For 10-kΩ loads, the THD+N performance is 0.01% at 1 kHz, and less than 0.02% across the audio band of 20 Hz to 20 kHz. D OR DGN PACKAGE (TOP VIEW) VO1 IN1− BYPASS GND 1 8 2 7 3 6 4 5 VDD VO2 IN2− SHUTDOWN TYPICAL APPLICATION CIRCUIT 320 kΩ RF Audio Input 320 kΩ VDD 8 VDD CS VDD/2 RI 2 IN1– 3 BYPASS 6 IN2– CI VO1 1 – + CC CB Audio Input RI CI From Shutdown Control Circuit 5 VO2 7 – + SHUTDOWN CC Bias Control 4 RF Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 1998–2004, Texas Instruments Incorporated TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. AVAILABLE OPTIONS PACKAGED DEVICES TA SMALL OUTLINE (1) (D) MSOP (1) (DGN) –40°C to 85°C TPA122D TPA122DGN (1) MSOP SYMBOLIZATION TI AAE The D and DGN packages are available in left-ended tape and reel only (e.g., TPA122DR, TPA122DGNR). Terminal Functions TERMINAL NAME I/O NO. DESCRIPTION BYPASS 3 I Tap to voltage divider for internal mid-supply bias supply. Connect to a 0.1 µF to 1 µF low ESR capacitor for best performance. GND 4 I GND is the ground connection. IN1- 2 I IN1- is the inverting input for channel 1. IN2- 6 I IN2- is the inverting input for channel 2. SHUTDOWN 5 I Puts the device in a low quiescent current mode when held high VDD 8 I VDD is the supply voltage terminal. VO1 1 O VO1 is the audio output for channel 1. VO2 7 O VO2 is the audio output for channel 2. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) UNIT VDD Supply voltage VI Input voltage 6V –0.3 V to VDD + 0.3 V Continuous total power dissipation Internally limited TJ Operating junction temperature range –40°C to 150°C Tstg Storage temperature range –65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds (1) 260°C Stresses beyond those listed under "absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATING TABLE PACKAGE (1) 2 TA ≤ 25°C POWER RATING DERATING FACTOR ABOVE TA = 25°C TA = 70°C POWER RATING TA = 85°C POWER RATING D 725 mW 5.8 mW/°C 464 mW 377 mW DGN 2.14 W (1) 17.1 mW/°C 1.37 W 1.11 W See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD of that document. TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 RECOMMENDED OPERATING CONDITIONS MIN MAX VDD Supply voltage 2.5 5.5 V TA Operating free-air temperature –40 85 °C VIH High-level input voltage, (SHUTDOWN) VIL Low-level input voltage, (SHUTDOWN) 0.80 × VDD UNIT V 0.40 × VDD V DC ELECTRICAL CHARACTERISTICS at TA = 25°C, VDD = 3.3 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP VOO Output offset voltage PSRR Power supply rejection ratio VDD = 3.2 V to 3.4 V 83 IDD Supply current VDD = 2.5, SHUTDOWN = 0 V IDD(SD) Supply current in SHUTDOWN mode VDD = 2.5, SHUTDOWN = VDD ZI Input impedance MAX UNIT 10 mV 1.5 3 mA 10 50 µA dB >1 MΩ AC OPERATING CHARACTERISTICS VDD = 3.3 V, TA = 25°C, RL = 8 Ω PARAMETER TEST CONDITIONS MIN TYP MAX 70 (1) UNIT PO Output power (each channel) THD≤ 0.1% THD+N Total harmonic distortion + noise PO = 70 mW, 20 Hz–20 kHz BOM Maximum output power BW G = 10, THD < 5% Phase margin Open loop 58° Supply ripple rejection f = 1 kHz 68 dB 86 dB Channel/channel output separation f = 1 kHz SNR Signal-to-noise ratio PO = 100 mW Vn Noise output voltage (1) mW 2% > 20 kHz 100 dB 9.5 µV(rms) Measured at 1 kHz DC ELECTRICAL CHARACTERISTICS at TA = 25°C, VDD = 5.5 V (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP VOO Output offset voltage PSRR Power supply rejection ratio VDD = 4.9 V to 5.1 V 76 IDD Supply current SHUTDOWN = 0 V IDD(SD) Supply current in SHUTDOWN mode SHUTDOWN = VDD |IIH| High-level input current (SHUTDOWN) |IIL| Low-level input current (SHUTDOWN) ZI Input impedance MAX UNIT 10 mV 1.5 3 mA 60 100 µA VDD = 5.5 V, VI = VDD 1 µA VDD = 5.5 V, VI = 0 V 1 >1 dB µA MΩ 3 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 AC OPERATING CHARACTERISTICS VDD = 5 V, TA = 25°C, RL = 8 Ω PARAMETER TEST CONDITIONS MIN TYP MAX 70 (1) UNIT PO Output power (each channel) THD≤ 0.1% THD+N Total harmonic distortion + noise PO = 150 mW, 20 Hz–20 kHz BOM Maximum output power BW G = 10, THD < 5% Phase margin Open loop 56° Supply ripple rejection ratio f = 1 kHz 68 dB 86 dB Channel/channel output separation f = 1 kHz SNR Signal-to-noise ratio PO = 150 mW Vn Noise output voltage (1) mW 2% > 20 kHz 100 dB 9.5 µV(rms) Measured at 1 kHz AC OPERATING CHARACTERISTICS VDD = 3.3 V, TA = 25°C, RL = 32 Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT PO Output power (each channel) THD≤ 0.1% 40 (1) THD+N Total harmonic distortion + noise PO = 30 mW, 20 Hz–20 kHz 0.5% BOM Maximum output power BW G = 10, THD < 2% > 20 Phase margin Open loop 58° Supply ripple rejection f = 1 kHz 68 dB Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO = 100 mW 100 dB Vn Noise output voltage 9.5 µV(rms) (1) mW kHz Measured at 1 kHz AC OPERATING CHARACTERISTICS VDD = 5 V, TA = 25°C, RL = 32 Ω PARAMETER TEST CONDITIONS MIN TYP MAX UNIT PO Output power (each channel) THD≤ 0.1% 40 (1) THD+N Total harmonic distortion + noise PO = 60 mW, 20 Hz–20 kHz 0.4% BOM Maximum output power BW G = 10, THD < 2% > 20 Phase margin Open loop 56° Supply ripple rejection f = 1 kHz 68 dB Channel/channel output separation f = 1 kHz 86 dB SNR Signal-to-noise ratio PO = 150 mW Vn Noise output voltage (1) 4 Measured at 1 kHz mW kHz 100 dB 9.5 µV(rms) TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 TYPICAL CHARACTERISTICS Table of Graphs FIGURE THD+N 1, 2, 4, 5, 7, 8, 10, 11, 13, 14, 16, 17, 34, 36 vs Frequency Total harmonic distortion plus noise vs Output power 3, 6, 9, 12, 15, 18 Supply ripple rejection vs Frequency Output noise voltage vs Frequency 21, 22 Crosstalk vs Frequency 23-26, 37, 38 Mute attenuation vs Frequency 27, 28 Open-loop gain and phase margin vs Frequency 29, 30 Output power vs Load resistance 31, 32 Phase vs Frequency 39-44 IDD Supply current vs Supply voltage SNR Signal-to-noise ratio vs Voltage gain Closed-loop gain vs Frequency 39-44 Power dissipation/amplifier vs Output power 45, 46 Vn 19, 20 33 35 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 1 10 VDD = 3.3 V PO = 30 mW CB = 1 µ F RL = 32 Ω AV = −5 V/V AV = −10 V/V 0.1 AV = −1 V/V 0.01 0.001 20 100 1k f − Frequency − Hz Figure 1. 10k 20k THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 1 0.1 VDD = 3.3 V AV = −1 V/V RL = 32 Ω CB = 1 µF PO = 15 mW PO = 10 mW 0.01 PO = 30 mW 0.001 20 100 1k 10k 20k f − Frequency − Hz Figure 2. 5 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 3.3 V RL = 32 Ω AV = −1 V/V CB = 1 µF 20 kHz 10 kHz 1 0.1 1 kHz 20 Hz 10 AV = −10 V/V 0.1 50 AV = −5 V/V 0.01 AV = −1 V/V 0.001 20 0.01 1 1 VDD = 5 V PO = 60 mW RL = 32 Ω CB = 1 µF 100 PO − Output Power − mW Figure 4. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 5 V RL = 32 Ω AV = −1 V/V CB = 1 µF 1 PO = 30 mW PO = 15 mW 0.01 PO = 60 mW 0.001 20 100 1k f − Frequency − Hz Figure 5. 6 10k 20k Figure 3. 10 0.1 1k f − Frequency − Hz 10k 20k VDD = 5 V AV = −1 V/V RL = 32 Ω CB = 1 µF 20 kHz 1 10 kHz 0.1 1 kHz 20 Hz 0.01 0.002 0.01 0.1 PO − Output Power − W Figure 6. 0.2 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VDD = 3.3 V RL = 10 kΩ PO = 100 µF CB = 1 µF THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 1 AV = −5 V/V 0.1 0.01 AV = −2 V/V 0.001 20 100 1k VDD = 3.3 V RL = 10 kΩ AV = −1 V/V CB = 1 µF 1 0.1 PO = 45 µW 0.01 0.001 20 10k 20k 100 f − Frequency − Hz 10k 20k Figure 7. Figure 8. TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VDD = 3.3 V RL = 10 kΩ AV = −1 V/V CB = 1 µF 1 0.1 20 Hz 10 kHz 0.01 20 Hz 1 kHz 0.001 5 1k f − Frequency − Hz THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 PO = 90 µW PO = 130 µW 10 100 PO − Output Power − µW Figure 9. 200 1 VDD = 5 V RL = 10 kΩ PO = 300 µW CB = 1 µF 0.1 AV = −5 V/V AV = −1 V/V 0.01 AV = −2 V/V 0.001 20 100 1k 10k 20k f − Frequency − Hz Figure 10. 7 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER 1 10 VDD = 5 V RL = 10 kΩ AV = −1 V/V CB = 1 µF PO = 300 µW 0.1 PO = 200 µW 0.01 PO = 100 µW 0.001 20 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 5 V RL = 10 kΩ AV = −1 V/V CB = 1 µ F 1 0.1 20 Hz 20 kHz 0.01 10 kHz 1 kHz 0.001 100 1k 10k 20k 5 500 Figure 12. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VDD = 3.3 V PO = 75 mW RL = 8 Ω CB = 1 µF 1 AV = −5 V/V AV = −2 V/V 0.1 AV = −1 V/V 0.01 0.001 100 1k f − Frequency − Hz Figure 13. 10k 20k THD+N −Total Harmonic Distortion + Noise − % THD+N − Total Harmonic Distortion Plus Noise − % 100 Figure 11. 2 20 8 10 PO − Output Power − µW f − Frequency − Hz VDD = 3.3 V RL = 8 Ω AV = −1 V/V PO = 30 mW 1 PO = 15 mW 0.1 0.01 PO = 75 mW 0.001 20 100 1k f − Frequency − Hz Figure 14. 10k 20k TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY VDD = 3.3 V RL = 8 Ω AV = −1 V/V THD+N − Total Harmonic Distortion Plus Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 20 kHz 10 kHz 1 1 kHz 0.1 20 Hz 0.01 10m 0.1 0.3 2 VDD = 5 V PO = 100 mW RL = 8 Ω CB = 1 µF 1 AV = −1 V/V 0.01 0.001 20 100 10k 20k Figure 15. Figure 16. TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 1k f − Frequency − Hz 10 VDD = 5 V RL = 8 Ω AV = −1 V/V PO = 30 mW 1 PO = 60 mW 0.01 PO = 10 mW 0.001 20 AV = −5 V/V 0.1 PO − Output Power − W 0.1 AV = −2 V/V 100 1k f − Frequency − Hz Figure 17. 10k 20k VDD = 5 V RL = 8 Ω AV = −1 V/V 20 kHz 1 10 kHz 1 kHz 0.1 20 Hz 0.01 10m 0.1 1 PO − Output Power − W Figure 18. 9 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 0 VDD = 3.3 V RL = 8 Ω to 10 kΩ −10 −20 CB = 0.1 µF −30 CB = 1 µF −40 −50 −60 CB = 2 µF −70 Bypass = 1.65 V −80 −90 −100 20 VDD = 5 V RL = 8 Ω to 10 kΩ −10 Supply Ripple Rejection Ratio − dB Supply Ripple Rejection Ratio − dB 0 −20 CB = 0.1 µF −30 CB = 1 µF −40 −50 −60 CB = 2 µF −70 −80 −90 100 1k −100 20 10k 20k Bypass = 2.5 V 100 f − Frequency − Hz Figure 19. Figure 20. OUTPUT NOISE VOLTAGE vs FREQUENCY OUTPUT NOISE VOLTAGE vs FREQUENCY Vn − Output Noise Voltage − µV Vn − Output Noise Voltage − µV 10 VDD = 3.3 V BW = 10 Hz to 22 kHz AV = −1 V/V RL = 8 Ω to 10 kΩ 100 1k f − Frequency − Hz Figure 21. 10 10k 20k 20 20 1 20 1k f − Frequency − Hz 10k 20k 10 VDD = 5 V BW = 10 Hz to 22 kHz RL = 8 Ω to 10 kΩ AV = −1 V/V 1 20 100 1k f − Frequency − Hz Figure 22. 10k 20k TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY −50 −60 −70 Crosstalk − dB −75 −60 −65 −80 IN2 TO OUT1 −85 −90 −95 −70 −75 IN2 TO OUT1 −80 −85 −100 IN1 TO OUT2 −90 IN1 TO OUT2 −105 −110 20 PO = 100 mW VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = −1 V/V −55 Crosstalk − dB −65 PO = 25 mW VDD = 3.3 V RL = 32 Ω CB = 1 µF AV = −1 V/V −95 −100 100 1k 10k 20k 20 100 f − Frequency − Hz Figure 23. Figure 24. CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY −60 10k 20k −50 VDD = 5 V PO = 25 mW CB = 1 µF RL = 32 Ω AV = −1 V/V −65 −75 −80 −85 −55 −60 −65 Crosstalk − dB −65 Crosstalk − dB 1k f − Frequency − Hz IN2 TO OUT1 −90 −95 VDD = 5 V PO = 100 mW CB = 1 µF RL = 8 Ω AV = −1 V/V −70 IN2 TO OUT1 −75 −80 −85 −100 −90 IN1 TO OUT2 IN1 TO OUT2 −105 −110 20 −95 100 1k 10k 20k −100 20 100 1k f − Frequency − Hz f − Frequency − Hz Figure 25. Figure 26. 10k 20k 11 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 MUTE ATTENUATION vs FREQUENCY MUTE ATTENUATION vs FREQUENCY 0 Mute Attenuation − dB −20 −20 −30 −40 −50 −60 −70 −30 −40 −50 −60 −70 −80 −80 −90 −90 −100 20 VDD = 5 V CB = 1 µF RL = 32 Ω −10 Mute Attenuation − dB −10 0 VDD = 3.3 V RL = 32 Ω CB = 1 µF 100 1k −100 20 10k 20k 100 f − Frequency − Hz 1k f − Frequency − Hz Figure 27. Figure 28. OPEN-LOOP GAIN AND PHASE MARGIN vs FREQUENCY 150° VDD = 3.3 V TA = 25°C No Load Open-Loop Gain − dB 80 Phase 60 40 90° 60° Gain 20 30° 0 0° −20 10 100 1k 10k f − Frequency − Hz Figure 29. 12 120° 100k −30° 10M φ m − Phase Margin 100 10k 20k TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 OPEN-LOOP GAIN AND PHASE MARGIN vs FREQUENCY 100 150° VDD = 5 V TA = 25°C No Load 120° Phase 60 40 90° 60° Gain 20 30° 0 0° −20 100 1k 10k 100k φ m − Phase Margin Open-Loop Gain − dB 80 −30° 10M 1M f − Frequency − Hz Figure 30. OUTPUT POWER vs LOAD RESISTANCE OUTPUT POWER vs LOAD RESISTANCE 300 120 THD+N = 1 % VDD = 3.3 V AV = −1 V/V 250 PO − Output Power − mW PO − Output Power − mW 100 THD+N = 1 % VDD = 5 V AV = −1 V/V 80 60 40 200 150 100 50 20 0 0 8 16 24 32 40 48 RL − Load Resistance − Ω Figure 31. 56 64 8 16 24 32 40 48 56 64 RL − Load Resistance − Ω Figure 32. 13 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 SUPPLY CURRENT vs SUPPLY VOLTAGE TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY THD+N − Total Harmonic Distortion Plus Noise − % 1.4 I DD − Supply Current − mA 1.2 1 0.8 0.6 0.4 0.2 0 2.5 3 3.5 4 4.5 5 1 VI = 1 V AV = −1 V/V RL = 10 kΩ CB = 1 µF 0.1 0.01 0.001 5.5 20 100 Figure 34. SIGNAL-TO-NOISE RATIO vs VOLTAGE GAIN TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY THD+N − Total Harmonic Distortion Plus Noise − % SNR − Signal-to-Noise Ratio − dB VI = 1 V 102 100 98 96 94 92 2 3 4 5 6 7 AV − Voltage Gain − V/V Figure 35. 14 10k 20k Figure 33. 104 1 1k f − Frequency − Hz VDD − Supply Voltage − V 8 9 10 1 VDD = 5 V AV = −1 V/V RL = 10 kΩ CB = 1 µF 0.1 0.01 0.001 20 100 1k f − Frequency − Hz Figure 36. 10k 20k TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 CROSSTALK vs FREQUENCY CROSSTALK vs FREQUENCY −60 −60 −70 −80 Crosstalk − dB −90 −100 IN2 to OUT1 −110 −90 −100 −120 IN2 to OUT1 −110 −120 −130 −130 IN1 to OUT2 −140 IN1 to OUT2 −140 −150 −150 100 1k 20 10k 20k 100 f − Frequency − Hz 1k 10k 20k f − Frequency − Hz Figure 37. Figure 38. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° Phase 160° 140° Phase 20 120° Closed-Loop Gain − dB Crosstalk − dB −80 VDD = 5 V VO = 1 V RL = 10 kΩ CB = 1 µF −70 VDD = 3.3 V VO = 1 V RL = 10 kΩ CB = 1 µF VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V 30 20 10 100° 80° Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 39. 15 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° 160° 140° Phase Phase Closed-Loop Gain − dB 120° VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 32 Ω CI = 1 µF AV = −1 V/V 30 100° 80° 20 10 Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 40. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° Phase 140° Closed-Loop Gain − dB 120° VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V 40 80° 60° Gain 20 0 −20 10 100 1k 10k f − Frequency − Hz Figure 41. 16 100° 100k 1M Phase 160° TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 160° 140° Phase 180° Phase Closed-Loop Gain − dB 120° VDD = 3.3 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V 30 20 10 100° 80° Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 42. CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 180° Phase Closed-Loop Gain − dB 140° VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 8 Ω CI = 1 µF AV = −1 V/V 120° Phase 160° 100° 80° 60° 40° Gain 20 0 −20 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 43. 17 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 200° 160° Closed-Loop Gain − dB 140° 120° VDD = 5 V RI = 20 kΩ RF = 20 kΩ RL = 10 kΩ CI = 1 µF AV = −1 V/V 30 Phase 180° Phase 100° 80° 20 10 Gain 0 −10 10 100 1k 10k 100k 1M f − Frequency − Hz Figure 44. POWER DISSIPATION/AMPLIFIER vs OUTPUT POWER POWER DISSIPATION/AMPLIFIER vs OUTPUT POWER 80 180 VDD = 3.3 V VDD = 5 V 8Ω 70 140 Amplifier Power − mW Amplifier Power − mW 60 50 40 16 Ω 30 32 Ω 20 120 100 16 Ω 80 60 32 Ω 40 64 Ω 10 64 Ω 20 0 0 0 20 40 60 80 100 120 140 160 180 Load Power − mW Figure 45. 18 8Ω 160 200 0 20 40 60 80 100 120 140 160 180 Load Power − mW Figure 46. 200 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 APPLICATION INFORMATION GAIN SETTING RESISTORS, RF and RI The gain for the TPA122 is set by resistors RF and RI according to Equation 1. Gain 
  RF RI (1) Given that the TPA122 is an MOS amplifier, the input impedance is high. Consequently, input leakage currents are not generally a concern, although noise in the circuit increases as the value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together, it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in Equation 2. R FR I Effective Impedance  RF
RI (2) As an example, consider an input resistance of 20 kΩ and a feedback resistor of 20 kΩ. The gain of the amplifier would be –1 and the effective impedance at the inverting terminal would be 10 kΩ, which is within the recommended range. For high-performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately 5 pF should be placed in parallel with RF. In effect, this creates a low-pass filter network with the cutoff frequency defined in Equation 3. 1 f c(lowpass)
2
R F CF (3) For example, if RF is 100 kΩ and CF is 5 pF, then fc(lowpass) is 318 kHz, which is well outside the audio range. INPUT CAPACITOR CI In the typical application, an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency determined in Equation 4. 1 f c(highpass)
2
R I CI (4) The value of CI is important to consider, as it directly affects the bass (low-frequency) performance of the circuit. Consider the example where RI is 20 kΩ and the specification calls for a flat bass response down to 20 Hz. Equation 4 is reconfigured as Equation 5. 1 CI
2
R I f c(highpass) (5) In this example, CI is 0.4 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high-gain applications (> 10). For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at VDD/2, which is likely higher than the source dc level. Note that it is important to confirm the capacitor polarity in the application. 19 TPA122 www.ti.com SLOS211E – AUGUST 1998 – REVISED JUNE 2004 APPLICATION INFORMATION (continued) POWER SUPPLY DECOUPLING, CS The TPA122 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF, placed as close as possible to the device VDD lead, works best. For filtering lower frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the power amplifier is recommended. MIDRAIL BYPASS CAPACITOR, CB The midrail bypass capacitor, CB, serves several important functions. During start-up, CB determines the rate at which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so low it can not be heard). The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capacitor is fed from a 160-kΩ source inside the amplifier. To keep the start-up pop as low as possible, the relationship shown in Equation 6 should be maintained. 1  1 C B
160 kΩ C I R I (6) As an example, consider a circuit where CB is 1 µF, CI is 1 µF, and RI is 20 kΩ. Inserting these values into Equation 6 results in: 6.25 ≤ 50 which satisfies the rule. Bypass capacitor, CB, values of 0.1-µF to 1-µF ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance. OUTPUT COUPLING CAPACITOR, CC In the typical single-supply, single-ended (SE) configuration, an output coupling capacitor (CC) is required to block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by Equation 7. 1 fc
2
R L CC (7) The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the low-frequency corner higher. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 68 µF is chosen and loads vary from 32 Ω to 47 kΩ. Table 1 summarizes the frequency response characteristics of each configuration. Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode RL CC LOWEST FREQUENCY 32 Ω 68 µF 73 Hz 10,000 Ω 68 µF 0.23 Hz 47,000 Ω 68 µF 0.05 Hz As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for example) is good. The output coupling capacitor required in single-supply, SE mode also places additional constraints on the selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following relationship: 20 TPA122 www.ti.com 1 C B
160 kΩ SLOS211E – AUGUST 1998 – REVISED JUNE 2004  1 C I R I  1 R LC C (8) USING LOW-ESR CAPACITORS Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor behaves like an ideal capacitor. 5-V VERSUS 3.3-V OPERATION The TPA122 was designed for operation over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V and 3.3-V operation because these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in the TPA122 can produce a maximum voltage swing of VDD – 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V, as opposed to VO(PP) = 4 V for 5-V operation. The reduced voltage swing subsequently reduces maximum output power into the load before distortion begins to become significant. 21 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) TPA122D ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 TPA122 TPA122DGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 AAE TPA122DGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 AAE TPA122DR ACTIVE SOIC D 8 2500 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 TPA122 (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of