TPA3002D2PHP

 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004  
        
   
 FEATURES D 9-W/Ch Into an 8-Ω Load From 12-V Supply D Efficient, Class-D Operation Eliminates D D D D D DESCRIPTION The TPA3002D2 is a 9-W (per channel) efficient, Class-D audio amplifier for driving bridged-tied stereo speakers. The TPA3002D2 can drive stereo speakers as low as 8 Ω. The high efficiency of the TPA3002D2 eliminates the need for external heatsinks when playing music. Heatsinks and Reduces Power Supply Requirements 32-Step DC Volume Control From −40 dB to 36 dB Line Outputs For External Headphone Amplifier With Volume Control Regulated 5-V Supply Output for Powering TPA6110A2 Space-Saving, Thermally-Enhanced PowerPAD Packaging Thermal and Short-Circuit Protection Stereo speaker volume is controlled with a dc voltage applied to the volume control terminal offering a range of gain from –40 dB to 36 dB. Line outputs, for driving external headphone amplifier inputs, are also dc voltage controlled with a range of gain from –56 dB to 20 dB. An integrated 5-V regulated supply is provided for powering an external headphone amplifier. APPLICATIONS D LCD Monitors and TVs D Powered Speakers 10 µF Cs 0.1 µF Cs 0.1 µF RINN RINP Crinn Crinp 1 µF 1 µF Clinp LINP LINN 1 µF C2p5 1 µF Clinn 10 nF Cbs VREF Ccpr VCLAMPR MODE_OUT RINN RINP MODE V2P5 AVCC LINP 1 µF MODE_OUT MODE VAROUTR LINN 1 µF BSRP PVCCR PVCCR ROUTN PGNDR Cs PGNDR PVCCR SD ROUTN BSRN SDZ PVCCR Cs PVCC ROUTP Cbs 10 µF ROUTP PVCC 10 nF RLINE_OUT VAROUTL TPA3002D2 AVDDREF AGND VREF AVDD VARDIFF VARDIFF COSC VARMAX VARMAX ROSC Cs Cbs 10 nF PVCC BSLP PVCCL PVCCL LOUTP LOUTP PGNDL PGNDL LOUTN LOUTN VCLAMPL PVCCL AGND REFGND PVCCL VOLUME BSLN VOL REFGND AVCC Cs 0.1 µF Cvcc 10 µF LLINE_OUT AVDD Cvdd Cosc 100 nF 220 pF Rosc 120 kΩ Ccpl 1 µF 10 kΩ 10 kΩ Cs 0.1 µF Cs 0.1 µF 10 µF 10 µF Cs Cbs 10 nF PVCC Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments.  
 
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 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam during storage or handling to prevent electrostatic damage to the MOS gates. AVAILABLE OPTIONS PACKAGED DEVICE TA 48-PIN HTQFP (PHP)(1) −40°C to 85°C TPA3002D2PHP (1) The PHP package is available taped and reeled. To order a taped and reeled part, add the suffix R to the part number (e.g., TPA3002D2PHPR). PIN ASSIGNMENTS PHP PACKAGE BSRP PVCCR ROUTP ROUTP PGNDR ROUTN ROUTN PGNDR 43 42 41 40 39 38 37 SD 1 36 VCLAMPR RINN 2 35 MODE_OUT RINP 3 34 MODE V2P5 4 33 AVCC LINP 5 32 VAROUTR LINN 6 31 VAROUTL AVDDREF 7 30 AGND VREF 8 29 AVDD VARDIFF 9 28 COSC VARMAX 10 27 ROSC VOLUME 11 26 AGND REFGND 12 25 VCLAMPL BSLP PVCCL LOUTP PVCCL PGNDL LOUTP 24 PGNDL 20 21 22 23 LOUTN 18 19 LOUTN PVCCL BSLN 15 16 17 PVCCL TPA3002D2 13 14 2 46 45 44 PVCCR 48 47 PVCCR PVCCR BSRN (TOP VIEW) 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 FUNCTIONAL BLOCK DIAGRAM V2P5 PVCC V2P5 VAROUTR VClamp Gen V2P5 VCLAMPR BSRN Gain Adj. PVCCR(2) Gate Drive Cint2 RINN Gain Adj. Rfdbk2 RINP Deglitch & PGNDR Modulation Logic BSRP PVCCR(2) Rfdbk2 V2P5 VREF Gate Drive Cint2 VOLUME Gain Control VARDIFF VARMAX PGNDR Blocks OC Detect V2P5 ROSC Ramp Generator Biases Startup Protection Logic & COSC References AVDDREF ROUTP(2) To Gain Adj. REFGND Thermal VDD VDDok AVCC AVDD VCCok AVDD 5V LDO PVCC TTL Input Buffer SD MODE MODE_OUT AVCC AGND VClamp Gen VCLAMPL BSLN Mode PVCCL(2) Control Gate Drive Cint2 V2P5 LINN ROUTN(2) PGNDL Deglitch & Gain Adj. Rfdbk2 BSLP Modulation Logic LINP LOUTN(2) PVCCL(2) Rfdbk2 V2P5 Cint2 Gain Adj. Gate Drive LOUTP(2) PGNDL VAROUTL 3 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Terminal Functions TERMINAL NO. NAME I/O DESCRIPTION AGND 26, 30 − Analog ground for digital/analog cells in core AVCC AVDD 33 − High-voltage analog power supply (8.5 V to 14 V) 29 O 5-V Regulated output capable of 100-mA output AVDDREF BSLN 7 O 5-V Reference output—provided for connection to adjacent VREF terminal. 13 I/O Bootstrap I/O for left channel, negative high-side FET BSLP 24 I/O Bootstrap I/O for left channel, positive high-side FET BSRN 48 I/O Bootstrap I/O for right channel, negative high-side FET BSRP 37 I/O Bootstrap I/O for right channel, positive high-side FET COSC 28 I/O I/O for charge/discharging currents onto capacitor for ramp generator triangle wave biased at V2P5 LINN 6 I Negative differential audio input for left channel LINP 5 I Positive differential audio input for left channel LOUTN 16, 17 O Class-D 1/2-H-bridge negative output for left channel LOUTP 20, 21 O Class-D 1/2-H-bridge positive output for left channel MODE 34 I Input for MODE control. A logic high on this pin places the amplifier in the variable output mode and the Class-D outputs are disabled. A logic low on this pin places the amplifier in the Class-D mode and Class-D stereo outputs are enabled. Variable outputs (VAROUTL and VAROUTR) are still enabled in Class-D mode to be used as line-level outputs for external amplifiers. MODE_OUT 35 O Output for control of the variable output amplifiers. When the MODE pin (34) is a logic high, the MODE_OUT pin is driven low. When the MODE pin (34) is a logic low, the MODE_OUT pin is driven high. This pin is intended for MUTE control of an external headphone amplifier. Leave unconnected when not used for headphone amplifier control. PGNDL 18, 19 − Power ground for left channel H-bridge PGNDR 42, 43 − Power ground for right channel H-bridge PVCCL 14, 15 − Power supply for left channel H-bridge (tied to pins 22 and 23 internally), not connected to PVCCR or AVCC. PVCCL 22, 23 − Power supply for left channel H-bridge (tied to pins 14 and 15 internally), not connected to PVCCR or AVCC. PVCCR 38,39 − Power supply for right channel H-bridge (tied to pins 46 and 47 internally), not connected to PVCCL or AVCC. PVCCR 46, 47 − Power supply for right channel H-bridge (tied to pins 38 and 39 internally), not connected to PVCCL or AVCC. REFGND 12 − Ground for gain control circuitry. Connect to AGND. If using a DAC to control the volume, connect the DAC ground to this terminal. RINP 3 I Positive differential audio input for right channel RINN 2 I Negative differential audio input for right channel ROSC 27 I/O Current setting resistor for ramp generator. Nominally equal to 1/8*VCC ROUTN 44, 45 O Class-D 1/2-H-bridge negative output for right channel ROUTP 40, 41 O Class-D 1/2-H-bridge positive output for right channel SD 1 I Shutdown signal for IC (low = shutdown, high = operational). TTL logic levels with compliance to VCC. VARDIFF 9 I DC voltage to set the difference in gain between the Class-D and VAROUT outputs. Connect to GND or AVDDREF if VAROUT outputs are unconnected. VARMAX 10 I DC voltage that sets the maximum gain for the VAROUT outputs. Connect to GND or AVDDREF if VAROUT outputs are unconnected. VAROUTL 31 O Variable output for left channel audio. Line level output for driving external HP amplifier. VAROUTR 32 O Variable output for right channel audio. Line level output for driving external HP amplifier. VCLAMPL 25 − Internally generated voltage supply for left channel bootstrap capacitors. VCLAMPR 36 − Internally generated voltage supply for right channel bootstrap capacitors. VOLUME 11 I DC voltage that sets the gain of the Class-D and VAROUT outputs. VREF 8 I Analog reference for gain control section. V2P5 4 O 2.5-V Reference for analog cells, as well as reference for unused audio input when using single-ended inputs. Thermal Pad − Connect to AGND and PGND—should be center point for both grounds. — 4 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range unless otherwise noted(1) UNIT Supply voltage range: AVCC, PVCC MODE, VREF, VARDIFF, VARMAX, VOLUME Input voltage range, VI SD −0.3 V to 15 V 0 V to 5.5 V −0.3 V to VCC + 0.3 V RINN, RINP, LINN, LINP −0.3 V to 7 V AVDD AVDDREF 120 mA Supply current Output current, VAROUTL, VAROUTR 20 mA 10 mA Continuous total power dissipation See Dissipation Rating Table Operating free-air temperature range, TA −40°C to 85°C Operating junction temperature range, T −40°C to 150°C Storage temperature range, Tstg −65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds 260°C (1) Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. PACKAGE DISSIPATION RATINGS PACKAGE PHP TA ≤ 25°C 4.3 W DERATING FACTOR 34.7 mW/°C(1) TA = 70°C 2.7 W TA = 85°C 2.2 W (1) The PowerPAD must be soldered to a thermal land on the printed circuit board. Please refer to the PowerPAD Thermally Enhanced Package application note (SLMA002). RECOMMENDED OPERATING CONDITIONS Supply voltage, VCC Volume reference voltage PVCC, AVCC VREF Volume control pins, input voltage VARDIFF, VARMAX, VOLUME SD High-level input voltage, VIH MODE MIN MAX UNIT 8.5 14 V 3.0 5.5 V 5.5 V 2 SD Low-level input voltage, VIL MODE High-level output voltage, VOH MODE_OUT, IOH = 1 mA Low-level output voltage, VOL MODE_OUT, IOL = −1 mA High-level input current, IIH Low-level input current, IIL V 3.5 0.8 2 AVDD−100mV MODE, VI= 5 V, VCC = 14 V SD, VI= 14 V, VCC = 14 V MODE, VI= 0 V, VCC = 14 V SD, VI= 0 V, VCC = 14 V V V AGND+100mV V 1 uA 30 uA 1 uA 1 uA Oscillator frequency, fOSC 225 275 kHz Operating free-air temperature, TA −40 85 °C 5 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 DC ELECTRICAL CHARACTERISTICS TA = 25°C, VCC = 12 V, RL = 8 Ω (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT | VOS | Class-D Output offset voltage (measured differentially) INN and INP connected together, Gain = 36 dB V2P5 (terminal 4) 2.5-V Bias voltage No load AVDD 5-V Regulated output PSRR Class-D power supply rejection ratio IO = 0 to 100 mA, SD = 2 V, VCC = 8 V to 14 V VCC = 11.5 V to 12.5 V ICC(class-D) ICC(varout) Class-D mode quiescent current MODE = 2 V, SD = 2 V 16 28.5 mA Variable output mode quiescent current MODE = 3.5 V, SD = 2 V 7 9 mA ICC(class-D – max power) Class-D mode RMS current at max power RL = 8 Ω, PO = 9 W 2 ICC(SD) Supply current in shutdown mode SD = 0.8 V 1 rds(on) Drain-source on-state resistance VCC = 12 V, IO = 1 A, TJ = 25°C 25 C 10 0.45x AVDD 4.5 65 0.5x 0.55x AVDD AVDD 5.0 5.5 −80 mV V V dB A ÑÑÑÑÑÑÑÑ ÑÑÑÑÑÑÑÑÑÑÑ ÑÑÑÑÑ ÑÑÑÑÑ ÑÑÑ ÑÑÑ ÑÑÑ ÑÑÑ ÑÑÑÑÑÑÑÑ ÑÑÑÑÑÑÑÑÑÑÑ ÑÑÑÑÑ ÑÑÑÑÑ ÑÑÑ ÑÑÑ ÑÑÑ ÑÑÑ ÑÑÑÑÑÑÑÑ ÑÑÑÑÑÑÑÑÑÑÑ ÑÑÑÑÑ ÑÑÑÑÑ ÑÑÑ Ñ ÑÑ ÑÑÑ ÑÑÑ ÑÑÑ High side 300 Low side 250 Total 550 10 uA mΩ 590 AC ELECTRICAL CHARACTERISTICS FOR CLASS-D OUTPUTS TA = 25°C, VCC = 12 V, RL = 8 Ω (unless otherwise noted) PARAMETER kSVR Supply ripple rejection ratio PO Continuous output power TEST CONDITIONS MIN SNR MAX UNITS VCC = 11.5 V to 12.5 V from 10 Hz to 1 kHz, Gain = 36 dB −67 dB THD+N = 1%, f = 1 kHz, RL = 8 Ω 7.5 W 9 W THD+N = 10%, f = 1 kHz, RL = 8 Ω 79 µV −82 dBV 20 Hz to 22 kHz, A-weighted filter, Gain = 13.2 dB 100 µV −80 dBV Crosstalk, Class-D−Left → Class-D−Right Gain = 13.2 dB, PO = 1 W, RL = 8 Ω −77 dB Crosstalk, Class-D → VAROUT Maximum output at THD < 0.5%, Gain = 36 dB −63 dB Signal-to-noise ratio Maximum output at THD+N < 0.5%, f= 1 kHz, Gain = 36 dB 96 dB Thermal trip point 150 °C Thermal hystersis 20 °C 20 Hz to 22 kHz, No filter, Gain = 0.5 dB Vn TYP Output integrated noise floor CHARACTERISTICS FOR VAROUT OUTPUTS PARAMETER TEST CONDITIONS Measured between V2P5 and VAROUT, Gain = 20 dB, RL = 10 kΩ MIN TYP UNITS |VOS| Output offset voltage THD+N Total harmonic distortion + noise PSRR DC power supply rejection ratio AV = 7.3 dB, f = 1 kHz, PO = 6 mW, RL = 32 Ω AV = 7.3 dB, f = 1 kHz, RL = 2 kΩ, VO = 1 Vrms Gain = 20 dB −74 dB kSVR Supply ripple rejection ratio Gain = 20 dB, f = 1 kHz −95 dB Crosstalk, VAROUTL → VAROUTR Maximum output at THD < 0.5%, Gain = 20 dB −60 dB Crosstalk, VAROUT → Class-D Maximum output at THD < 0.5%, Gain = 20 dB −74 dB Vn 6 Output integrated noise floor 10 MAX mV 0.025% 0.002% 20 Hz to 22 kHz, Gain = 20 dB 75 20 Hz to 22 kHz, Gain = −0.3 dB 15 µV V 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Table 1. DC Volume Control for Class-D Outputs VOLTAGE ON THE VOLUME PIN AS A PERCENTAGE OF VREF (INCREASING VOLUME OR FIXED GAIN) VOLTAGE ON THE VOLUME PIN AS A PERCENTAGE OF VREF (DECREASING VOLUME) GAIN OF CLASS-D AMPLIFIER % % dB 0 − 4.5 0 − 2.9 −75(1) 4.5 − 6.7 2.9 − 5.1 −40.0 6.7 − 8.91 5.1 − 7.2 −37.5 8.9 − 11.1 7.2 − 9.4 −35.0 11.1 − 13.3 9.4 − 11.6 −32.4 13.3 − 15.5 11.6 − 13.8 −29.9 15.5 − 17.7 13.8 − 16.0 −27.4 17.7 − 19.9 16.0 − 18.2 −24.8 19.9 − 22.1 18.2 − 20.4 −22.3 22.1 − 24.3 20.4 − 22.6 −19.8 24.3 − 26.5 22.6 − 24.8 −17.2 26.5 − 28.7 24.8 − 27.0 −14.7 28.7 − 30.9 27.0 − 29.1 −12.2 30.9 − 33.1 29.1 − 31.3 −9.6 33.1 − 35.3 31.3 − 33.5 −7.1 35.3 − 37.5 33.5 − 35.7 −4.6 37.5 − 39.7 35.7 − 37.9 39.7 − 41.9 37.9 − 40.1 −2.0 0.5† 41.9 − 44.1 40.1 − 42.3 3.1 44.1 − 46.4 42.3 − 44.5 5.6 46.4 − 48.6 44.5 − 46.7 8.1 48.6 − 50.8 46.7 − 48.9 10.7 50.8 − 53.0 48.9 − 51.0 13.2 53.0 − 55.2 51.0 − 53.2 15.7 55.2 − 57.4 53.2 − 55.4 18.3 57.4 − 59.6 55.4 − 57.6 20.8 59.6 − 61.8 57.6 − 59.8 23.3 61.8 − 64.0 59.8 − 62.0 25.9 64.0 − 66.2 62.0 − 64.2 28.4 66.2 − 68.4 64.2 − 66.4 30.9 68.4 − 70.6 66.4 − 68.6 > 70.6 >68.6 33.5 36.0(1) (1) Tested in production. Remaining steps are specified by design. 7 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Table 2. DC Volume Control for VAROUT Outputs VAROUT_VOLUME (V) − FROM FIGURE 35 − AS A PERCENTAGE OF VREF (INCREASING VOLUME OR FIXED GAIN) VAROUT_VOLUME (V) − FROM FIGURE 35 − AS A PERCENTAGE OF VREF (DECREASING VOLUME) GAIN OF VAROUT AMPLIFIER % % dB 0 − 4.5 0 − 2.9 −66(1) 4.5 − 6.7 2.9 − 5.1 −56.0 6.7 − 8.91 5.1 − 7.2 −53.5 8.9 − 11.1 7.2 − 9.4 −50.9 11.1 − 13.3 9.4 − 11.6 −48.4 13.3 − 15.5 11.6 − 13.8 −45.9 15.5 − 17.7 13.8 − 16.0 −43.3 17.7 − 19.9 16.0 − 18.2 −40.8 19.9 − 22.1 18.2 − 20.4 −38.3 22.1 − 24.3 20.4 − 22.6 −35.7 24.3 − 26.5 22.6 − 24.8 −33.2 26.5 − 28.7 24.8 − 27.0 −30.7 28.7 − 30.9 27.0 − 29.1 −28.1 30.9 − 33.1 29.1 − 31.3 −25.6 33.1 − 35.3 31.3 − 33.5 −23.1 35.3 − 37.5 33.5 − 35.7 −20.5 37.5 − 39.7 35.7 − 37.9 −18.0 39.7 − 41.9 37.9 − 40.1 41.9 − 44.1 40.1 − 42.3 −15.5 −13.0(1) 44.1 − 46.4 42.3 − 44.5 −10.4 46.4 − 48.6 44.5 − 46.7 −7.9 48.6 − 50.8 46.7 − 48.9 −5.3 50.8 − 53.0 48.9 − 51.0 −2.8 53.0 − 55.2 51.0 − 53.2 −0.3 55.2 − 57.4 53.2 − 55.4 2.3 57.4 − 59.6 55.4 − 57.6 4.8 59.6 − 61.8 57.6 − 59.8 7.3 61.8 − 64.0 59.8 − 62.0 9.9 64.0 − 66.2 62.0 − 64.2 12.4 66.2 − 68.4 64.2 − 66.4 14.9 68.4 − 70.6 66.4 − 68.6 17.5 20.0(1) > 70.6 >68.6 (1) Tested in production. Remaining steps are specified by design. 8 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE Class-D Efficiency vs Output power 1 PO Class-D Output power vs Load resistance 2 vs Supply voltage 3 ICC Class-D Supply current vs Supply voltage 4 vs Output Power 5 Shutdown supply current vs Supply voltage 6 Class-D Input resistance vs Gain IO(sd) vs Frequency THD+N Class-D Total harmonic distortion + noise kSVR Class-D Supply ripple rejection ratio vs Output power vs Frequency Class-D Closed loop response 12 14 Class-D Input offset voltage vs Common-mode input voltage 15 Class-D Crosstalk vs Frequency 16 Class-D Mute attenuation Class-D Shutdown attenuation 17 vs Frequency 18 Class-D Common-mode rejection ratio vs Frequency 19 VAROUT Input resistance vs Gain 20 VAROUT Noise vs Frequency 21 VAROUT Closed Loop Response kSVR 10, 11 13 Class-D Intermodulation performance THD+N 7 8, 9 22 VAROUT Common-mode rejection ratio vs Frequency 23 VAROUT Crosstalk vs Frequency 24 vs Output power 25 vs Output voltage 26 vs Frequency 27 vs Frequency 28 VAROUT Total harmonic distortion + noise VAROUT Supply ripple rejection ratio 9 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 EFFICIENCY vs OUTPUT POWER OUTPUT POWER vs LOAD RESISTANCE 100 16 RL = 8 Ω 90 f = 1 kHz, LC Filter, Class-D, Resistive Load, TA = 25°C 14 80 12 PO − Output Power − W Efficiency − % 70 60 50 40 30 VCC = 12 V, Class-D, LC Filter, Resistive Load 20 10 2 4 6 PO − Output Power − W 8 VCC = 12 V, THD = 1% 10 8 6 4 2 0 0 VCC = 12 V, THD = 10% VCC = 8.5 V, THD = 10% 0 10 8 10 12 14 RL − Load Resistance − Ω Figure 1 16 Figure 2 OUTPUT POWER vs SUPPLY VOLTAGE SUPPLY CURRENT vs SUPPLY VOLTAGE 17 14 16 I CC− Supply Current − mA 12 PO − Output Power − W VCC = 8.5 V, THD = 1% 10 8 Ω Speaker 10% THD+N 8 6 8 Ω Speaker 1% THD+N 4 SD = 2 V, MODE = 2 V, Class-D, No Load 15 14 13 12 11 TA = 25°C 2 8.5 9 10 11 12 VCC − Supply Voltage − V Figure 3 10 13 14 10 8.5 9 10 11 12 VCC − Supply Voltage − V Figure 4 13 14 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 SUPPLY CURRENT vs OUTPUT POWER SHUTDOWN SUPPLY CURRENT vs SUPPLY VOLTAGE 2.5 2.2 I CC − Supply Current − A 2.0 I CC(sd)− Shutdown Supply Current − µ A VCC = 12 V, MODE = 2 V, Class-D, Stereo, TA = 25°C 1.5 8Ω 1.0 16 Ω 0.5 0 0 5 10 15 SD = 0 V No Load 2 1.8 1.6 1.4 1.2 1 0.8 0.6 0.4 8.5 20 PO − Output Power − W 13 14 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY INPUT RESISTANCE vs GAIN 1 THD+N − Total Harmonic Distortion + Noise − % 120 Class-D 100 RL − Input Resistance − k Ω 12 10 11 VCC − Supply Voltage − V Figure 6 Figure 5 80 60 40 20 0 −50 9 −30 −10 10 Gain − dB Figure 7 30 50 VCC = 8 V RL = 8 Ω Gain = +36 dB Class-D PO = 3 W 0.1 PO = 0.25 W PO = 1.5 W 0.01 20 100 1k 10k f − Frequency − Hz Figure 8 11 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 10 VCC = 12 V RL = 8 Ω Gain = +36 dB Class-D PO = 0.5 W PO = 5 W 0.1 THD+N − Total Harmonic Distortion + Noise − % THD+N − Total Harmonic Distortion + Noise − % 1 PO = 2.5 W 0.01 10 100 1k 10k VCC = 8 V RL = 8 Ω Gain = +13.2 dB Class-D 1 f = 1 kHz 0.1 0.01 10 m f = 20 Hz 100 m Figure 9 SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY −40 VCC = 12 V RL = 8 Ω Gain = +13.2 dB Class-D 1 f = 20 Hz f = 1 kHz 0.1 1 PO − Output Power − W Figure 11 12 k SVR − Supply Ripple Rejection Ratio − dB THD+N − Total Harmonic Distortion + Noise − % 10 100 m 10 Figure 10 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER 0.01 10 m 1 PO − Output Power − W f − Frequency − Hz 10 −45 RL = 8 Ω, C2P5 = 1 µF, Class-D −50 −55 VCC = 8 V −60 −65 VCC = 12 V −70 −75 −80 20 100 1k f − Frequency − Hz Figure 12 10 k 20 k 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 INTERMODULATION PERFORMANCE CLOSED LOOP RESPONSE 100 Gain 50 −20 50 0 −40 −50 −100 −100 −150 −150 VCC = 12 V, Gain = +36 dB, RL = 8 Ω Class-D −200 −250 10 100 FFT − dBr −50 Phase − Deg 0 Phase Gain − dB 0 100 100 k −60 −80 −100 −120 −200 1k 10 k f − Frequency − Hz VCC = 12 V, 19 kHz, 20 kHz, 1:1, PO = 1 W, RL = 8 Ω Gain = +13.2 dB, BW =20 Hz to 22 kHz, Class-D No Filter −140 50 −250 1M 100 1k f − Frequency − Hz Figure 13 Figure 14 INPUT OFFSET VOLTAGE vs COMMON-MODE INPUT VOLTAGE CROSSTALK vs FREQUENCY 6 0 VCC = 12 V Class-D 5 −10 −20 4 Crosstalk − dB VIO − Input Offset Voltage − mV 10 k 3 2 −30 VCC = 12 V, C2P5 = 1 µF, PO = 1 W, Gain = +13.2 dB, Class-D, RL = 8 Ω −40 −50 −60 1 −70 0 −80 −1 0 1 2 3 4 VICM − Common-Mode Input Voltage − V Figure 15 5 −90 20 100 1k 10 k 20 k f − Frequency − Hz Figure 16 13 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 MUTE ATTENUATION vs FREQUENCY SHUTDOWN ATTENUATION vs FREQUENCY −30 Mute Attenuation − dB −50 −85 Shutdown Attenuation − dB −40 −80 VCC = 12 V, RL = 8 Ω, VI = 1 Vrms Class-D, VOLUME = 0 V −60 −70 −80 −90 −100 −90 −95 −100 −105 −110 −115 −110 −120 −120 −125 −130 10 100 1k VCC = 12 V, RL = 8 Ω, VI = 1 Vrms Gain = +13.2 dB, Class-D −130 10 10 k 100 f − Frequency − Hz f − Frequency − Hz Figure 17 Figure 18 INPUT RESISTANCE vs GAIN COMMON-MODE REJECTION RATIO vs FREQUENCY 160 VAROUT VCC = 12 V, RL = 8 Ω, C2P5 = 1 µF, Class-D 140 RL − Input Resistance − k Ω CMRR − Common-Mode Rejection Ratio − dB −40 −50 −60 −70 −80 −90 −100 10 120 100 80 60 40 20 0 −50 100 10 k 1k f − Frequency − Hz Figure 19 14 10 k 1k 100 k −30 −10 Gain − dB Figure 20 10 30 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 NOISE vs FREQUENCY CLOSED LOOP RESPONSE 0 12.9 VCC = 12 V, Gain = +20 dB, RL = 8 Ω Inputs AC Coupled to GND, VAROUT, No Filter −40 150 125 Gain − dB −80 −100 −120 −140 5.8 100 2.2 75 −1.3 50 25 Phase −4.8 0 −8.4 −25 −50 −11.9 −75 VCC = 12 V, Gain = +7.9 dB, RL = 8 Ω VAROUT −160 −15.4 −180 −19.0 −200 −22.5 10 20 100 1k f − Frequency − Hz 10 k 100 −175 10 k CROSSTALK (VAROUTL-TO-VAROUTR) vs FREQUENCY 0 VCC = 12 V, RL = 8 Ω , C2P5 = 1 µF, VAROUT VO = 1 Vrms, RL = 10 kΩ, VAROUT −10 −20 −46 G = 20 dB G = 10 dB G = 0 dB Crosstalk − dB −30 −48 −50 −52 −40 G = −10 dB −50 −60 −54 −70 −56 −80 −58 −90 −60 20 −150 Figure 22 −40 −44 −125 f − Frequency − Hz COMMON-MODE REJECTION RATIO vs FREQUENCY −42 −100 1k Figure 21 CMRR − Common-Mode Rejection Ratio − dBv Noise FFT − dBV −60 9.3 Phase − Deg −20 175 Gain −100 100 1k f− Frequency − Hz Figure 23 10 k 20 100 1k 10 k f − Frequency − Hz Figure 24 15 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT POWER TOTAL HARMONIC DISTORTION + NOISE vs OUTPUT VOLTAGE VCC = 12 V RL = 32 Ω, Gain = +6 dB, VAROUT 10 2 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 20 1 0.2 f = 1 kHz 0.1 f = 20 kHz 0.02 0.01 f = 20 Hz 0.001 20 µ 100 µ 200 µ 1m 2m 10 m 20 m 20 10 2 VCC = 12 V RL = 10 kΩ, Gain = +6 dB VAROUT 1 0.2 0.1 f = 1 kHz 0.02 0.01 0.001 20 m 100 m PO − Output Power − W Figure 25 k SVR − Supply Ripple Rejection Ratio − dB THD+N −Total Harmonic Distortion + Noise − % 0.1 0.02 0.01 100 1k f − Frequency − Hz Figure 27 16 −40 VCC = 12 V RL = 32 Ω, PO = 5 mW, Gain = +7.9 dB, VAROUT 0.2 0.005 20 2 SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 10 1 1 Figure 26 TOTAL HARMONIC DISTORTION + NOISE vs FREQUENCY 2 100 m VO − Output Voltage − VRMS 10 k −50 VCC = 12 V VAROUT −60 −70 −80 −90 −100 −110 20 100 1k f − frequency − Hz Figure 28 10 k 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 C22 1 nF VCC ROUT+ GND VCC ROUT− APPLICATION INFORMATION C23 1 nF L1 (Bead) L2 (Bead) 10 µF PGND 10 nF C15 0.1uF 0.1uF C9 C10 10 nF 1 µF 1 µF C5 C3 1 µF 1 µF C4 LIN− 1 µF MODEB BSRP PVCCR ROUTP ROUTP PGNDR PGNDR ROUTN ROUTN MODE V2P5 AVCC LINP VAROUTR VAROUTR LINN VAROUTL VAROUTL TPA3002D2 AVDDREF AGND VREF AVDD VARDIFF COSC VARMAX ROSC T6 T5 VOLUME AGND REFGND VCLAMPL MODE C13 0.1 µF C16 10 µF C11 AVDD 220pF 100 nF 10 µF R1 120 kΩ AGND BSLP PVCCL PVCCL LOUTP 10 nF PGND L3 (Bead) L4 (Bead) C25 1nF GND C24 1nF VCC C21 0.1 µF 10 µF LOUT− PGND VCC 10 nF GND 1 µF C12 0.1 µF C17 LOUT+ C20 LOUTP PGNDL PGNDL LOUTN LOUTN PVCCL AGND VCC C14 C6 C8 GND PVCCL P1 50 kΩ GND RINP T7 P2 50 kΩ PGND 1 µF MODE_OUT BSLN P3 50k C7 VCLAMPR RINN C2 AGND PVCCR SD C1 RIN− PVCCR BSRN SHUTDOWN PVCCR C19 C18 Figure 29. Stereo Class-D With Single-Ended Inputs 17 18 AGND REFGND VOLUME VARMAX VARDIFF VREF AVDDREF LINN BSRN T5 1 µF LINP V2P5 PGNDL LOUTN LOUT− VCC PVCCR PVCCL VCC PVCCR PVCCL 10 nF LOUTN C24 1 nF L3 (Bead) C25 1 nF L4 (Bead) PGND PGNDL 10 µ F 0.1 µ F C12 LOUTP C11 VCLAMPR VCLAMPL AGND ROSC COSC AVDD AGND VAROUTL VAROUTR AVCC MODE MODE_OUT 10 nF C21 PVCCL 0.1 µ F C17 C19 PVCCR C20 C10 10 nF VCC PVCCR PVCCL VCC ROUT+ ROUTP LOUTP LOUT+ TPA3002D2 ROUTN GND C4 1 µF RINP RINN PGNDR P1 50 kΩ T6 C5 1 µF 1 µF PGNDR P2 50 kΩ T7 C3 C1 1 µF C9 0.1 µ F PGND ROUTP P3 50 kΩ LIN− AGND C2 SD ROUTN C18 C15 0.1 µ F 10 µ F L2 (Bead) C23 1 nF BSRP RIN− SHUTDOWN ROUT− 10 nF GND L1 (Bead) C22 1 nF 1 µF C8 220 pF C6 120 kΩ R1 100 nF C14 C16 10 µ F PGND C13 0.1µ F 1 µF AVDD AVCC C7 1 µF Cin2 1 µF Cin1 10 kΩ (T3) Rhps2 0.47µ F Cvcc 10 kΩ Rhps1 Rout1 1 kΩ Rout2 1 kΩ R3 120 kΩ AVDD Cout2 BYP IN1 10 kΩ Rhpf2 IN2 Vo2 SD VDD GND Vo1 Rhpf1 10 kΩ 220 µF TPA6110A2 Cout1 AVDD (T4) 220 µ F
 SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 www.ti.com 10 µF BSLP GND BSLN Figure 30. Stereo Class-D With Single-Ended Inputs and Stereo Headphone Amplifier Interface 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 CLASS-D OPERATION This section focuses on the class-D operation of the TPA3002D2. Traditional Class-D Modulation Scheme The traditional class-D modulation scheme, which is used in the TPA032D0x family, has a differential output where each output is 180 degrees out of phase and changes from ground to the supply voltage, VCC. Therefore, the differential prefiltered output varies between positive and negative VCC, where filtered 50% duty cycle yields 0 V across the load. The traditional class-D modulation scheme with voltage and current waveforms is shown in Figure 31. Note that even at an average of 0 V across the load (50% duty cycle), the current to the load is high, causing high loss, thus causing a high supply current. OUTP OUTN +12 V Differential Voltage Across Load 0V −12 V Current Figure 31. Traditional Class-D Modulation Scheme’s Output Voltage and Current Waveforms Into an Inductive Load With No Input TPA3002D2 Modulation Scheme The TPA3002D2 uses a modulation scheme that still has each output switching from 0 to the supply voltage. However, OUTP and OUTN are now in phase with each other with no input. The duty cycle of OUTP is greater than 50% and OUTN is less than 50% for positive output voltages. The duty cycle of OUTP is less than 50% and OUTN is greater than 50% for negative output voltages. The voltage across the load sits at 0 V throughout most of the switching period, greatly reducing the switching current, which reduces any I2R losses in the load. 19 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 OUTP OUTN Differential Voltage Across Load Output = 0 V +12 V 0V −12 V Current OUTP OUTN Differential Voltage Output > 0 V +12 V 0V Across Load −12 V Current Figure 32. The TPA3002D2 Output Voltage and Current Waveforms Into an Inductive Load Efficiency: LC Filter Required With the Traditional Class-D Modulation Scheme The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple current is large for the traditional modulation scheme, because the ripple current is proportional to voltage multiplied by the time at that voltage. The differential voltage swing is 2 × VCC, and the time at each voltage is half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from each half cycle for the next half cycle, while any resistance causes power dissipation. The speaker is both resistive and reactive, whereas an LC filter is almost purely reactive. The TPA3002D2 modulation scheme has very little loss in the load without a filter because the pulses are very short and the change in voltage is VCC instead of 2 × VCC. As the output power increases, the pulses widen, making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for most applications the filter is not needed. An LC filter with a cutoff frequency less than the class-D switching frequency allows the switching current to flow through the filter instead of the load. The filter has less resistance than the speaker, which results in less power dissipation, therefore increasing efficiency. Effects of Applying a Square Wave Into a Speaker Audio specialists have advised for years not to apply a square wave to speakers. If the amplitude of the waveform is high enough and the frequency of the square wave is within the bandwidth of the speaker, the square wave could cause the voice coil to jump out of the air gap and/or scar the voice coil. A 250-kHz switching frequency, however, does not significantly move the voice coil, as the cone movement is proportional to 1/f2 for frequencies beyond the audio band. 20 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Damage may occur if the voice coil cannot handle the additional heat generated from the high-frequency switching current. The amount of power dissipated in the speaker may be estimated by first considering the overall efficiency of the system. If the on-resistance (rds(on)) of the output transistors is considered to cause the dominant loss in the system, then the maximum theoretical efficiency for the TPA3002D2 with an 8-Ω load is as follows: ǒ Efficiency (theoretical, %) + R ń R ) r L ds(on) L Ǔ 100% + 8ń(8 ) 0.58) 100% + 93.24% (1) The maximum measured output power is approximately 7.5 W with an 12-V power supply. The total theoretical power supplied (P(total)) for this worst-case condition would therefore be as follows: P (total) + P ńEfficiency + 7.5 W ń 0.9324 + 8.04 W O (2) The efficiency measured in the lab using an 8-Ω speaker was 89%. The power not accounted for as dissipated across the rds(on) may be calculated by simply subtracting the theoretical power from the measured power: Other losses + P (total) (measured) * P (total) (theoretical) + 8.43 * 8.04 + 0.387 W (3) The quiescent supply current at 14 V is measured to be 14.3 mA. It can be assumed that the quiescent current encapsulates all remaining losses in the device, i.e., biasing and switching losses. It may be assumed that any remaining power is dissipated in the speaker and is calculated as follows: P (dis) + 0.387 W * (14 V 14.3 mA) + 0.19 W (4) Note that these calculations are for the worst-case condition of 7.5 W delivered to the speaker. Since the 0.19 W is only 2.5% of the power delivered to the speaker, it may be concluded that the amount of power actually dissipated in the speaker is relatively insignificant. Furthermore, this power dissipated is well within the specifications of most loudspeaker drivers in a system, as the power rating is typically selected to handle the power generated from a clipping waveform. When to Use an Output Filter Design the TPA3002D2 without the filter if the traces from amplifier to speaker are short (< 1 inch). Powered speakers, where the speaker is in the same enclosure as the amplifier, is a typical application for class-D without a filter. Most applications require a ferrite bead filter. The ferrite filter reduces EMI around 1 MHz and higher (FCC and CE only test radiated emissions greater than 30 MHz). When selecting a ferrite bead, choose one with high impedance at high frequencies, but very low impedance at low frequencies. Use a LC output filter if there are low frequency ( YES (VOLUME−VARDIFF) VAROUT_VOLUME (V) = VOLUME (V) − VARDIFF (V) ? NO VAROUT_VOLUME (V) = VARMAX (V) Figure 35. Block Diagram of VAROUT Volume Control Decreasing Voltage on VOLUME Terminal 5.6 Class-D Gain − dB + 3.1 Increasing Voltage on VOLUME Terminal 0.5 2.00 (40.1%*VREF) 2.21 2.10 2.11 (44.1%*VREF) (41.9%*VREF) (42.3%*VREF) Voltage on VOLUME Pin − V Figure 36. DC Volume Control Operation, VREF = 5 V 23 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 MODE OPERATION The MODE pin is an input for controlling the output mode of the TPA3002D2. A logic HIGH on this pin disables the Class-D outputs. A logic LOW on this pin enables the class-D outputs. The VAROUT outputs are active in both modes and can be used as line level inputs to an external powered subwoofer while driving internal stereo speakers with the class-D outputs. The trip levels are defined in the specifications table. For interfacing with an external headphone amplifier like the TPA6110A2, the MODE pin can be connected to the switch on a headphone jack. When configured like Figure 30, the class-D outputs will be disabled when a headphone plug is inserted into the headphone jack. MODE_OUT OPERATION for controlling the SHUTDOWN pin on an external headphone amplifier like the TPA6110A2 or for interfacing with other logic. The output voltages for a given load condition are given in the specifications table. This output is controlled by the MODE pin logic. When the MODE input is driven to a logic low, the MODE_OUT output drives to a logic high. Conversely, when the MODE pin is driven to a logic high, the MODE_OUT output drives LOW. The MODE_OUT output is simply the inverted state of the MODE input. It is designed in this manner because the TPA6110A2 SHUTDOWN input is active high. This allows the TPA3002D2 to place the TPA6110A2 into the shutdown state when driving internal speakers in the Class-D mode. Conversely, the MODE_OUT pin drives low to enable the TPA6110A2 headphone amplifier when headphones are plugged into the headphone jack and the MODE input is driven high. SELECTION OF COSC AND ROSC The switching frequency is determined using the values of the components connected to ROSC (pin 27) and COSC (pin 28) and may be calculated with the following equation: fOSC = 6.6 / (ROSC * COSC) INTERNAL 2.5-V BIAS GENERATOR CAPACITOR SELECTION The internal 2.5-V bias generator (V2P5) provides the internal bias for the preamplifier stages on both the class-D amplifiers and the variable amplifiers. The external input capacitors and this internal reference allow the inputs to be biased within the optimal common-mode range of the input preamplifiers. The selection of the capacitor value on the V2P5 terminal is critical for achieving the best device performance. During startup or recovery from the shutdown state, the V2P5 capacitor determines the rate at which the amplifier starts up. When the voltage on the V2P5 capacitor equals 0.75xV2P5, or 75% of its final value, the device turns on and the class-D outputs start switching. The startup time is not critical for the best depop performance since any pop sound that is heard is the result of the class-D outputs switching on and not the startup time. However, at least a 0.47-µF capacitor is recommended for the V2P5 capacitor. A secondary function of the V2P5 capacitor is to filter high frequency noise on the internal 2.5-V bias generator. INPUT RESISTANCE Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest value to over six times that value. As a result, if a single capacitor is used in the input high-pass filter, the −3 dB or cutoff frequency also changes by over six times. Zf Ci Input Signal IN Zi The −3-dB frequency can be calculated using equation 5. Input impedance (Zi) vs Gain can be found in Figure 7. 24 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 f *3dB + 1 2p Z iC i (5) INPUT CAPACITOR, CI In the typical application an input capacitor (Ci) is required to allow the amplifier to bias the input signal to the proper dc level (V2P5)for optimum operation. In this case, Ci and the input impedance of the amplifier (Zi) form a high-pass filter with the corner frequency determined in equation 6. −3 dB (6) 1 fc + 2 p Zi C i fc The value of Ci is important, as it directly affects the bass (low frequency) performance of the circuit. Consider the example where Zi is 20 kΩ and the specification calls for a flat bass response down to 20 Hz. Equation 6 is reconfigured as equation 7. Ci + 1 2p Z i f c (7) In this example, Ci is 0.4 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. If the gain is known and will be constant, use Zi from Figure 7 (Input Impedance vs Gain) to calculate Ci. Calculations for Ci should be based off the impedance at the lowest gain step intended for use in the system. A further consideration for this capacitor is the leakage path from the input source through the input network (Ci) and the feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications as the dc level there is held at 2.5 V, which is likely higher than the source dc level. Note that it is important to confirm the capacitor polarity in the application. Power Supply Decoupling, CS The TPA3002D2 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VCC lead works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio power amplifier is recommended. The 10-µF capacitor also serves as local storage capacitor for supplying current during large signal transients on the amplifier outputs. BSN and BSP Capacitors The full H-bridge output stages use only NMOS transistors. They therefore require bootstrap capacitors for the high side of each output to turn on correctly. A 10-nF ceramic capacitor, rated for at least 25 V, must be connected from each output to its corresponding bootstrap input. Specifically, one 10-nF capacitor must be connected from xOUTP to xBSP, and one 10-nF capacitor must be connected from xOUTN to xBSN. (See the application circuit diagram in Figure 29.) 25 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 The bootstrap capacitors connected between the BSxx pins and corresponding output function as a floating power supply for the high-side N-channel power MOSFET gate drive circuitry. During each high-side switching cycle, the bootstrap capacitors attempt to hold the gate-to-source voltage high enough to keep the high-side MOSFETs turned on. However, there is a leakage path and the voltage on the bootstrap capacitors slowly decrease while the high-side is conducting. By driving the outputs into heavy clipping with a sine wave of less than 50 Hz, the bootstrap voltage can decrease below the minimum Vgs required to keep the high-side output MOSFET turned on. When this occurs, the output transistor becomes a source-follower and the output drops from VCC to approximately Vclamp (voltage on pins 25 and 36). For the majority of applications, driving a square wave at low frequencies is not a design consideration and the recommended bootstrap capacitor value of 10 nF is acceptable. However, if this is a concern, increasing the bootstrap capacitors holds the gate voltage for a longer period of time and the drop in the output voltage does not occur. A value of 220 nF is recommended with a 51 Ω resistor placed in series between the outputs and bootstrap pins. The 51 Ω series resistor is necessary to limit the current charging and discharging the bootstrap capacitors. VCLAMP Capacitors To ensure that the maximum gate-to-source voltage for the NMOS output transistors is not exceeded, two internal regulators clamp the gate voltage. Two 1-µF capacitors must be connected from VCLAMPL (pin 25) and VCLAMPR (pin 36) to ground and must be rated for at least 25 V. The voltages at the VCLAMP terminals vary with VCC and may not be used for powering any other circuitry. Internal Regulated 5-V Supply (AVDD) The AVDD terminal (pin 29) is the output of an internally-generated 5-V supply, used for the oscillator, preamplifier, and volume control circuitry. It requires a 0.1-µF to 1-µF capacitor, placed very close to the pin, to ground to keep the regulator stable. The regulator may be used to power an external headphone amplifier or other circuitry, up to a current limit specified in the specification table. When powering external circuitry, like the TPA6110A2 headphone amplifier, an additional 10 µF or larger capacitor should be added to the AVDD terminal. AVDD − POWER-UP RESPONSE Power−Up Ch1 (AVDD) AVDD (pin 29) Ch2 (AVCC) AVCC (pin 33) Ch1 2 V/div Ch2 5 V/div M 10.0 µs Figure 37. Power-Up Response Differential Input The differential input stage of the amplifier cancels any noise that appears on both input lines of the channel. To use the TPA3002D2 EVM with a differential source, connect the positive lead of the audio source to the INP input and the negative lead from the audio source to the INN input. To use the TPA3002D2 with a single-ended source, ac ground the INP input through a capacitor equal in value to the input capacitor on INN and apply the audio source to the INN input. In a single-ended input application, the INP input should be ac-grounded at the audio source instead of at the device input for best noise performance. 26 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 SD OPERATION The TPA3002D2 employs a shutdown mode of operation designed to reduce supply current (ICC) to the absolute minimum level during periods of nonuse for power conservation. The SD input terminal should be held high (see specification table for trip point)during normal operation when the amplifier is in use. Pulling SD low causes the outputs to mute and the amplifier to enter a low-current state, ICC(SD) = 10 µA. SD should never be left unconnected, because amplifier operation would be unpredictable. POWER-OFF POP REDUCTION For the best power-off pop performance, the amplifier should be placed in the shutdown mode prior to removing the power supply voltage. Another method to reduce power-off pop can be implemented in the hardware. A 100-µF − 150-µF capacitor can be added to the AVDD terminal in parallel with the 100-nF capacitor shown in Figure 29. The additional capacitance holds up the regulator voltage for a longer period of time and results in smaller power-off pop. USING LOW-ESR CAPACITORS Low-ESR capacitors are recommended throughout this application section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the real capacitor behaves like an ideal capacitor. SHORT-CIRCUIT PROTECTION The TPA3002D2 has short circuit protection circuitry on the outputs that prevents damage to the device during output-to-output shorts, output-to-GND shorts, and output-to-VCC shorts. When a short-circuit is detected on the outputs, the part immediately disables the output drive. This is a latched fault and must be reset by cycling the voltage on the SD pin to a logic low and back to the logic high state for normal operation. This will clear the short-circuit flag and allow for normal operation if the short was removed. If the short was not removed, the protection circuitry will again activate. The trip-point for the short-circuit protection is nominally set at 8 A. However, this trip point can vary with PCB layout and the separation of AVCC and PVCC. It is important to connect the AVCC pin as close as possible to all of the PVCC pins with a wide (>20 mils) trace. This minimizes the inductance between the two pins and allows the short-circuit protection to trip at the nominal current. If the inductance between these two pins is large, the short-circuit protection may inadvertently trip when drive low impedance loads into heavy clipping. THERMAL PROTECTION Thermal protection on the TPA3002D2 prevents damage to the device when the internal die temperature exceeds 150°C. There is a ±15 degree tolerance on this trip point from device to device. Once the die temperature exceeds the thermal set point, the device enters into the shutdown state and the outputs are disabled. This is not a latched fault. The thermal fault is cleared once the temperature of the die is reduced by 20°C. The device begins normal operation at this point with no external system interaction. THERMAL CONSIDERATIONS: OUTPUT POWER AND MAXIMUM AMBIENT TEMPERATURE To calculate the maximum ambient temperature, the following equation may be used: TAmax = TJmax – ΘJAPDissipated where: TJmax = 150°C ΘJA = 19°C/W (2-Layer PCB, 5 sq. in. copper, see Figure 38) (8) To estimate the power dissipation, the following equation may be used: PDissipated = PO(average) x ((1 / Efficiency) – 1) Efficiency = ~85% for an 8-Ω load = ~75% for a 4-Ω load (9) 27 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Example. What is the maximum ambient temperature for an application that requires the TPA3002D2 to drive 7.5 W into stereo 8-Ω speakers? PDissipated = 15 W × ((1 / 0.85) – 1) = 2.65 W (PO = 7.5 W × 2) TAmax = 150°C – (19°C/W × 2.65 W) = 99.65°C This calculation shows that the TPA3002D2 can drive 7.5 W into an 8-Ω speaker up to the absolute maximum ambient temperature rating of 85°C, which must never be exceeded. Figure 38 and Figure 39 show the results of several thermal experiments conducted with the TPA3002D2. Both figures show that the best thermal performance can be achieved with more copper area for heat dissipation and an adequate number of thermal vias. Figure 38 shows two curves for a 2-layer and 4-layer PCB. The 2-layer PCB layout was tightly controlled with a fixed amount of 2 oz. copper on the bottom layer of the PCB. The amount of copper is shown on the x-axis. Nine thermal vias of 13 mil (0,33 mm) diameter were drilled under the PowerPad and connected to the bottom layer. The top layer only consisted of traces for signal routing. The 4-layer PCB layout was also tightly controlled with a fixed amount of 2 oz. copper in middle GND layer. The top layer only consisted of traces for signal routing. The bottom and other middle layer were left blank. Nine thermal vias of 0,33 mm diameter were drilled under the PowerPAD and connected to the middle layer. Figure 39 shows the effect of the number of thermal vias drilled under the PowerPAD on the thermal performance of the PCB. The experiment was conducted with a 2-layer PCB and 3 square inches of copper on the bottom layer. For the best thermal performance, at least 16 vias in a 4x4 pattern should be used under the PowerPAD. Refer to the TPA3002D2 EVM User’s Manual (SLOU151), for an example layout with a 4x4 via pattern. PCB gerber files are available at request. 28 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 PRINTED CIRCUIT BOARD (PCB) LAYOUT Because the TPA3002D2 is a class-D amplifier that switches at a high frequency, the layout of the printed circuit board (PCB) should be optimized according to the following guidelines for the best possible performance. D Decoupling capacitors — As described on page 28, the high-frequency 0.1-µF decoupling capacitors should be placed as close to the PVCC (pin 14, 15, 22, 23, 38, 39, 46, 47) and AVCC (pin 33) terminals as possible. The V2P5 (pin 4) capacitor, AVDD (pin 29) capacitor, and VCLAMP (pins 25, 36) capacitor should also be placed as close to the device as possible. Large (10 µF or greater) bulk power supply decoupling capacitors should be placed near the TPA3002D2 on the PVCCL, PVCCR, and AVCC terminals. D Grounding — The AVCC (pin 33) decoupling capacitor, AVDD (pin 29) capacitor, V2P5 (pin 4) capacitor, COSC (pin 28) capacitor, and ROSC (pin 27) resistor should each be grounded to analog ground (AGND, pin 26 and pin 30). The PVCC (pin 9 and pin 16) decoupling capacitors should each be grounded to power ground (PGND, pins 18, 19, 42, 43). Analog ground and power ground may be connected at the PowerPAD, which should be used as a central ground connection or star ground for the TPA3002D2. Basically, an island should be created with a single connection to PGND at the PowerPAD. D Output filter — The ferrite EMI filter (Figure 34) should be placed as close to the output terminals as possible for the best EMI performance. The LC filter (Figure 33 should be placed close to the outputs. The capacitors used in both the ferrite and LC filters should be grounded to power ground. D PowerPAD — The PowerPAD must be soldered to the PCB for proper thermal performance and optimal reliability. The dimensions of the PowerPAD thermal land should be 5 mm by 5 mm (197 mils by 197 mils). The PowerPAD size measures 4,55 x 4,55 mm. Four rows of solid vias (four vias per row, 0,3302 mm or 13 mils diameter) should be equally spaced underneath the thermal land. The vias should connect to a solid copper plane, either on an internal layer or on the bottom layer of the PCB. The vias must be solid vias, not thermal relief or webbed vias. For additional information, please refer to the PowerPAD Thermally Enhanced Package application note (SLMA002). For an example layout, refer to the TPA3002D2 Evaluation Module (TPA3002D2EVM) User Manual (SLOU151). Both the EVM user manual and the PowerPAD application note are available on the TI web site at http://www.ti.com. THERMAL RESISTANCE vs COPPER AREA 2-LAYER PCB THERMAL RESISTANCE vs COPPER AREA 4-LAYER PCB 35 θ JA − Thermal Resistance − °C/W θ JA − Thermal Resistance − °C/W 35 30 25 20 15 1 1.5 2 2.5 3 3.5 4 Copper Area − sq. Inches 4.5 5 30 25 20 15 1 2 3 4 Copper Area − sq. Inches 5 Figure 38. Thermal Resistance 29 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 THERMAL RESISTANCE vs THERMAL VIA QUANTITY 2-LAYER PCB θ JA − Thermal Resistance − °C/W 25 24 23 22 21 20 4 6 8 10 12 14 Thermal Via Quantity (13 Mil Diameter) 16 Figure 39. Thermal Resistance BASIC MEASUREMENT SYSTEM This application note focuses on methods that use the basic equipment listed below: D Audio analyzer or spectrum analyzer D Digital multimeter (DMM) D Oscilloscope D Twisted pair wires D Signal generator D Power resistor(s) D Linear regulated power supply D Filter components D EVM or other complete audio circuit Figure 40 shows the block diagrams of basic measurement systems for class-AB and class-D amplifiers. A sine wave is normally used as the input signal since it consists of the fundamental frequency only (no other harmonics are present). An analyzer is then connected to the APA output to measure the voltage output. The analyzer must be capable of measuring the entire audio bandwidth. A regulated dc power supply is used to reduce the noise and distortion injected into the APA through the power pins. A System Two audio measurement system (AP-II) (Reference 1) by Audio Precision includes the signal generator and analyzer in one package. The generator output and amplifier input must be ac-coupled. However, the EVMs already have the ac-coupling capacitors, (CIN), so no additional coupling is required. The generator output impedance should be low to avoid attenuating the test signal, and is important since the input resistance of APAs is not very high (about 10 kΩ). Conversely the analyzer-input impedance should be high. The output impedance, ROUT, of the APA is normally in the hundreds of milliohms and can be ignored for all but the power-related calculations. Figure 40(a) shows a class-AB amplifier system. They take an analog signal input and produce an analog signal output. These amplifier circuits can be directly connected to the AP-II or other analyzer input. This is not true of the class-D amplifier system shown in Figure 40(b), which requires low pass filters in most cases in order to measure the audio output waveforms. This is because it takes an analog input signal and converts it into a pulse-width modulated (PWM) output signal that is not accurately processed by some analyzers. 30 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Power Supply Signal Generator APA RL Analyzer 20 Hz − 20 kHz (a) Basic Class-AB Power Supply Low-Pass RC Filter Signal Generator Class-D APA RL† Low-Pass RC Filter Analyzer 20 Hz − 20 kHz † For efficiency measurements with filter-free class-D amplifiers, RL should be an inductive load like a speaker. (b) Filter-Free and Traditional Class-D Figure 40. Audio Measurement Systems The TPA3002D2 uses a modulation scheme that does not require an output filter for operation, but they do sometimes require an RC low-pass filter when making measurements. This is because some analyzer inputs cannot accurately process the rapidly changing square-wave output and therefore record an extremely high level of distortion. The RC low-pass measurement filter is used to remove the modulated waveforms so the analyzer can measure the output sine wave. DIFFERENTIAL INPUT AND BTL OUTPUT All of the class-D APAs and many class-AB APAs have differential inputs and bridge-tied load (BTL) outputs. Differential inputs have two input pins per channel and amplify the difference in voltage between the pins. Differential inputs reduce the common-mode noise and distortion of the input circuit. BTL is a term commonly used in audio to describe differential outputs. BTL outputs have two output pins providing voltages that are 180 degrees out of phase. The load is connected between these pins. This has the added benefits of quadrupling the output power to the load and eliminating a dc blocking capacitor. A block diagram of the measurement circuit is shown in Figure 41. The differential input is a balanced input, meaning the positive (+) and negative (−) pins will have the same impedance to ground. Similarly, the BTL output equates to a balanced output. 31 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Evaluation Module Audio Power Amplifier Generator Analyzer Low-Pass RC Filter CIN VGEN RGEN RIN ROUT RIN ROUT CIN RGEN RL Low-Pass RC Filter Twisted-Pair Wire RANA CANA RANA CANA Twisted-Pair Wire Figure 41. Differential Input—BTL output Measurement Circuit The generator should have balanced outputs and the signal should be balanced for best results. An unbalanced output can be used, but it may create a ground loop that will affect the measurement accuracy. The analyzer must also have balanced inputs for the system to be fully balanced, thereby cancelling out any common mode noise in the circuit and providing the most accurate measurement. The following general rules should be followed when connecting to APAs with differential inputs and BTL outputs: D D D D D Use a balanced source to supply the input signal. Use an analyzer with balanced inputs. Use twisted-pair wire for all connections. Use shielding when the system environment is noisy. Ensure the cables from the power supply to the APA, and from the APA to the load, can handle the large currents (see Table 3). Table 3 shows the recommended wire size for the power supply and load cables of the APA system. The real concern is the dc or ac power loss that occurs as the current flows through the cable. These recommendations are based on 12-inch long wire with a 20-kHz sine-wave signal at 25°C. Table 3. Recommended Minimum Wire Size for Power Cables POUT (W) RL (Ω) AWG SIZE 10 4 2 4 18 1 8 22 < 0.75 8 18 22 22 DC POWER LOSS (MW) AC POWER LOSS (MW) 16 40 18 42 22 3.2 8.0 3.7 8.5 28 2.0 8.0 2.1 8.1 1.5 6.1 1.6 6.2 28 CLASS-D RC LOW-PASS FILTER An RC filter is used to reduce the square-wave output when the analyzer inputs cannot process the pulse-width modulated class-D output waveform. This filter has little effect on the measurement accuracy because the cutoff frequency is set above the audio band. The high frequency of the square wave has negligible impact on measurement accuracy because it is well above the audible frequency range and the speaker cone cannot respond at such a fast rate. The RC filter is not required when an LC low-pass filter is used, such as with the class-D APAs that employ the traditional modulation scheme (TPA032D0x, TPA005Dxx). The component values of the RC filter are selected using the equivalent output circuit as shown in Figure 42. RL is the load impedance that the APA is driving for the test. The analyzer input impedance specifications should be available and substituted for RANA and CANA. The filter components, RFILT and CFILT, can then be derived for the system. The filter should be grounded to the APA near the output ground pins or at the power supply ground pin to minimize ground loops. 32 
 www.ti.com SLOS402C − DECEMBER 2002 − REVISED JANUARY 2004 Load RC Low-Pass Filters RFILT AP Analyzer Input CFILT VL= VIN RL CANA RANA CANA RANA VOUT RFILT CFILT To APA GND Figure 42. Measurement Low-Pass Filter Derivation Circuit—Class-D APAs The transfer function for this circuit is shown in equation (10) where ωO = REQCEQ, REQ = RFILTRANA and CEQ = (CFILT + CANA). The filter frequency should be set above fMAX, the highest frequency of the measurement bandwidth, to avoid attenuating the audio signal. Equation (11) provides this cutoff frequency, fC. The value of RFILT must be chosen large enough to minimize current that is shunted from the load, yet small enough to minimize the attenuation of the analyzer-input voltage through the voltage divider formed by RFILT and RANA. A rule of thumb is that RFILT should be small (~100 Ω) for most measurements. This reduces the measurement error to less than 1% for RANA ≥ 10 kΩ. ǒ Ǔ V OUT V IN f C + Ǹ2 ǒ R R + f ANA )R ANA FILT Ǔ ǒ Ǔ 1 ) j ww O (10) MAX (11) An exception occurs with the efficiency measurements, where RFILT must be increased by a factor of ten to reduce the current shunted through the filter. CFILT must be decreased by a factor of ten to maintain the same cutoff frequency. See Table 4 for the recommended filter component values. Once fC is determined and RFILT is selected, the filter capacitance is calculated using equation (12). When the calculated value is not available, it is better to choose a smaller capacitance value to keep fC above the minimum desired value calculated in equation (11). C FILT + 1 2p f C R FILT (12) Table 4 shows recommended values of RFILT and CFILT based on common component values. The value of fC was originally calculated to be 28 kHz for an fMAX of 20 kHz. CFILT, however, was calculated to be 57000 pF, but the nearest values of 56000 pF and 51000 pF were not available. A 47000 pF capacitor was used instead, and fC is 34 kHz, which is above the desired value of 28 kHz. Table 4. Typical RC Measurement Filter Values MEASUREMENT Efficiency All other measurements RFILT CFILT 1 000 Ω 5 600 pF 100 Ω 56 000 pF 33 PACKAGE OPTION ADDENDUM www.ti.com 14-Oct-2022 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) Samples (4/5) (6) TPA3002D2PHP ACTIVE HTQFP PHP 48 250 RoHS & Green NIPDAU Level-4-260C-72 HR -40 to 85 TPA3002D2 Samples TPA3002D2PHPR ACTIVE HTQFP PHP 48 1000 RoHS & Green NIPDAU Level-4-260C-72 HR -40 to 85 TPA3002D2 Samples (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of