TPA711DR

    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 D Fully Specified for 3.3-V and 5-V Operation D Wide Power Supply Compatibility D D D D 2.5 V − 5.5 V D Output Power − 700 mW at VDD = 5 V, BTL, RL = 8 Ω − 85 mW at VDD = 5 V, SE, RL = 32 Ω − 250 mW at VDD = 3.3 V, BTL, RL = 8 Ω − 37 mW at VDD = 3.3 V, SE, RL = 32 Ω Shutdown Control − IDD = 7 µA at 3.3 V − IDD = 50 µA at 5 V D BTL to SE Mode Control Integrated Depop Circuitry Thermal and Short-Circuit Protection Surface-Mount Packaging − SOIC − PowerPAD MSOP D OR DGN PACKAGE (TOP VIEW) SHUTDOWN BYPASS SE/BTL IN description 1 8 2 7 3 6 4 5 VO − GND VDD VO + The TPA711 is a bridge-tied load (BTL) or single-ended (SE) audio power amplifier developed especially for low-voltage applications where internal speakers and external earphone operation are required. Operating with a 3.3-V supply, the TPA711 can deliver 250-mW of continuous power into a BTL 8-Ω load at less than 0.6% THD+N throughout voice band frequencies. Although this device is characterized out to 20 kHz, its operation is optimized for narrower band applications such as wireless communications. The BTL configuration eliminates the need for external coupling capacitors on the output in most applications, which is particularly important for small battery-powered equipment. A unique feature of the TPA711 is that it allows the amplifier to switch from BTL to SE on the fly when an earphone drive is required. This eliminates complicated mechanical switching or auxiliary devices just to drive the external load. This device features a shutdown mode for power-sensitive applications with special depop circuitry to eliminate speaker noise when exiting shutdown mode. The TPA711 is available in an 8-pin SOIC and the surface-mount PowerPAD MSOP package, which reduces board space by 50% and height by 40%. VDD 6 VDD RF VDD/2 Audio Input RI 4 IN 2 BYPASS CS − CI VO+ 5 + CB − VO− 8 + From System Control From HP Jack 1 SHUTDOWN 3 SE/BTL 700 mW 7 GND Bias Control Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. Copyright  2002, Texas Instruments Incorporated 
 
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       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 AVAILABLE OPTIONS PACKAGED DEVICES TA SMALL OUTLINE† (D) MSOP SYMBOLIZATION MSOP‡ (DGN) −40°C to 85°C TPA711D TPA711DGN ABB † In the SOIC package, the maximum RMS output power is thermally limited to 350 mW; 700 mW peaks can be driven, as long as the RMS value is less than 350 mW. ‡ The D and DGN packages are available taped and reeled. To order a taped and reeled part, add the suffix R to the part number (e.g., TPA311DR). Terminal Functions TERMINAL NAME NO. I/O DESCRIPTION I BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to a 0.1-µF to 2.2-µF capacitor when used as an audio amplifier. BYPASS 2 GND 7 IN 4 I IN is the audio input terminal. SE/BTL 3 I When SE/BTL is held low, the TPA711 is in BTL mode. When SE/BTL is held high, the TPA711 is in SE mode. SHUTDOWN 1 I SHUTDOWN places the entire device in shutdown mode when held high (IDD = 7 µA). VDD VO+ 6 5 O VDD is the supply voltage terminal. VO+ is the positive output for BTL and SE modes. VO− 8 O VO− is the negative output in BTL mode and a high-impedance output in SE mode. GND is the ground connection. absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§ Supply voltage, VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V Input voltage, VI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −0.3 V to VDD +0.3 V Continuous total power dissipation . . . . . . . . . . . . . . . . . . . . . Internally limited (see Dissipation Rating Table) Operating free-air temperature range, TA (see Table 3) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 85°C Operating junction temperature range, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −40°C to 150°C Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . −65°C to 150°C Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C § Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. DISSIPATION RATING TABLE DERATING FACTOR D TA ≤ 25°C 725 mW DGN 2.14 W¶ 17.1 mW/°C PACKAGE 5.8 mW/°C TA = 70°C 464 mW TA = 85°C 377 mW 1.37 W 1.11 W ¶ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended Board for PowerPAD on page 33 of that document. 2 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 recommended operating conditions ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ Supply voltage, VDD SHUTDOWN High-level voltage, VIH MIN MAX 2.5 5.5 0.9VDD 0.9VDD SE/BTL SE/BTL Operating free-air temperature, TA (see Table 3) V V SHUTDOWN Low-level voltage, VIL UNIT −40 0.1VDD 0.1VDD V 85 °C electrical characteristics at specified free-air temperature, VDD = 3.3 V, TA = 25°C (unless otherwise noted) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ PARAMETER TEST CONDITIONS MIN VOO Output offset voltage (measured differentially) SHUTDOWN = 0 V, SE/BTL = 0 V, RL = 8 Ω, RF = 10 kΩ PSRR Power supply rejection ratio VDD = 3.2 V to 3.4 V IDD IDD(SD) Supply current (see Figure 6) Supply current, shutdown mode (see Figure 7) |IIH| |IIL| TYP MAX 20 BTL mode 85 SE mode 83 UNIT mV dB BTL mode, SHUTDOWN = 0 V, SE/BTL = 0.33 V, RF = 10 kΩ 1.25 2.5 SE mode, SHUTDOWN = 0 V, SE/BTL = 2.97 V, RF = 10 kΩ 0.65 1.25 7 50 mA SE/BTL = 2.97 V, SHUTDOWN = VDD, RF = 10 kΩ SHUTDOWN, VDD = 3.3 V, VI = VDD 1 SE/BTL, VDD = 3.3 V, VI = VDD SHUTDOWN, VDD = 3.3 V, VI = 0 V 1 SE/BTL, 1 1 VDD = 3.3 V, VI = 0 V µA A µA µA A operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 8 Ω ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ PARAMETER TEST CONDITIONS MIN THD = 0.2%, BTL mode, See Figure 14 TYP MAX UNIT 250 PO Output power, see Note 1 THD = 0.1%, See Figure 22 SE mode, RL = 32 Ω, THD + N Total harmonic distortion plus noise See Figure 12 0.55% Maximum output power bandwidth PO = 250 mW, Gain = 2, f = 200 Hz to 4 kHz, BOM B1 THD = 2%, See Figure 12 20 kHz Unity-gain bandwidth Open Loop, See Figure 36 1.4 MHz f = 1 kHz, See Figure 5 CB = 1 µF, BTL mode, f = 1 kHz, See Figure 3 CB = 1 µF, SE mode, Gain = 1, CB = 0.1 µF, See Figure 42 Supply ripple rejection ratio Vn Noise output voltage 37 mW 79 dB 70 17 µV(rms) NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 3 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise noted) ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ PARAMETER TEST CONDITIONS MIN VOO Output offset voltage (measured differentially) SHUTDOWN = 0 V, SE/BTL = 0 V, RL = 8 Ω, RF = 10 kΩ PSRR Power supply rejection ratio VDD = 4.9 V to 5.1 V IDD IDD(SD) Supply current (see Figure 6) Supply current, shutdown mode (see Figure 7) |IIH| |IIL| TYP MAX 20 BTL mode 78 SE mode 76 UNIT mV dB BTL mode, SHUTDOWN = 0 V, SE/BTL = 0.5 V, RF = 10 kΩ 1.25 2.5 SE mode, SHUTDOWN = 0 V, SE/BTL = 4.5 V, RF = 10 kΩ 0.65 1.25 50 100 mA SE/BTL = 0 V, SHUTDOWN = VDD, RF = 10 kΩ SHUTDOWN, VDD = 5.5 V, VI = VDD 1 SE/BTL, VDD = 5.5 V, VI = VDD SHUTDOWN, VDD = 5.5 V, VI = 0 V 1 SE/BTL, 1 1 VDD = 5.5 V, VI = 0 V µA A µA µA A operating characteristics, VDD = 5 V, TA = 25°C, RL = 8 Ω ÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁÁÁ ÁÁÁÁ ÁÁÁÁÁÁÁÁÁÁ ÁÁÁ ÁÁÁ ÁÁÁ PARAMETER TEST CONDITIONS MIN THD = 0.3%, BTL mode, See Figure 18 THD = 0.1%, See Figure 26 SE mode, RL = 32 Ω, TYP 700† MAX UNIT PO Output power, see Note 1 THD + N Total harmonic distortion plus noise PO = 700 mW, f = 200 Hz to 4 kHz, See Figure 16 0.5% Maximum output power bandwidth Gain = 2, THD = 2%, See Figure 16 20 kHz Unity-gain bandwidth Open Loop, See Figure 37 1.4 MHz f = 1 kHz, See Figure 5 CB = 1 µF, BTL mode, f = 1 kHz, See Figure 4 CB = 1 µF, SE mode, BOM B1 Supply ripple rejection ratio 85 mW 80 dB 73 Vn Noise output voltage Gain = 1, CB = 0.1 µF, See Figure 43 17 µV(rms) † The DGN package, properly mounted, can conduct 700 mW RMS power continuously. The D package, can only conduct 350 mW RMS power continuously, with peaks to 700 mW. NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz. 4 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 PARAMETER MEASUREMENT INFORMATION VDD 6 RF VDD/2 Audio Input RI VDD CS 4 IN 2 BYPASS − CI VO+ 5 + CB RL = 8 Ω − VO− 8 + 1 SHUTDOWN 3 SE/BTL 7 GND Bias Control Figure 1. BTL Mode Test Circuit VDD 6 RF VDD/2 Audio Input RI VDD CS 4 IN 2 BYPASS − CI VO+ 5 + CO CB RL = 32 Ω − VO− 8 + VDD 1 SHUTDOWN 3 SE/BTL 7 GND Bias Control Figure 2. SE Mode Test Circuit POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 5 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS Table of Graphs FIGURE IDD PO Supply ripple rejection ratio vs Frequency Supply current vs Supply voltage Output power THD + N Vn PD 8, 9 vs Load resistance 10, 11 vs Frequency 12, 13, 16, 17, 20, 21, 24, 25, 28, 29, 32, 33 vs Output power 14, 15, 18, 19, 22, 23, 26, 27, 30, 31, 34, 35 Total harmonic distortion plus noise Open loop gain and phase vs Frequency 36, 37 Closed loop gain and phase vs Frequency 38, 39, 40, 41 Output noise voltage vs Frequency Power dissipation vs Output power 0 VDD = 3.3 V RL = 8 Ω SE −10 −20 CB = 0.1 µF −40 CB = 1 µF −60 −70 −80 42, 43 BYPASS = 1/2 VDD −90 VDD = 5 V RL = 8 Ω SE −10 −20 CB = 0.1 µF −30 −40 −50 CB = 1 µF −60 −70 −80 BYPASS = 1/2 VDD −90 −100 20 100 1k 10k 20k −100 20 f − Frequency − Hz 100 1k f − Frequency − Hz Figure 3 6 44, 45, 46, 47 SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY Supply Ripple Rejection Ratio − dB Supply Ripple Rejection Ratio − dB 0 −50 6, 7 vs Supply voltage SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY −30 3, 4, 5 Figure 4 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 10k 20k 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS SUPPLY CURRENT vs SUPPLY VOLTAGE SUPPLY RIPPLE REJECTION RATIO vs FREQUENCY 1.8 RL = 8 Ω CB = 1 µF BTL −10 −20 1.6 SHUTDOWN = 0 V RF = 10 kΩ I DD − Supply Current − mA 1.4 −30 −40 −50 −60 −70 VDD = 3.3 V −80 VDD = 5 V −90 20 100 BTL (SE/BTL = 0.1 VDD) 1.2 1 0.8 SE (SE/BTL = 0.9 VDD) 0.6 0.4 0.2 0 2.5 −100 10k 1k 20k 3 3.5 4 4.5 5 5.5 VDD − Supply Voltage − V f − Frequency − Hz Figure 5 Figure 6 SUPPLY CURRENT vs SUPPLY VOLTAGE 90 80 SHUTDOWN = VDD SE/BTL = 0 V RF = 10 kΩ 70 I DD − Supply Current − µ A Supply Ripple Rejection Ratio − dB 0 60 50 40 30 20 10 0 2.5 3 3.5 4 4.5 5 5.5 VDD − Supply Voltage − V Figure 7 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 7 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS OUTPUT POWER vs SUPPLY VOLTAGE OUTPUT POWER vs SUPPLY VOLTAGE 1000 350 THD+N 1% f = 1 kHz BTL THD+N = 1% f = 1 kHz SE 300 PO − Output Power − mW PO − Output Power − mW 800 600 RL = 8 Ω 400 RL = 32 Ω 250 200 RL = 8 Ω 150 100 RL = 32 Ω 200 50 0 2.5 3 3.5 4 4.5 5 0 2.5 5.5 3 3.5 VDD − Supply Voltage − V Figure 8 5 5.5 OUTPUT POWER vs LOAD RESISTANCE 800 350 THD+N = 1% f = 1 kHz BTL 700 600 VDD = 5 V 500 400 300 THD+N = 1% f = 1 kHz SE 300 PO − Output Power − mW PO − Output Power − mW 4.5 Figure 9 OUTPUT POWER vs LOAD RESISTANCE VDD = 3.3 V 200 250 200 VDD = 5 V 150 100 50 100 VDD = 3.3 V 0 8 16 24 32 40 48 56 64 0 8 14 RL − Load Resistance − Ω 20 26 32 Figure 11 POST OFFICE BOX 655303 38 44 50 RL − Load Resistance − Ω Figure 10 8 4 VDD − Supply Voltage − V • DALLAS, TEXAS 75265 56 62 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10 VDD = 3.3 V PO = 250 mW RL = 8 Ω BTL THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 AV =− 20 V/V 1 AV = −10 V/V AV = −2 V/V 0.1 0.01 20 100 1k 10k VDD = 3.3 V RL = 8 Ω AV = −2 V/V BTL PO = 50 mW 1 0.1 PO = 125 mW PO = 250 mW 0.01 20 20k 1k 100 f − Frequency − Hz Figure 12 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10 VDD = 3.3 V f = 1 kHz AV = −2 V/V BTL THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 1 RL = 8 Ω 0.1 0.01 0.05 0.1 20k Figure 13 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 0 10k f − Frequency − Hz 0.15 0.2 0.25 0.3 0.35 0.4 f = 20 kHz 1 f = 10 kHz f = 1 kHz 0.1 f = 20 Hz 0.01 0.01 PO − Output Power − W VDD = 3.3 V RL = 8 Ω CB = 1 µF AV = −2 V/V BTL 0.1 1 PO − Output Power − W Figure 14 Figure 15 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 9 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10 VDD = 5 V PO = 700 mW RL = 8 Ω BTL THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 AV = −20 V/V 1 AV = −10 V/V AV = −2 V/V 0.1 0.01 20 100 1k 10k 20k VDD = 5 V RL = 8 Ω AV = −2 V/V BTL 1 PO = 700 mW 0.1 PO = 350 mW 0.01 20 100 f − Frequency − Hz 20k TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10k Figure 17 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER VDD = 5 V f = 1 kHz AV = −2 V/V BTL 1 RL = 8 Ω 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1 f = 20 kHz 1 f = 10 kHz f = 1 kHz f = 20 Hz 0.1 0.01 0.01 VDD = 5 V RL = 8 Ω CB = 1 µF AV = −2 V/V BTL PO − Output Power − W 0.1 PO − Output Power − W Figure 18 10 1k f − Frequency − Hz Figure 16 0.01 0.1 PO = 50 mW Figure 19 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 1 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 3.3 V PO = 30 mW RL = 32 Ω SE 1 AV = −10 V/V 0.1 AV = −1 V/V 0.01 AV = −5 V/V 0.001 20 100 1k 10k VDD = 3.3 V RL = 32 Ω AV = −1 V/V SE 1 0.1 PO = 10 mW 0.01 PO = 15 mW PO = 30 mW 0.001 20 20k 1k 100 f − Frequency − Hz Figure 20 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 20k Figure 21 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER VDD = 3.3 V f = 1 kHz RL = 32 Ω AV = −1 V/V SE 1 0.1 0.01 0.02 10k f − Frequency − Hz 0.025 0.03 0.035 0.04 0.045 0.05 VDD = 3.3 V RL = 32 Ω AV = −1 V/V SE 1 f = 20 kHz f = 10 kHz 0.1 f = 20 Hz f = 1 kHz 0.01 0.002 PO − Output Power − W 0.01 0.1 PO − Output Power − W Figure 22 Figure 23 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 11 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 10 VDD = 5 V PO = 60 mW RL = 32 Ω SE THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 AV = −10 V/V 1 AV = −5 V/V 0.1 0.01 AV = −1 V/V 0.001 20 100 1k 10k VDD = 5 V RL = 32 Ω AV = −1 V/V SE 1 PO = 15 mW 0.1 PO = 30 mW 0.01 PO = 60 mW 0.001 20 20k 1k 100 f − Frequency − Hz Figure 24 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10 10 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 20k Figure 25 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER VDD = 5 V f = 1 kHz RL = 32 Ω AV = −1 V/V SE 1 0.1 0.01 0.02 0.04 0.06 0.08 0.1 0.12 0.14 VDD = 5 V RL = 32 Ω AV = −1 V/V SE 1 f = 20 kHz f = 10 kHz 0.1 f = 20 Hz f = 1 kHz 0.01 0.002 PO − Output Power − W 0.01 PO − Output Power − W Figure 26 12 10k f − Frequency − Hz Figure 27 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 0.1 0.2 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 1 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 1 VDD = 3.3 V PO = 0.1 mW RL = 10 kΩ SE 0.1 AV = −5 V/V 0.01 AV = −2 V/V AV = −1 V/V 0.001 20 100 1k 10k VDD = 3.3 V RL = 10 kΩ CB = 1 µF AV = −1 V/V SE 0.1 PO = 0.13 mW PO = 0.05 mW 0.01 PO = 0.1 mW 0.001 20 20k 100 1k f − Frequency − Hz Figure 28 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 10 VDD = 3.3 V f = 1 kHz RL = 10 kΩ AV = −1 V/V SE 0.1 0.01 0.001 50 75 100 125 20 k Figure 29 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 1 10 k f − Frequency − Hz 150 175 200 10 VDD = 3.3 V RL = 10 kΩ AV = −1 V/V SE 1 f = 20 Hz 0.1 f = 20 kHz 0.01 f = 10 kHz f = 1 kHz 0.001 5 PO − Output Power − µW 10 100 500 PO − Output Power − µW Figure 30 Figure 31 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 13 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY TOTAL HARMONIC DISTORTION PLUS NOISE vs FREQUENCY 1 VDD = 5 V PO = 0.3 mW RL = 10 kΩ SE THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % 1 0.1 AV = −5 V/V 0.01 AV = −2 V/V AV = −1 V/V 0.001 20 100 1k 10k VDD = 5 V RL = 10 kΩ AV = −1 V/V SE 0.1 PO = 0.3 mW PO = 0.2 mW 0.01 PO = 0.1 mW 0.001 20 20k 100 f − Frequency − Hz Figure 32 THD+N −Total Harmonic Distortion + Noise − % THD+N −Total Harmonic Distortion + Noise − % VDD = 5 V f = 1 kHz RL = 10 kΩ AV = −1 V/V SE 0.1 0.01 150 200 250 300 350 400 450 500 10 VDD = 5 V RL = 10 kΩ AV = −1 V/V SE 1 f = 20 Hz 0.1 f = 20 kHz 0.01 f = 10 kHz 5 10 100 PO − Output Power − µW Figure 34 Figure 35 POST OFFICE BOX 655303 f = 1 kHz 0.001 PO − Output Power − µW 14 20k TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 10 0.001 50 100 10k Figure 33 TOTAL HARMONIC DISTORTION PLUS NOISE vs OUTPUT POWER 1 1k f − Frequency − Hz • DALLAS, TEXAS 75265 500 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS OPEN-LOOP GAIN AND PHASE vs FREQUENCY 80 180° VDD = 3.3 V RL = Open BTL 70 140° Phase 100° 50 60° 40 20° 30 Gain 20 −20° 10 Phase Open-Loop Gain − dB 60 −60° 0 −100° −10 −140° −20 −30 1 101 102 103 104 −180° f − Frequency − kHz Figure 36 OPEN-LOOP GAIN AND PHASE vs FREQUENCY 80 180° VDD = 5 V RL = Open BTL 70 60 140° 100° 60° 40 20° 30 Gain 20 −20° 10 Phase Open-Loop Gain − dB Phase 50 −60° 0 −100° −10 −140° −20 −30 1 101 102 f − Frequency − kHz 103 104 −180° Figure 37 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 15 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 1 180° Phase 0.75 170° 0.25 0 160° Gain −0.25 150° −0.5 −0.75 140° −1 −1.25 −1.5 −1.75 −2 101 Phase Closed-Loop Gain − dB 0.5 VDD = 3.3 V RL = 8 Ω PO = 250 mW BTL 130° 102 103 104 105 106 120° f − Frequency − Hz Figure 38 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 1 180° Phase 0.75 170° 0.25 0 160° Gain −0.25 150° −0.5 −0.75 140° −1 −1.25 −1.5 −1.75 −2 101 VDD = 5 V RL = 8 Ω PO = 700 m W BTL 102 130° 103 104 105 f − Frequency − Hz Figure 39 16 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 120° 106 Phase Closed-Loop Gain − dB 0.5 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 7 180° Phase 6 170° 5 160° 4 150° 3 140° 2 0 −1 −2 101 130° VDD = 3.3 V RL = 32 Ω AV = 2 V/V PO = 30 mW SE 1 102 103 104 Phase Closed-Loop Gain − dB Gain 120° 110° 105 106 100° f − Frequency − Hz Figure 40 CLOSED-LOOP GAIN AND PHASE vs FREQUENCY 7 180° Phase 6 170° Gain 160° 4 150° 3 140° 2 Phase Closed-Loop Gain − dB 5 130° 1 VDD = 5 V RL = 32 Ω AV = 2 V/V PO = 60 mW SE 0 −1 −2 101 102 103 104 120° 110° 105 106 100° f − Frequency − Hz Figure 41 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 17 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS OUTPUT NOISE VOLTAGE vs FREQUENCY 100 VDD = 3.3 V BW = 22 Hz to 22 kHz RL = 8 Ω or 32 Ω AV = 1 Vn − Output Noise Voltage − µV (rms) Vn − Output Noise Voltage − µV (rms) 100 OUTPUT NOISE VOLTAGE vs FREQUENCY VO BTL VO+ 10 1 20 100 1k 10 k VDD = 5 V BW = 22 Hz to 22 kHz RL = 8 Ω or 32 Ω AV = 1 VO BTL VO+ 10 1 20 20 k 100 f − Frequency − Hz Figure 42 20 k POWER DISSIPATION vs OUTPUT POWER 350 100 90 PD − Power Dissipation − mW RL = 8 Ω 300 PD − Power Dissipation − mW 10 k Figure 43 POWER DISSIPATION vs OUTPUT POWER 250 200 150 RL = 32 Ω 100 VDD = 3.3 V BTL 50 0 200 400 80 RL = 8 Ω 70 60 50 40 30 RL = 32 Ω 20 VDD = 3.3 V SE 10 0 600 0 0 PD − Output Power − mW 50 Figure 45 POST OFFICE BOX 655303 100 PD − Output Power − W Figure 44 18 1k f − Frequency − Hz • DALLAS, TEXAS 75265 150 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 TYPICAL CHARACTERISTICS POWER DISSIPATION vs OUTPUT POWER POWER DISSIPATION vs OUTPUT POWER 800 200 180 RL = 8 Ω PD − Power Dissipation − mW PD − Power Dissipation − mW 700 600 500 400 300 RL = 32 Ω 200 VDD = 5 V BTL 100 RL = 8 Ω 160 140 120 100 80 60 RL = 32 Ω 40 VDD = 5 V SE 20 0 0 200 400 600 800 1000 0 0 PD − Output Power − mW 50 100 150 200 250 300 PD − Output Power − mW Figure 46 Figure 47 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 19 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION bridged-tied load versus single-ended mode Figure 48 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA711 BTL amplifier consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive configuration but initially consider power to the load. The differential drive to the speaker means that as one side is slewing up, the other side is slewing down, and vice versa. This, in effect, doubles the voltage swing on the load as compared to a ground referenced load. Plugging 2 × VO(PP) into the power equation, where voltage is squared, yields 4× the output power from the same supply rail and load impedance (see equation 1). V V (rms) + O(PP) 2 Ǹ2 V Power + (1) 2 (rms) R L VDD VO(PP) RL 2x VO(PP) VDD −VO(PP) Figure 48. Bridge-Tied Load Configuration In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an 8-Ω speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is a 6-dB improvement, which is loudness that can be heard. In addition to increased power there are frequency response concerns. Consider the single-supply SE configuration shown in Figure 49. A coupling capacitor is required to block the dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 µF to 1000 µF) so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the system. This frequency-limiting effect is due to the high pass filter network created with the speaker impedance and the coupling capacitance and is calculated with equation 2. fc + 20 1 2p R C L C (2) POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION bridged-tied load versus single-ended mode (continued) For example, a 68-µF capacitor with an 8-Ω speaker would attenuate low frequencies below 293 Hz. The BTL configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency performance is then limited only by the input network and speaker response. Cost and PCB space are also minimized by eliminating the bulky coupling capacitor. VDD −3 dB VO(PP) CC RL VO(PP) fc Figure 49. Single-Ended Configuration and Frequency Response Increasing power to the load does carry a penalty of increased internal power dissipation. The increased dissipation is understandable considering that the BTL configuration produces 4× the output power of the SE configuration. Internal dissipation versus output power is discussed further in the thermal considerations section. BTL amplifier efficiency Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from VDD. The internal voltage drop multiplied by the RMS value of the supply current, IDDrms, determines the internal power dissipation of the amplifier. An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 50). IDD VO IDD(RMS) V(LRMS) Figure 50. Voltage and Current Waveforms for BTL Amplifiers POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 21 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION BTL amplifier efficiency (continued) Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different. Keep in mind that for most of the waveform, both the push and pull transistors are not on at the same time, which supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform. The following equations are the basis for calculating amplifier efficiency. Efficiency + where P P L (3) SUP 2 Vp V rms 2 L P + + L R 2R L L V V rms + P L Ǹ2 P I SUP + V rms + DD DD I 2V P rms + DD pR L V 2V DD P pR L ǒ p 2P R L L P + Efficiency of a BTL configuration + 4V 4V DD DD pV Ǔ 1ń2 (4) Table 1 employs equation 4 to calculate efficiencies for three different output power levels. The efficiency of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting in a nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper power supply design. Table 1. Efficiency Vs Output Power in 3.3-V 8-Ω BTL Systems OUTPUT POWER (W) EFFICIENCY (%) PEAK VOLTAGE (V) INTERNAL DISSIPATION (W) 0.125 33.6 1.41 0.26 0.25 47.6 2.00 2.45† 0.29 0.375 58.3 † High-peak voltage values cause the THD to increase. 0.28 A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the efficiency equation to utmost advantage when possible. In equation 4, VDD is in the denominator. This indicates that as VDD goes down, efficiency goes up. 22 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION application schematic Figure 51 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of −10 V/V. CF 5 pF RF 50 kΩ VDD 6 VDD VDD/2 Audio Input RI 10 kΩ CI 0.47 µF 4 IN 2 BYPASS − VO+ 5 CC 330 µF CS 1 µF + 1 kΩ CB 2.2 µF − VO− 8 + From System Control 0.1 µF 1 SHUTDOWN 3 SE/BTL 7 GND Bias Control 100 kΩ VDD 100 kΩ Figure 51. TPA711 Application Circuit The following sections discuss the selection of the components used in Figure 51. component selection gain setting resistors, RF and RI The gain for each audio input of the TPA711 is set by resistors RF and RI according to equation 5 for BTL mode. ǒ Ǔ BTL gain + * 2 R F R I (5) BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the voltage swing across the load. Given that the TPA711 is a MOS amplifier, the input impedance is very high; consequently input leakage currents are not generally a concern, although noise in the circuit increases as the value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5 kΩ and 20 kΩ. The effective impedance is calculated in equation 6. Effective impedance + R R F I R )R F I POST OFFICE BOX 655303 (6) • DALLAS, TEXAS 75265 23 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION component selection (continued) As an example consider an input resistance of 10 kΩ and a feedback resistor of 50 kΩ. The BTL gain of the amplifier would be −10 V/V and the effective impedance at the inverting terminal would be 8.3 kΩ, which is well within the recommended range. For high performance applications, metal film resistors are recommended because they tend to have lower noise levels than carbon resistors. For values of RF above 50 kΩ, the amplifier tends to become unstable due to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than 50 kΩ. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7. −3 dB f c(lowpass) + 1 2 pR C F F (7) fc For example, if RF is 100 kΩ and CF is 5 pF, then fc is 318 kHz, which is well outside of the audio range. input capacitor, CI In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency determined in equation 8. −3 dB f c(highpass) + 1 2pR C I I (8) fc The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit. Consider the example where RI is 10 kΩ and the specification calls for a flat bass response down to 40 Hz. Equation 8 is reconfigured as equation 9. 1 C + I 2p R f c I 24 (9) POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION component selection (continued) In this example, CI is 0.40 µF, so one would likely choose a value in the range of 0.47 µF to 1 µF. A further consideration for this capacitor is the leakage path from the input source through the input network (RI, CI) and the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most applications, as the dc level there is held at VDD/2, which is likely higher than the source dc level. It is important to confirm the capacitor polarity in the application. power supply decoupling, CS The TPA711 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved by using two capacitors of different types that target different types of noise on the power supply leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR) ceramic capacitor, typically 0.1 µF placed as close as possible to the device VDD lead, works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 µF or greater placed near the audio power amplifier is recommended. midrail bypass capacitor, CB The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During start-up or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR THD + N. The capacitor is fed from a 250-kΩ source inside the amplifier. To keep the start-up pop as low as possible, the relationship shown in equation 10 should be maintained. This insures the input capacitor is fully charged before the bypass capacitor is fully charged and the amplifier starts up. ǒCB 10 v 1 ǒRF ) RIǓ CI 250 kΩǓ (10) As an example, consider a circuit where CB is 2.2 µF, CI is 0.47 µF, RF is 50 kΩ, and RI is 10 kΩ. Inserting these values into the equation 10 we get: 18.2 v 35.5 which satisfies the rule. Bypass capacitor, CB, values of 0.1 µF to 2.2 µF ceramic or tantalum low-ESR capacitors are recommended for the best THD and noise performance. single-ended operation In SE mode (see Figure 51), the load is driven from the primary amplifier output (VO+, terminal 5). In SE mode the gain is set by the RF and RI resistors and is shown in equation 11. Since the inverting amplifier is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included. SE Gain + * ǒ Ǔ R F R I (11) POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 25 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION component selection (continued) The output coupling capacitor required in single-supply SE mode also places additional constraints on the selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of the following relationship: ǒCB 10 250 kΩ 1 v Ǔ ǒRF ) RIǓ CI Ơ 1 R C (12) L C output coupling capacitor, CC In the typical single-supply SE configuration, an output coupling capacitor (CC) is required to block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by equation 13. −3 dB f c(high) + 1 2 pR C L C (13) fc The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives the low-frequency corner higher, degrading the bass response. Large values of CC are required to pass low frequencies into the load. Consider the example where a CC of 330 µF is chosen and loads vary from 4 Ω, 8 Ω, 32 Ω, and 47 kΩ. Table 2 summarizes the frequency response characteristics of each configuration. Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode RL 8Ω CC 330 µF LOWEST FREQUENCY 32 Ω 330 µF Ą15 Hz 47,000 Ω 330 µF 0.01 Hz 60 Hz As Table 2 indicates, an 8-Ω load is adequate, earphone response is good, and drive into line level inputs (a home stereo for example) is exceptional. 26 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION SE/BTL operation The ability of the TPA711 to easily switch between BTL and SE modes is one of its most important cost-saving features. This feature eliminates the requirement for an additional earphone amplifier in applications where internal speakers are driven in BTL mode but external earphone or speaker must be accommodated. Internal to the TPA711, two separate amplifiers drive VO+ and VO−. The SE/BTL input (terminal 3) controls the operation of the follower amplifier that drives VO− (terminal 8). When SE/BTL is held low, the amplifier is on and the TPA711 is in the BTL mode. When SE/BTL is held high, the VO− amplifier is in a high output impedance state, which configures the TPA711 as an SE driver from VO+ (terminal 5). IDD is reduced by approximately one-half in SE mode. Control of the SE/BTL input can be from a logic-level TTL source or, more typically, from a resistor divider network as shown in Figure 52. 4 IN 2 BYPASS − VO+ 5 + 1 kΩ − VO− 8 + 0.1 µF 1 SHUTDOWN 3 SE/BTL CC 7 GND Bias Control 100 kΩ VDD 100 kΩ Figure 52. TPA711 Resistor Divider Network Circuit Using a readily available 1/8-in. (3.5 mm) mono earphone jack, the control switch is closed when no plug is inserted. When closed, the 100-kΩ/1-kΩ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kΩ resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the VO− amplifier is shut down causing the BTL speaker to mute (virtually open-circuits the speaker). The VO+ amplifier then drives through the output capacitor (CC ) into the earphone jack. using low-ESR capacitors Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal) capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the real capacitor behaves like an ideal capacitor. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 27 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION 5-V versus 3.3-V operation The TPA711 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability. The most important consideration is that of output power. Each amplifier in TPA711 can produce a maximum voltage swing of VDD − 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed to VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into an 8-Ω load before distortion becomes significant. Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes approximately two-thirds the supply power of operation from 5-V supplies for a given output-power level. headroom and thermal considerations Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions. A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion as compared with the average power output. From the TPA711 data sheet, one can see that when the TPA711 is operating from a 5-V supply into a 8-Ω speaker that 700 mW peaks are available. Converting watts to dB: P dB + 10 Log ǒ Ǔ P P W ref ǒ Ǔ + 10Log 700 mW + –1.5 dB 1W Subtracting the headroom restriction to obtain the average listening level without distortion yields: −1.5 dB − 15 dB = −16.5 (15 dB headroom) −1.5 dB − 12 dB = −13.5 (12 dB headroom) −1.5 dB − 9 dB = −10.5 (9 dB headroom) −1.5 dB − 6 dB = −7.5 (6 dB headroom) −1.5 dB − 3 dB = −4.5 (3 dB headroom) Converting dB back into watts: P W + 10 PdBń10 P ref + 22 mW (15 dB headroom) + 44 mW (12 dB headroom) + 88 mW (9 dB headroom) + 175 mW (6 dB headroom) + 350 mW (3 dB headroom) 28 POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 
    
       SLOS230D − NOVEMBER 1998 − REVISED OCTOBER 2002 APPLICATION INFORMATION headroom and thermal considerations (continued) This is valuable information to consider when attempting to estimate the heat dissipation requirements for the amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the system. Using the power dissipation curves for a 5-V, 8-Ω system, the internal dissipation in the TPA711 and maximum ambient temperatures is shown in Table 3. Table 3. TPA711 Power Rating, 5-V, 8-Ω, BTL D PACKAGE (SOIC) DGN PACKAGE (MSOP) MAXIMUM AMBIENT TEMPERATURE MAXIMUM AMBIENT TEMPERATURE 675 34°C 110°C 350 mW (3 dB) 595 47°C 115°C 176 mW (6 dB) 475 68°C 122°C 700 88 mW (9 dB) 350 89°C 125°C 700 44 mW (12 dB) 225 111°C 125°C PEAK OUTPUT POWER (mW) AVERAGE OUTPUT POWER POWER DISSIPATION (mW) 700 700 mW 700 700 Table 3 shows that the TPA711 can be used to its full 700-mW rating without any heat sinking in still air up to 110°C and 34°C for the DGN package (MSOP) and D package (SOIC) respectively. POST OFFICE BOX 655303 • DALLAS, TEXAS 75265 29 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) TPA711D ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 711 TPA711DG4 ACTIVE SOIC D 8 75 RoHS & Green NIPDAU Level-1-260C-UNLIM -40 to 85 711 TPA711DGN ACTIVE HVSSOP DGN 8 80 RoHS & Green NIPDAU | NIPDAUAG Level-1-260C-UNLIM -40 to 85 ABB TPA711DGNR ACTIVE HVSSOP DGN 8 2500 RoHS & Green NIPDAU | NIPDAUAG Level-1-260C-UNLIM -40 to 85 ABB TPA711DR ACTIVE SOIC D 8 2500 RoHS & Green Level-1-260C-UNLIM -40 to 85 711 NIPDAU (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of