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TPS23754, TPS23754-1, TPS23756
SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
TPS2375x IEEE 802.3at, High Efficiency, PoE Interface and DC-DC Controller
1
1 Features
•
•
•
•
•
•
•
•
•
•
The PoE interface features the new extended
hardware classification necessary for compatibility
with high-power midspan power sourcing equipment
(PSE) per IEEE 802.3at. The detection signature pin
can also be used to force power from the PoE source
off. Classification can be programmed to any of the
defined types with a single resistor.
Powers up to 30-W (Input) PDs
DC-DC Control Optimized for Isolated Converters
Supports High-Efficiency Topologies
Complete PoE Interface
Enhanced Classification per IEEE 802.3at With
Status Flag
Adapter ORing Support
Programmable Frequency With Synchronization
Robust 100-V, 0.5-Ω Hotswap MOSFET
–40°C to 125°C Junction Temperature Range
Industry Standard PowerPAD™ HTSSOP-20
The DC-DC controller features two complementary
gate drivers with programmable dead time. This
simplifies the design of active-clamp forward
converters or optimized gate drive for highly-efficient
flyback topologies. The second gate driver may be
disabled if desired for single MOSFET topologies.
The controller also features internal soft start,
bootstrap
start-up
source,
current-mode
compensation, and a 78% maximum duty cycle. A
programmable and synchronizable oscillator allows
design optimization for efficiency and eases use of
the controller to upgrade existing power supply
designs. Accurate programmable blanking, with a
default period, simplifies the usual current-sense filter
design trade-offs.
2 Applications
•
•
•
•
•
IEEE 802.3at Compliant Devices
Video and VoIP Telephones
RFID Readers
Multiband Access Points
Security Cameras
The TPS23754 device has a 15-V converter start-up
while the TPS23756 device has a 9-V converter startup. The TPS23754-1 replaces the PPD pin with a noconnect for increased pin spacing.
3 Description
The TPS23754 and TPS23756 devices have a
combined power-over-ethernet (PoE), powereddevice (PD) interface, and current-mode DC-DC
controller
optimized
specifically
for
isolated
converters. The PoE interface supports the IEEE
802.3at standard.
Device Information(1)
PART NUMBER
PACKAGE
BODY SIZE (NOM)
TPS23754
The TPS23754 and TPS23756 support a number of
input voltage ORing options including highest voltage,
external adapter preference, and PoE preference.
These features allow the designer to determine which
power source will carry the load under all conditions.
TPS23754-1
HTSSOP (20)
6.50 mm × 4.40 mm
TPS23756
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
From Ethernet
Pairs 1,2
High-Efficiency Converter Using TPS23754
CIO
VOUT
M4
DVC1
RT2P
VC
LVC
COUT
M3
DVC2
LOUT
M2
VB
RCTL
GAT2
VT2P-OUT
CCTL
ROB
CIZ
RFBU
CC
RT2P-OUT
Type 2 PSE
Indicator
M1
CVC
RCS
CCL
DCL
TLV431
RFBL
VB
GATE
CS
RBLNK
RFRS
RAPD2
Adapter
Optional
Interface
T2P
CVB
VSS
APD
CTL
FRS
COM
ARTN
RTN
DT
BLNK
RCLS
C1
From Ethernet
Pairs 3,4
D1
DEN
CLS
N/C OR PPD
PAD
DA
RAPD1
VDD
VDD1
RDEN
T1
CIN
RDT
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS23754, TPS23754-1, TPS23756
SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
7
1
1
1
2
3
4
Absolute Maximum Ratings ...................................... 4
ESD Ratings.............................................................. 4
Recommended Operating Conditions....................... 5
Thermal Information .................................................. 5
Electrical Characteristics........................................... 6
Electrical Characteristics: PoE and Control .............. 7
Switching Characteristics .......................................... 9
Typical Characteristics ............................................ 10
Detailed Description ............................................ 13
7.1 Overview ................................................................. 13
7.2 Functional Block Diagram ....................................... 13
7.3 Feature Description................................................. 13
7.4 Device Functional Modes........................................ 17
8
Application and Implementation ........................ 27
8.1 Application Information............................................ 27
8.2 Typical Application ................................................. 27
9 Power Supply Recommendations...................... 37
10 Layout................................................................... 37
10.1 Layout Guidelines ................................................. 37
10.2 Layout Example .................................................... 37
10.3 ESD....................................................................... 37
11 Device and Documentation Support ................. 38
11.1
11.2
11.3
11.4
11.5
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
38
38
38
38
38
12 Mechanical, Packaging, and Orderable
Information ........................................................... 38
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (September 2015) to Revision I
•
Page
Changed title to TPS2375x IEEE 802.3at, High Efficiency, PoE Interface and DC-DC Controller ........................................ 1
Changes from Revision G (October 2013) to Revision H
Page
•
Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device Functional
Modes, Application and Implementation section, Power Supply Recommendations section, Layout section, Device
and Documentation Support section, and Mechanical, Packaging, and Orderable Information section ............................... 1
•
Changed "Operating junction temperature" row to "Maximum junction temperature", and updated the MIN and MAX
values to Internally limited ...................................................................................................................................................... 4
2
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SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
5 Pin Configuration and Functions
PWP Package
20-Pin HTSSOP
Top View
TPS23754/6
CTL
VB
CS
COM
GATE
VC
GAT2
ARTN
RTN
VSS
20
19
18
17
16
15
14
13
12
11
1
2
3
4
5
6
7
8
9
10
TPS23754-1
T2P
FRS
BLNK
APD
DT
CLS
PPD
DEN
VDD
VDD1
CTL
VB
CS
COM
GATE
VC
GAT2
ARTN
RTN
VSS
PAD = V SS
20
19
18
17
16
15
14
13
12
11
1
2
3
4
5
6
7
8
9
10
T2P
FRS
BLNK
APD
DT
CLS
N/C
DEN
VDD
VDD1
PAD = V SS
N/C = Leave Pin Unused
Pin Functions
PIN
TPS23754
and
TPS23756
TPS23754-1
APD
17
17
I
ARTN
8
8
—
BLNK
18
18
I
Connect to ARTN to use the internally set current-sense blanking period, or connect
a resistor from BLNK to ARTN to program a more accurate period.
CLS
15
15
I
Connect a resistor from CLS to VSS to program classification current. 2.5 V is applied
to the program resistor during classification to set class current.
COM
4
4
—
Gate driver return, connect to ARTN and RTN.
CS
3
3
I/O
DC-DC converter switching MOSFET current sense input. See RCS in Figure 27.
CTL
1
1
I
The control loop input to the pulse-width modulator (PWM), typically driven by output
regulation feedback (for example, optocoupler). Use VB as a pullup for CTL.
DEN
13
13
I/O
Connect a 24.9-kΩ resistor from DEN to VDD to provide the PoE detection signature.
Pulling this pin to VSS during powered operation causes the internal hotswap
MOSFET to turn off.
DT
16
16
I
Connect a resistor from DT to ARTN to set the GATE to GAT2 dead time. Tie DT to
VB to disable GAT2 operation.
FRS
19
19
I
Connect a resistor from FRS to ARTN to program the converter switching frequency.
FRS may be used to synchronize the converter to an external timing source.
GATE
5
5
O
Gate drive output for the main DC-DC converter switching MOSFET.
GAT2
7
7
O
Gate drive output for a second DC-DC converter switching MOSFET (see Figure 27).
NC
—
14
—
Float this no-connect pin.
PAD
—
—
—
Connect to VSS.
PPD
14
—
I
RTN
9
9
—
RTN is the output of the PoE hotswap MOSFET.
T2P
20
20
O
Active low output that indicates a PSE has performed the IEEE 802.3at type 2
hardware classification, PPD is active, or APD is active.
Copyright © 2008–2017, Texas Instruments Incorporated
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NAME
TYPE
DESCRIPTION
Raising VAPD – VARTN above 1.5 V disables the internal hotswap switch, turns class
off, and forces T2P active. This forces power to come from a external VDD1-RTN
adapter. Tie APD to ARTN when not used.
ARTN is the DC-DC converter analog return. Tie to RTN and COM on the circuit
board.
Raising VPPD-VSS above 1.55 V enables the hotswap MOSFET and activates T2P.
Connecting PPD to VDD enables classification when APD is active. Tie PPD to VSS
or float when not used.
Product Folder Links: TPS23754 TPS23754-1 TPS23756
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TPS23754, TPS23754-1, TPS23756
SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
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Pin Functions (continued)
PIN
TPS23754
and
TPS23756
TPS23754-1
VB
2
2
O
5.1-V bias rail for DC-DC control circuits and the feedback optocoupler. Typically
bypass with a 0.1 μF to ARTN.
VC
6
6
I/O
DC-DC converter bias voltage. Connect a 0.47 μF (minimum) ceramic capacitor to
ARTN at the pin, and a larger capacitor to power start-up.
VDD
12
12
I
Connect to the positive PoE input power rail. VDD powers the PoE interface circuits.
Bypass with a 0.1-μF capacitor and protect with a TVS.
VDD1
11
11
I
Source of DC-DC converter start-up current. Connect to VDD for many applications.
VSS
10
10
—
NAME
TYPE
DESCRIPTION
Connect to the negative power rail derived from the PoE source.
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2). Voltage with respect to VSS unless otherwise noted.
MIN
MAX
ARTN (2), COM (2), DEN, PPD, RTN (3), VDD, VDD1
–0.3
100
CLS (4)
–0.3
6.5
–0.3
6.5
–0.3
VB
(4)
Input voltage
[APD, BLNK , CTL, DT
to [ARTN, COM]
(4)
(4)
(4)
, FRS , VB ]
CS to [ARTN,COM]
[ARTN, COM] to RTN
Voltage
UNIT
V
–2
2
VC, T2P, to [ARTN, COM]
–0.3
19
GATE (4), GAT2 (4) to [ARTN, COM]
–0.3
VC+ 0.3
V
Sinking current
RTN
Internally limited
mA
Sourcing current
VB
Internally limited
mA
Average Sourcing
or sinking current
GATE, GAT2
25
Maximum junction temperature, TJ(MAX)
Internally limited
Storage temperature, Tstg
(1)
(2)
(3)
(4)
mArms
–65
°C
150
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
ARTN and COM must be tied to RTN.
IRTN = 0 for VRTN > 80 V.
Do not apply voltage to these pins
6.2 ESD Ratings
VALUE
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
V(ESD)
(1)
(2)
(3)
4
Electrostatic
discharge
(1)
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
System level (contact/air) at RJ-45 (3)
UNIT
2000
500
V
8000 / 15000
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
ESD per EN61000-4-2. A power supply containing the TPS23754 was subjected to the highest test levels in the standard. See the ESD
section.
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SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted). Voltage with respect to VSS (unless otherwise noted) (1)
MIN
Input voltage range
UNIT
0
57
V
T2P, VC to [ARTN, COM]
0
18
V
APD, CTL, DT to [ARTN, COM]
0
VB
V
CS to [ARTN, COM]
0
2
V
(2)
VB
0
VB capacitance
2.5
825
mA
5
mA
0.08
RBLNK
μF
0
Synchronization pulse width input (when used)
Operating junction temperature range, TJ
(1)
(2)
MAX
ARTN, COM, PPD, RTN, VDD, VDD1
Continuous RTN current (TJ ≤ 125°C)
Sourcing current
NOM
350
kΩ
125
°C
25
ns
–40
ARTN and COM tied to RTN.
This is the minimum current limit value. Viable systems are designed for maximum currents less than this value with reasonable margin.
IEEE 802.3at permits 600-mA continuous loading.
6.4 Thermal Information
TPS23754,
TPS23754-1,
TPS23756
THERMAL METRIC (1)
UNIT
PWP (HTSSOP)
20 PINS
RθJA
Junction-to-ambient thermal resistance
44.4
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
23.4
°C/W
RθJB
Junction-to-board thermal resistance
20.1
°C/W
ψJT
Junction-to-top characterization parameter (2)
0.7
°C/W
ψJB
Junction-to-board characterization parameter
19.9
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
Thermal resistance junction to case top for devices mounted per SLMA002. TJ = TTOP + (ΨJT × PJ). Use ΨJT to validate TJ from
measurements.
Copyright © 2008–2017, Texas Instruments Incorporated
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6.5 Electrical Characteristics
Unless otherwise noted: CS = COM = APD = CTL = RTN = ARTN, GATE and GAT2 float, RFRS= 68.1 kΩ, RBLNK= 249 kΩ, DT
= VB, PPD = VSS, T2P open, CVB = CVC = 0.1 μF, RDEN = 24.9 kΩ, RCLS open, 0 V ≤ (VDD, VDD1) ≤ 57 V, 0 V ≤ VC ≤ 18 V,
–40°C ≤ TJ ≤ 125°C. Typical specifications are at 25°C.
CONTROLLER SECTION ONLY
[VSS = RTN and VDD = VDD1] or [VSS = RTN = VDD], all voltages referred to [ARTN, COM]
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
VC
VCUV
UVLO
VCUVH
Operating current
Bootstrap start-up time,
CVC = 22 μF
tST
Start-up current source is IVC
VC rising ‘754
14.3
15
15.7
VC rising ‘756
8.7
9
9.3
6.5
6.8
3.7
Hysteresis ‘754
(1)
6.2
Hysteresis ‘756
(1)
3.3
3.5
VC = 12 V, CTL = VB, RDT = 68.1 kΩ
0.7
0.92
1.2
TPS23756, VDD1 = 10.2 V, VC(0) = 0 V
50
85
175
TPS23756, VDD1 = 35 V, VC(0) = 0 V
27
45
92
TPS23754, VDD1 = 19.2 V, VC(0) = 0 V
49
81
166
TPS23754, VDD1 = 35 V, VC(0) = 0 V
44
75
158
TPS23754, VDD1 = 19.2 V, VC = 13.9 V
1.7
3.4
5.5
TPS23756, VDD1 = 10.2 V, VC = 8.6 V
0.44
1.06
1.8
TPS23754, TPS23756, VDD1 = 48 V, VC = 0 V
2.7
4.8
6.8
6.5 V ≤ VC ≤ 18 V, 0 ≤ IVB ≤ 5 mA
4.8
5.1
5.25
227
253
278
76%
78%
80%
2
2.2
2.4
V
mA
ms
mA
VB
Voltage
V
FRS
CTL = VB, measure GATE
Switching frequency
RFRS = 68.1 kΩ
DMAX
Duty cycle
CTL= VB, measure GATE
VSYNC
Synchronization
Input threshold
0% duty cycle threshold
VCTL ↓ until GATE stops
kHz
V
CTL
VZDC
Input resistance
1.3
1.5
1.7
V
70
100
145
kΩ
CS
VCSMAX
Maximum threshold voltage
VCTL = VB, VCS rising until GATE duty cycle drops
0.5
0.55
0.6
V
VSLOPE
Internal slope compensation
voltage
Peak voltage at maximum duty cycle, referenced to CS
120
155
185
mV
ISL_EX
Peak slope compensation
current
VCTL = VB, ICS at maximum duty cycle
30
42
54
μA
Bias current (sourcing)
DC component of ICS
1
2.5
4.3
μA
Source current
VCTL = VB, VC = 12 V, GATE high, pulsed measurement
0.37
0.6
0.95
A
Sink current
VCTL = VB, VC = 12 V, GATE low, pulsed measurement
0.7
1
1.4
A
Source current
VCTL = VB, VC = 12 V, GAT2 high, RDT = 24.9 kΩ, pulsed
measurement
0.37
0.6
0.95
A
Sink current
VCTL = VB, VC = 12 V, GAT2 low, RDT = 24.9 kΩ, pulsed
measurement
0.7
1
1.4
A
GATE
GAT2
(1)
6
The hysteresis tolerance tracks the rising threshold for a given device.
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SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
Electrical Characteristics (continued)
Unless otherwise noted: CS = COM = APD = CTL = RTN = ARTN, GATE and GAT2 float, RFRS= 68.1 kΩ, RBLNK= 249 kΩ, DT
= VB, PPD = VSS, T2P open, CVB = CVC = 0.1 μF, RDEN = 24.9 kΩ, RCLS open, 0 V ≤ (VDD, VDD1) ≤ 57 V, 0 V ≤ VC ≤ 18 V,
–40°C ≤ TJ ≤ 125°C. Typical specifications are at 25°C.
CONTROLLER SECTION ONLY
[VSS = RTN and VDD = VDD1] or [VSS = RTN = VDD], all voltages referred to [ARTN, COM]
PARAMETER
TEST CONDITIONS
MIN
TYP MAX
UNIT
APD / PPD
VAPDEN
VAPDH
VAPD rising
APD threshold voltage
Hysteresis
(1)
VPPDEN
VPPD- VVSS rising, UVLO disable
VPPDH
Hysteresis
VPPD2
PPD threshold voltage
VPPD- VVSS rising, Class enable
VPPD2H
IPPD
(1)
Hysteresis
(1)
1.43
1.5
1.57
0.29
0.31
0.33
1.45
1.55
1.65
0.29
0.31
0.33
7.4
8.3
9.2
0.5
0.6
0.7
APD leakage current
(source or sink)
VC = 12 V, VAPD = VB
PPD sink current
VPPD-VSS = 1.5 V
2.5
TJ rising
135
V
V
V
1
μA
5
7.5
μA
145
155
°C
THERMAL SHUTDOWN
turnon temperature
Hysteresis (2)
(2)
20
°C
These parameters are provided for reference only, and do not constitute part of TI's published specifications for purposes of TI's product
warranty.
6.6 Electrical Characteristics: PoE and Control
[VDD = VDD1] or [VDD1 = RTN], VC = RTN, COM = RTN = ARTN, all voltages referred to VSS unless otherwise noted
PARAMETER
TEST CONDITIONS
DETECTION (DEN)
MIN
TYP MAX
UNIT
(VDD = VDD1 = RTN = VSUPPLY positive)
Measure ISUPPLY
Detection current
62
64.3
66.5
VDD = 10 V
399
406
414
5.6
10
μA
4
5
V
0.1
5
μA
1.8
2.1
2.4
RCLS = 243 Ω
9.9
10.4
10.9
RCLS = 137 Ω
17.6
18.5
19.4
RCLS = 90.9 Ω
26.5
27.7
29.3
42
VDD = 10 V, float DEN, measure ISUPPLY,
Note: Not during Mark state
Detection bias current
VPD_DIS
VDD = 1.6 V
Hotswap disable threshold
3
DEN leakage current
VDEN = VDD = 57 V, float VDD1 and RTN, measure IDEN
CLASSIFICATION (CLS)
μA
(VDD = VDD1 = RTN = VSUPPLY positive)
13 V ≤ VDD ≤ 21 V, Measure ISUPPLY
RCLS = 1270 Ω
Classification current,
applies to both cycles
ICLS
mA
RCLS = 63.4 Ω
38
39.7
Classification mark resistance
5.6 V ≤ VDD ≤ 9.4 V
7.5
9.7
12
Classification regulator lower
threshold
Regulator turns on, VDD rising
11.2
11.9
12.6
Hysteresis (1)
1.55
1.65
1.75
Classification regulator upper
threshold
Regulator turns off, VDD rising
21
22
23
VCU_H
Hysteresis (1)
0.5
0.75
1
VMSR
Mark state reset
VDD falling
3
4
5
V
Leakage current
VDD = 57 V, VCLS = 0 V, DEN = VSS, measure ICLS
1
μA
VCL_ON
VCL_H
VCU_OFF
(1)
kΩ
V
V
The hysteresis tolerance tracks the rising threshold for a given device.
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Electrical Characteristics: PoE and Control (continued)
[VDD = VDD1] or [VDD1 = RTN], VC = RTN, COM = RTN = ARTN, all voltages referred to VSS unless otherwise noted
PARAMETER
TEST CONDITIONS
PASS DEVICE (RTN)
MIN
TYP MAX
UNIT
(VDD1 = RTN)
On resistance
0.75
Ω
0.25
0.43
Current limit
VRTN = 1.5 V, VDD = 48 V, pulsed measurement
850
970 1100
mA
Inrush limit
VRTN = 2 V, VDD: 0 V → 48 V, pulsed measurement
100
140
180
mA
Foldback voltage threshold
VDD rising
11
12.3
13.6
V
33.9
35
36.1
4.4
4.55
4.76
UVLO
VUVLO_R
VUVLO_H
VDD rising
UVLO threshold
Hysteresis
(1)
V
T2P
ON characteristic
Perform classification algorithm, VT2P-RTN = 1 V,
CTL = ARTN
Leakage current
VT2P = 18 V, CTL = VB
2
mA
10
μA
155
°C
THERMAL SHUTDOWN
Turnoff temperature
Hysteresis
(2)
8
TJ rising
135
(2)
145
20
°C
These parameters are provided for reference only, and do not constitute part of TI's published specifications for purposes of TI's product
warranty.
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6.7 Switching Characteristics
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
Interval from switching start to
VCSMAX
1.9
3.9
6.2
BLNK = RTN
35
55
78
RBLNK = 49.9 kΩ
38
55
70
RDT = 24.9 kΩ, GAT2 ↑ to GATE ↑
40
50
62.5
RDT = 24.9 kΩ, GATE ↓ to GAT2 ↓
40
50
62.5
UNIT
CTL
Soft-start period
ms
BLNK
Blanking delay
(In addition to t1)
ns
DT
CTL = VB, CGATE = 1 nF,
CGAT2 = 1 nF, measure GATE,
GAT2
tDT1
tDT2
Dead time
See Figure 1 for tDTx definition
ns
tDT1
RDT = 75 kΩ, GAT2 ↑ to GATE ↑
120
150
188
tDT2
RDT = 75 kΩ, GATE ↓ to GAT2 ↓
120
150
188
24
40
70
ns
5
9
15
ms
CS
t1
Turnoff delay
VCS = 0.65 V
PoE AND CONTROL - T2P
Delay
From start of switching to T2P active
GATE
tT2P
hi
50%
GAT2
lo
hi
50%
time
lo
tDT1
tDT2
Figure 1. GATE and GAT2 Timing and Phasing
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6.8 Typical Characteristics
970
8
Pulsed Current Measurement
7
960
PoE − Current Limit − mA
IVDD − Bias Current − µA
6
25°C
5
125°C
4
3
2
950
940
930
−40°C
920
1
910
−40
0
0
2
4
6
8
10
(VVDD − VVSS) − PoE Voltage − V
−20
0
20
40
60
G001
Figure 2. Detection Bias Current vs Voltage
100
120
G002
Figure 3. PoE Current Limit vs Temperature
160
160
CVC = 22 µF
VVDD1 = 10.2 V
140
140
CVC = 22 mF
VVDD1 = 19.2 V
Converter Start Time − ms
80
TJ − Junction Temperature − °C
120
Start Time − ms
120
100
80
VVDD1 = 35 V
100
80
60
60
40
40
VVDD1 = 35 V
20
−40
−20
0
20
40
60
80
100
TJ − Junction Temperature − °C
20
−40
120
−20
0
20
40
60
G003
120
Figure 5. '756 Converter Start Time vs Temperature
6
6
VVC = 13.9V
VVC = 8.6V
o
TJ = -40oC
TJ = -40 C
5
5
IVC − Source Current − mA
IVC − Source Current − mA
100
o
Figure 4. '754 Converter Start Time vs Temperature
o
TJ = 25 C
4
o
TJ = 125 C
3
2
1
o
TJ = 25 C
4
TJ = 125oC
3
2
1
0
0
5
10
80
TJ - Junction Temperature - C
10
15 20
25
30
35
40
45 50
55 60
5
10
15 20
25
30
35
40
45 50
55 60
VVDD1-RTN − V
VVDD1-RTN − V
Figure 6. '754 Converter Start-Up Current vs VVDD1
Figure 7. '756 Converter Start-Up Current vs VVDD1
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Typical Characteristics (continued)
3000
3500
GATE and GAT2 Open
VVC = 12 V
IC − Controller Bias Current − mA
2000
937 kHz
484 kHz
1500
245 kHz
100 kHz
1000
937 kHz
2500
484 kHz
2000
245 kHz
100 kHz
1500
1000
500
500
50 kHz
0
−40
VCTL = 0 V
50 kHz
VCTL = 0 V
0
−20
0
20
40
60
80
100
9
120
TJ - Junction Temperature - °C
10
11
12
13
14
15
16
17
G005
Figure 8. Controller Bias Current vs Temperature
G006
Figure 9. '754 Controller Bias Current vs Voltage
1200
600
4000
GATE, GAT2 open
o
TJ = 25 C
3500
RFRS = 34.6 kΩ (484 kHz)
Switching Frequency − kHz
3000
2500
480 kHz
2000
100 kHz
50 kHz
1100
500
960 kHz
1500
18
VC − Controller Bias Voltage − V
250 kHz
1000
400
1000
RFRS = 17.35 kΩ (937 kHz)
300
900
RFRS = 69.8 kΩ (245 kHz)
RFRS = 347 kΩ (50 kHz)
200
800
RFRS = 173 kΩ (100 kHz)
Switching Frequency − kHz
IVC − Sinking −mA
2500
VC − Bias Current − mA
GATE and GAT2 Open
TJ = 25°C
3000
700
100
500
VCTL = 0 V
0
6
8
10
12
14
16
0
−40
18
600
−20
VC − Controller Bias Voltage − V
0
20
40
60
80
100
120
TJ - Junction Temperature - °C
Figure 10. '756 Controller Bias Current vs Voltage
G007
Figure 11. Switching Frequency vs Temperature
79
1200
78
RFRS = 347 kW (50 kHz)
Ideal
77
Maximum Duty Cycle − %
Switching Frequency − kHz
1000
800
600
Typical
400
RFRS = 69.8 kW (245 kHz)
76
75
RFRS = 34.6 kW (484 kHz)
74
RFRS = 26.7 kW (623 kHz)
73
RFRS = 21.5 kW (766 kHz)
72
200
RFRS = 17.3 kW (937 kHz)
71
0
0
10
20
30
40
50
Programmed Resistance (106 / RFRS) − Ω−1
60
G008
Figure 12. Switching Frequency vs Program Conductance
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70
−40
−20
0
20
40
60
80
TJ - Junction Temperature - °C
100
120
G009
Figure 13. Maximum Duty Cycle vs Temperature
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Typical Characteristics (continued)
50
155
154
45
ISLOPE − µAPP
VSLOPE − mVPP
153
152
40
151
35
150
−20
0
20
40
60
80
100
30
−40
120
105
260
255
RBLNK = 100 kΩ
250
85
RBLNK = 249 kΩ
245
75
RBLNK = 49.9 kΩ
55
235
230
0
20
40
60
80
100
120
G011
450
18
400
14
350
10
300
6
250
2
200
−2
150
−6
100
−10
50
−14
240
RBLNK = RTN
−20
Blanking Period − ns
265
45
−40
20
Figure 15. Current Slope Compensation Current vs
Temperature
Blanking Period − ns
Blanking Period − ns
115
65
0
G010
Figure 14. Current Slope Compensation Voltage vs
Temperature
95
−20
TJ − Junction Temperature − °C
TJ − Junction Temperature − °C
40
60
80
100
0
120
0
TJ - Junction Temperature - °C
50
100
150
250
300
350
−18
400
RBLNK − kΩ
G012
Figure 16. Blanking Period vs Temperature
200
Difference From Computed − ns
149
−40
G013
Figure 17. Blanking Period vs Blanking Resistance (RBLNK)
11
900
800
10
Ideal
T2P Delay Time - ms
Dead Time - ns
700
600
500
400
Typical
300
200
9
8
7
100
0
0
50
100 150 200 250 300
Dead Time Resistance - kW
350
400
Figure 18. Dead Time vs Dead Time Resistance (RDT )
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6
-40
-20
0
20
40
60
80
Temperature - °C
100
120
Figure 19. T2P Delay Time vs Temperature
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7 Detailed Description
7.1 Overview
The TPS23754 and TPS23756 devices have a PoE that contains all of the features needed to implement an
IEEE802.3at type-2 powered device (PD) such as Detection, Classification, Type 2 Hardware Classification, and
140-mA inrush current limit during start-up. It combines a current mode DC-DC controller optimized specifically
for isolated converters.
The TPS23754 and TPS23756 devices integrate a low 0.5-Ω internal switch to allow for up to 0.85 A of
continuous current through the PD during normal operation.
The TPS23754 and TPS23756 devices contain several protection features such as thermal shutdown, current
limit foldback, and a robust 100-V internal switch.
7.2 Functional Block Diagram
VC
VDD1
f
f
Oscillator
FRS
CTL
CONV.
OFF
4ms
Softstart
50kW
Control
CLRB
ARTN 40mA
(pk)
DT
COM
GAT2
f
3.75kW
ARTN
t2
Converter
Thermal
Monitor
+
ss
CTL
-
0.55V
T2P Logic
Switch
Matrix
T2P
ARTN
BLNK
ARTN
VDD
GATE
Deadtime
D Q
CK
1
0.75V
CS
Ref
Global Cvtr.
Enable enb
enb
+
+
50kW
VB
Reg
uvlo, fpd
11V &
9V
pa, sa, den
2.5V
Class
Logic &
Regulator
CLS
uvlo
T2
State
Eng.
t2
22V &
21.25V
12.5V
& 1V
5V
& 4V
VSS
DEN
CONV.
OFF
400ms
35V &
30.5V
R Q
uvlo
S
7.8V
PPD
1.55V
&1.25V
H
L
1
ILIM
+
0
-
EN
pa
Common
Circuits and
PoE Thermal
Monitor
fpd
Hotswap
MOSFET
RTN
VSS
50mW
sa
den
1.5V
&1.2V
ARTN
APD
4V
7.3 Feature Description
See Figure 27 for component reference designators (RCS for example), and the Electrical Characteristics for
values denoted by reference (VCSMAX for example). Electrical characteristic values take precedence over any
numerical values used in the following sections.
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Feature Description (continued)
7.3.1 APD
APD forces power to come from an external adapter connected from VDD1 to RTN by opening the hotswap
switch, disabling the CLS output (see PPD pin description), and enabling the T2P output. TI recommends a
resistor divider on APD when it is connected to an external adapter. The divider provides ESD protection,
leakage discharge for the adapter ORing diode, and input voltage qualification. Voltage qualification assures the
adapter output voltage is high enough that it can support the PD before the PoE current is cut off.
Select the APD divider resistors per Equation 1 where VADPTR-ON is the desired adapter voltage that enables the
APD function as adapter voltage rises.
RAPD1 = RAPD2 ´
VADPTR_OFF =
(VADPTR_ON
R APD1 + R APD2
R APD2
- VAPDEN
´
(VAPDEN
) VAPDEN
- VAPDH )
(1)
Place the APD pulldown resistor adjacent to the APD pin.
APD should be tied to ARTN when not used.
7.3.2 BLNK
Blanking provides an interval between GATE going high and the current-control comparators on CS actively
monitoring the input. This delay allows the normal turnon current transient (spike) to subside before the
comparators are active, preventing undesired short duty cycles and premature current limiting.
Connect BLNK to ARTN to obtain the internally set blanking period. Connect a resistor from BLNK to ARTN for a
more accurate, programmable blanking period. The relationship between the desired blanking period and the
programming resistor is defined by Equation 2.
RBLNK (kW ) = tBLNK (ns )
(2)
Place the resistor adjacent to the BLNK pin when it is used.
7.3.3 CLS
A resistor from CLS to VSS programs the classification current per the IEEE standard. The PD power ranges and
corresponding resistor values are listed in Table 1. The power assigned should correspond to the maximum
average power drawn by the PD during operation.
High-power PSEs may perform two classification cycles if Class 4 is presented on the first cycle. The TPS23754
device presents the same (resistor programmed) class each cycle per the standard.
Table 1. Class Resistor Selection
POWER AT PD
RESISTOR
(Ω)
CLASS
MINIMUM
(W)
MAXIMUM
(W)
0
0.44
13
1270
1
0.44
3.84
243
2
3.84
6.49
137
3
6.49
13
90.9
4
13
25.5
63.4
NOTES
Minimum may be reduced by pulsed loading. Serves as a catch-all default class.
Not allowed before IEEE 802.3at. Use to indicate a Type 2 PD (high power)
device per IEEE 802.3at.
7.3.4 Current Sense (CS)
The CS input for the DC-DC converter should be connected to the high side of the switching MOSFET’s current
sense resistor (RCS). The current limit threshold, VCSMAX, defines the voltage on CS above which the GATE ON
time will be terminated regardless of the voltage on CTL.
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The TPS23754 device provides internal slope compensation (150 mV, VSLOPE), an output current for additional
slope compensation, a peak current limiter, and an off-time pulldown to this pin.
Routing between the current-sense resistor and the CS pin should be short to minimize cross-talk from noisy
traces such as the gate drive signal.
7.3.5 Control (CTL)
CTL is the voltage-control loop input to the pulse-width modulator (PWM). Pulling VCTL below VZDC causes GATE
to stop switching. Increasing VCTL above VZDC (0 duty cycle voltage) raises the switching MOSFET programmed
peak current. The maximum (peak) current is requested at approximately VZDC + (2 × VCSMAX). The AC gain from
CTL to the PWM comparator is 0.5. The internal divider from CTL to ARTN is approximately 100 kΩ.
Use VB as a pullup source for CTL.
7.3.6 Detection and Enable (DEN)
DEN is a multifunction pin for PoE detection and inhibiting operation from PoE power. Connect a 24.9-kΩ resistor
from DEN to VDD to provide the PoE detection signature. DEN goes to a high-impedance state when VVDD-VSS is
outside of the detection range. Pulling DEN to VSS during powered operation causes the internal hotswap
MOSFET and class regulator to turn off, while the reduced detection resistance prevents the PD from properly
redetecting.
7.3.7 DT
Dead-time programming sets the delay between GATE and GAT2 to prevent overlap of MOSFET ON times as
shown in Figure 1. GAT2 turns the second MOSFET off when it transitions high. Both MOSFETs should be off
between GAT2 going high to GATE going high, and GATE going low to GAT2 going low. The maximum GATE
ON time is reduced by the programmed dead-time period. The dead time period is specified with 1 nF of
capacitance on GATE and GAT2. Different loading on these pins changes the effective dead time.
A resistor connected from DT to ARTN sets the delay between GATE and GAT2 per Equation 3.
RDT (kW ) =
tDT (ns )
2
(3)
Connect DT to VB to set the dead time to 0 and turn GAT2 off.
7.3.8 Frequency and Synchronization (FRS)
Connect a resistor from FRS to ARTN to program the converter switching frequency. Select the resistor per the
following relationship.
RFRS (kW) =
17250
fSW (kHz)
(4)
The converter may be synchronized to a frequency above its maximum free-running frequency by applying short
AC-coupled pulses into the FRS pin per Figure 30.
The FRS pin is high impedance. Keep the connections short and apart from potential noise sources. Take
special care to avoid crosstalk when synchronizing circuits are used.
7.3.9 GATE
Gate drive output for the DC-DC converter’s main switching MOSFET. GATE’s phase turns the main switch on
when it transitions high, and off when it transitions low. GATE is held low when the converter is disabled.
7.3.10 GAT2
GAT2 is the second gate drive output for the DC-DC converter. GAT2’s phase turns the second switch off when
it transitions high, and on when it transitions low. This drives active-clamp PMOS devices per Figure 27, and
driven flyback synchronous rectifiers per Figure 27. See the DT pin description for GATE to GAT2 timing.
Connecting DT to VB disables GAT2 in a high-impedance condition. GAT2 is low when the converter is disabled.
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7.3.11 PPD
PPD is a multifunction pin that has two voltage thresholds, PPD1 and PPD2.
PPD1 permits power to come from an external low voltage adapter, that is, 24 V, connected from VDD to VSS by
overriding the normal hotswap UVLO. Voltage on PPD more than 1.55 V (VPPDEN) enables the hotswap
MOSFET, inhibits class current, and enables T2P. A resistor divider per Figure 34 provides ESD protection,
leakage discharge for the adapter ORing diode, reverse adapter protection, and input voltage qualification.
Voltage qualification assures the adapter output voltage is high enough that it can support the PD before it
begins to draw current.
æ
ö
çV
÷
V
ADPTR_ON
PPDEN ÷
RPPD1 = ç
VPPDEN
ç
÷
+ IPPD
ç
÷
R
PPD2
è
ø
é
æ (VPPDEN - VPPDH )
öù
VADPTR_OFF = (VPPDEN - VPPDH )+ êRPPD1 ´ ç
+ IPPD ÷ ú
ç
÷
RPPD2
è
ø ûú
ëê
(5)
PPD2 enables normal class regulator operation when VPPD is more than 8.3 V to permit type 2 classification
when APD is used in conjunction with diode DVDD (see Figure 33). Tie PPD to VDD when PPD2 operation is
desired.
The PPD pin has a 5-μA internal pulldown current.
Locate the PPD pulldown resistor adjacent to the pin when used.
PPD may be tied to VSS or left open when not used.
7.3.12 RTN, ARTN, COM
RTN is internally connected to the drain of the PoE hotswap MOSFET, while ARTN is the quiet analog reference
for the DC-DC controller return. COM serves as the return path for the gate drivers and should be tied to ARTN
on the circuit board. The ARTN / COM / RTN net should be treated as a local reference plane (ground plane) for
the DC-DC control and converter primary. RTN and (ARTN/COM) may be separated by several volts for special
applications.
7.3.13 T2P
T2P is an active low output that indicates [ (VAPD > 1.5 V) OR (1.55 V ≤ VPPD ≤ 8.3 V) OR (type 2 hardware
classification observed) ]. T2P is valid after both a delay of tT2P from the start of converter switching, and [VCTL ≤
(VB – 1 V)]. Once T2P is valid, VCTL does not affect it. T2P becomes invalid if the converter goes back into soft
start, overtemperature, or is held off by the PD during CIN recharge (inrush). T2P is referenced to ARTN and is
intended to drive the diode side of an optocoupler. T2P should be left open or tied to ARTN if not used.
7.3.14 VB
VB is an internal 5.1-V regulated DC-DC controller supply rail that is typically bypassed by a 0.1-μF capacitor to
ARTN. VB should be used to bias the feedback optocoupler.
7.3.15 VC
VC is the bias supply for the DC-DC controller. The MOSFET gate drivers run directly from VC. VB is regulated
down from VC, and is the bias voltage for the rest of the converter control. A start-up current source from VDD1 to
VC is controlled by a comparator with hysteresis to implement the converter bootstrap start-up. VC must be
connected to a bias source, such as a converter auxiliary output, during normal operation.
A minimum 0.47 μF-capacitor, located adjacent to the VC pin, should be connected from VC to COM to bypass
the gate driver. A larger total capacitance is required for start-up to provide control power between the time the
converter starts switching and the availability of the converter auxiliary output voltage.
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7.3.16 VDD
VDD is the positive input power rail that is derived from the PoE source (PSE). VDD should be bypassed to VSS
with a 0.1-μF capacitor as required by the IEEE standard. A transient suppressor diode (TVS), a special type of
Zener diode, such as SMAJ58A should be connected from VDD to VSS to protect against overvoltage transients.
7.3.17 VDD1
VDD1 is the DC-DC converter start-up supply. Connect to VDD for many applications. VDD1 may be isolated by a
diode from VDD to support PoE priority operation.
7.3.18 VSS
VSS is the PoE input-power return side. It is the reference for the PoE interface circuits and has a current limited
hotswap switch that connects it to RTN. VSS is clamped to a diode drop above RTN by the hotswap switch.
A local VSS reference plane should be used to connect the input bypass capacitor, TVS, RCLS, and the
PowerPAD. This plane becomes the main heatsink for the TPS23754.
VSS is internally connected to the PowerPAD.
7.3.19 PowerPAD
The PowerPAD is internally connected to VSS. It should be tied to a large VSS copper area on the PCB to provide
a low-resistance thermal path to the circuit board. TI recommends that a clearance of 0.025” be maintained
between VSS, RTN, and various control signals to high-voltage signals such as VDD and VDD1.
7.4 Device Functional Modes
The following text is intended as an aid in understanding the operation of the TPS23754, but not as a substitute
for the IEEE 802.3at standard. The IEEE 802.3at standard is an update to IEEE 802.3-2008 clause 33 (PoE),
adding high-power options and enhanced classification. Generally speaking, a device compliant to IEEE 802.32008 is referred to as a type 1 device, and devices with high power and enhanced classification are referred to
as type 2 devices. Standards change and should always be referenced when making design decisions.
7.4.1 PoE Overview
The IEEE 802.3at standard defines a method of safely powering a PD over a cable by power sourcing equipment
(PSE), and then removing power if a PD is disconnected. The process proceeds through an idle state and three
operational states of detection, classification, and operation. The PSE leaves the cable unpowered (idle state)
while it periodically looks to see if something has been plugged in; this is referred to as detection. The low-power
levels used during detection are unlikely to damage devices not designed for PoE. If a valid PD signature is
present, the PSE my inquire how much power the PD requires; this is referred to as classification. The PSE may
then power the PD if it has adequate capacity.
Type 2 PSEs are required to do type 1 hardware classification, plus a (new) data-layer classification, or an
enhanced type 2 hardware classification. Type 1 PSEs are not required to do hardware or data link layer (DLL)
classification. A type 2 PD must do type 2 hardware classification as well as DLL classification. The PD may
return the default, 13W current-encoded class, or one of four other choices. DLL classification occurs after
power-on and the ethernet data link has been established.
Once started, the PD must present the maintain power signature (MPS) to assure the PSE that it is still present.
The PSE monitors its output for a valid MPS, and turns the port off if it loses the MPS. Loss of the MPS returns
the PSE to the idle state. Figure 20 shows the operational states as a function of PD input voltage. The upper
half is for IEEE 802.3-2008, and the lower half shows specific differences for IEEE 802.3at. The dashed lines in
the lower half indicate these are the same (for example, Detect and Class) for both.
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Maximum Input
Voltage
Must Turn On byVoltage Rising
Shutdown
Classify
Detect
6.9
Lower Limit Operating Range
Must Turn Off by Voltage Falling
Classification
Upper Limit
Classification
Lower Limit
Detection
Upper Limit
Detection
Lower Limit
IEEE 802.3-2005
Device Functional Modes (continued)
Normal Operation
42.5
0
20.5
30
37
Lower Limit 13W Op.
Class-Mark
Transition
57 PI Voltage (V)
42
Normal Operation
250ms
Transient
10.1 14.5
Mark
T2 Reset
Range
IEEE 802.3at
2.7
Figure 20. Operational States for PD
The PD input, typically an RJ-45 eight-lead connector, is referred to as the power interface (PI). PD input
requirements differ from PSE output requirements to account for voltage drops and operating margin. The
standard allots the maximum loss to the cable regardless of the actual installation to simplify implementation.
IEEE 802.3-2008 was designed to run over infrastructure including ISO/IEC 11801 class C (CAT3 per TIA/EIA568) that may have had AWG 26 conductors. IEEE 802.3at type 2 cabling power loss allotments and voltage
drops have been adjusted for 12.5-Ω power loops per ISO/IEC11801 class D (CAT5 or higher per TIA/EIA-568,
typically AWG number 24 conductors). Table 2 shows key operational limits broken out for the two revisions of
the standard.
Table 2. Comparison of Operational Limits
STANDARD
POWER LOOP
RESISTANCE
(max)
PSE
OUTPUT POWER
(min)
PSE STATIC
OUTPUT VOLTAGE
(min)
PD INPUT
POWER
(max)
POWER ≤
13 W
STATIC PD INPUT VOLTAGE
POWER >
13 W
IEEE 802.3-2008
802.3at (Type 1)
20 Ω
15.4 W
44 V
13 W
37 – 57 V
N/A
802.3at (Type 2)
12.5 Ω
30 W
50 V
25.5 W
37 – 57 V
42.5 – 57 V
The PSE can apply voltage either between the RX and TX pairs (pins 1 to 2 and 3 to 6 for 10baseT or
100baseT), or between the two spare pairs (4 to 5 and 7 to 8). Power application to the same pin combinations
in 1000baseT systems is recognized in IEEE 802.3at. 1000baseT systems can handle data on all pairs,
eliminating the spare pair terminology. The PSE may only apply voltage to one set of pairs at a time. The PD
uses input diode bridges to accept power from any of the possible PSE configurations. The voltage drops
associated with the input bridges create a difference between the standard limits at the PI and the TPS23754
specifications.
A compliant type 2 PD has power management requirements not present with a type 1 PD. These requirements
include the following:
1. Must interpret type 2 hardware classification
2. Must present hardware class 4
3. Must implement DLL negotiation
4. Must behave like a type 1 PD during inrush and start-up
5. Must not draw more than 13 W for 80 ms after PSE applies operating voltage (power up)
6. Must not draw more than 13 W if it has not received a type 2 hardware classification or received permission
through DLL
7. Must meet various operating and transient templates
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8. Optionally monitor for the presence or absence of an adapter (assume high power)
As a result of these requirements, the PD must be able to dynamically control its loading and monitor T2P for
changes. In cases where the design must know specifically if an adapter is plugged in and operational, the
adapter should be individually monitored, typically with an optocoupler.
7.4.1.1 Threshold Voltages
The TPS23754 has a number of internal comparators with hysteresis for stable switching between the various
states. Figure 21 relates the parameters in the Electrical Characteristics to the PoE states. The mode labeled idle
between classification and operation implies that the DEN, CLS, and RTN pins are all high impedance. The state
labeled Mark, which is drawn in dashed lines, is part of the new type 2 hardware class state machine.
Functional
State
PD Powered
Idle
Classification
Mark
VDD-VSS
Detection
VCU_H
VCL_H
VMSR
VUVLO_H
VCU_OFF
VCL_ON
VUVLO_R
Note: Variable names refer to Electrical Characteristic
Table parameters
Figure 21. Threshold Voltages
7.4.1.2 PoE Start-Up Sequence
50 mA/div
The waveforms of Figure 22 demonstrate detection, classification, and start-up from a PSE with type 2 hardware
classification. The key waveforms shown are VVDD-VVSS, VRTN-VVSS, and IPI. IEEE 802.3at requires a minimum of
two detection levels, two class and mark cycles, and start-up from the second mark event. VRTN to VSS falls as
the TPS23754 charges CIN following application of full voltage. Subsequently, the converter starts up, drawing
current as seen in the IPI waveform.
Cvtr. Starts
Inrush
IPI
Class
VVDD-VSS
Mark
10 V/div
Detect
VRTN-VSS
t - Time - 25 ms/div
Figure 22. Start-Up
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7.4.1.3 Detection
The TPS23754 drives DEN to VSS whenever VVDD-VVSS is below the lower classification threshold. When the
input voltage rises above VCL-ON, the DEN pin goes to an open-drain condition to conserve power. While in
detection, RTN is high impedance, and almost all the internal circuits are disabled. An RDEN of 24.9 kΩ (1%),
presents the correct signature. It may be a small, low-power resistor because it only sees a stress of about 5
mW. A valid PD detection signature is an incremental resistance ( ΔV / ΔI ) from 23.7 kΩ to 26.3 kΩ at the PI.
The detection resistance seen by the PSE at the PI is the result of the input bridge resistance in series with the
parallel combination of RDEN and internal VDD loading. The input diode bridge’s incremental resistance may be
hundreds of ohms at the very low currents drawn when 2.7 V is applied to the PI. The input bridge resistance is
partially cancelled by the TPS23754's effective resistance during detection.
The type 2 hardware classification protocol of IEEE 802.3at specifies that a type 2 PSE drops its output voltage
into the detection range during the classification sequence. The PD is required to have an incorrect detection
signature in this condition, which is referred to as the mark event (see Figure 22). After the first mark event, the
TPS23754 will present a signature less than 12 kΩ until it has experienced a VVDD-VVSS voltage below the mark
reset (VMSR). This is explained more fully in Hardware Classification.
7.4.1.4 Hardware Classification
Hardware classification allows a PSE to determine a PD’s power requirements before powering, and helps with
power management once power is applied. Type 2 hardware classification permits high power PSEs and PDs to
determine whether the connected device can support high-power operation. A type 2 PD presents class 4 in
hardware to indicate it is a high-power device. A type 1 PSE will treat a class 4 device like a class 0 device,
allotting 13 W if it chooses to power the PD. A PD that receives a 2 event class understands that it is powered
from a high-power PSE and it may draw up to 25.5 W immediately after the 80-ms start-up period completes. A
type 2 PD that does not receive a 2-event hardware classification may choose to not start, or must start in a 13
W condition and request more power through the DLL after start-up. The standard requires a type 2 PD to
indicate that it is underpowered if this occurs. Start-up of a high-power PD under 13 W implicitly requires some
form of powering down sections of the application circuits.
The maximum power entries in Table 1 determine the class the PD must advertise. The PSE may disconnect a
PD if it draws more than its stated class power, which may be the hardware class or a lower DLL-derived power
level. The standard permits the PD to draw limited current peaks that increase the instantaneous power above
the Table 1 limit; however, the average power requirement always applies.
The TPS23754 implements two-event classification. Selecting an RCLS of 63.4 Ω provides a valid type 2
signature. TPS23754 may be used as a compatible type 1 device simply by programming class 0–3 per Table 1.
DLL communication is implemented by the ethernet communication system in the PD and is not implemented by
the TPS23754.
The TPS23754 disables classification above VCU_OFF to avoid excessive power dissipation. CLS voltage is turned
off during PD thermal limit or when APD or DEN are active. The CLS output is inherently current limited, but
should not be shorted to VSS for long periods of time.
Figure 23 shows how classification works for the TPS23754. Transition from state-to-state occurs when
comparator thresholds are crossed (see Figure 20 and Figure 21). These comparators have hysteresis, which
adds inherent memory to the machine. Operation begins at idle (unpowered by PSE) and proceeds with
increasing voltage from left to right. A 2-event classification follows the (heavy-lined) path towards the bottom,
ending up with a latched type 2 decode along the lower branch that is highlighted. This state results in a low T2P
during normal operation. Once the valid path to type 2 PSE detection is broken, the input voltage must transition
below the mark reset threshold to start anew.
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Mark
Reset
Idle
Class
UVLO
Falling
Class
Between
Ranges
Mark
Class
Between
Ranges
Mark
Class
Between
Ranges
Mark
Detect
Mark
Reset
TYPE 2 PSE
Hardware Class
UVLO
Rising
Operating
T2P
open-drain
TYPE 1 PSE
Hardware Class
UVLO
Rising
Operating
T2P low
UVLO
Falling
Figure 23. Two-Event Class Internal States
7.4.1.5 Inrush and Start-Up
802.3at has a start-up current and time limitation, providing type 2 PSE compatibility for type 1 PDs. A type 2
PSE limits output current to from 400 mA to 450 mA for up to 75 ms after power up (applying 48 V to the PI) to
mirror type 1 PSE functionality. The type 2 PSE supports higher output current after 75 ms. The TPS23754
implements a 140-mA inrush current, which is compatible with all PSE types. A high-power PD must control its
converter start-up peak and operational currents drawn to less than 400 mA for 80 ms. The TPS23754 device's
internal soft-start permits control of the converter start-up; however, the application circuits must assure that their
power draw does not cause the PD to exceed the current/time limitation. This requirement implicitly requires
some form of powering down sections of the application circuits. T2P becomes valid within tT2P after switching
starts, or if an adapter is plugged in while the PD is operating from a PSE.
7.4.1.6 Maintain Power Signature
The MPS is an electrical signature presented by the PD to assure the PSE that it is still present after operating
voltage is applied. A valid MPS consists of a minimum DC current of 10 mA (or a 10-mA pulsed current for at
least 75 ms every 325 ms) and an AC impedance lower than 26.3 kΩ in parallel with 0.05 μF. The AC impedance
is usually accomplished by the minimum operating CIN requirement of 5 μF. When either APD or DEN is used to
force the hotswap switch off, the DC MPS will not be met. A PSE that monitors the DC MPS will remove power
from the PD when this occurs. A PSE that monitors only the AC MPS may remove power from the PD.
7.4.1.7 Start-Up and Converter Operation
The internal PoE undervoltage lockout (UVLO) circuit holds the hotswap switch off before the PSE provides full
voltage to the PD. This prevents the converter circuits from loading the PoE input during detection and
classification. The converter circuits discharge CIN, CVC, and CVB while the PD is unpowered. Thus VVDD-VRTN will
be a small voltage just after full voltage is applied to the PD, as seen in Figure 22. The PSE drives the PI voltage
to the operating range once it has decided to power up the PD. When VVDD rises above the UVLO turnon
threshold (VUVLO-R, about 35 V) with RTN high, the TPS23754 device enables the hotswap MOSFET with an
approximately 140-mA (inrush) current limit as seen in Figure 24. Converter switching is disabled while CIN
charges and VRTN falls from VVDD to nearly VVSS, however the converter start-up circuit is allowed to charge CVC
(the bootstrap start-up capacitor). Additional loading applied between VVDD and VRTN during the inrush state may
prevent successful PD and subsequent converter start-up. Converter switching is allowed if the PD is not in
inrush, OTSD is not active, and the VC UVLO permits it. Once the inrush current falls about 10% less than the
inrush current limit, the PD current limit switches to the operational level (about 970 mA). Continuing the start-up
sequence shown in Figure 24, VVC continues to rise until the start-up threshold (VCUV, about 15 V or about 9 V) is
exceeded, turning the start-up source off and enabling switching. The VB regulator is always active, powering the
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internal converter circuits as VVC rises. There is a slight delay between the removal of charge current and the
start of switching as the soft-start ramp sweeps above the VZDC threshold. VVC falls as it powers both the internal
circuits and the switching MOSFET gates. If the converter control bias output rises to support VVC before it falls
to VCUV – VCUVH (about 8.5 V or about 5.5 V), a successful start-up occurs. T2P in Figure 22 (Figure 27, VT2P-OUT)
becomes active within tT2P from the start of switching, indicating that a type 2 PSE or an adapter is plugged in.
10
5 V/div
99
88
200 mA/div
T2P @ output
Inrush
I PI
7
66
10 V/div
5
PI Powered
V C -RTN
Switching starts
44
2 V/div
33
VOUT
2
11
50 V/div
V DD -RTN
0
t - Time - 10 ms/div
Figure 24. Power Up and Start
If VVDD- VVSS drops below the lower PoE UVLO (VUVLO-R - VUVLO-H, about 30.5 V), the hotswap MOSFET is turned
off, but the converter will still run. The converter will stop if VVC falls below the converter UVLO (VCUV – VCUVH,
about 8.5 V or about 5.5 V), the hotswap is in inrush current limit, 0% duty cycle is demanded by VCTL (VCTL <
VZDC, about 1.5 V), or the converter is in thermal shutdown.
7.4.1.8 PD Hotswap Operation
IEEE 802.3at has taken a new approach to PSE output limiting. A type 2 PSE must meet an output current
versus time template with specified minimum and maximum sourcing boundaries. The peak output current may
be as high as 50 A for 10 μs or 1.75 A for 75 ms. This makes robust protection of the PD device even more
important than it was in IEEE 802.3-2008.
The internal hotswap MOSFET is protected against output faults and input voltage steps with a current limit and
deglitched (time-delay filtered) foldback. An overload on the pass MOSFET engages the current limit, with VRTNVVSS rising as a result. If VRTN rises above about 12 V for longer than about 400 μs, the current limit reverts to
the inrush value, and turns the converter off. The 400 μs deglitch feature prevents momentary transients from
causing a PD reset, provided that recovery lies within the bounds of the hotswap and PSE protection. Figure 25
shows an example of recovery from a 16 V PSE rising voltage step. The hotswap MOSFET goes into current
limit, overshooting to a relatively low current, recovers to about 950-mA full current limit, and charges the input
capacitor while the converter continues to run. The MOSFET did not go into foldback because VRTN-VVSS was
less than 12 V after the 400 μs deglitch.
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500 mA/div
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I PI
10 V/div
CIN completes charge
while converter operates
V RTN-VSS
20 V/div
16 V Input step
VRTN < 12 V @ 400 ms
Recovery from PI dropout
V VDD-VSS
t - Time - 200 ms/div
Figure 25. Response to PSE Step Voltage
The PD control has a thermal sensor that protects the internal hotswap MOSFET. Conditions like start-up or
operation into a VDD to RTN short cause high power dissipation in the MOSFET. An overtemperature shutdown
(OTSD) turns off the hotswap MOSFET and class regulator, which are restarted after the device cools. The
hotswap MOSFET will be re-enabled with the inrush current limit when exiting from an overtemperature event.
Pulling DEN to VSS during powered operation causes the internal hotswap MOSFET to turn off. This feature
allows a PD with Option three ORing per Figure 26 to achieve adapter priority. Take care with synchronous
converter topologies that can deliver power in both directions.
The hotswap switch will be forced off under the following conditions:
1. VAPD above VAPDEN (about 1.5 V)
2. VDEN < VPD-DIS when VVDD– VVSS is in the operational range
3. PD over-temperature
4. (VVDD– VVSS) < PoE UVLO (about 30.5 V)
7.4.1.9 Converter Controller Features
The TPS23754 DC-DC controller implements a typical current-mode control as shown in the Functional Block
Diagram. Features include oscillator, overcurrent and PWM comparators, current-sense blanker, dead-time
control, soft start, and gate driver. In addition, an internal slope-compensation ramp generator, frequency
synchronization logic, thermal shutdown, and start-up current source with control are provided.
The TPS23754 is optimized for isolated converters, and does not provide an internal error amplifier. Instead, the
optocoupler feedback is directly fed to the CTL pin which serves as a current-demand control for the PWM.
There is an offset of VZDC (about 1.5 V) and 2:1 resistor divider between the CTL pin and the PWM. A VCTL below
VZDC will stop converter switching, while voltages above (VZDC + (2 × VCSMAX)) will not increase the requested
peak current in the switching MOSFET. Optocoupler biasing design is eased by this limited control range.
7.4.1.10 Bootstrap Topology
The internal start-up current source and control logic implement a bootstrap-type start-up as discussed in StartUp and Converter Operation. The start-up current source charges CVC from VDD1 when the converter is disabled
(either by the PD control or the VC control) to store enough energy to start the converter. Steady-state operating
power must come from a converter (bias winding) output or other source. Loading on VC and VB must be minimal
while CVC charges, otherwise the converter may never start. The optocoupler will not load VB when the converter
is off for most situations; however take care in ORing topologies where the output is powered when PoE is off.
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The converter will shut off when VC falls below its lower UVLO. This can happen when power is removed from
the PD, or during a fault on a converter output rail. When one output is shorted, all the output voltages fall
including the one that powers VC. The control circuit discharges VC until it hits the lower UVLO and turns off. A
restart will initiate, as described in Start-Up and Converter Operation, if the converter turns off and there is
sufficient VDD1 voltage. This type of operation is sometimes referred to as hiccup mode, which provides robust
output short protection by providing time-average heating reduction of the output rectifier.
The bootstrap control logic disables most of the converter controller circuits except the VB regulator and internal
reference. Both GATE and GAT2 (assuming GAT2 is enabled) will be low when the converter is disabled. FRS,
BLNK, and DT will be at ARTN while the VC UVLO disables the converter. While the converter runs, FRS, BLNK,
and DT will be about 1.25 V.
The start-up current source transitions to a resistance as (VVDD1 – VVC) falls less than 7 V, but will start the
converter from adapters within tST. The lower test voltage for tST was chosen based on an assumed adapter
tolerance, but is not meant to imply a hard cutoff exists. start-up takes longer and eventually will not occur as
VDD1 decreases below the test voltage. The bootstrap source provides reliable start-up from widely varying input
voltages, and eliminates the continual power loss of external resistors. The start-up current source will not charge
above the maximum recommended VVC if the converter is disabled and there is sufficient VDD1 to charge higher.
7.4.1.11 Current Slope Compensation and Current Limit
Current-mode control requires the addition of a compensation ramp to the sensed inductive (transformer or
inductor) current for stability at duty cycles near and over 50%. The TPS23754 device has a maximum duty cycle
limit of 78%, permitting the design of wide input-range flyback and active clamp converters with a lower voltage
stress on the output rectifiers. While the maximum duty cycle is 78%, converters may be designed that run at
duty cycles well below this for a narrower, 36-V to 57-V PI range. The TPS23754 device provides a fixed internal
compensation ramp that suffices for most applications.
The TPS23754 device provides internal, frequency independent, slope compensation (150 mV, VSLOPE) to the
PWM comparator input for current-mode control-loop stability. This voltage is not applied to the current limit
comparator whose threshold is 0.55 V (VCSMAX). If the provided slope is not sufficient, the effective slope may be
increased by addition of RS per Figure 31. The additional slope voltage is provided by (ISL-EX × RS). There is also
a small DC offset caused by the about 2.5-μA pin current. The peak current limit does not have duty cycle
dependency unless RS is used. This makes it easier to design the current limit to a fixed value. See Current
Slope Compensation for more information.
The internal comparators monitoring CS are isolated from the IC pin by the blanking circuits while GATE is low,
and for a short time (blanking period) just after GATE switches high. A 440 Ω (maximum) equivalent pulldown on
CS is applied while GATE is low.
7.4.1.12 Blanking – RBLNK
The TPS23754 device provides a choice between internal fixed and programmable blanking periods. The
blanking period is specified as an increase in the minimum GATE on time over the inherent gate driver and
comparator delays. The default period (see the Electrical Characteristics) is selected by connecting BLNK to
RTN, and the programmable period is set with RBLNK.
The TPS23754 device blanker timing is precise enough that the traditional R-C filters on CS can be eliminated.
This avoids current-sense waveform distortion, which tends to get worse at light output loads. There may be
some situations or designers that prefer an R-C approach. The TPS23754 device provides a pulldown on CS
during the GATE off time to improve sensing when an R-C filter must be used. The CS input signal should be
protected from nearby noisy signals like GATE drive and the switching MOSFET drain.
7.4.1.13 Dead Time
The TPS23754 device features two switching MOSFET gate drivers to ease implementation of high-efficiency
topologies. Specifically, these include active (primary) clamp topologies and those with synchronous drivers that
are hard-driven by the control circuit. In all cases, there is a need to assure that both driven MOSFETs are not
on at the same time. The DT pin programs a fixed time period delay between the turnon of one gate driver until
the turnon of the next. This feature is an improvement over the repeatability and accuracy of discrete solutions
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while eliminating a number of discrete parts on the board. Converter efficiency is easily tuned with this one
repeatable adjustment. The programmed dead time is the same for both GATE-to-GAT2 and GAT2-to-GATE
transitions. The dead time is triggered from internal signals that are several stages back in the driver to eliminate
the effects of gate loading on the period; however, the observed and actual dead-time will be somewhat
dependent on the gate loading. The turnoff of GAT2 coincides with the start of the internal clock period.
DT may be used to disable GAT2, which goes to a high-impedance state.
GATE’s phase turns the main switch on when it transitions high, and off when it transitions low. GAT2’s phase
turns the second switch off when it transitions high, and on when it transitions low. Both switches should be off
when GAT2 is high and GATE is low. The signal phasing is shown in Figure 1. Many topologies that use
secondary-side synchronous rectifiers also use N-Channel MOSFETs driven through a gate-drive transformer.
The proper signal phase for these rectifiers may be achieved by inverting the phasing of the secondary winding
(swapping the leads). Use of the two gate drives is shown in Figure 27 and Figure 27.
7.4.1.14 FRS and Synchronization
The FRS pin programs the (free-running) oscillator frequency, and may also be used to synchronize the
TPS23754 device converter to a higher frequency. The internal oscillator sets the maximum duty cycle at 78%
and controls the slope-compensation ramp circuit. Synchronization may be accomplished by applying a short
pulse (TSYNC) of magnitude VSYNC to FRS as shown in Figure 30. The synchronization pulse terminates the
potential on-time period, and the off-time period does not begin until the pulse terminates.
7.4.1.15 T2P, Start-Up, and Power Management
T2P (type 2 PSE) is an active-low multifunction pin that indicates if
[(PSE = Type_2) + (1.5 V < VAPD) + (1.55 V < VPPD< 8.3 V)] × (VCTL < 4 V) × (pd current limit ≠ Inrush).
The term with VCTL prevents an optocoupler connected to the secondary-side from loading VC before the
converter is started. The APD and PPD terms allow the PD to operate from an adapter at high-power if a type 2
PSE is not present, assuming the adapter has sufficient capacity. Applications must monitor the state of T2P to
detect power source transitions. Transitions could occur when a local power supply is added or dropped or when
a PSE is enabled on the far end. The PD may be required to adjust the load appropriately. The usage of T2P is
demonstrated in Figure 27.
For a type 2 PD to operate at less than 13 W the first 80 ms after power application, the various delays must be
estimated and used by the application controller to meet the requirement. The bootup time of many applications
processors may be long enough to eliminate the need to do any timing.
7.4.1.16 Thermal Shutdown
The DC-DC controller has an OTSD that can be triggered by heat sources including the VB regulator, GATE
driver, bootstrap current source, and bias currents. The controller OTSD turns off VB, the GATE driver, and
forces the VC control into an undervoltage state.
7.4.1.17 Adapter ORing
Many PoE-capable devices are designed to operate from either a wall adapter or PoE power. A local power
solution adds cost and complexity, but allows a product to be used if PoE is not available in a particular
installation. While most applications only require that the PD operate when both sources are present, the
TPS23754 device supports forced operation from either of the power sources. Figure 26 illustrates three options
for diode ORing external power into a PD. Only one option would be used in any particular design. Option 1
applies power to the TPS23754 device PoE input, option 2 applies power between the TPS23754 device PoE
section and the power circuit, and option 3 applies power to the output side of the converter. Each of these
options has advantages and disadvantages. Many of the basic ORing configurations and discussion contained in
application note, Advanced Adapter ORing Solutions using the TPS23753 (SLVA306), apply to the TPS23754
device.
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VSS
VDD1
VDD
DEN
CLS
Low Voltage
Output
Power
Circuit
TPS23754
RCLS
58V
From Spare
Pairs or
Transformers
0.1uF
RDEN
From Ethernet
Transformers
Optional for PoE Priority
RTN
Adapter
Option 2
Adapter
Option 1
Adapter
Option 3
Figure 26. ORing Configurations
The IEEE standards require that the Ethernet cable be isolated from ground and all other system potentials. The
adapter must meet a minimum 1500 Vac dielectric withstand test between the output and all other connections
for ORing options 1 and 2. The adapter only needs this isolation for option 3 if it is not provided by the converter.
Adapter ORing diodes are shown for all the options to protect against a reverse voltage adapter, a short on the
adapter input pins, and damage to a low-voltage adapter. ORing is sometimes accomplished with a MOSFET in
option 3.
7.4.1.18 PPD ORing Features
The TPS23754 device provides several additional features to ease ORing based on the multifunction PPD pin
(not available on TPS23754-1 device). These include T2P signaling of an option 1 adapter, use of a 24-V
adapter (reduced output power) for option 1, and use of PoE as a power backup in conjunction with option 2.
See the Advanced ORing Techniques.
7.4.1.19 Using DEN to Disable PoE
The DEN pin may be used to turn the PoE hotswap switch OFF by pulling it to VSS while in the operational state,
or to prevent detection when in the idle state. A low on DEN forces the hotswap MOSFET OFF during normal
operation. Additional information is available in the Advanced Adapter ORing Solutions using the TPS23753
(SLVA306) application report.
7.4.1.20 ORing Challenges
Preference of one power source presents a number of challenges. Combinations of adapter output voltage
(nominal and tolerance), power insertion point, and which source is preferred determine solution complexity.
Several factors adding to the complexity are the natural high-voltage selection of diode ORing (the simplest
method of combining sources), the current limit implicit in the PSE, and PD inrush and protection circuits
(necessary for operation and reliability). Creating simple and seamless solutions is difficult if not impossible for
many of the combinations. However, the TPS23754 device offers several built-in features that simplify some
combinations.
Several examples will demonstrate the limitations inherent in ORing solutions. Diode ORing, a 48-V adapter with
PoE (option 1), presents the problem that either source might be higher. A blocking switch would be required to
assure which source was active. A second example is combining a 12-V adapter with PoE using option 2. The
converter will draw approximately four times the current at 12 V from the adapter than it does from PoE at 48 V.
Transition from adapter power to PoE may demand more current than can be supplied by the PSE. The
converter must be turned off while CIN capacitance charges, with a subsequent converter restart at the higher
voltage and lower input current. A third example is use of a 12-V adapter with ORing option 1. The PD hotswap
would have to handle four times the current, and have 1/16 the resistance (be 16 times larger) to dissipate equal
power. A fourth example is that MPS is lost when running from the adapter, causing the PSE to remove power
from the PD. If AC power is then lost, the PD will stop operating until the PSE detects and powers the PD.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS23754 device will support many power supply topologies that require a single PWM gate drive or two
complementary gate drives and will operate with current-mode control. Figure 27 provides an example of an
active clamp forward converter that uses the second gate driver to control M2, the active element in the clamp.
GAT2 may also be used to drive a synchronous rectifier as demonstrated in Figure 27. The TPS23754 may be
used in topologies that do not require GAT2, which may be disabled to reduce its idling loss.
From Ethernet
Pairs 1,2
8.2 Typical Application
DVC1
TLV431
RFBU
CIZ
RFBL
RCTL
CCTL
ROB
RCS
CVC
VT2P_OUT
M2
T2
CVB
RT2P_OUT
VB
GAT2
RBLNK
BLNK
VB
GATE
CS
COUT1
VDD1
VDD
RT2P
M1
RTN, COM
ARTN
DT
RCLS
RFRS
RAPD1
RAPD2
Adapter
DA
T2P
VC
RDT
58V
0.1uF
From Ethernet
Pairs 3,4
DEN
PPD
CLS
PAD
VSS
APD
CTL
FRS
VOUT
LOUT
COUT2
RDEN
T1
CIN
CIO
Figure 27. Driven Synchronous Flyback
8.2.1 Design Requirements
Selecting a converter topology along with a design procedure is beyond the scope of this applications section.
Examples to help in programming the TPS23754 are shown in Device and Documentation Support. Additional
special topics are included to explain the ORing capabilities, frequency dithering, and other design
considerations.
8.2.2 Detailed Design Procedure
8.2.2.1 Input Bridges and Schottky Diodes
Using Schottky diodes instead of PN junction diodes for the PoE input bridges and DVDD will reduce the loss of
this function by about 30%. However, there are some things to consider when using them.
The IEEE standard specifies a maximum backfeed voltage of 2.8 V. A 100-kΩ resistor is placed between the
unpowered pairs and the voltage is measured across the resistor. Schottky diodes often have a higher reverse
leakage current than PN diodes, making this a harder requirement to meet. Use conservative design for diode
operating temperature, select lower-leakage devices where possible, and match leakage and temperatures by
using packaged bridges to help with this.
Schottky diode leakage current and lower dynamic resistance can impact the detection signature. Setting
reasonable expectations for the temperature range over which the detection signature is accurate is the simplest
solution. Increasing RDEN slightly may also help meet the requirement.
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Typical Application (continued)
Schottky diodes have proven less robust to the stresses of ESD transients, failing as a short or becoming leaky.
Take care to provide adequate protection in line with the exposure levels. This protection may be as simple as
ferrite beads and capacitors.
A general recommendation for the input rectifiers are 1 A or 2 A, 100-V rated discrete or bridge diodes.
8.2.2.2 Protection, D1
A TVS, D1, across the rectified PoE voltage per Figure 28 must be used. TI recommends a SMAJ58A, or a part
with equal to or better performance, for general indoor applications. If an adapter is connected from VDD1 to RTN,
as in ORing option 2 above, voltage transients caused by the input cable inductance ringing with the internal PD
capacitance can occur. Adequate capacitive filtering or a TVS must limit this voltage to be within the absolute
maximum ratings. Outdoor transient levels or special applications require additional protection.
Use of diode DVDD for PoE priority may dictate the use of additional protection around the TPS23754. ESD
events between the PD power inputs, or the inputs and converter output, cause large stresses in the hotswap
MOSFET if DVDD becomes reverse biased and transient current around the TPS23754 is blocked. The use of
CVDD and DRTN in Figure 28 provides additional protection should over-stress of the TPS23754 device be an
issue. A SMAJ58A would be a good initial selection for DRTN. Individual designs may have to tune the value of
CVDD.
CIN
PPD
DEN
CLS
VDD1
VDD
VSS
RTN
COM
ARTN
RDEN
RCLS
D1 58V
C1 0.1mF
DVDD
DRTN
58V
From Spare
Pairs or
Transformers
From Ethernet
Transformers
CVDD
0.01mF
Figure 28. Example of Added ESD Protection for PoE Priority
8.2.2.3 Capacitor, C1
The IEEE 802.3at standard specifies an input bypass capacitor (from VDD to VSS) of 0.05 μF to 0.12 μF. Typically
a 0.1-μF, 100-V, 10% ceramic capacitor is used.
8.2.2.4 Detection Resistor, RDEN
The IEEE 802.3at standard specifies a detection signature resistance, RDEN from 23.7 kΩ to 26.3 kΩ, or 25 kΩ ±
5%. Choose an RDEN of 24.9 kΩ.
8.2.2.5 Classification Resistor, RCLS
Connect a resistor from CLS to VSS to program the classification current according to the IEEE 802.3at standard.
The class power assigned should correspond to the maximum average power drawn by the PD during operation.
Select RCLS according to Table 1.
For a high-power design, choose class 4 and RCLS = 63.4 Ω.
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Typical Application (continued)
8.2.2.6 Dead Time Resistor, RDT
The required dead-time period depends on the specific topology and parasitics. The easiest technique to obtain
the optimum timing resistor is to build the supply and tune the dead time to achieve the best efficiency after
considering all corners of operation (load, input voltage, and temperature). A good initial value is 100 ns.
Program the dead time with a resistor connected from DT to ARTN per Equation 3.
1. Choose RDT as follows assuming a tDT of 100 ns:
t (ns)
100
RDT (kW) = DT
=
= 50
2
2
a.
b. Choose RDT = 49.9 kΩ
8.2.2.7 Switching Transformer Considerations and RVC
Care in design of the transformer and VC bias circuit is required to obtain hiccup overload protection. Leadingedge voltage overshoot on the bias winding may cause VC to peak-charge, preventing the expected tracking with
output voltage. Some method of controlling this is usually required. This may be as simple as a series resistor, or
an R-C filter in front of DVC1. Good transformer bias-to-output-winding coupling results in reduced overshoot and
better voltage tracking.
RVC as shown in Figure 29 helps to reduce peak charging from the bias winding. This becomes especially
important when tuning hiccup mode operation during output overload. Typical values for RVC will be from 10 Ω to
100 Ω.
RVC
VC
DVC1
CVC
T1
Bias Winding
ARTN
Figure 29. RVC Usage
8.2.2.8 Special Switching MOSFET Considerations
Special care must be used in selecting the converter switching MOSFET. The TPS23756 minimum switching
MOSFET VGATE is about 5.5 V, which is due to the VC lower threshold. This will occur during an output overload,
or toward the end of a (failed) bootstrap start-up. The MOSFET must be able to carry the anticipated peak fault
current at this gate voltage.
8.2.2.9 Thermal Considerations and OTSD
Sources of nearby local PCB heating should be considered during the thermal design. Typical calculations
assume that the TPS23754 device is the only heat source contributing to the PCB temperature rise. It is possible
for a normally operating TPS23754 device to experience an OTSD event if it is excessively heated by a nearby
device.
8.2.2.10 APD Pin Divider Network, RAPD1, RAPD2
The APD pin can be used to disable the TPS23754 device internal hotswap MOSFET giving the adapter source
priority over the PoE source. An example calculation is provided, see SLVA306.
8.2.2.11 PPD Pin Divider Network, RPPD1, RPPD2
The PPD pin can be used to override the internal hotswap MOSFET UVLO (VUVLO_R and VUVLO_H) when using
low voltage adapters connected between VDD and VSS. The PPD pin has an internal 5-μA pulldown current
source. As an example, consider the choice of RPPD1 and RPPD2, for a 24-V adapter.
1. Select the start-up voltage, VADPTR-ON approximately 75% of nominal for a 24-V adapter. Assuming that the
adapter output is 24 V ± 10%, this provides 15% margin below the minimum adapter operating voltage.
2. Choose VADPTR-ON = 24 V × 0.75 = 18 V.
3. Choose RPPD2 = 3.01 kΩ.
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Typical Application (continued)
4. Calculate RPPD1.
RPPD1
æ
ö æ
ö
çV
÷ ç
V
18 V - 1.55 V ÷
ADPTR_ON
PPDEN ÷
ç
=
=ç
÷ = 31.64 kW
VPPDEN
ç
÷ ç 1.55 V
÷÷
+
5
m
A
+
I
PPD
ç
÷ ç
RPPD2
ø
è
ø è 3.01 kW
a.
b. Choose RPPD1 = 32.4 kΩ.
5. Check PPD turnon and PPD turnoff voltages.
é
æV
öù
VADPTR_ON = VPPDEN + êRPPD1 ´ ç PPDEN + IPPD ÷ ú = 18.4 V
è RPPD2
ø ûú
ëê
a.
é
æ (VPPDEN - VPPDH )
öù
+ IPPD ÷ ú = 14.75 V
VADPTR_OFF = (VPPDEN - VPPDH )+ êRPPD1 ´ ç
ç
÷ú
RPPD2
êë
è
øû
b.
c. Voltages look acceptable.
6. Check PPD resistor power consumption.
2
PRPPD =
(24 V ´ 1.1)
(VDD - VSS )2
=
= 19.6 mW
RPPD1 + RPPD2
3.01 kW + 32.4 kW
a.
b. Power is acceptable, but resistor values could be increased to reduce the power loss.
The PPD pin can also be used to modify the internal MOSFET UVLO for use with a lower output voltage
PSE (within certain limits). Connect the RPPD1 and RPPD2 dividers directly between VDD and VSS with
the midpoint connected to PPD. For this case and to allow classification, target the minimum PSE OFF
voltage (VADPTR_OFF) > VCU_OFF = 23 V. Then follow the procedure outlined above to select RPPD1, RPPD2,
and determine the PSE ON (VADPTR_ON) and PSE OFF (VADPTR_OFF) voltages ensuring that PSE OFF >
23 V. Lastly, because the RPPD1 and RPPD2 divider is in parallel with RDEN during detection, RDEN must be
increased such that the equivalent detection resistance is 25 kΩ nominal.
8.2.2.12 Setting Frequency (RFRS) and Synchronization
The converter switching frequency is set by connecting RFRS from the FRS pin to ARTN. The frequency may be
set as high as 1 MHz with some loss in programming accuracy as well as converter efficiency. Synchronization
at high duty cycles may become more difficult above 500 kHz due to the internal oscillator delays reducing the
available on-time. As an example:
1. Assume a desired switching frequency (fSW) of 250 kHz.
2. Compute RFRS:
17250
17250
RFRS (k W ) =
=
= 69
f
(kHz)
250
SW
a.
b. Select 69.8 kΩ.
The TPS23754 device may be synchronized to an external clock to eliminate beat frequencies from a sampled
system, or to place emission spectrum away from an RF input frequency. Synchronization may be accomplished
by applying a short pulse (TSYNC) of magnitude VSYNC to FRS as shown in Figure 30. RFRS should be chosen so
that the maximum free-running frequency is just below the desired synchronization frequency. The
synchronization pulse terminates the potential on-time period, and the off-time period does not begin until the
pulse terminates. The pulse at the FRS pin should reach between 2.5 V and VB, with a minimum width of 22 ns
(above 2.5 V) and rise and fall times less than 10 ns. The FRS node should be protected from noise because it is
high-impedance. An RT on the order of 100 Ω in the isolated example reduces noise sensitivity and jitter.
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VSYNC
FRS
47pF
VSYNC
TSYNC
1000pF
RFRS
TSYNC
RFRS
47pF
Synchronization
Pulse
RT
FRS
RTN
ARTN
COM
Synchronization
Pulse
RTN
ARTN
COM
Typical Application (continued)
1:1
Figure 30. Synchronization
8.2.2.13 Current Slope Compensation
The TPS23754 device provides a fixed internal compensation ramp that suffices for most applications. RS (see
Figure 31) may be used if the internally provided slope compensation is not enough.
Most current-mode control papers and application notes define the slope values in terms of VPP / TS (peak ramp
voltage / switching period). However, the electrical characteristics table specifies the slope peak (VSLOPE) based
on the maximum (78%) duty cycle. Assuming that the desired slope, VSLOPE-D (in mV/period), is based on the full
period, compute RS per the following equation where VSLOPE, DMAX, and ISL-EX are from the electrical
characteristics table with voltages in mV, current in μA, and the duty cycle is unitless (for example, DMAX = 0.78).
é
æ VSLOPE (mV) ö ù
ê VSLOPE_D (mV) - ç
÷ú
DMAX
è
ø ûú
ëê
´ 1000
RS (W) =
ISL_EX (mA)
(6)
RTN
COM
ARTN
GATE
CS
RS
CS
RCS
Figure 31. Additional Slope Compensation
CS may be required if the presence of RS causes increased noise, due to adjacent signals like the gate drive, to
appear at the CS pin.
8.2.2.14 Blanking Period, RBLNK
Selection of the blanking period is often empirical because it is affected by parasitics and thermal effects of every
device between the gate-driver and output capacitors. The minimum blanking period prevents the current limit
and PWM comparators from being falsely triggered by the inherent current spike that occurs when the switching
MOSFET turns on. The maximum blanking period is bounded by the output rectifier's ability to withstand the
currents experienced during a converter output short.
If blanking beyond the internal default is desired choose RBLNK using RBLNK (kΩ) = tBLNK (ns).
1. For a 100 ns blanking interval:
a. RBLNK (kΩ) = 100
b. Choose RBLNK = 100 kΩ.
The blanking interval can also be chosen as a percentage of the switching period.
1. Compute RBLNK as follows for 2% blanking interval in a switcher running at 250 kHz.
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Typical Application (continued)
RBLNK (k W ) =
BIanking_Interval(%)
2
´ 10 4 =
´ 10 4 = 80
fSW (kHz)
250
a.
b. Select RBLNK = 80.6 kΩ.
8.2.2.15 Estimating Bias Supply Requirements and CVC
The bias supply (VC) power requirements determine the CVC sizing and frequency of hiccup during a fault. The
first step is to determine the power and current requirements of the power supply control, then use this to select
CVC. The control current draw will be assumed constant with voltage to simplify the estimate, resulting in an
approximate value.
First determine the switching MOSFET gate drive power.
1. Let VQG be the gate voltage swing that the MOSFET QG is rated to (often 10 V).
æ
VC
PGATE = VC ´ fSW ´ ç QGATE ´
ç
VQG
è
ö
æ
VC
÷÷ PGAT2 = VC ´ fSW ´ çç QGATE2 ´
VQG
ø
è
a.
b. Compute gate drive power if VC is 12 V, QGATE is 17 nC, and QGAT2 is 8 nC.
12
PGATE = 12 V ´ 250 kHz ´ 17 nC ´
= 61.2 mW
10
c.
PG AT2 = 12 V ´ 250 kHz ´ 8 nC ´
ö
÷÷
ø
12
= 28.8 mW
10
PDRIVE = 61.2 mW + 28.8 mW = 90 mW
d. This illustrates why MOSFET QG should be an important consideration in selecting the switching
MOSFETs.
2. Estimate the required bias current at some intermediate voltage during the CVC discharge. For the TPS23754
device, 12 V provides a reasonable estimate. Add the operating bias current to the gate drive current.
P
90 mW
IDRIVE = DRIVE =
= 7.5 mA
VC
12 V
a.
b. ITOTAL = IDRIVE + IOPERATING = 7.5 mA + 0.92 mA = 8.42 mA
3. Compute the required CVC based on start-up within the typical soft-start period of 4 ms.
CVC1 + CVC2 =
TSTARTUP ´ ITOTAL
4 ms ´ 8.42 mA
=
= 5.18 mF
VCUVH
6.5 V
a.
b. For this case, a standard 10-μF electrolytic plus a 0.47 μF should be sufficient.
4. Compute the initial time to start the converter when operating from PoE.
a. Using a typical bootstrap current of 4 mA, compute the time to start-up.
C
´ VCUV
10.47 mF ´ 15 V
TST = VC1
=
= 39 ms
I
4 mA
VC
b.
5. Compute the fault duty cycle and hiccup frequency:
(C VC1 + CVC2 ) ´ VCUVH
(10 mF + 0.47 mF) ´ 6.5 V
TRECHARGE =
=
= 17 m s
IVC
4 mA
a.
TDISCHARGE =
b.
(C VC1 + CVC2 ) ´ VCUVH
ITOTAL
=
(10 mF + 0.47 mF) ´ 6.5V
= 8.08 ms
8.42 m A
i. The optocoupler current is 0 mA because the output is in current limit.
ii. Also, it is assumed IT2P is 0 mA.
TDISCHARGE
8.08 ms
Duty Cycle: D =
=
= 32%
T
+
T
8.08
ms + 17 ms
DISCHARGE
RECHARGE
c.
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Typical Application (continued)
Hiccup Frequency: F =
1
1
=
= 39.9 Hz
TDISCHARGE + TRECHARGE
8.08 ms + 17 ms
d.
6. With the TPS23754 device, the voltage rating of CVC1 and CVC2 should be 25 V minimum while with the
TPS23756 rating can be 16 V.
8.2.2.16 T2P Pin Interface
The T2P pin is an active low, open-drain output indicating a high-power source is available. An optocoupler is
typically used to interface with the T2P pin to signal equipment on the secondary side of the converter of T2P
status. Optocoupler current-gain is referred to as current transfer ratio (CTR), which is the ratio of transistor
collector current to LED current. To preserve efficiency, TI recommends a high-gain optocoupler ( 250% ≤ CTR ≤
500%, or 300% ≤ CTR ≤ 600% ) along with a high-impedance (for example, CMOS) receiver. Design of the T2P
optocoupler interface can be accomplished as follows:
VC
RT2P
VOUT
RT2P_OUT
Type 2 PSE
Indicator
Low = T2
T2P From
TPS23754
Figure 32. T2P Interface
1. T2P ON characteristic: IT2P = 2 mA minimum, VT2P = 1 V
2. Let VC = 12 V, VOUT = 5 V, RT2P-OUT = 10 kΩ, VT2P-OUT (low) = 400 mV maximum
V
- VT2P-OUT (low)
5 - 0.4
IRT2P-OUT = OUT
=
= 0.46 m A
R
10000
T2P-OUT
a.
3. The optocoupler CTR will be needed to determine RT2P. A device with a minimum CTR of 300% at 5-mA
LED bias current is selected. CTR will also vary with temperature and LED bias current. The strong variation
of CTR with diode current makes this a problem that requires some iteration using the CTR versus IDIODE
curve on the optocoupler data sheet.
a. Using the (normalized) curves, a current of 0.4 to 0.5 mA is required to support the output current at the
minimum CTR at 25°C.
i. Pick an IDIODE. For example one around the desired load current.
ii. Use the optocoupler data sheet curve to determine the effective CTR at this operating current. It is
usually necessary to apply the normalized curve value to the minimum specified CTR. It might be
necessary to ratio or offset the curve readings to obtain a value that is relative to the current that the
CTR is specified at.
iii. If IDIODE × CTRI_DIODE is substantially different from IRT2P_OUT, choose another IDIODE and repeat.
b. This manufacturer’s curves also indicate a –20% variation of CTR with temperature. The approximate
forward voltage of the optocoupler diode is 1.1 V from the data sheet.
100
100
IRT2P @ IMIN ´
= 0.5 mA ´
= 0.625 mA
100 - D CTRTEMP
100 - 20
c. VFLED ≉ 1.1 V
V - VT2P - VFLED
12 - 1 - 1.1
RT2P = C
=
= 15.48 kW
IRT2P
0.625 mA
d. Select a 15.4-kΩ resistor. Even though the minimum CTR and temperature variation were considered,
the designer might choose a smaller resistor for a little more margin.
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Typical Application (continued)
8.2.2.17 Advanced ORing Techniques
See Advanced Adapter ORing Solutions using the TSP23753, TI document number SLVA306A for ORing
applications that also work with the TPS23754 device. The material in sections Adapter ORing and Protection,
D1 are important to consider as well. The following applications are unique to the TPS23754 device with the
introduction of PPD.
CVDD
10nF
VDD1
VDD
PPD
DEN
CLS
DA
30V
26.7kW
APD
3.01kW
VSS
RAPD2 RAPD1 DAPD
CIN
RTN
COM
ARTN
RDEN
3.3MW
RVDD1
RAPD2
DRTN
RTN
COM
ARTN
APD
RAPD1
RCLS
VSS
58V
CIN
1.8KW
RHLD
For 48V
Adapter
24V
VDD1
PPD
DEN
CLS
DA
Adapter
DVDD
DHLD
RCLS
VDD
RDEN
1.8kW
24V
D1 58V
DVDD
C1 0.1uF
From Spare
Pairs or
Transformers
From Ethernet
Transformers
Option 2 ORing with PoE acting as a hot backup is eased by connecting PPD to VDD per Figure 33. This PPD
connection enables the class regulator even when APD is high. The R-Zener network (1.8 kΩ – 24 V) is the
simplest circuit that will satisfy MPS requirements, keeping the PSE online. This network may be switched out
when the APD is not powered with an optocoupler. This works best with a 48-V adapter and the APDprogrammed threshold as high as possible. An example of an adapter priority application with smooth switchover
between a 48-V adapter and PoE is shown on the right side of Figure 33. DAPD is used to reduce the effective
APD hysteresis, allowing the PSE to power the load before VVDD1-VRTN falls too low and causes a hotswap
foldback.
Adapter
Figure 33. Option 2 PoE Backup ORing
VDD
VDD1
DEN
PPD
CLS
RCLS
D1 58V
From Spare
Pairs or
Transformers
C1 0.1uF
RDEN
From Ethernet
Transformers
Option 1 ORing of a low voltage adapter (for example, 24 V) is possible by connecting a resistor divider to PPD
as in Figure 34. When 1.55 V ≤ VPPD ≤ 8.3 V, the hotswap MOSFET is enabled, T2P is activated, and the class
feature is disabled. The hotswap current limit is unaffected, limiting the available power. For example, the
maximum input power from a 24-V adapter would be 19.3 W [(24 V – 0.6 V) × 0.825 A].
DA
Adapter
RPPD2
RPPD1
APD
RTN
COM
ARTN
VSS
Figure 34. Low-Voltage Option 1 ORing
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Typical Application (continued)
8.2.2.18 Soft Start
Converters require a soft start on the voltage error amplifier to prevent output overshoot on start-up. Figure 35
shows a common implementation of a secondary-side soft start that works with the typical TL431 error amplifier.
The soft-start components consist of DSS, RSS, and CSS. They serve to control the output rate-of-rise by pulling
VCTL down as CSS charges through ROB, the optocoupler, and DSS. This has the added advantage that the TL431
output and CIZ are preset to the proper value as the output voltage reaches the regulated value, preventing
voltage overshoot due to the error amplifier recovery. The secondary-side error amplifier will not become active
until there is sufficient voltage on the secondary. The TPS23754 device provides a primary-side soft start, which
persists long enough (about 4 ms) for secondary side voltage-loop soft start to take over. The primary-side
current-loop soft start controls the switching MOSFET peak current by applying a slowly rising ramp voltage to a
second PWM control input. The PWM is controlled by the lower of the soft-start ramp or the CTL-derived current
demand. The actual output voltage rise time is usually much shorter than the internal soft-start period. Initially the
internal soft-start ramp limits the maximum current demand as a function of time. Either the current limit,
secondary-side soft start, or output regulation assume control of the PWM before the internal soft-start period is
over. Figure 24 shows a smooth handoff between the primary and secondary-side soft start with minimal output
voltage overshoot.
From Regulated
Output Voltage
ROB
RSS
CIZ
DSS
CSS
RFBU
RFBL
TLV431
Figure 35. Error Amplifier Soft Start
8.2.2.19 Frequency Dithering for Conducted Emissions Control
The international standard CISPR 22 (and adopted versions) are often used as a requirement for conducted
emissions. Ethernet cables are covered as a telecommunication port in section 5.2 for conducted emissions.
Meeting EMI requirements is often a challenge, with the lower limits of Class B being especially hard. Circuit
board layout, filtering, and snubbing various nodes in the power circuit are the first layer of control techniques. A
more detailed discussion of EMI control is presented in Practical Guidelines to Designing an EMI Compliant PoE
Powered Device With Isolated Flyback, (SLUA469). Additionally, IEEE802.3at sections 33.3 and 33.4 have
requirements for noise injected onto the Ethernet cable based on compatibility with data transmission.
Occasionally, a technique referred to as frequency dithering is used to provide additional EMI measurement
reduction. The switching frequency is modulated to spread the narrowband individual harmonics across a wider
bandwidth, thus lowering peak measurements. The circuit of Figure 36 modulates the switching frequency by
feeding a small AC signal into the FRS pin. These values may be adapted to suit individual needs.
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SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
www.ti.com
Typical Application (continued)
10kW
49.9kW
VB
+
-
6.04kW
TL331IDBV
4.99kW
0.01mF
10kW
301kW
1mF
To
FRS
ARTN
Figure 36. Frequency Dithering
10
5 V/div
99
88
200 mA/div
50 mA/div
8.2.3 Application Curves
T2P @ output
Inrush
I PI
Cvtr. Starts
Inrush
IPI
7
66
10 V/div
5
PI Powered
V C -RTN
Class
VVDD-VSS
Switching starts
2 V/div
33
VOUT
2
11
50 V/div
V DD -RTN
Mark
10 V/div
44
Detect
VRTN-VSS
0
t - Time - 10 ms/div
t - Time - 25 ms/div
Figure 37. Power Up and Start
36
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Figure 38. Start-Up
Copyright © 2008–2017, Texas Instruments Incorporated
Product Folder Links: TPS23754 TPS23754-1 TPS23756
TPS23754, TPS23754-1, TPS23756
www.ti.com
SLVS885I – OCTOBER 2008 – REVISED DECEMBER 2017
9 Power Supply Recommendations
The TPS23754/TPS23756 converter should be designed such that the input voltage of the converter is capable
of operating within the IEEE802.3at recommended input voltage as shown in Table 2 and the minimum operating
voltage of the adapter if applicable.
10 Layout
10.1 Layout Guidelines
Printed-circuit-board layout recommendations are provided in the evaluation module (EVM) documentation
available for these devices.
10.2 Layout Example
Figure 39. TPS23754EVM-383 EVM Parts Placement and Example Layout
10.3 ESD
The TPS23754 device has been tested to EN61000-4-2 using a power supply based on Figure 27. The levels
used were 8-kV contact discharge and 15-kV air discharge. Surges were applied between the PoE input and the
DC output, between the adapter input and the DC output, between the adapter and the PoE inputs, and to the
DC output with respect to earth. Tests were done both powered and unpowered. No TPS23754 device failures
were observed and operation was continuous. See Figure 28 for additional protection for some test
configurations.
ESD requirements for a unit that incorporates the TPS23754 have a much broader scope and operational
implications than are used in TI’s testing. Unit-level requirements should not be confused with reference design
testing that only validates the ruggedness of the TPS23754.
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Product Folder Links: TPS23754 TPS23754-1 TPS23756
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www.ti.com
11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
For more specific converter design examples see the following application notes:
• Designing with the TPS23753 Powered Device and Power Supply Controller, SLVA305
• Understanding and Designing an Active Clamp Current Mode Controlled Converter Using the UCC2897A.
SLUA535
• Advanced Adapter ORing Solutions using the TPS23753, SLVA306
• TPS23754EVM-420 EVM: Evaluation Module for TPS23754, SLVU301
• TPS23754EVM-383 EVM: Evaluation Module for TPS23754, SLVU304
11.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.3 Trademarks
PowerPAD, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
38
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Copyright © 2008–2017, Texas Instruments Incorporated
Product Folder Links: TPS23754 TPS23754-1 TPS23756
PACKAGE OPTION ADDENDUM
www.ti.com
19-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS23754PWP
ACTIVE
HTSSOP
PWP
20
70
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
TPS23754
Samples
TPS23754PWP-1
ACTIVE
HTSSOP
PWP
20
70
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
23754-1
Samples
TPS23754PWPR
ACTIVE
HTSSOP
PWP
20
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
TPS23754
Samples
TPS23754PWPR-1
ACTIVE
HTSSOP
PWP
20
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
23754-1
Samples
TPS23756PWP
ACTIVE
HTSSOP
PWP
20
70
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
TPS23756
Samples
TPS23756PWPR
ACTIVE
HTSSOP
PWP
20
2000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
TPS23756
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of