TPS40090-Q1
PW
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
HIGH-FREQUENCY MULTIPHASE CONTROLLER
Check for Samples: TPS40090-Q1
FEATURES
APPLICATIONS
•
•
•
•
•
•
•
•
1
2
•
•
•
•
•
•
•
•
•
•
(1)
Qualified for Automotive Applications
Two-, Three-, or Four-Phase Operation
5-V to 15-V Operating Range
Programmable Switching Frequency Up to
1-MHz/Phase
Current Mode Control With Forced Current
Sharing (1)
1% Internal 0.7-V Reference
Resistive Divider Set Output Voltage
True Remote Sensing Differential Amplifier
Resistive or DCR Current Sensing
Current Sense Fault Protection
Programmable Load Line
Compatible with UCC37222 Predictive Gate
Drive™ Technology Drivers
24-Pin Space-Saving TSSOP Package
Binary Outputs
Internet Servers
Network Equipment
Telecommunications Equipment
DC Power Distributed Systems
PW PACKAGE
(TOP VIEW)
CS1
CS2
CS3
CS4
CSCN
ILIM
DROOP
REF
COMP
FB
DIFFO
VOUT
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
EN/SYNC
VIN
BP5
PWM1
PWM2
PWM3
PWM4
GND
RT
SS
PGOOD
GNDS
Patent pending
DESCRIPTION
The TPS40090 is a two-, three-, or four-phase programmable synchronous buck controller that is optimized for
low-voltage, high-current applications powered by a 5-V to 15-V distributed supply. A multi-phase converter offers
several advantages over a single power stage including lower current ripple on the input and output capacitors,
faster transient response to load steps, improved power handling capabilities, and higher system efficiency.
Each phase can be operated at a switching frequency up to 1-MHz, resulting in an effective ripple frequency of
up to 4-MHz at the input and the output in a four-phase application. A two-phase design operates 180° out of
phase, a three-phase design operates 120° out of phase, and a four-phase design operates 90° out of phase, as
shown in Figure 1.
The number of phases is programmed by connecting the deactivated phase PWM output to the output of the
internal 5-V LDO. In two-phase operation the even phase outputs should be deactivated.
The TPS40090 uses fixed frequency, peak current mode control with forced phase current balancing. When
compared to voltage mode control, current mode results in a simplified feedback network and reduced input line
sensitivity. Phase current is sensed by using either current sense resistors installed in series with output
inductors or, for improved efficiency, by using the DCR (direct current resistance) of the filter inductors. The latter
method involves generation of a current proportional signal with an R-C circuit (shown in Figure 11).
The R-C values are selected by matching the time constants of the R-C circuit and the inductor; R-C = L/DCR.
With either current sense method, the current signal is amplified and superimposed on the amplified voltage error
signal to provide current mode PWM control.
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Predictive Gate Drive is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
An output voltage droop can be programmed to improve the transient window and reduce size of the output filter.
Other features include a single voltage operation, a true differential sense amplifier, a programmable current
limit, soft-start, and a power good indicator.
ORDERING INFORMATION (1)
PACKAGE (2)
TJ
–40°C to 125°C
(1)
(2)
TSSOP – PW
ORDERABLE PART NUMBER
Reel of 2000
TPS40090QPWRQ1
TOP-SIDE MARKING
TPS40090Q
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
Package drawings, thermal data, and symbolization are available at www.ti.com/packaging.
SIMPLIFIED TWO-PHASE APPLICATION DIAGRAM
TPS40090PW
CBP5
2
CS2
4
CS4
14
PGOOD
22
BP5
6
ILIM
17
GND
CS3
RCS3
3
CCS3
CSCN
5
CCS1
R CS1
CSS
15
SS
16
RT
7
DROOP
CS1
1
VIN
23
VIN (4.5 V to 15 V)
CIN
R ILIM2
L1
RRT
R DROOP
R ILIM1
8
R FB2
2
21
BP5
REF
CREF
R FB3
PWM1
24
EN/SYNC
9
COMP
10
FB
CFB1
TI
Synchronous
Buck
Driver
R FB1
PWM2
20
PWM4
18
L2
PWM3
11
DIFFO
13
GNDS
12
VOUT
19
TI
Synchronous
Buck
Driver
VOUT
(0.7 V to 3.5 V)
COUT
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
ABSOLUTE MAXIMUM RATING
over operating free-air temperature range unless otherwise noted (1)
EN/SYNC, VIN,
16.5 V
VIN
Input voltage range
VOUT
Output voltage range
TJ
Operating virtual-junction temperature range
–40°C to 125°C
Tstg
Storage temperature
–65°C to 150°C
ESD
Electrostatic discharge protection, Human-Body Model (HBM)
(1)
–0.3 V to 6 V
CS1, CS2, CS3, CS4, CSCN, DROOP, FB, GNDS, ILIM, VOUT
–0.3 V to 6 V
REF, COMP, DIFFO, PGOOD, SS, RT, PWM1, PWM2, PWM3, PWM4, BP5
1500 V
Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under "recommended operating
conditions" is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
MIN
MAX
UNIT
VIN
Input voltage
4.5
15
V
TJ
Operating virtual-junction temperature
–40
125
°C
MAX
UNIT
ELECTRICAL CHARACTERISTICS
TJ = –40°C to 125°C, VIN = 12 V, R(RT) = 64.9 kΩ, TJ = TA (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
INPUT SUPPLY
VIN
Operating voltage range, VIN
VIN
UVLO
VIN
UVLO (1)
IIN
Shutdown current, VIN
IIN
Quiescent current switching
4.5
15
Rising VIN
4.25
4.45
Falling VIN
4.1
4.35
Four channels, 400 kHz each, no load
V
2
10
μA
4
6
mA
OSCILLATOR/SYNCHRONIZATION
Phase frequency accuracy
Four channels, RRT = 64.9 kΩ
350
Phase frequency set range (1)
Four channels
100
Four channels
800
Synchronization frequency range
(1)
Synchronization input threshold (1)
Four channels
415
455
1200
kHz
9600
VBP5/2
V
PWM
Maximum duty cycle per channel
4-phase operation
87.5
2- and 3-phase operation
83.3
Minimum duty cycle per channel (1)
Minimum controllable on-time (1)
%
0
%
50
100
ns
0.700
0.707
V
25
150
nA
ERROR AMPLIFIER
Feedback input voltage
0.690
Feedback input bias current
VFB = 0.7 V
VOH
High-level output voltage
ICOMP = –1 mA
VOL
low-level output voltage
ICOMP = 1 mA
GBW
Gain bandwidth (1)
5
MHz
AVOL
(1)
90
dB
Open loop gain
2.5
2.9
0.5
0.8
V
SOFT START
ISS
Soft-start source current
3.5
5
6
μA
VSS
Soft-start clamp voltage
0.95
1.00
1.05
V
(1)
Specified by design
Copyright © 2008–2011, Texas Instruments Incorporated
3
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
TJ = –40°C to 125°C, VIN = 12 V, R(RT) = 64.9 kΩ, TJ = TA (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
0.8
2
MAX
UNIT
ENABLE
Enable threshold voltage
Enable voltage capability (2)
2.5
VIN(max)
V
PWM OUTPUT
PWM pullup resistance
IOH = 5 mA
27
45
PWM pulldown resistance
IOL = 10 mA
27
45
Ω
5-V REGULATOR
VOUT
Output voltage
External ILOAD = 2 mA on BP5
Pass device voltage drop
VIN = 4.5 V, No external load on BP5
4.8
Short circuit current
5
8
5.2
V
200
mV
30
mA
5.9
V/V
CURRENT SENSE AMPLIFIER
Gain transfer
–100 mV ≤ V(CS) ≤ 100 mV, VCSRTN = 1.5 V
4.7
Gain variance between phases
VCS = 100 mV
–5
Input offset variance at zero current
VCS = 0 V
–7
Input common mode (2)
5.4
0
0
Bandwidth (2)
5
%
8
mV
4
18
V
MHz
Maximum VCS in regulation
200
mV
DIFFERENTIAL AMPLIFIER
Gain
1
Gain tolerance
CMRR
Common mode rejection ratio
VOUT 4 V vs 0.7 V, VGNDS = 0 V
(2)
0.7 V ≤ VOUT ≤ 4 V
Bandwidth (2)
–0.5
V/V
0.5
%
60
dB
5
MHz
RAMP
Ramp amplitude (2)
0.4
0.5
0.6
V
POWER GOOD
PGOOD high threshold
Reference to VREF
10
14
%
PGOOD low threshold
Reference to VREF
–14
–10
%
VOL
Low-level output voltage
IPGOOD = 4 mA
Ilkg
PGOOD output leakage
VPGOOD = 5 V
0.35
0.60
V
50
80
μA
OUTPUT OVERVOLTAGE/UNDERVOLTAGE FAULT
VOV
Overvoltage threshold voltage
VFBK relative to VREF
15
19
%
VUV
Undervoltage threshold voltage
VFBK relative to VREF
–18
–14
%
LOAD LINE PROGRAMMING
IDROOP
(2)
4
Pulldown current on DROOP
4-phase, VCS = 100 mV
40
μA
Specified by design
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
BP5
22
O
Output of an internal 5-V regulator. A 4.7-μF capacitor should be connected from this pin to ground.
For 5-V applications, this pin should be connected to VDD.
COMP
9
O
Output of the error amplifier. The voltage at this pin determines the duty cycle for the PWM.
CS1
1
I
CS2
2
I
CS3
3
I
CS4
4
I
CSCN
5
I
Common point of current sense resistors or filter inductors
DIFFO
11
O
Output of the differential amplifier. The voltage at this pin represents the true output voltage without
drops that result from high current in the PCB traces
DROOP
7
I
Used to program droop function. A resistor between this pin and the REF pin sets the desired droop
value.
Used to sense the inductor current in the phases. Inductor current can be sensed with an external
current sense resistor or by using an external circuit and the inductor's DC resistance. They are also
used for overcurrent protection and forced current sharing between the phases.
EN/SYNC
24
I
A logic high signal on this input enables the controller operation. A pulsing signal to this pin
synchronizes the main oscillator to the rising edge of an external clock source. These pulses must be
of higher frequency than the free running frequency of the main oscillator set by the resistor from the
RT pin.
FB
10
I
Inverting input of the error amplifier. In closed loop operation, the voltage at this pin is the internal
reference level of 700 mV. This pin is also used for the PGOOD and OVP comparators.
GND
17
GNDS
13
Ground connection to the device.
I
Inverting input of the differential amplifier. This pin should be connected to ground at the point of load.
ILIM
6
I
Used to set the cycle-by-cycle current limit threshold. If ILIM threshold is reached, the PWM cycle is
terminated and the converter delivers limited current to the output. Under these conditions the
undervoltage threshold is reached eventually and the controller enters the hiccup mode. The controller
stays in hiccup mode for seven consecutive cycles. At the eighth cycle the controller attempts a full
start-up sequence.
PGOOD
14
O
Power good indicator of the output voltage. This open-drain output connects to the supply via an
external resistor.
PWM1
21
O
PWM2
20
O
PWM3
19
O
PWM4
18
O
REF
8
O
Output of an internal 0.7-V reference voltage.
RT
16
I
Connecting a resistor from this pin to ground sets the oscillator frequency.
VIN
23
I
Power input for the chip. Decoupling of this pin is required.
VOUT
12
I
Noninverting input of the differential amplifier. This pin should be connected to VOUT at the point of
load.
SS
15
I
Provides user programmable soft-start by means of a capacitor connected to the pin.
Phase shifted PWM outputs which control the external drivers. The high output signal commands a
PWM cycle. The low output signal commands controlled conduction of the synchronous rectifiers.
These pins are also used to program various operating modes as follows: for three-phase mode,
PWM4 is connected to 5 V; for two-phase mode, PWM2 and PWM4 are connected to 5 V.
Copyright © 2008–2011, Texas Instruments Incorporated
5
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
FUNCTIONAL BLOCK DIAGRAM
RT
16
TPS40090PW
COMP
9
CLOCK
FB 10
+
5 mA
DROOP 7
REF
8
01
A = -(K +Y)
B = +1
A
SS 15
+
21 PWM1
IDROOP
B
1/N
+ 700 mV
02
+
20 PWM2
PH2
PH4
A
DIFFO 11
GNDS 13
VOUT 12
CSCN
1
CS2
2
CS3
CS4
6
3
4
+
19 PWM3
A
+
04
B
5
CS1
03
B
+
18 PWM4
A
gM
PHDET
IPH1
+
gM
+
IPH3
gM
+
PH2
IPH2
B
Σ IPH x K
PH2
gM
+
IPH1
IPH4
IPH2
POWER
GOOD
IPH3
CURRENT
LIMIT
5V
REG
IPH4
PH4
23 VIN
22 BP5
17 GND
PH4
14
6
24
PGOOD
ILIM
EN/SYNC
18
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
APPLICATION INFORMATION
Functional Description
The TPS40090 is a multiphase, synchronous, peak current mode, buck controller. The controller uses external
gate drivers to operate N-channel power MOSFETs. The controller can be configured to operate in a two-, three-,
or four-phase power supply.
The controller accepts current feedback signals from either current sense resistors placed in series with the filter
inductors or current proportional signals derived from the inductors' DCR.
Other features include an LDO regulator with UVLO to provide single voltage operation, a differential input
amplifier for precise output regulation, user programmable operation frequency for design flexibility, external
synchronization capability, programmable pulse-by-pulse overcurrent protection, output overvoltage protection,
and output undervoltage shutdown.
Differential Amplifier
The unity gain differential amplifier with high bandwidth allows improved regulation at a user-defined point and
eases layout constraints. The output voltage is sensed between the VOUT and GNDS pins. The output voltage
programming divider is connected to the output of the amplifier (DIFFO). The differential amplifier can be used
only for output voltages lower then 3.3 V.
If there is no need for a differential amplifier, or if the output voltage required is higher than 3.3 V, the differential
amplifier can be disabled by connecting the GNDS pin to the BP5 pin. The voltage programming divider in this
case should be connected directly to the output of the converter.
Current Sensing and Balancing
The controller employs a peak current-mode control scheme, which naturally provides a certain degree of current
balancing. With current mode, the level of current feedback should comply with certain guidelines depending on
duty factor, known as slope compensation to avoid subharmonic instability. This requirement can prohibit
achieving a higher degree of phase current balance. To avoid the controversy, a separate current loop that
forces phase currents to match is added to the proprietary control scheme. This effectively provides high degree
of current sharing independently of properties of controller's small signal response.
High-bandwidth current amplifiers can accept as an input voltage either voltage drop across dedicated precise
current-sense resistors, or inductor's DCR voltage derived by an R-C network, or thermally compensated voltage
derived from the inductor's DCR. The wide range of current-sense settings eases the cost and complexity
constraints and provides performance superior to those found in controllers using low-side MOSFET current
sensing.
Copyright © 2008–2011, Texas Instruments Incorporated
7
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
Setting Controller Configuration
By default, the controller operates at four-phase configuration. The alternate number of active phases is
programmed by connecting unused PWM outputs to BP5. (See Figure 1) For example, for three-phase
operation, the unused fourth phase output, PWM4, should be connected to BP5. For two-phase operation, the
second, PWM2, and the fourth, PWM4, outputs should be connected to BP5.
Power Up
Capacitors connected to the BP5 pin and the soft-start pin set the power-up time. When EN is high, the capacitor
connected to the BP5 pin gets charged by the internal LDO as shown in Figure 2.
4.5 C BP5
t BPS +
8 10 *3
(1)
1
4-Phase
Operation
2
3
4
1
3-Phase
Operation
2
3
4
BP5
1
2-Phase
Operation
2
BP5
3
4
BP5
Figure 1. Programming Controller Configuration
8
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
EN
BP5
SS
1.0
0.7
VOUT
PGOOD
t - Time
Figure 2. Power-Up Waveforms
When the BP5 pin voltage crosses its lower undervoltage threshold and the power-on reset function is cleared,
the calibrated current source starts charging the soft start capacitor. The PGOOD pin is held low during the start
up. The rising voltage across the capacitor serves as a reference for the error amplifier assuring start-up in a
closed loop manner. When the soft start pin voltage reaches the level of the reference voltage VREF = 0.7 V, the
converter's output reaches the regulation point and further rise of the soft start voltage has no effect on the
output.
0.7 C SS
t SS +
5 10 *6
(2)
When the soft-start voltage reaches level of 1 V, the power good (PGOOD) function is cleared and reported on
the PGOOD pin. Normally, the PGOOD pin goes high at this moment. The time from when SS begins to rise to
when PGOOD is reported is:
t PG + 1.43 T SS
(3)
Output Voltage Programming
The converter output voltage is programmed by the R1/R2 divider from the output of the differential amplifier. The
center point of the divider is connected to the inverting output of the error amplifier (FB), as shown in Figure 5.
V OUT + 0.7 V
ǒR1
) 1Ǔ
R2
(4)
Current Sense Fault Protection
Multiphase controllers with forced current sharing are inherently sensitive to failure of a current sense
component. In the event of such failure, the whole load current can be steered with catastrophic consequences
into a single channel where the fault has happened. The dedicated circuit in the TPS40090 controller prevents it
from starting up if any current sense pin is open or shorted to ground. The current-sense fault detection circuit is
active only during device initialization, and it does not provide protection should a current-sense failure happen
during normal operation.
Copyright © 2008–2011, Texas Instruments Incorporated
9
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
Overvoltage Protection
If the voltage at the FB pin (VFB) exceeds VREF by more than 16%, the TPS40090 enters into an overvoltage
state. In this condition, the output signals from the controller to the external drivers is pulled low, causing the
drivers to force all of the upper MOSFETs to the OFF position and all the lower MOSFETs to the ON position. As
soon as VFB returns to regulation, the normal operating state resumes.
Overcurrent Protection
The overcurrent function monitors the voltage level separately on each current sense input and compares it to
the voltage on the ILIM pin set by a divider from the controller's reference. In case a threshold of V(ILIM)/2.7 is
exceeded the PWM cycle on the associated phase is terminated. The voltage level on the ILIM pin is determined
by the following expression:
V ILIM + 2.7 I PH(max) R CS
(5)
I PH(max) + I OUT )
ǒVIN * VOUTǓ
2
L
f SW
V OUT
V IN
where:
• IPH(max) is a maximum value of the phase current allowed
• RCS is a value of the current sense resistor used
(6)
If the overcurrent condition continues, each phase's PWM cycle is terminated by the overcurrent signals. This
puts a converter in a constant current mode with the output current programmed by the ILIM voltage. Eventually,
the supply and demand equilibrium on the converter output fails and the output voltage declines. When the
undervoltage threshold is reached, the converter enters a hiccup mode. The controller is stopped and the output
is not regulated any more, the soft-start pin function changes. It now serves as a timing capacitor for a fault
control circuit. The soft-start pin is periodically charged and discharged by the fault control circuit. After seven
hiccup cycles expire, the controller attempts to restore normal operation. If the overload condition is not cleared,
the controller stays in the hiccup mode indefinitely. In such conditions, the average current delivered to the load
is roughly 1/8 of the set overcurrent value.
Undervoltage Protection
If the FB pin voltage falls lower than the undervoltage protection threshold (84.5%), the controller enters the
hiccup mode as it is described in the Overcurrent Protection section.
Fault-Free Operation
If the SS pin voltage is prevented from rising above the 1-V threshold, the controller does not execute nor report
most faults and the PGOOD output remains low. Only the overcurrent function and current-sense fault remain
active. The overcurrent protection continues to terminate PWM cycle every time when the threshold is exceeded
but the hiccup mode is not entered.
Setting the Switching Frequency
The clock frequency is programmed by the value of the timing resistor connected from the RT pin to ground.
R RT + KPH
ǒ39.2
10 3
f *1.041
* 7Ǔ
PH
(7)
where:
KPH is a coefficient that depends on the number of active phases. For two-phase and three-phase
configurations, KPH= 1.333. For four-phase configurations, KPH= 1. fPH is a single phase frequency, kHz. The
RT resistor value is returned by the last expression in kΩ.
To calculate the output ripple frequency, use the following equation:
F RPL + NPH f PH
where:
•
10
NPH is a number of phases used in the converter.
(8)
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
The switching frequency of the controller can be synchronized to an external clock applied to the EN/SYNC pin.
The external frequency should be somewhat higher than the free-running clock frequency for synchronization to
take place.
SWITCHING FREQUENCY
vs
TIMING RESISTANCE
fSW - Switching Frequency - kHz
10000
100
0
50
100
150
200
250
300
RT - Timing Resistance - kΩ
Figure 3.
Setting the Output Voltage Droop
In many applications, the output voltage of the converter is intentionally allowed to droop as load current
increases. This approach (sometimes referred to as active load line programming) allows for better use of the
regulation window and reduces the amount of the output capacitors required to handle the same load current
step. A resistor from the REF pin to the DROOP pin sets the desired value of the output voltage droop.
2500 NPH VDROOP
2500 NPH VDROOP
VREF
R2
R DROOP +
+
VOUT
I OUT RCS
VCS1 ) VCS2 ) VCS3 ) VCS4 R1 ) R2
•
•
•
•
•
(1)
where:
VDROOP is the value of droop at maximum load current IOUT
NPH is number of phases
RCS is the current-sense resistor value
2500 Ω is the inversed value of transconductance from the current sense pins to DROOP (1)
VCSx, are the average voltages on the current sense pins
(9)
IDROOP is relatively linear vs VCS and is typically 40 μA at VCS = 100 mV. Above VCS = 100 mV, IDROOP becomes nonlinear, rolls off, and
saturates to approximately 50 μA to 65 μA when VCS > 200 mV (see Figure 6). Thus, above 100 mV, Equation 9 is not accurate.
Copyright © 2008–2011, Texas Instruments Incorporated
11
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
OUTPUT VOLTAGE
vs
OUTPUT CURRENT
VOUT
VOUT - Output Voltage - V
VDROOP
0
IOUT(max)
IOUT - Output Current - A
Figure 4.
GNDS
Differential
Amplifier
13
VOUT
+
12
DIFFO
11
COMP
9
R1
I DROOP
C1
Error
Amplifier
R3 FB
10
+
DROOP
R2
7
I DROOP
RDROOP
REF
+
8
700 mV
Figure 5.
12
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
DROOP CURRENT
vs
CS VOLTAGE
70
60
IDROOP - µA
50
40
30
20
10
0
0
0.05
0.1
0.15
0.2
0.25
0.3
VCS - V
Figure 6.
Feedback Loop Compensation
The TPS40090 operates in a peak current mode and the converter exhibits a single pole response with ESR
zero for which Type II compensation network is usually adequate, as shown in Figure 8.
The following equations show where the load pole and ESR zero calculations are situated.
1
1
f OP +
f ESRZ +
2p R OUT C OUT
2p R ESR C OUT
(10)
To achieve desired bandwidth the error amplifier must compensate for modulator gain loss on the crossover
frequency and this is facilitated by placing the zero over the load pole. The ESR zero alters the modulator's -1
slope at higher frequencies. To compensate for that alteration, the pole in-error amplifier transfer function should
be added at frequency of the ESR zero as shown in Figure 7.
Copyright © 2008–2011, Texas Instruments Incorporated
13
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
Figure 7.
The following equations help in choosing components of the error amplifier compensation network. Fixing the
value of the resistor R1 first is recommended as it simplifies further adjustments of the output voltage without
altering the compensation network.
R2 + R1
10
ǒ
*GOMAG
20
Ǔ;
C1 +
1
ǒ2p
F OP
R2Ǔ
;
C2 +
1
ǒ2p
F ESRZ
R2Ǔ
where:
•
GOMAG is the control to output gain at desired system crossover frequency.
(11)
Introduction of output voltage droop as a measure to reduce amount of filter capacitors changes the transfer
function of the modulator as it is shown in the Figure 9.
14
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
GAIN AND PHASE
vs
FREQUENCY WITHOUT DROOP
80
Converter Overall
EA
60
G - Gain - dB
40
Type II
Modulator
20
0
Load Pole
-20
ESR Zero
-40
200
150
Phase
Phase - °
100
50
0
-50
-100
10
100
1k
10 k
f - Frequency - Hz
100 k
1M
Figure 8.
Copyright © 2008–2011, Texas Instruments Incorporated
15
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
GAIN AND PHASE
vs
FREQUENCY WITH DROOP
80
60
G - Gain - dB
40
Droop Zero
20
0
Load Pole
ESR Zero
-20
-40
200
150
Phase - °
100
50
0
-50
-100
10
100
1k
10 k
f - Frequency - Hz
100 k
1M
Figure 9.
The droop function, as well as the output capacitor ESR, introduces zero on some frequency left of the crossover
point.
1
F
+
DROOPZ
2p
ǒ
Ǔ
VDROOP
I OUT(max)
COUT
(12)
To compensate for this zero, pole on the same frequency should be added to the error amplifier transfer function.
With Type II compensation network a new value for the capacitor C2 is required compared to the case without
droop.
C1
C2 +
2p R2 C1 ǒF DROOPZ * 1Ǔ
(13)
16
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
When attempting to close the feedback loop at frequency that is near the theoretical limit, use the above
considerations as a first approximation and perform on bench measurements of closed loop parameters as
effects of switching frequency proximity and finite bandwidth of voltage and current amplifiers may substantially
alter them as it is shown in Figure 10.
GAIN AND PHASE
vs
FREQUENCY
60
Phase
80
50
60
30
40
20
10
Gain
Phase - °
G - Gain - dB
40
20
0
0
-10
VIN = 12 V
VOUT = 1.5 V
IOUT= 100 A
-20
100
1k
10 k
100 k
f - Frequency - Hz
-20
1M
Figure 10.
Thermal Compensation of DCR Current Sensing
Inductor DCR current sensing is a known lossless technique to retrieve a current proportional signal. Equation 14
and Equation 15 show the calculation used to determine the DCR voltage drop for any given frequency. (See
Figure 11)
DCR
V DCR + ǒVIN * VOUTǓ
DCR ) w L
(14)
1
V C + ǒVIN * VOUTǓ
w C
R) 1
w C
(15)
ǒ
Ǔ
Voltage across the capacitor is equal to voltage drop across the inductor DCR, VC = VDCR when time constant of
the inductor and the time constant of the R-C network are equal:
DCR
1
L + R C; t
VC +
+
;
DCRL + t RC
DCR ) w L DCR
1
w C
R)
w C
(16)
ǒ
Ǔ
The output signal generated by the network shown in Figure 11 is temperature dependant due to positive thermal
coefficient of copper specific resistance as determined using Equation 17. The temperature variation of the
inductor coil can exceed 100°C in a practical application leading to approximately 40% variation in the output
signal and in turn, respectively move the overcurrent threshold and the load line.
K(T) + 1 ) 0.0039 (T * 25)
(17)
The relatively simple network shown in Figure 12 (made of passive components including one NTC resistor) can
provide almost complete compensation for copper thermal variations.
Copyright © 2008–2011, Texas Instruments Incorporated
17
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
Figure 11.
L
DCR
C
R
R2
R1
RNTC
RTHE
Figure 12.
The following algorithm and expressions help to determine components of the network.
1. Calculate the equivalent impedance of the network at 25°C that matches the inductor parameters in
Equation 18. Use of COG type capacitors for this application is recommended. For example, for L = 0.4 μH,
DCR = 1.22 mΩ, C = 10 nF; RE = 33.3 kΩ. It is recommended to keep RE < 50 kΩ as higher values may
produce false triggering of the current sense fault protection.
L
DCR
RE +
C
(18)
2. It is necessary to set the network attenuation value KDIV(25) at 25°C. For example, KDIV(25) = 0.85. The
attenuation values KDIV(25) > 0.9 produces higher values for NTC resistors that are harder to get from
suppliers. Attenuation values lower 0.7 substantially reduce the network output signal.
3. Based on calculated RE and KDIV(25) values, calculate and pick the closest standard value for the resistor R
= RE/KDIV(25). For the given example R = 33 kΩ/ 0.85 = 38.8 kΩ. The closest standard value from 1% line is
R = 39.2 kΩ.
4. Pick two temperature values at which curve fitting is made. For example T1 = 50°C and T2 = 90°C.
5. Find the relative values of RTHE required on each of these temperatures.
R
(T1)
R
(T2)
R THE1 + THE
R THE2 + THE
R THE(25)
R THE(25)
(19)
RT +
K DIV(T)
1 * K DIV(T)
R
K DIV(T) +
K DIV(25)
1 ) 0.0039 (t * 25)
(20)
For the given example RTHE1= 0.606, RTHE2=0.372.
18
Copyright © 2008–2011, Texas Instruments Incorporated
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
6. From the NTC resistor datasheet get the relative resistance for resistors with desired curve. For the given
example and curve 17 for NTHS NTC resistors from Vishay RNTC1= 0.3507 and RNTC2= 0.08652.
7. Calculate relative values for network resistors including the NTC resistor.
R1 R +
ǒRNTC1 * RNTC2Ǔ
RNTC1
RE1
RE1
R E2 * R NTC1
R E2
ǒ1 * R NTC2Ǔ * RNTC2
ǒ1 * RNTC2Ǔ ) RNTC2 R E1 ǒ1 * RNTC1Ǔ
RE2 ǒ1 * R NTC1Ǔ * ǒR NTC1 * R NTC2Ǔ
(21)
RNTC R +
ƫ
ƪ
R NTC1
1
*
1 * R1R RE1 * R1 R
R2 R + ǒ1 * R NTC1Ǔ
ƪǒ1 * R1 Ǔ
R
*1
* ǒR2RǓ
ƫ
*1
*1
(22)
*1
(23)
For the given example R1R= 0.281, R2R = 2.079, and RNTCR = 1.1.
8. Calculate the absolute value of the NTC resistor as RTHE(25). In given example RNTC = 244.3 kΩ.
9. Find a standard value for the NTC resistor with chosen curve type. In case the close value does not exist in a
desired form factor or curve type. Chose a different type of the NTC resistor and repeat steps 6 to 9. In the
example, the NTC resistor with the part number NTHS0402N17N2503J with RNTCS(25) = 250 kΩ is close
enough to the calculated value.
10. Calculate a scaling factor for the chosen NTC resistor as a ratio between selected and calculated NTC value
and. In the example k = 1.023.
RNTC S
k+
RNTC C
(24)
11. Calculate values of the remaining network resistors.
R1 C + RTHE(25)
ƪǒ(1 * k) ) k
R1 RǓƫ
(25)
For the given example, R1C= 58.7 kΩ and R2C = 472.8 kΩ. Pick the closest available 1% standard values:
R1 = 39.2 kΩ, and R2 = 475 kΩ, thus completing the design of the thermally compensated network for the
DCR current sensor.
Figure 13 illustrates the fit of the designed network to the required function.
Copyright © 2008–2011, Texas Instruments Incorporated
19
TPS40090-Q1
SLUS845C – JUNE 2008 – REVISED MAY 2011
www.ti.com
CURRENT SENSE IMPEDANCE
vs
AMBIENT TEMPERATURE
40
r
Measured
RTHE (T5C) - Current Sense Impedance - kΩ
Acquired
30
r
20
10
r
r
10
20
40
60
80
100
TA - Ambient Temperature - °C
120
Figure 13.
Operation With Output Voltages Higher Than 3.3 V
The TPS40090 controllers are designed to operate in power supplies with output voltages ranging from 0.7 V to
3.3 V. To support higher output voltages, mainly in 12-V to 5-V power supplies, the BP5 voltage needs to be
increased slightly to provide enough headroom to ensure linearity of current sense amplifiers. The simple circuit
on Figure 14 shows a configuration that generates a 6-V voltage source to power the controller with increased
bias voltage. Both the VIN and BP5 pins should be connected to this voltage source. The differential amplifier
normally excessive for higher-output voltages can be disabled by connecting GNDS pin to the BP5 pin.
12 V
TPS40090
1.1 kW
EN/SYNC
24
13.7 kW
VIN 23
6V
BP5 22
4.7 mF
TLA431
10 kW
Figure 14. Biasing the TPS40090 With a 5-V Power Supply
Design Example
A design example is available in the TPS40090EVM-001 user’s guide (SLUU175).
20
Copyright © 2008–2011, Texas Instruments Incorporated
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS40090QPWRQ1
ACTIVE
TSSOP
PW
24
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
TPS40090Q
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
工商网监
湘ICP备2023018690号
