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TPS50601-SP
SLVSD45 – DECEMBER 2015
TPS50601-SP Radiation Hardened 1.6- to 6.3-V Input, 6-A Synchronous Buck Converter
1 Features
2 Applications
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5962R10221:
– Radiation Hardness Assurance (RHA) up to
TID 100 krad (Si)
– ELDRS Free 100 krad (Si) – 10 mRAD(Si)/s
– Single Event Latchup (SEL) Immune to
LET = 85 MeV-cm2/mg (See Radiation Report)
– SEB and SEGR Immune to 85 MeV-cm2/mg,
SOA Curve Available (See Radiation Report)
– SET/SEFI Cross-Section Plot Available (See
Radiation Report)
Peak Efficiency: 95% (VO = 3.3 V)
Integrated 55-mΩ/50-mΩ MOSFETs
Split Power Rail: 1.6 to 6.3 V on PVIN
Power Rail: 3 to 6.3 V on VIN
6-A Maximum Output Current
Flexible Switching Frequency Options:
– 100-kHz to 1-MHz Adjustable Internal
Oscillator
– External Sync Capability: 100 kHz to 1 MHz
– Sync Pin can be Configured as a 500-kHz
Output for Master/Slave Applications
0.795-V ±1.258% Voltage Reference at 25°C
Monotonic Start-Up into Prebiased Outputs
Adjustable Soft Start Through External Capacitor
Input Enable and Power-Good Output for Power
Sequencing
Power Good Output Monitor for Undervoltage and
Overvoltage
Adjustable Input Undervoltage Lockout (UVLO)
20-Pin Thermally-Enhanced Ceramic Flatpack
Package (HKH)
See www.ti.com/swift for SWIFT™ Documentation
See the Tools & Software Tab
•
•
•
•
Space Satellite Point of Load Supply for FPGAs,
Microcontrollers, and ASICs
Space Satellite Payloads
Radiation-Tolerant Applications
Available in Military (–55°C to 125°C)
Temperature Range
Engineering Evaluation (/EM) Samples are
Available(1)
3 Description
The TPS50601-SP is a radiation hardened, 6.3-V, 6A synchronous step-down converter, which is
optimized for small designs through high efficiency
and integrating the high-side and low-side MOSFETs.
Further space savings are achieved through current
mode control, which reduces component count, and a
high switching frequency, reducing the inductor's
footprint. The devices are offered in a thermally
enhanced 20-pin ceramic, dual in-line flatpack
package.
Device Information(2)
PART NUMBER
TPS50601-SP
PACKAGE
BODY SIZE (NOM)
CFP (20)
7.38 × 12.70 mm
KGD(3)
N/A(4)
(1) These units are intended for engineering evaluation only.
They are processed to a non-compliant flow (that is no burnin, and so forth) and are tested to temperature rating of 25°C
only. These units are not suitable for qualification, production,
radiation testing or flight use. Parts are not warranted for
performance on full MIL specified temperature range of –55°C
to 125°C or operating life.
(2) For all available packages, see the orderable addendum at
the end of the data sheet.
(3) Known good die
(4) Bare die in waffle pack
SPACE
SPACE
Efficiency (p.u.)
Efficiency vs Load Current, Vin = 5 V
1.00
0.95
0.90
0.85
0.80
0.75
0.70
0.65
0.60
0.55
0.50
0.45
0.40
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0.00
Vo = 3.3 V
Vo = 1.2 V
0.00
1.00
2.00
3.00
4.00
5.00
6.00
IL- Load Current (A)
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS50601-SP
SLVSD45 – DECEMBER 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Description (continued).........................................
Pin Configuration and Functions .........................
Specifications.........................................................
7.1
7.2
7.3
7.4
7.5
7.6
7.7
8
8.3 Feature Description................................................. 15
8.4 Device Functional Modes........................................ 27
1
1
1
2
3
3
7
9
Application and Implementation ........................ 28
9.1 Application Information............................................ 28
9.2 Typical Application ................................................. 28
10 Power Supply Recommendations ..................... 34
11 Layout................................................................... 34
11.1 Layout Guidelines ................................................. 34
11.2 Layout Example .................................................... 35
Absolute Maximum Ratings ...................................... 7
ESD Ratings.............................................................. 7
Recommended Operating Conditions....................... 7
Thermal Information .................................................. 8
Electrical Characteristics........................................... 8
Dissipation Ratings ................................................. 10
Typical Characteristics ............................................ 11
12 Device and Documentation Support ................. 36
12.1
12.2
12.3
12.4
12.5
Detailed Description ............................................ 14
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
36
36
36
36
36
13 Mechanical, Packaging, and Orderable
Information ........................................................... 36
8.1 Overview ................................................................. 14
8.2 Functional Block Diagram ....................................... 15
13.1 Device Nomenclature............................................ 36
4 Revision History
DATE
REVISION
December 2015
*
NOTES
•
•
2
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Initial release.
Created separate data sheet for this part
number
Removed the ψJT thermal metric
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SLVSD45 – DECEMBER 2015
5 Description (continued)
The output voltage startup ramp is controlled by the SS/TR pin which allows operation as either a stand alone
power supply or in tracking situations. Power sequencing is also possible by correctly configuring the enable and
the open drain power good pins.
Cycle-by-cycle current limiting on the high-side FET protects the device in overload situations and is enhanced
by a low-side sourcing current limit which prevents current runaway. There is also a low-side sinking current limit
which turns off the low-side MOSFET to prevent excessive reverse current. Thermal shutdown disables the part
when die temperature exceeds thermal shutdown temperature.
6 Pin Configuration and Functions
HKH Package
20-Pin CFP
Bottom View
GND
1
20
PWRGD
EN
2
19
SS/TR
RT
3
18
COMP
SYNC
4
17
VSENSE
16
BOOT
Thermal Pad
(Bottom Side)
21
VIN
5
PVIN
6
15
PH
PVIN
7
14
PH
PGND
8
13
PH
PGND
9
12
PH
PGND
10
11
PH
Pin Functions
PIN
NO.
DESCRIPTION
NAME
1
GND
Return for control circuitry/thermal pad (1)
2
EN
EN pin has an internal pullup thus EN pin can be floated to enable the device. As an option external pullup can also
be added if desired. Adjust the input undervoltage lockout (UVLO) with two resistors.
3
RT
In internal oscillation mode, a resistor is connected between the RT pin and GND to set the switching frequency.
4
SYNC
Optional 1-MHz external system clock input. The device operates with an internal oscillator if this pin is left open.
5
VIN
Supplies the power to the output FET controllers
PVIN
Power input. Supplies the power switches of the power converter
PGND
Return for low-side power MOSFET
6
7
8
9
10
(1)
Thermal pad (analog ground) must be connected to PGND external to the package.
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Pin Functions (continued)
PIN
NO.
DESCRIPTION
NAME
11
12
13
PH
Switch node
16
BOOT
A bootstrap capacitor is required between BOOT and PH. The voltage on this capacitor carries the gate drive voltage
for the high-side MOSFET.
17
VSENSE
Inverting input of the gm error amplifier
18
COMP
Error amplifier output and input to the output switch current comparator. Connect frequency compensation to this pin.
19
SS/TR
Slow-start and tracking. An external capacitor connected to this pin sets the internal voltage reference rise time. The
voltage on this pin overrides the internal reference. It can be used for tracking and sequencing.
20
PWRGD
Power Good fault pin is an open-drain connection. Power Good fault pin. Asserts low if output voltage is low due to
thermal shutdown, dropout, overvoltage, or EN shutdown, or during slow start.
14
15
Bare Die Information
4
DIE THICKNESS
BACKSIDE FINISH
BACKSIDE
POTENTIAL
15 mils.
Silicon with backgrind
Ground
BOND PAD METALLIZATION
COMPOSITION
BOND PAD THICKNESS
Al5TiN
557.5 nm
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Bond Pad Coordinates in Microns
6
DESCRIPTION
PAD NUMBER
X MIN
Y MIN
X MAX
Y MAX
GND
1
400.77
5039.325
578.07
5216.625
EN
2
44.19
4169.79
221.49
4347.09
RT
3
44.19
3894.21
221.49
4071.51
SYNC
4
44.19
3618.63
221.49
3795.93
VIN
5
47.565
2952.27
224.865
3129.57
PVIN
6
280.215
2414.115
457.515
2591.415
PVIN
7
280.215
2170.665
457.515
2347.965
PVIN
8
280.215
1928.115
457.515
2105.415
PVIN
9
280.215
1684.665
457.515
1861.965
PGND
10
254.52
1236.285
431.82
1413.585
PGND
11
254.52
1008.315
431.82
1185.615
PGND
12
254.52
780.345
431.82
957.645
PGND
13
254.52
552.375
431.82
729.675
PGND
14
254.52
324.405
431.82
501.705
PGND
15
254.52
96.435
431.82
273.735
PH
16
1590.12
99.405
1767.42
276.705
PH
17
1590.12
321.435
1767.42
498.735
PH
18
1590.12
555.345
1767.42
732.645
PH
19
1590.12
777.375
1767.42
954.675
PH
20
1590.12
1011.285
1767.42
1188.585
PH
21
1590.12
1233.315
1767.42
1410.615
PH
22
1564.335
1684.665
1741.635
1861.965
PH
23
1564.335
1928.115
1741.635
2105.415
PH
24
1564.335
2170.665
1741.635
2347.965
PH
25
1564.335
2414.115
1741.635
2591.415
BOOT
26
1801.71
3352.14
1979.01
3529.44
VSENSE
27
1801.71
3644.145
1979.01
3821.445
COMP
28
1801.71
3940.92
1979.01
4118.22
SS/TR
29
1801.71
4216.5
1979.01
4393.8
PWRGD
30
1463.67
5039.325
1640.97
5216.625
GND
31
1251.09
5039.325
1428.39
5216.625
GND
32
1038.51
5039.325
1215.81
5216.625
GND
33
825.93
5039.325
1003.23
5216.625
GND
34
613.35
5039.325
790.65
5216.6
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SLVSD45 – DECEMBER 2015
7 Specifications
7.1 Absolute Maximum Ratings
over operating temperature (unless otherwise noted) (1)
Input voltage
Output voltage
MIN
MAX
VIN
–0.3
7
PVIN
–0.3
7
EN
–0.3
5.5
BOOT
–0.3
14
VSENSE
–0.3
3.3
COMP
–0.3
3.3
PWRGD
–0.3
5.5
SS/TR
–0.3
5.5
SYNC
–0.3
7
BOOT-PH
0
7
PH
–1
7
PH 10-ns transient
Vdiff
(GND to exposed thermal pad)
Sink current
V
V
–3
7
–0.2
0.2
V
6
A
±100
µA
Output current
Source current
UNIT
PH
Current limit
RT
PH
Current limit
PVIN
Current limit
COMP
PWRGD
A
A
A
±200
µA
–0.1
5
mA
Operating junction temperature
–55
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic
discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins
(1)
±1000
Charged device model (CDM), per JEDEC specification JESD22-C101, all pins (2)
±1000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
TJ
Junction operating temperature range
–55
NOM
MAX
UNIT
125
°C
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7.4 Thermal Information
TPS50601-SP
THERMAL METRIC (1)
HKH (CFP)
UNIT
20 PINS
RθJC(bot)
(1)
Junction-to-case (bottom) thermal resistance
0.514
°C/W
Taken per Mil Standard 883 method 1012.1
7.5 Electrical Characteristics
TJ = –55°C to 125°C, VIN = 3 V to 6.3 V, PVIN = 1.6 V to 6.3 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN AND PVIN PINS)
PVIN operating input voltage
VIN operating input voltage
VIN internal UVLO threshold
1.6
6.3
V
3
6.3
V
3
V
VIN rising
2.75
VIN shutdown supply current
VEN = 0 V
2.5
5.9
mA
VIN operating – non switching supply current
VSENSE = VBG
5
10
mA
1.13
1.18
VIN internal UVLO hysteresis
50
mV
ENABLE AND UVLO (EN PIN)
Rising
Enable threshold
Falling
1.05
1.09
V
Input current
VEN = 1.1 V
3.2
μA
Hysteresis current
VEN = 1.3 V
3
μA
VOLTAGE REFERENCE
0 A ≤ Iout ≤ 6 A
Voltage reference
–55°C
0.767
0.795
0.804
25°C
0.785
0.795
0.804
125°C
0.785
0.795
0.815
V
MOSFET
High-side switch resistance
BOOT-PH = 2.2 V
55
mΩ
High-side switch resistance (1)
BOOT-PH = 6.3 V
50
mΩ
Low-side switch resistance (1)
VIN = 6.3 V
50
mΩ
ERROR AMPLIFIER
Error amplifier transconductance (gm) (2)
–2 μA < ICOMP < 2 μA, V(COMP) = 1 V
Error amplifier dc gain (2)
VSENSE = 0.792 V
Error amplifier source/sink (2)
Start switching threshold
V(COMP) = 1 V, 40-mV input overdrive
(2)
1300
μS
39000
V/V
±125
μA
0.25
COMP to Iswitch gm (2)
V
18
A/V
CURRENT LIMIT
High-side switch current limit threshold
Low-side switch sourcing current limit
(3)
(3)
Low-side switch sinking current limit
VIN = 6.3 V
8
11
A
VIN = 6.3 V
7
10
A
3
A
175
°C
10
°C
VIN = 6.3 V
THERMAL SHUTDOWN
Thermal shutdown
Thermal shutdown hysteresis
INTERNAL SWITCHING FREQUENCY
Internally set frequency
(1)
(2)
(3)
8
RT = Open
395
500
585
kHz
Measured at pins
Ensured by design only. Not tested in production.
Parameter is not tested in production.
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Electrical Characteristics (continued)
TJ = –55°C to 125°C, VIN = 3 V to 6.3 V, PVIN = 1.6 V to 6.3 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
RT = 100 kΩ (1%)
Externally set frequency
TYP
MAX
UNIT
480
RT = 485 kΩ (1%)
100
RT = 47 kΩ (1%)
1000
kHz
EXTERNAL SYNCHRONIZATION
SYNC out low-to-high rise time (10%/90%)
Cload = 25 pF
25
111
ns
SYNC out high-to-low fall time (90%/10%)
Cload = 25 pF
3
15
ns
Falling edge delay time (4)
180
SYNC out high level threshold
IOH = 50 µA
SYNC out low level threshold
IOL = 50 µA
SYNC in low level threshold
2
V
600
800
1.85
Percent of program frequency
mV
mV
SYNC in high level threshold
SYNC in frequency range (5)
°
V
–5%
5%
100
1000
kHz
175
ns
PH (PH PIN)
Minimum on time
Measured at 10% to 90% of VIN,
25°C, IPH = 2 A
Minimum off time
BOOT-PH ≥ 3 V
94
500
ns
BOOT (BOOT PIN)
BOOT-PH UVLO
2.2
3
V
90
mV
SLOW START AND TRACKING (SS/TR PIN)
SS charge current
SS/TR to VSENSE matching
μA
2.5
V(SS/TR) = 0.4 V
30
VSENSE falling (fault)
91
% Vref
VSENSE rising (good)
94
% Vref
VSENSE rising (fault)
109
% Vref
VSENSE falling (good)
106
% Vref
POWER GOOD (PWRGD PIN)
VSENSE threshold
Output high leakage
VSENSE = Vref, V(PWRGD) = 5 V
Output low
I(PWRGD) = 2 mA
Minimum VIN for valid output
V(PWRGD) < 0.5 V at 100 μA
Minimum SS/TR voltage for PWRGD
(4)
(5)
30
0.6
181
nA
0.3
V
1
V
1.4
V
Bench verified. Not tested in production.
Parameter is production tested at nominal voltage with VIN = PVIN = 5V.
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7.6 Dissipation Ratings
See
(1)
(2)
(3)
(4)
(1) (2) (3) (4)
PACKAGE
RθJA THERMAL IMPEDANCE,
JUNCTION TO AMBIENT
RθJC THERMAL IMPEDANCE,
JUNCTION TO CASE (THERMAL PAD)
RθJB THERMAL IMPEDANCE,
JUNCTION TO BOARD
HKH
39.9°C/W
0.52°C/W
43.1°C/W
Maximum power dissipation may be limited by overcurrent protection
Power rating at a specific ambient temperature, TA, should be determined with a junction temperature of 150°C. This is the point where
distortion starts to substantially increase. Thermal management of the PCB should strive to keep the junction temperature at or below
150°C for best performance and long-term reliability. See power dissipation estimate in Application and Implementation for more
information.
Test board conditions:
(a) 2.5 inches × 2.5 inches, 4 layers, thickness: 0.062 inch
(b) 2-oz. copper traces located on the top of the PCB
(c) 2-oz. copper ground planes on the 2 internal layers and bottom layer
(d) 40.010-inch thermal vias located under the device package
For information on thermal characteristics, see SPRA953.
10000000
Life (Hours)
1000000
Duty Cycle
100
80
60
40
20
100000
10000
1000
95
105
115
125
135
145
Operating Junction Temperature (°C)
A.
See data sheet for absolute maximum and minimum recommended operating conditions.
B.
Product operating life design goal is >15 years for 65°C ≤ TJ ≤ 95°C based on silicon technology characterization per
MIL-PRF-38535.
C.
The predicted operating lifetime versus junction temperature is based on reliability modeling using electromigration as
the dominant failure mechanism affecting device wearout for the specific device process and design characteristics.
Figure 1. 6-A Continuous Current Estimated Device Life
10
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7.7 Typical Characteristics
60.00
0.83
VIN = 3 V
58.00
0.82
VIN = 6.3 V
Voltage Reference (V)
Current Sharing (%)
56.00
54.00
POL 2
52.00
50.00
48.00
POL 1
46.00
0.81
0.80
0.79
0.78
44.00
0.77
42.00
0.76
40.00
0.00
2.00
4.00
6.00
8.00
10.00
±75
12.00
±50
0
±25
75
100
125
C003
6000
Nom
Min
Max
VIN = 3 V
Shutdown Quiescent Current ( A)
Oscillator Frequency (kHz)
50
Figure 3. Voltage Reference vs Temperature
Figure 2. Current Sharing vs Load Current
1250
1000
750
500
250
VIN = 6.3 V
5000
4000
3000
2000
1000
0
0
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
125
±75
±25
0
25
50
75
100
Junction Temperature (ƒC)
Figure 4. Oscillator Frequency vs Temperature
125
C005
Figure 5. Shutdown Quiescent Current vs Temperature
±5
VIN = 3 V
En Pin Pull-Up Current ( A)
VIN = 6.3 V
4
±50
C004
5
EN Pin Hysteresis Current ( A)
25
Junction Temperature (ƒC)
IL- Current Load (A)
3
2
1
0
VIN = 3 V
VIN = 6.3 V
±6
±7
±8
±9
±10
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
125
±75
±50
Figure 6. EN Pin Hysteresis Current vs Temperature
±25
0
25
50
75
100
Junction Temperature (ƒC)
C006
125
C007
Figure 7. EN Pin Pullup Current vs Temperature
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1.170
VIN = 3 V
En Pin UVLO Threshold (V)
1.165
VIN = 6.3 V
1.160
1.155
1.150
1.145
1.140
1.135
1.130
±75
±50
0
±25
25
50
75
100
125
Junction Temperature (ƒC)
C008
Figure 8. EN Pin UVLO Threshold vs Temperature
Non-Switching Operating Quiescent Current ( A)
Typical Characteristics (continued)
8000
VIN = 6.3 V
7000
6500
6000
5500
5000
4500
4000
3500
3000
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
125
C009
Figure 9. Non-Switching Operating Quiescent Current (VIN)
vs Temperature
3.2
0.05
VIN = 3 V
VIN = 6.3 V
3.0
(SS - Vsense) Offset (V)
Iss Slow Start Charge Current ( A)
VIN = 3 V
7500
2.8
2.6
2.4
2.2
VIN = 6.3 V
0.04
0.03
0.02
0.01
0.00
±75
±50
±25
0
25
50
75
100
125
Junction Temperature (ƒC)
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
C010
Figure 10. Slow Start Charge Current vs Temperature
125
C011
Figure 11. (SS-VSENSE) Offset vs Temperature
170
12
11
On-State Resistance (m
Current Limit Threshold (A)
VIN = 3 V
150
10
9
8
7
VIN = 6.3 V
130
110
90
70
50
30
VIN = 6.3 V
6
10
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
C012
Figure 12. High-Side Current Limit Threshold vs
Temperature
12
125
125
C002
Figure 13. Low-Side RDS(On) vs Temperature
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Typical Characteristics (continued)
180
250
On-State Resistance (m
160
Minimum Controllable On Time (ns)
VIN = 3 V
VIN = 6.3 V
140
120
100
80
60
40
20
VIN = 3 V
VIN = 6.3 V
200
150
100
50
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
125
±75
±50
±25
0
25
50
75
100
Junction Temperature (ƒC)
C001
Figure 14. High-Side RDS(On) vs Temperature
125
C013
Figure 15. Minimum Controllable On-Time vs Temperature
Minimum Controllable Duty Ratio (%)
15
VIN = 3 V
VIN = 6.3 V
10
5
0
±75
±50
±25
0
25
50
75
Junction Temperature (ƒC)
100
125
C014
Figure 16. Minimum Controllable Duty Ratio vs Temperature
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8 Detailed Description
8.1 Overview
The device is a 6.3-V or 6-A synchronous step-down (buck) converter with two integrated N-channel MOSFETs.
To improve performance during line and load transients, the device implements a constant frequency, peak
current mode control, which also simplifies external frequency compensation. The wide switching frequency, 100
kHz to 1 MHz, allows for efficiency and size optimization when selecting the output filter components.
The device is designed for safe monotonic startup into prebiased loads. The default start up is when VIN is
typically 3 V. The EN pin has an internal pullup current source that can be used to adjust the input voltage UVLO
with two external resistors. In addition, the EN pin can be floating for the device to operate with the internal
pullup current. The total operating current for the device is approximately 5 mA when not switching and under no
load. When the device is disabled, the supply current is typically less than 2.5 mA.
The integrated MOSFETs allow for high-efficiency power supply designs with continuous output currents up to 6
A. The MOSFETs have been sized to optimize efficiency for lower duty cycle applications.
The device reduces the external component count by integrating the boot recharge circuit. The bias voltage for
the integrated high-side MOSFET is supplied by a capacitor between the BOOT and PH pins. The boot capacitor
voltage is monitored by a BOOT to PH UVLO (BOOT-PH UVLO) circuit allowing the PH pin to be pulled low to
recharge the boot capacitor. The device can operate over duty cycle range per Equation 2 and Equation 3 as
long as the boot capacitor voltage is higher than the preset BOOT-PH UVLO threshold, which is typically 2.2 V.
The output voltage can be stepped down to as low as the 0.795-V voltage reference (Vref).
The device has a power good comparator (PWRGD) with hysteresis which monitors the output voltage through
the VSENSE pin. The PWRGD pin is an open-drain MOSFET which is pulled low when the VSENSE pin voltage
is less than 91% or greater than 109% of the reference voltage Vref and asserts high when the VSENSE pin
voltage is 94% to 106% of the Vref.
The SS/TR (slow start/tracking) pin is used to minimize inrush currents or provide power-supply sequencing
during power-up. A small-value capacitor or resistor divider should be coupled to the pin for slow start or critical
power-supply sequencing requirements.
The device is protected from output overvoltage, overload, and thermal fault conditions. The device minimizes
excessive output overvoltage transients by taking advantage of the overvoltage circuit power good comparator.
When the overvoltage comparator is activated, the high-side MOSFET is turned off and prevented from turning
on until the VSENSE pin voltage is lower than 106% of the Vref. The device implements both high-side MOSFET
overload protection and bidirectional low-side MOSFET overload protections, which help control the inductor
current and avoid current runaway. The device also shuts down if the junction temperature is higher than thermal
shutdown trip point. The device is restarted under control of the slow-start circuit automatically when the junction
temperature drops 10°C typical below the thermal shutdown trip point.
14
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8.2 Functional Block Diagram
PWRGD
VIN
EN
Shutdown
Ip
Enable
Comparator
Ih
Thermal
Shutdown
PVIN PVIN
UVLO
Shutdown
UV
Shutdown
Logic
Logic
Enable
Threshold
OV
Boot
Charge
Current
Sense
Minimum Clamp
Pulse Skip
ERROR
AMPLIFIER
VSENSE
BOOT
Boot
UVLO
V/I
SS/TR
HS MOSFET
Current
Comparator
Voltage
Reference
Power Stage
& Deadtime
Control
Logic
PH
PH
Slope
Compensation
VIN
Overload Recovery
and
Clamp
Oscillator
Regulator
RT Bias
LS MOSFET
Current Limit
Current
Sense
PGND
SYNC
Detect
COMP
SYNC
PGND
RT
Thermal Pad/GND
8.3 Feature Description
8.3.1 VIN and Power VIN Pins (VIN and PVIN)
The device allows for a variety of applications by using the VIN and PVIN pins together or separately. The VIN
pin voltage supplies the internal control circuits of the device. The PVIN pin voltage provides the input voltage to
the power converter system.
If tied together, the input voltage for VIN and PVIN can range from 3 to 6.3 V. If using the VIN separately from
PVIN, the VIN pin must be between 3 and 6.3 V, and the PVIN pin can range from as low as 1.6 to 6.3 V. A
voltage divider connected to the EN pin can adjust the input voltage UVLO appropriately. Adjusting the input
voltage UVLO on the PVIN pin helps to provide consistent power-up behavior.
8.3.2 PVIN vs Frequency
With VIN tied to PVIN, minimum off-time determines what output voltage is achievable over frequency range.
8.3.3 Voltage Reference
The voltage reference system produces a precise voltage reference as indicated in Electrical Characteristics.
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Feature Description (continued)
8.3.4 Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output (VOUT) to the VSENSE pin. TI recommends to
use 1% tolerance or better divider resistors. Start with a 10 kΩ for R15 (top resistor) and use Equation 1 to
calculate R38 (bottom resistor divider). To improve efficiency at light loads, consider using larger-value resistors.
If the values are too high, the regulator is more susceptible to noise and voltage errors from the VSENSE input
current are noticeable.
Vref
R38 =
R15
Vo - Vref
where
•
Vref = 0.795 V
(1)
The minimum output voltage and maximum output voltage can be limited by the minimum on-time of the highside MOSFET and bootstrap voltage (BOOT-PH voltage) respectively. For more information, see Bootstrap
Voltage (BOOT) and Low Dropout Operation.
8.3.5 Maximum Duty Cycle Limit
The TPS50601-SP can operate at duty cycle per Equation 2 and Equation 3 as long as the boot capacitor
voltage is higher than the preset BOOT-PH UVLO threshold, which is typically 2.2 V.
Duty cycle can be calculated based on Equation 2.
VOUT + IOUT_max · RTesr + IOUT_max · Rds_low
D(VIN) =
VIN - IOUT_max · Rds_high + IOUT_max · Rds_low
where
•
•
•
•
•
RTesr = Rdcr + Rtrace
Rdcr is the dc resistance of the inductor.
Rtrace is the dc trace resistance (miscellaneous drop).
Rds_high is the maximum RDS of the high-side MOSFET.
Rds_low is the maximum RDS of the low-side MOSFET.
(2)
8.3.6 PVIN vs Frequency
With VIN tied to PVIN, minimum off-time determines the output voltage that is achievable over frequency range.
For VIN = PVIN must be ≥ 3 V. For VIN = 3 V, PVIN can vary from 1.6 to 6.3 V as highlighted in Electrical
Characteristics.
This is given by Equation 3.
VO + IO(Rds _ onLS + Rmisc )
PVin _ min(fSW ) =
1 - Toff _ min· fSW
where
•
•
•
Rds_onLS = Low-side Rds-on
Rmisc = Miscellaneous trace drops
Toff_min = Minimum off time
(3)
Using this approach, the designer can calculate minimum PVIN required for specific VOUT as indicated in the
example in Figure 17.
16
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Feature Description (continued)
VO = 1.5 V
PVin_min(100 kHz = 1.889 V)
PVin_min(1000 kHz = 3.396 V)
fsw - Switching Frequency - Hz
Figure 17. PVIN vs Frequency
8.3.7 Safe Start-Up into Prebiased Outputs
The device is designed to prevent the low-side MOSFET from discharging a prebiased output. During monotonic
prebiased startup, the low-side MOSFET is not allowed to sink current until the SS/TR pin voltage is higher than
1.4 V.
8.3.8 Error Amplifier
The device uses a transconductance error amplifier. The error amplifier compares the VSENSE pin voltage to the
lower of the SS/TR pin voltage or the internal 0.795-V voltage reference. The transconductance of the error
amplifier is 1300 μA/V during normal operation. The frequency compensation network is connected between the
COMP pin and ground. Error amplifier dc gain is typically 39000 V/V with minimum value of 22000 V/V per
design.
8.3.9 Slope Compensation
The device adds a compensating ramp to the switch current signal. This slope compensation prevents
subharmonic oscillations. The available peak inductor current remains constant over the full duty cycle range.
Minimum peak-to-peak inductor current should be greater than 1 A.
8.3.10 Enable and Adjust UVLO
The EN pin provides electrical on and off control of the device. When the EN pin voltage exceeds the threshold
voltage, the device starts operation. If the EN pin voltage is pulled below the threshold voltage, the regulator
stops switching and enters low Iq state. If an external Schottky diode is used from VIN to boot, then a bleeder
may be required 500 mV.
The EN pin has a small pullup current, Ip, which sets the default state of the pin to enable when no external
components are connected. The pullup current is also used to control the voltage hysteresis for the UVLO
function because it increases by Ih after the EN pin crosses the enable threshold. Calculate the UVLO thresholds
with Equation 4 and Equation 5.
TPS50601-SP
VIN
ip
ih
R1
EN
R2
Figure 18. Adjustable VIN UVLO
TPS50601-SP
PVIN
ip
ih
R1
EN
R2
Figure 19. Adjustable PVIN UVLO, VIN ≥ 3 V
TPS50601-SP
PVIN
VIN
ip
ih
R1
EN
R2
Figure 20. Adjustable VIN and PVIN UVLO
æV
ö
VSTART ç ENFALLING ÷ - VSTOP
è VENRISING ø
R1 =
æ V
ö
Ip ç1 - ENFALLING ÷ + Ih
V
ENRISING ø
è
18
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Feature Description (continued)
R1´ VENFALLING
VSTOP - VENFALLING + R1(Ip + Ih )
R2 =
where
•
•
•
•
Ih = 3 μA
Ip = 3.2 μA
VENRISING = 1.131 V
VENFALLING = 1.09 V
(5)
8.3.11 Adjustable Switching Frequency and Synchronization (SYNC)
The switching frequency of the device supports three modes of operations. The modes of operation are set by
the conditions on the RT and SYNC pins. At a high level, these modes can be described as master, internal
oscillator, and external synchronization modes.
In master mode, the RT pin should be left floating; the internal oscillator is set to 500 kHz, and the SYNC pin is
set as an output clock. The SYNC output is in phase with respect to the internal oscillator. SYNC out signal level
is the same as VIN level with 50% duty cycle. SYNC signal feeding the slave module—which is in phase with the
master clock—gets internally inverted (180° out of phase with the master clock) internally in the slave module.
In internal oscillator mode, a resistor is connected between the RT pin and GND. The SYNC pin requires a 10kΩ resistor to GND for this mode to be effective. The switching frequency of the device is adjustable from 100
kHz to 1 MHz by placing a maximum of 510 kΩ and a minimum of 47 kΩ respectively. To determine the RT
resistance for a given switching frequency, use Equation 6 or the curve in Figure 21. To reduce the solution size,
the designer should set switching frequency as high as possible, but consider the tradeoffs of supply efficiency
and minimum controllable on-time.
-1.0549
RT(FSW) = 67009 x FSW
where
•
•
RT in kΩ
FSW in kHz
(6)
3
1.2x10
3
1.08x10
RT(FSW)
960
840
RT - kW
720
600
RT = 500 kW
480
360
240
120
0
0
100
200
300
400
500
600
700
800
900
1x 10
3
Switching Frequency (FSW) - kHz
Figure 21. RT vs Switching Frequency
When operating the converter in internal oscillator mode (internal oscillator determines the switching frequency
(500 kHz) default), the synchronous pin becomes the output and there is a phase inversion. When trying to
parallel with another converter, the RT pin of the second (slave) converter must have its RT pin populated such
that the converter frequency of the slave converter must be within ±5% of the master converter. This is required
because the RT pin also sets the proper operation of slope compensation.
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Feature Description (continued)
In external synchronization mode, a resistor is connected between the RT pin and GND. The Sync pin requires a
toggling signal for this mode to be effective. The switching frequency of the device goes 1:1 with that of Sync pin.
External system clock-user supplied sync clock signal determines the switching frequency. If no external clock
signal is detected for 20 µs, then TPS50601-SP transitions to its internal clock, which is typically 500 kHz. An
external synchronization using an inverter to obtain phase inversion is necessary. RT values of the master and
slave converter must be within ±5% of the external synchronization frequency. This is necessary for proper slope
compensation. A resistance in the RT pin is required for proper operation of the slope compensation circuit. To
determine the RT resistance for a given switching frequency, use Equation 6 or the curve in Figure 21. To reduce
the solution size, the designer should set switching frequency as high as possible, but consider the tradeoffs of
supply efficiency and minimum controllable on-time.
These modes are described in Table 1.
Table 1. Switching Frequency, SYNC and RT Pin Usage Table
RT PIN
Float
SWITCHING
FREQUENCY
SYNC PIN
DESCRIPTION AND NOTES
Generates an output signal
500 kHz
SYNC pin behaves as an output. SYNC output signal
is 180° out of phase to the internal 500-kHz switching
frequency.
10-kΩ resistor to AGND
100 kHz to 1 MHz
Internally generated switching frequency is based
upon the resistor value present at the RT pin.
Internally synchronized to
external clock
Set value of RT that corresponds to the externally
supplied sync frequency.
47- to 485-kΩ resistor
User-supplied sync clock or
to AGND
TPS50601-SP master device
sync output
8.3.12 Slow Start (SS/TR)
The device uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as the reference
voltage and regulates the output accordingly. A capacitor on the SS/TR pin to ground implements a slow-start
time. The device has an internal pullup current source of 5 mA that charges the external slow-start capacitor.
Equation 7 shows the calculations for the slow-start time (Tss, 10% to 90%) and slow-start capacitor (Css). The
voltage reference (Vref) is 0.795 V and the slow-start charge current (Iss) is 2.5 μA.
t SS (ms) =
Css (nF) ´ Vref (V)
Iss (µA)
(7)
When the input UVLO is triggered, the EN pin is pulled below 1.032 V, or a thermal shutdown event occurs the
device stops switching and enters low current operation. At the subsequent power-up, when the shutdown
condition is removed, the device does not start switching until it has discharged its SS/TR pin to ground ensuring
proper soft-start behavior.
8.3.13 Power Good (PWRGD)
The PWRGD pin is an open-drain output. When the VSENSE pin is between 94% and 106% of the internal
voltage reference, the PWRGD pin pull-down is deasserted and the pin floats. TI recommends to use a pullup
resistor between 10 to 100 kΩ to a voltage source that is 5.5 V or less. The PWRGD is in a defined state when
the VIN input voltage is greater than 1 V but has reduced current sinking capability. The PWRGD achieves full
current sinking capability when the VIN input voltage is above 3 V.
The PWRGD pin is pulled low when VSENSE is lower than 91% or greater than 109% of the nominal internal
reference voltage. Also, the PWRGD is pulled low, if the input UVLO or thermal shutdown are asserted, the EN
pin is pulled low or the SS/TR pin is below 1.4 V.
8.3.14 Bootstrap Voltage (BOOT) and Low Dropout Operation
The device has an integrated boot regulator, and requires a small ceramic capacitor between the BOOT and PH
pins to provide the gate drive voltage for the high-side MOSFET. The boot capacitor is charged when the BOOT
pin voltage is less than VIN and BOOT-PH voltage is below regulation. The value of this ceramic capacitor
should be 0.1 μF. TI recommends a ceramic capacitor with an X7R- or X5R-grade dielectric with a voltage rating
of 10 V or higher because of the stable characteristics over temperature and voltage.
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To improve dropout, the device is designed to operate at a high duty cycle as long as the BOOT to PH pin
voltage is greater than the BOOT-PH UVLO threshold, which is typically 2.1 V. When the voltage between BOOT
and PH drops below the BOOT-PH UVLO threshold, the high-side MOSFET is turned off and the low-side
MOSFET is turned on allowing the boot capacitor to be recharged. In applications with split-input voltage rails,
high duty cycle operation can be achieved as long as (VIN – PVIN) > 4 V.
Maximum switching frequency is also limited by minimum on-time (specified in Electrical Characteristics) as
indicated by Equation 8. Switching frequency will be worst case at no load conditions.
1
VO + Rds_on · (IO)
FSW =
=
T
VIN · (Ton_max)
(8)
8.3.15 Sequencing (SS/TR)
Many of the common power-supply sequencing methods can be implemented using the SS/TR, EN, and
PWRGD pins.
The sequential method is shown in Figure 22 using two TPS50601-SP devices. The power good of the first
device is coupled to the EN pin of the second device, which enables the second power supply after the primary
supply reaches regulation.
TPS50601-SP
TPS50601-SP
PWRGD
EN
EN
SS/TR
SS/TR
PWRGD
Figure 22. Sequential Start-Up Sequence
Figure 23 shows the method implementing ratiometric sequencing by connecting the SS/TR pins of two devices
together. The regulator outputs ramp up and reach regulation at the same time. When calculating the slow-start
time, the pullup current source must be doubled in Equation 7.
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TPS50601-SP
EN
SS/TR
PWRGD
TPS50601-SP
EN
SS/TR
PWRGD
Figure 23. Ratiometric Start-Up Sequence
Ratiometric and simultaneous power-supply sequencing can be implemented by connecting the resistor network
of R1 and R2 (shown in Figure 24) to the output of the power supply that needs to be tracked or another voltage
reference source. Using Equation 9 and Equation 10, the tracking resistors can be calculated to initiate the Vout2
slightly before, after, or at the same time as Vout1. Equation 11 is the voltage difference between Vout1 and
Vout2.
To design a ratiometric start-up in which the Vout2 voltage is slightly greater than the Vout1 voltage when Vout2
reaches regulation, use a negative number in Equation 9 and Equation 10 for ΔV. Equation 11 results in a
positive number for applications where the Vout2 is slightly lower than Vout1 when Vout2 regulation is achieved.
The ΔV variable is 0 V for simultaneous sequencing. To minimize the effect of the inherent SS/TR to VSENSE
offset (Vssoffset, 29 mV) in the slow-start circuit and the offset created by the pullup current source (Iss, 2 μA)
and tracking resistors, the Vssoffset and Iss are included as variables in the equations.
To ensure proper operation of the device, the calculated R1 value from Equation 9 must be greater than the
value calculated in Equation 12.
R1 =
Vout2 + D V
Vssoffset
´
Vref
Iss
(9)
Vref ´ R1
R2 =
Vout2 + DV - Vref
DV = Vout1 - Vout2
R1 > 2800 ´ Vout1- 180 ´ DV
22
(10)
(11)
(12)
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TPS50601-SP
EN
VOUT1
SS/TR
PWRGD
TPS50601-SP
EN
VOUT 2
R1
SS/TR
R2
PWRGD
R4
R3
Figure 24. Ratiometric and Simultaneous Start-Up Sequence
8.3.16 Output Overvoltage Protection (OVP)
The device incorporates an output OVP circuit to minimize output voltage overshoot. For example, when the
power supply output is overloaded, the error amplifier compares the actual output voltage to the internal
reference voltage. If the VSENSE pin voltage is lower than the internal reference voltage for a considerable time,
the output of the error amplifier demands maximum output current. After the condition is removed, the regulator
output rises and the error amplifier output transitions to the steady-state voltage. In some applications with small
output capacitance, the power supply output voltage can respond faster than the error amplifier. This leads to the
possibility of an output overshoot. The OVP feature minimizes the overshoot by comparing the VSENSE pin
voltage to the OVP threshold. If the VSENSE pin voltage is greater than the OVP threshold, the high-side
MOSFET is turned off, preventing current from flowing to the output and minimizing output overshoot. When the
VSENSE voltage drops lower than the OVP threshold, the high-side MOSFET is allowed to turn on at the next
clock cycle.
8.3.17 Overcurrent Protection
The device is protected from overcurrent conditions by cycle-by-cycle current limiting on both the high-side and
low-side MOSFET.
8.3.17.1 High-Side MOSFET Overcurrent Protection
The device implements current mode control which uses the COMP pin voltage to control the turn off of the highside MOSFET and the turn on of the low-side MOSFET on a cycle-by-cycle basis. Each cycle the switch current
and the current reference generated by the COMP pin voltage are compared, when the peak switch current
intersects the current reference, the high-side switch is turned off.
8.3.17.2 Low-Side MOSFET Overcurrent Protection
While the low-side MOSFET is turned on its conduction current is monitored by the internal circuitry. During
normal operation the low-side MOSFET sources current to the load. At the end of every clock cycle, the low-side
MOSFET sourcing current is compared to the internally set low-side sourcing current limit. If the low-side
sourcing current is exceeded, the high-side MOSFET is not turned on and the low-side MOSFET stays on for the
next cycle. The high-side MOSFET is turned on again when the low-side current is below the low-side sourcing
current limit at the start of a cycle.
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The low-side MOSFET may also sink current from the load. If the low-side sinking current limit is exceeded, the
low-side MOSFET is turned off immediately for the rest of that clock cycle. In this scenario, both MOSFETs are
off until the start of the next cycle.
When the low-side MOSFET turns off, the switch node increases and forward biases the high-side MOSFET
parallel diode (the high-side MOSFET is still off at this stage).
8.3.18 TPS50601-SP Thermal Shutdown
The internal thermal shutdown circuitry forces the device to stop switching if the junction temperature exceeds
175°C typically. The device reinitiates the power-up sequence when the junction temperature drops below 165°C
typically.
8.3.19 Turn-On Behavior
Minimum on-time specification determines the maximum operating frequency of the design. As the unit starts up
and goes through its soft-start process, the required duty-cycle is less than the minimum controllable on-time.
This can cause the converter to skip pulses. Thus, instantaneous output pulses can be higher or lower than the
desired voltage. This behavior is only evident when operating at high frequency with high bandwidth. When the
minimum on-pulse is greater than the minimum controllable on-time, the turn-on behavior is normal. When
operating at low frequencies (100 kHz or less), the turn-on behavior does not exhibit any ringing at initial startup.
8.3.20 Small Signal Model for Loop Response
Figure 25 shows an equivalent model for the device control loop, which can be modeled in a circuit simulation
program to check frequency response and transient responses. The error amplifier is a transconductance
amplifier with a gm of 1300 μA/V. The error amplifier can be modeled using an ideal voltage-controlled current
source. The resistor, Roea (30 MΩ), and capacitor, Coea (20.7 pF), model the open-loop gain and frequency
response of the error amplifier. The 1-mV ac voltage source between the nodes a and b effectively breaks the
control loop for the frequency response measurements. Plotting a/c and c/b show the small signal responses of
the power stage and frequency compensation respectively. Plotting a/b shows the small signal response of the
overall loop. The dynamic loop response can be checked by replacing the RL with a current source with the
appropriate load-step amplitude and step rate in a time domain analysis.
PH
VOUT
Power Stage
18 A/V
a
b
c
0.8 V
R3 Coea
C2
R1
RESR
VSENSE
CO
COMP
C1
Roea
gm
1300 mA/V
RL
R2
Figure 25. Small Signal Model For Loop Response
8.3.21 Simple Small Signal Model for Peak Current Mode Control
Figure 26 is a simple small signal model that can be used to understand how to design the frequency
compensation. The device power stage can be approximated to a voltage-controlled current source (duty cycle
modulator) supplying current to the output capacitor and load resistor. Equation 13 shows the control to output
transfer function, which consists of a dc gain, one dominant pole, and one ESR zero. The quotient of the change
in switch current and the change in COMP pin voltage (node c in Figure 25) is the power stage transconductance
(gmps), which is 18 A/V for the device. The dc gain of the power stage is the product of gmps and the load
24
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resistance (RL) as shown in Equation 14 with resistive loads. As the load current increases, the dc gain
decreases. This variation with load may seem problematic at first glance, but fortunately, the dominant pole
moves with load current (see Equation 15). The combined effect is highlighted by the dashed line in Figure 27.
As the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB crossover
frequency the same for the varying load conditions, which makes it easier to design the frequency compensation.
VOUT
VC
RESR
RL
gm ps
CO
Figure 26. Simplified Small Signal Model for Peak Current Mode Control
VOUT
Adc
VC
RESR
fp
RL
gm ps
CO
fz
Figure 27. Simplified Frequency Response for Peak Current Mode Control
æ
ç1+
2p
VOUT
= Adc ´ è
VC
æ
ç1+
è 2p
ö
s
÷
´ ¦z ø
ö
s
÷
´ ¦p ø
(13)
Adc = gmps ´ RL
(14)
1
¦p =
C O ´ R L ´ 2p
(15)
1
CO ´ RESR ´ 2p
¦z =
where
•
•
•
•
gmea is the GM amplifier gain (1300 μA/V).
gmps is the power stage gain (18 A/V).
RL is the load resistance.
CO is the output capacitance.
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RESR is the equivalent series resistance of the output capacitor.
(16)
8.3.22 Small Signal Model for Frequency Compensation
The device uses a transconductance amplifier for the error amplifier and readily supports two of the commonly
used frequency compensation circuits shown in Figure 28. In Type 2A, one additional high-frequency pole is
added to attenuate high-frequency noise.
The following design guidelines are provided for advanced users who prefer to compensate using the general
method. The step-by-step design procedure described in Detailed Design Procedure may also be used.
VOUT
R1
VSENSE
COMP Type 2A
Vref
R2
gm ea
Roea
R3
Coea
C2
Type 2B
R3
C1
C1
Figure 28. Types of Frequency Compensation
The general design guidelines for device loop compensation are as follows:
1. Determine the crossover frequency ƒc. A good starting point is one-tenth of the switching frequency, ƒSW.
2. R3 can be determined by:
2p ´ ¦ c ´ VOUT ´ Co
R3 =
gmea ´ Vref ´ gmps
where
•
•
•
gmea is the GM amplifier gain ( 1300 μA/V).
gmps is the power stage gain (18 A/V).
Vref is the reference voltage (0.795 V)
(17)
æ
ö
1
ç ¦p =
÷
CO ´ RL ´ 2p ø .
3. Place a compensation zero at the dominant pole è
C1 can be determined by
R ´ Co
C1 = L
R3
(18)
4. C2 is optional. It can be used to cancel the zero from the equivalent series resistance (ESR) of the output
capacitor Co.
R
´ Co
C2 = ESR
R3
(19)
NOTE
For PSpice models and WEBENCH design tool, see the Tools & Software tab.
1. PSpice average model (stability – bode plot)
2. PSpice transient model (switching waveforms)
3. WEBENCH design tool www.ti.com/product/TPS50601-SP/toolssoftware
26
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8.4 Device Functional Modes
8.4.1 Fixed-Frequency PWM Control
The device uses fixed frequency, peak current mode control. The output voltage is compared through external
resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives the COMP pin.
An internal oscillator initiates the turn on of the high-side power switch. The error amplifier output is converted
into a current reference which compares to the high-side power switch current. When the power switch current
reaches the current reference generated by the COMP voltage level, the high-side power switch is turned off and
the low-side power switch is turned on.
8.4.2 Continuous Current Mode (CCM) Operation
As a synchronous buck converter, the device normally works in CCM under all load conditions.
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The TPS50601-SP device is a highly-integrated synchronous step-down DC-DC converter. The device is used to
convert a higher DC-DC input voltage to a lower DC output voltage with a maximum output current of 6 A.
The TPS50601-SP user's guide is available on the TI website, SLVU499. The guide highlights standard EVM test
results, schematic, and BOM for reference. (Basic design equations in following sections are provided for
reference only)
9.2 Typical Application
PVIN
VIN
TPS50601-SP
BOOT
VIN
Cin
Cboot
VOUT
Lo
EN
PH
Co
PWRGD
R1
VSENSE
SS/TR
RT/CLK
COMP
Css
R3
Rrt C2
C1
R2
GND
Exposed
Thermal
Pad
Figure 29. Typical Application Schematic
9.2.1 Design Requirements
This example details the design of a high frequency switching regulator design using ceramic output capacitors.
A few parameters must be known in order to start the design process. These parameters are typically determined
at the system level. For this example, we start with the following known parameters:
Table 2. Design Parameters
DESIGN PARAMETER
3.3 V
Output current
6A
Transient response 1-A load step
Input voltage
ΔVout = 5%
5 V nominal, 4.5 to 6.3 V
Output voltage ripple
Start input voltage (rising Vin)
28
EXAMPLE VALUE
Output voltage
33 mV p-p
4.425V
Stop input voltage (falling Vin)
4.234V
Switching frequency
480 kHz
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9.2.2 Detailed Design Procedure
9.2.2.1 Operating Frequency
The first step is to decide on a switching frequency for the regulator. There is a trade off between higher and
lower switching frequencies. Higher switching frequencies may produce smaller a solution size using lower
valued inductors and smaller output capacitors compared to a power supply that switches at a lower frequency.
However, the higher switching frequency causes extra switching losses, which hurt the converter’s efficiency and
thermal performance. In this design, a moderate switching frequency of 480 kHz is selected to achieve both a
small solution size and a high efficiency operation.
9.2.2.2 Output Inductor Selection
To calculate the value of the output inductor, use Equation 20. KIND is a coefficient that represents the amount
of inductor ripple current relative to the maximum output current. The inductor ripple current is filtered by the
output capacitor. Therefore, choosing high inductor ripple currents impact the selection of the output capacitor
since the output capacitor must have a ripple current rating equal to or greater than the inductor ripple current. In
general, the inductor ripple value is at the discretion of the designer; however, KIND is normally from 0.1 to 0.4
for the majority of applications.
L1 =
Vinm ax - Vout
Vout
×
Io × Kind
Vinm ax × f sw
(20)
For this design example, use KIND = 0.1 and the inductor value is calculated to be 2.78 µH. For this design, a
nearest standard value was chosen: 3.3 µH. For the output filter inductor, it is important that the RMS current
and saturation current ratings not be exceeded. The RMS and peak inductor current can be found from
Equation 22 and Equation 23.
Vinmax - Vout
Vout
×
Iripple =
L1
Vinmax × f sw
(21)
ILrms = Io2 +
1 æ Vo × (Vinmax - Vo ) ö
×ç
÷
12 çè Vinmax × L1× f sw ÷ø
2
(22)
Iripple
ILpeak = Iout +
2
(23)
For this design, the RMS inductor current is 6.02 A and the peak inductor current is 6.84 A. The chosen inductor
is a Coilcraft MSS1048 series 3.3 µH. It has a saturation current rating of 7.38 A and a RMS current rating of
7.22 A.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
9.2.2.3 Output Capacitor Selection
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the more stringent of these three criteria
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator can not. This situation would occur if there are desired hold-up
times for the regulator where the output capacitor must hold the output voltage above a certain level for a
specified amount of time after the input power is removed. The regulator is also temporarily not able to supply
sufficient output current if there is a large, fast increase in the current needs of the load such as a transition from
no load to full load. The regulator usually needs two or more clock cycles for the control loop to see the change
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in load current and output voltage and adjust the duty cycle to react to the change. The output capacitor must be
sized to supply the extra current to the load until the control loop responds to the load change. The output
capacitance must be large enough to supply the difference in current for 2 clock cycles while only allowing a
tolerable amount of droop in the output voltage. Equation 24 shows the minimum output capacitance necessary
to accomplish this.
2 × DIout
Co >
f sw × DVout
(24)
Where ΔIout is the change in output current, Fsw is the regulators switching frequency and ΔVout is the
allowable change in the output voltage. For this example, the transient load response is specified as a 5%
change in Vout for a load step of 1A. For this example, ΔIout = 1.0 A and ΔVout = 0.05 x 3.3 = 0.165 V. Using
these numbers gives a minimum capacitance of 25 μF. This value does not take the ESR of the output capacitor
into account in the output voltage change. For ceramic capacitors, the ESR is usually small enough to ignore in
this calculation.
Equation 25 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
Where fsw is the switching frequency, Vripple is the maximum allowable output voltage ripple, and Iripple is the
inductor ripple current. In this case, the maximum output voltage ripple is 33mV. Under this requirement,
Equation 25 yields 13.2 µF.
1
1
Co >
×
8 × f sw Voripple
Iripple
(25)
Equation 26 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 26 indicates the ESR should be less than 19.7 mΩ. In this case, the ceramic caps’ ESR is
much smaller than 19.7 mΩ.
Voripple
Resr <
Iripple
(26)
Additional capacitance de-ratings for aging, temperature and DC bias should be factored in which increases this
minimum value. For this example, a 47 μF 6.3V X5R ceramic capacitor with 3 mΩ of ESR is be used. Capacitors
generally have limits to the amount of ripple current they can handle without failing or producing excess heat. An
output capacitor that can support the inductor ripple current must be specified. Some capacitor data sheets
specify the RMS (Root Mean Square) value of the maximum ripple current. Equation 27 can be used to calculate
the RMS ripple current the output capacitor needs to support. For this application, Equation 27 yields 485mA.
Vout × (Vinmax - Vout )
Icorms =
12 × Vinmax × L1× f sw
(27)
9.2.2.4 Input Capacitor Selection
The TPS50601-SP requires a high quality ceramic, type X5R or X7R, input decoupling capacitor of at least 4.7
µF of effective capacitance on the PVIN input voltage pins and 4.7 µF on the Vin input voltage pin. In some
applications additional bulk capacitance may also be required for the PVIN input. The effective capacitance
includes any DC bias effects. The voltage rating of the input capacitor must be greater than the maximum input
voltage. The capacitor must also have a ripple current rating greater than the maximum input current ripple of the
TPS50601-SP. The input ripple current can be calculated using Equation 28.
Icirms = Iout ×
Vout (Vinmin - Vout )
×
Vinmin
Vinmin
(28)
The value of a ceramic capacitor varies significantly over temperature and the amount of DC bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the DC bias taken into account. The capacitance value of a capacitor
decreases as the DC bias across a capacitor increases. For this example design, a ceramic capacitor with at
least a 25-V voltage rating is required to support the maximum input voltage. For this example, one 10 μF and
30
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one 4.7-µF 25-V capacitors in parallel have been selected as the VIN and PVIN inputs are tied together so the
TPS50601-SP may operate from a single supply. The input capacitance value determines the input ripple voltage
of the regulator. The input voltage ripple can be calculated using Equation 29. Using the design example values,
Ioutmax = 6 A, Cin = 14.7 μF, FSW = 480 kHz, yields an input voltage ripple of 213 mV and a RMS input ripple
current of 2.95 A.
Ioutmax × 0.25
DVin =
Cin × f sw
(29)
9.2.2.5 Slow Start Capacitor Selection
The slow start capacitor determines the minimum amount of time it takes for the output voltage to reach its
nominal programmed value during power up. This is useful if a load requires a controlled voltage slew rate. This
is also used if the output capacitance is very large and would require large amounts of current to quickly charge
the capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the
TPS50601-SP reach the current limit or excessive current draw from the input power supply may cause the input
voltage rail to sag. Limiting the output voltage slew rate solves both of these problems. The soft start capacitor
value can be calculated using Equation 30. For the example circuit, the soft start time is not too critical since the
output capacitor value is 47 μF which does not require much current to charge to 3.3 V. The example circuit has
the soft start time set to an arbitrary value of 3.5 ms which requires a 10-nF capacitor. In TPS50601-SP, Iss is
2.5 µA typical, and Vref is 0.795 V.
Tss(ms) x Iss(μA)
C5(nF) =
Vref(V)
(30)
9.2.2.6 Bootstrap Capacitor Selection
A 0.1-µF ceramic capacitor must be connected between the BOOT to PH pin for proper operation. TI
recommends to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have a voltage
rating of 10 V or higher.
9.2.2.7 Undervoltage Lockout (UVLO) Set Point
The UVLO can be adjusted using the external voltage divider network of R6a and R7a. R6a is connected
between VIN and the EN pin of the TPS50601-SP and R7a is connected between EN and GND . The UVLO has
two thresholds, one for power up when the input voltage is rising and one for power down or brown outs when
the input voltage is falling. For the example design, the supply should turn on and start switching once the input
voltage increases above selected voltage (UVLO start or enable). After the regulator starts switching, it should
continue to do so until the input voltage falls below (UVLO stop or disable) voltage. Equation 4 and Equation 5
can be used to calculate the values for the upper and lower resistor values. For the stop voltages specified the
nearest standard resistor value for R6a is 10.0 kΩ and for R7a is 3.4 kΩ.
9.2.2.8 Output Voltage Feedback Resistor Selection
The resistor divider network R5 and R6 is used to set the output voltage. For the example design, 10 kΩ was
selected for R6. Using Equation 31, R5 is calculated as 31.25 kΩ. The nearest standard 1% resistor is 31.6 kΩ.
V re f
R5
9 R ± 9 UH I
u R6
(31)
9.2.2.8.1 Minimum Output Voltage
Due to the internal design of the TPS50601-SP, there is a minimum output voltage limit for any given input
voltage. The output voltage can never be lower than the internal voltage reference of 0.8 V. Above 0.8 V, the
output voltage may be limited by the minimum controllable on time. The minimum output voltage in this case is
given by Equation 32
spacer
VOUTmin = Ontimemin × ƒsmax (VINmax + IOUTmin (RDS2min - RDS1min)) - IOUTmin (RL + RDS2min)
where
•
•
VOUTmin = Minimum achievable output voltage
Ontimemin = Minimum controllable on-time (175 ns maximum)
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•
•
•
•
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ƒsmax = Maximum switching frequency including tolerance
VINmax = Maximum input voltage
IOUTmin = Minimum load current
RDS1min = Minimum high-side MOSFET on-resistance (36-32 mΩ typical)
RDS2min = Minimum low-side MOSFET on-resistance (19 mΩ typical)
RL = Series resistance of output inductor
(32)
9.2.2.9 Compensation Component Selection
There are several industry techniques used to compensate DC/DC regulators. The method presented here is
easy to calculate and yields high phase margins. For most conditions, the regulator has a phase margin between
60 and 90 degrees. The method presented here ignores the effects of the slope compensation that is internal to
the TPS50601-SP. Since the slope compensation is ignored, the actual cross over frequency is usually lower
than the cross over frequency used in the calculations. Use WEBENCH, Pspice model for simulation.
First, the modulator pole, fpmod, and the esr zero, fzmod must be calculated using Equation 33 and Equation 34.
For Cout, use a derated value of 22.4 µF. use Equation 35 and Equation 36 to estimate a starting point for the
closed loop crossover frequency fco. Then the required compensation components may be derived. For this
design example, fpmod is 12.9 kHz and fzmod is 2730 kHz. Equation 35 is the geometric mean of the modulator
pole and the esr zero and Equation 36 is the geometric mean of the modulator pole and one half the switching
frequency. Use a frequency near the lower of these two values as the intended crossover frequency fco. In this
case Equation 35 yields 175 kHz and Equation 36 yields 55.7 kHz. The lower value is 55.7 kHz. A slightly higher
frequency of 60.5 kHz is chosen as the intended crossover frequency.
Iout
f pmod =
2 × p × Vout × Cout
(33)
f zm od =
1
2 × p × RESR × Cout
f co =
f pmod × f zmod
f co =
f pmod ×
(34)
(35)
f sw
2
(36)
Now the compensation components can be calculated. First calculate the value for R2 which sets the gain of the
compensated network at the crossover frequency. Use Equation 37 to determine the value of R2.
2p × f c × Vout × Cout
R2 =
gmea × Vref × gmps
(37)
Next calculate the value of C3. Together with R2, C3 places a compensation zero at the modulator pole
frequency. Equation 38 to determine the value of C3.
Vout × Cout
C3 =
Iout × R2
(38)
Using Equation 37 and Equation 38 the standard values for R2 and C3 are 1.69 kΩ and 8200 pF.
An additional high frequency pole can be used if necessary by adding a capacitor in parallel with the series
combination of R2 and C3. The pole frequency is given by Equation 39. This pole is not used in this design.
1
fp =
2 × p × R2 × Cp
(39)
32
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9.2.3 Application Curve
A.
Per EVM - for additional details see the User's Guide, SLVU499
Figure 30. Typical Switching Waveform for 100-kHz Switching Operation
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10 Power Supply Recommendations
The TPS50601-SP is designed to operate from an input voltage supply range between 3.0V and 6.3V. This
supply voltage must be well regulated. Power supplies must be well bypassed for proper electrical performance.
This includes a minimum of one 4.7 µF (after de-rating) ceramic capacitor, type X5R or better from PVIN to GND,
and from VIN to GND. Additional local ceramic bypass capacitance may be required in systems with small input
ripple specifications, in addition to bulk capacitance if the TPS50601-SP device is located more than a few
inches away from its input power supply. In systems with an auxiliary power rail available, the power stage input,
PVIN, and the analog power input, VIN, may operate from separate input supplies. See Layout Example (layout
recommendation) for recommended bypass capacitor placement.
11 Layout
11.1 Layout Guidelines
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
34
Layout is a critical portion of good power supply design. See Layout Example for a PCB layout example.
The top layer contains the main power traces for VIN, VOUT, and VPHASE. Also on the top layer are
connections for the remaining pins of the TPS50601-SP and a large top side area filled with ground.
The top layer ground area should be connected to the internal ground layer(s) using vias at the input bypass
capacitor, the output filter capacitor and directly under the TPS50601-SP device to provide a thermal path
from the exposed thermal pad land to ground
The GND pin should be tied directly to the power pad under the IC and the power pad.
For operation at full rated load, the top side ground area together with the internal ground plane, must provide
adequate heat dissipating area.
There are several signals paths that conduct fast changing currents or voltages that can interact with stray
inductance or parasitic capacitance to generate noise or degrade the power supplies performance.
To help eliminate these problems, the PVIN pin should be bypassed to ground with a low ESR ceramic
bypass capacitor with X5R or X7R dielectric.
Care should be taken to minimize the loop area formed by the bypass capacitor connections, the PVIN pins,
and the ground connections.
The VIN pin must also be bypassed to ground using a low ESR ceramic capacitor with X5R or X7R dielectric.
Make sure to connect this capacitor to the quite analog ground trace rather than the power ground trace of
the PVIn bypass capacitor.
Since the PH connection is the switching node, the output inductor should be located close to the PH pins,
and the area of the PCB conductor minimized to prevent excessive capacitive coupling.
The output filter capacitor ground should use the same power ground trace as the PVIN input bypass
capacitor.
Try to minimize this conductor length while maintaining adequate width.
The small signal components should be grounded to the analog ground path as shown.
The RT pin is sensitive to noise so the RT resistor should be located as close as possible to the IC and
routed with minimal lengths of trace.
It may be possible to obtain acceptable performance with alternate PCB layouts, however this layout has
been shown to produce good results and is meant as a guideline.
Land pattern and stencil information is provided in the data sheet addendum.
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11.2 Layout Example
Bond pad connected to
ground for thermal path
Figure 31. PCB Layout Example
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12 Device and Documentation Support
12.1 Documentation Support
12.1.1 Related Documentation
For related documentation see the following:
TPS50601SP EVM, 6-A/12-A, SWIFT ™ Regulator Evaluation Module, SLVU499
12.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.3 Trademarks
SWIFT, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
12.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
13.1 Device Nomenclature
KGD
Known good die
RHA
Radiation hardness assurance for space systems
5962R10221 Same device as TPS50601-SP, shown with standard microcircuit drawing (SMD)
TPS50601-SP Same device as 5962R10221, shown with TI package drawing
36
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PACKAGE OPTION ADDENDUM
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13-Sep-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
(3)
Device Marking
Samples
(4/5)
(6)
5962-1022101VSC
ACTIVE
CFP
HKH
20
1
RoHS-Exempt
& Green
AU
N / A for Pkg Type
-55 to 125
5962-1022101VS
C
TPS50601MHKHV
5962R1022101V9A
ACTIVE
XCEPT
KGD
0
25
RoHS & Green
Call TI
N / A for Pkg Type
-55 to 125
5962R1022101VSC
ACTIVE
CFP
HKH
20
1
RoHS-Exempt
& Green
AU
N / A for Pkg Type
-55 to 125
5962R1022101VS
C
TPS50601-RHA
TPS50601HKH/EM
ACTIVE
CFP
HKH
20
1
RoHS-Exempt
& Green
AU
N / A for Pkg Type
25 to 25
TPS50601HKH/EM
EVAL ONLY
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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