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TPS51117
SLVS631C – DECEMBER 2005 – REVISED MAY 2015
TPS51117 Single Synchronous Step-Down Controller
1 Features
3 Description
•
The TPS51117 device is a cost-effective,
synchronous buck controller for POL voltage
regulation in notebook PC applications. The controller
is dedicated for the operation of the Adaptive OnTime D-CAP mode. This mode provides ease-of-use,
low external component count, and fast transient
response. Auto-skip mode for high efficiency down to
the milliampere load range, or PWM-only mode for
low-noise operation is selectable.
1
•
•
•
•
•
•
•
•
•
•
•
•
•
High Efficiency, Low Power Consumption, 4.5-μA
Typical Shutdown Current
Fixed Frequency Emulated On-Time Control,
Adjustable from 100 kHz to 550 kHz
D-CAP™ Mode With 100-ns Load Step Response
< 1% Initial Reference Accuracy
Output Voltage Range: 0.75 V to 5.5 V
Wide Input Voltage Range: 1.8 V to 28 V
Selectable Auto-Skip/PWM-Only Operation
Temperature Compensated (4500 ppm/°C)
Low-Side RDS(on) Overcurrent Sensing
Negative Overcurrent Limit
Integrated Boost Diode
Integrated OVP/UVP and Thermal Shutdown
Powergood Signal
Internal 1.2-ms Voltage Soft-Start
Integrated Output Discharge (Soft-Stop)
The current-sensing scheme for positive overcurrent
and negative overcurrent protection is loss-less lowside RDS(on) sensing plus temperature compensation.
The device receives a 5-V (4.5 V to
5.5 V) supply from another regulator such as the
TPS51120 or TPS51020. The conversion input can
be either VBAT or a 5-V rail, ranging from 1.8 V to
28 V, and the output voltage range is from 0.75 V to
5.5 V.
The TPS51117 is available in a 14-pin VQFN or a 14pin TSSOP package and is specified from –40°C to
85°C.
2 Applications
•
•
•
Device Information(1)
Notebook Computers
I/O Supplies
System Power Supplies
PART NUMBER
PACKAGE
TPS51117
BODY SIZE (NOM)
TSSOP (14)
4.40 mm × 5.00 mm
VQFN (14)
3.50 mm × 3.50 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Typical Application Circuit
+5V
+
+
TPS51117RGY
EN_PSV
R3
R5
TON
3
VOUT
5
C3
PGOOD
14
VBST
2
4
R6
C4
1
EN_PSV
V5FILT
13
LL
12
TRIP
V5DRV
VFB
Q1
DRVH
C2
L1
+
R4
PGOOD
DRVL
GND
PGND
7
8
VOUT
0.75V~5.5V
R1
11
GND
C1
10
Q2
6
VIN
1.8V~28V
R2
9
-
PGND
GND
GND
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS51117
SLVS631C – DECEMBER 2005 – REVISED MAY 2015
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
4
4
5
7
Detailed Description ............................................ 10
7.1
7.2
7.3
7.4
8
Absolute Maximum Ratings .....................................
Recommended Operating Conditions.......................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
10
10
11
14
Application and Implementation ........................ 16
8.1 Application Information............................................ 16
8.2 Typical Application .................................................. 16
8.3 System Examples ................................................... 20
9 Power Supply Recommendations...................... 21
10 Layout................................................................... 21
10.1 Layout Guidelines ................................................. 21
10.2 Layout Example .................................................... 22
10.3 Thermal Considerations ........................................ 22
11 Device and Documentation Support ................. 23
11.1
11.2
11.3
11.4
11.5
Device Support......................................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
23
23
23
23
23
12 Mechanical, Packaging, and Orderable
Information ........................................................... 23
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision B (September 2009) to Revision C
•
Page
Added Feature Description section, Device Functional Modes, Application and Implementation section, Power
Supply Recommendations section, Layout section, Device and Documentation Support section, and Mechanical,
Packaging, and Orderable Information section. .................................................................................................................... 1
Changes from Revision A (June 2009) to Revision B
Page
•
Added Start-Up Sequence section ....................................................................................................................................... 13
•
Added Start-Up Timing Sequence diagram.......................................................................................................................... 13
2
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SLVS631C – DECEMBER 2005 – REVISED MAY 2015
5 Pin Configuration and Functions
14
DRVH
PW Package
14-Pin TSSOP
Top View
EN_PSV
VBST
RGY Package
14-Pin VQFN
Bottom View
EN_PSV
1
14
VBST
TON
2
13
DRVH
VOUT
3
12
LL
V5FILT
4
11
TRIP
VFB
5
10
V5DRV
PGOOD
6
9
DRVL
GND
7
8
PGND
1
13
2
TON
LL
12
3
VOUT
TRIP
11
4
V5FILT
V5DRV
10
5
VFB
DRVL
9
6
PGOOD
8
7
PGND
GND
Pin Functions
PIN
I/O
DESCRIPTION
NAME
NO.
DRVH
13
O
High-side NFET gate driver output. Source 5 Ω, sink 1.5-Ω LL-node referenced driver. Drive voltage
corresponds to VBST to LL voltage.
DRVL
9
O
Rectifying (low-side) NFET gate driver output. Source 5 Ω, sink 1.5-Ω PGND referenced driver. Drive voltage
is V5DRV voltage.
EN_PSV
1
I
Enable / power save pin. Connect to ground to disable SMPS. Connect to 3.3 V or 5 V to turn on SMPS and
activate skip mode. Float to turn on SMPS but disable skip mode (forced continuous conduction mode).
GND
7
I
Signal ground pin.
LL
12
I/O
High-side NFET gate driver return. Also serves as anode of overcurrent comparator.
PGND
8
I/O
Ground return for rectifying NFET gate driver. Also cathode of overcurrent protection and source node of the
output discharge switch.
PGOOD
6
O
Powergood window comparator, open-drain, output. Pull up to 5-V rail with a pullup resistor. Current capability
is 7.5 mA.
TON
2
I
On-time / frequency adjustment pin. Connect to LL with 100-kΩ to 600-kΩ resistor.
TRIP
11
I
Overcurrent trip point set input. Connect resistor from this pin to signal ground to set threshold for both
overcurrent and negative overcurrent limit.
VBST
14
I
Supply input for high-side NFET gate driver (boost terminal). Connect capacitor from this pin to LL-node. An
internal PN diode is connected between V5DRV to this pin. Designer can add external Schottky diode if
forward drop is critical to drive the power NFET.
VFB
5
I
SMPS voltage feedback input. Connect the resistor divider here for adjustable output.
VOUT
3
I
Connect to SMPS output. This terminal serves two functions: output voltage monitor for on-time adjustment,
and input for the output discharge switch.
V5DRV
10
I
5-V Power supply input for FET gate drivers. Internally connected to VBST by a PN diode. Connect 1 μF or
more between this pin and PGND to support instantaneous current for gate drivers.
V5FILT
4
I
5-V Power supply input for all the control circuitry except gate drivers. Supply 5-V ramp rate should be 17
mV/μs or less and Tj < 85°C to secure safe start-up of the internal reference circuit. Apply RC filter consists of
300 Ω + 1 μF or 100 Ω + 4.7 μF at the pin input.
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6 Specifications
6.1 Absolute Maximum Ratings (1)
Input voltage
MIN
MAX
VBST
–0.3
36
VBST (with respect to LL)
–0.3
6
EN_PSV, TRIP, V5DRV, V5FILT
–0.3
6
VOUT
–0.3
6
TON
–0.3
6
–1
36
DRVH
DRVH (with respect to LL)
Output voltage
–0.3
6
–1
30
PGOOD, DRVL
–0.3
6
PGND
–0.3
0.3
LL
UNIT
V
V
TA
Operating free-air temperature
–40
85
°C
TJ
Junction temperature
–40
125
°C
260
°C
150
°C
Lead temperature 1.6 mm (1/16 inch) from case for 10 seconds
Tstg Storage temperature
(1)
–55
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
Supply input voltage
Input voltage
Output voltage
MAX
4.5
5.5
VBST
4.5
34
VBST (with respect to LL)
4.5
5.5
EN_PSV, TRIP, V5DRV, V5FILT
–0.1
5.5
VOUT
–0.1
5.5
TON
–0.1
5.5
DRVH
–0.8
34
DRVH (with respect to LL)
–0.1
5.5
LL
–0.8
28
PGOOD, DRVL
–0.1
5.5
PGND
–0.1
0.1
–40
85
Operating free-air temperature, TA
4
MIN
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UNIT
V
V
V
°C
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6.3 Electrical Characteristics
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY CURRENT
IV5FILTPWM
Supply current
V5FILT + V5DRV current, PWM, EN_PSV =
float, VFB = 0.77 V, LL = –0.1 V
400
750
μA
IV5FILTSKIP
Supply current
V5FILT + V5DRV current, auto-skip, EN_PSV =
5 V, VFB = 0.77 V, LL = 0.5 V
250
470
μA
IV5DRVSDN
V5DRV shutdown current
V5DRV current, EN_PSV = 0 V
0
1
μA
IV5FILTSDN
V5FILT shutdown current
V5FILT current, EN_PSV = 0 V
4.5
7.5
μA
VOUT AND VFB VOLTAGES
VOUT
Output voltage
VVFB
VFB regulation voltage
VVFB_TOL
VFB regulation voltage
tolerance
Adjustable output range
0.75
5.5
750
V
mV
TA = 25°C, bandgap initial accuracy
–0.9%
0.9%
TA = 0°C to 85°C
–1.3%
1.3%
TA = –40°C to 85°C
–1.6%
1.6%
IVFB
VFB input current
VFB = 0.75 V, absolute value
0.02
0.1
μA
RDischg
VOUT discharge resistance
EN_PSV = 0 V, VOUT = 0.5 V
20
32
Ω
ON-TIME TIMER AND INTERNAL SOFT-START
TONN
Nominal on-time
VLL = 12 V, VOUT = 2.5 V, RTON = 250 kΩ
TONF
Fast on-time
VLL = 12 V, VOUT = 2.5 V, RTON = 100 kΩ
TONS
Slow on-time
VLL = 12 V, VOUT = 2.5 V, RTON = 400 kΩ
TON(MIN)
Minimum on-time
VOUT = 0.75 V, RTON = 100 kΩ to 28 V (1)
TOFF(MIN)
Minimum off-time
VFB = 0.7 V, LL = –0.1 V,
TRIP = open
TSS
Internal soft-start time
Time from EN_PSV > 3 V to VFB regulation
value = 0.735 V
750
264
330
ns
396
1169
80
110
ns
140
440
0.82
1.2
ns
ns
ns
1.5
ms
OUTPUT DRIVERS
RDRVH
DRVH resistance
RDRVL
DRVL resistance
TD
Dead time
Source, VVBST-DRVH = 0.5 V
Sink, VDRVH-LL = 0.5 V
Source, VV5DRV-DRVL = 0.5 V
Sink, VDRVL-PGND = 0.5 V
5
7
1.5
2.5
5
7
1.5
2.5
Ω
Ω
DRVH-low (DRVH = 1 V) to DRVL-high
(DRVL = 4 V), LL = –0.05 V
10
20
50
ns
DRVL-low (DRVL = 1 V) to DRVH-high
(DRVH = 4 V), LL = –0.05 V
30
40
60
ns
0.7
0.8
0.9
V
0.1
1
μA
INTERNAL BST DIODE
VFBST
Forward voltage
VV5DRV-VBST, IF = 10 mA, TA = 25°C
IVBSTLK
VBST leakage current
VBST = 34 V, LL = 28 V
UVLO/LOGIC THRESHOLD
VUVLO
VEN_PSV
IEN_PSV
(1)
(2)
V5FILT UVLO Threshold
EN_PSV logic input voltage
EN_PSV source current
Wake up
3.7
3.9
4.1
V
Hysteresis
200
300
400
mV
EN_PSV low
0.7
1.0
1.3
V
Hysteresis
150
200
250
mV
EN_PSV float (set PWM_only mode)
1.7
1.95
2.25
V
EN_PSV high (set Auto_skip mode)
2.4
2.65
2.9
V
Hysteresis
100
175
250
mV
EN_PSV = GND, absolute value (2)
1
μA
Design constraint, ensure actual on-time is larger than the maximum value (that is, design RTON such that the minimum tolerance is 100
kΩ).
Ensured by design. Not production tested.
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Electrical Characteristics (continued)
over operating free-air temperature range (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
POWERGOOD COMPARATOR
VTHPG
PG threshold
PG in from lower (PGOOD goes high)
92.5%
95%
97.5%
PG low hysteresis (PGOOD goes low)
–4%
–5.5%
–7%
PG in from higher (PGOOD goes high)
102%
105%
107%
PG high hysteresis (PGOOD goes low)
4%
5.5%
7%
IPGMAX
PG sink current
PGOOD = 0.5 V
2.5
7.5
TPGDEL
PG delay
Delay for PGOOD in
45
63
85
mA
10
11
μs
CURRENT SENSE
ITRIP
TRIP source current
VTRIP < 0.3 V, TA = 25°C
TCITRIP
ITRIP temperature coefffecient
On the basis of 25°C
9
VRtrip
Current limit threshold range
setting range
VTRIP-GND voltage (2), all temperatures
VOCLoff
Overcurrent limit comparator
offset
(VTRIP-GND-VPGND-LL) voltage VTRIP-GND = 60 mV
–10
VUCLoff
Negative overcurrent limit
comparator offset
(VTRIP-GND-VLL-PGND) voltage VTRIP-GND = 60 mV,
EN_PSV = float
VZCoff
Zero crossing comparator offset VPGND-LL voltage, EN_PSV = 3.3 V
4500
30
μA
ppm/°C
200
mV
0
10
mV
–9.5
0.5
10.5
mV
–9.5
0.5
10.5
mV
111%
115%
119%
UNDERVOLTAGE AND OVERVOLTAGE PROTECTION
VOVP
VFB OVP trip threshold
OVP detect
TOVPDEL
VFB OVP propagation delay
See
VUVP
VFB UVP trip threshold
TUVPDEL
VFB UVP delay
TUVPEN
UVP enable delay
(2)
UVP detect
65%
Hysteresis
After 1.7 × TSS, UVP protection engaged
μs
1.5
70%
75%
10%
22
32
42
μs
1.4
2
2.6
ms
THERMAL SHUTDOWN
TSDN
6
Thermal shutdown threshold
Shutdown temperature (2)
Hysteresis (2)
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160
12
°C
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800
8
700
7
IV5FILT_SHDN - Shutdown Current - mA
IV5FILTPWM - Supply Current - mA
6.4 Typical Characteristics
600
500
400
300
200
100
0
-50
0
50
100
TJ - Junction Temperature - ºC
4
3
2
1
VOVP, VUVP - OVP/UVP Threshold - %
12
10
8
6
4
-50
0
50
110
100
90
80
UVP
70
60
50
-50
150
100
OVP
120
TJ - Junction Temperature - º C
Figure 3. Trip Current vs Junction Temperature
0
50
100
TJ - Junction Temperature - ºC
150
Figure 4. OVP/UVP Threshold vs Junction Temperature
800
500
VI = 15 V,
PWM Mode
600
500
400
300
VO = 2.5 V
200
IO = 2 A,
PWM Mode
450
fsw - Switching Frequency - kHz
700
fsw - Switching Frequency - kHz
150
130
14
100
VO = 1.05 V
400
350
300
VO = 2.5 V
250
200
150
100
50
VO = 1.05 V
0
100
0
50
100
TJ - Junction Temperature - ºC
Figure 2. V5FILT Shutdown Current vs Junction
Temperature
16
ITRIP - TRIP Source Current - mA
5
0
-50
150
Figure 1. PWM Supply Current vs Junction Temperature
6
0
200
300
400
500
600
5
700
RTON - TON Resistance - kW
Figure 5. Measured Switching Frequency vs Ton Resistance
9
13
17
VI - Input Voltage - V
21
25
Figure 6. Switching Frequency vs Input Voltage
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450
450
400
400
350
fsw - Switching Frequency - kHz
fsw - Switching Frequency - kHz
Typical Characteristics (continued)
PWM Only
300
250
200
150
100
50
PWM Only
300
250
200
150
100
Auto Skip
50
Auto Skip
0
0.001
350
0.010
0.100
1.000
0
0.001
10.000
IO - Output Current - A
Figure 7. Switching Frequency vs Output Current (1.05 V)
2.54
VO - Output Voltage - V
VO - Output Voltage - V
2.52
1.06
PWM Only
1.05
Auto Skip
1.04
1.03
PWM Only
2.50
Auto Skip
2.48
2.46
0
2
4
6
IO - Output Current - A
8
10
Figure 9. 1.05-V Output Voltage vs Output Current
0
1.06
2.52
VO - Output Voltage - V
2.54
IO = 10 A
1.05
IO = 0 A
2
4
6
IO - Output Current - A
8
10
Figure 10. 2.5-V Output Voltage vs Output Current
1.07
IO = 10 A
2.50
IO = 0 A
2.48
1.04
Auto Skip
Auto Skip
1.03
2.46
5
9
13
17
VI - Input Voltage - V
21
5
25
Figure 11. 1.05-V Output Voltage vs Input Voltage
8
10
Figure 8. Switching Frequency vs Output Current (2.5 V)
1.07
VO - Output Voltage - V
0.010
0.1
1
IO - Output Current - A
9
13
17
VI - Input Voltage - V
21
25
Figure 12. 2.5-V Output Voltage vs Input Voltage
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Typical Characteristics (continued)
100
100
Auto Skip
90
90
VI = 8 V
80
70
h - Efficiency - %
h - Efficiency - %
80
VI = 8 V
VI = 12 V
60
50
VI = 20 V
VI = 8 V
VI = 12 V
40
70
60
VI = 12 V
VI = 8 V
VI = 20 V
VI = 12 V
50
40
30
VI = 20 V
30
VI = 20 V
20
0
0.001
20
PWM Only
fsw = 350 kHz
10
0.01
0.1
1
IO - Output Current - A
PWM Only
fsw = 300 kHz
10
0
0.001
10
0.01
0.1
1
10
IO - Output Current - A
Figure 13. 1.05-V Efficiency vs Output Current
Figure 14. 2.5-V Efficiency vs Output Current
VO (50 mV/div)
VO (50 mV/div)
IIND (5 A/div)
IIND (5 A/div)
IO (5 A/div)
IO (5 A/div)
t - Time - 10 ms/div
t - Time - 10 ms/div
Figure 15. 1.05-V Load Transient Response
Figure 16. 2.5-V Load Transient Response
VO (20 mV/div)
VO (20 mV/div)
LL (10 V/div)
LL (10 V/div)
DRVL (5 V/div)
DRVL (5 V/div)
EN_PSV (5 V/div)
EN_PSV (5 V/div)
Figure 17. Mode Transition
Auto-Skip to PWM
Figure 18. Mode Transition
PWM to Auto-Skip
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7 Detailed Description
7.1 Overview
The TPS51117 is a synchronous buck controller for POL voltage regulation in notebook PC applications. The
controller is dedicated for the operation of the Adaptive On-Time D-CAP mode. This mode provides ease-of-use,
low external component count, and fast transient response. Auto-skip mode for high efficiency down to the
milliampere load range, or PWM-only mode for low-noise operation is selectable.
7.2 Functional Block Diagram
2.9
3.9 /3.6
48
10
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7.3 Feature Description
7.3.1 PWM Frequency and Adaptive On-Time Control
The TPS51117 employs an adaptive on-time control scheme and does not have a dedicated oscillator onboard.
However, the device emulates a constant frequency by feed-forwarding the input and output voltages into the ontime one-shot timer. The ON time is controlled inverse proportional to the input voltage, and proportional to the
output voltage, so that the duty ratio is kept as VOUT/VIN technically with the same cycle time. Equation 1 shows a
simplified calculation of the ON time.
æ
ö
+ 100 mV ÷
2/3 V
TON = 19 ´ 10-12 ´ RTON çç ( ) OUT
+ 50 ns
÷
VIN
è
ø
(1)
Here, RTON is the external resistor connected from TON pin to the LL node. In the equation, 19 pF represents the
internal timing capacitor with some typical parasitic capacitance at the TON pin. Also, 50 ns is the turnoff delay
time contributed by the internal circuit and that of the high-side MOSFET. Although this equation provides a good
approximation with which to begin, the accuracy depends on each design and selection of the high-side
MOSFET. Figure 19 shows the relationship of RTON to the switching frequency.
700
VIN = 15 V,
VOUT = 2.5 V,
PWM
600
f - Frequency - kHz
500
400
300
200
100
0
100
200
300
400
RTON - kW
500
600
Figure 19. Switching Frequency vs RTON
The TPS51117 does not have a pin connected to VIN, but the input voltage information comes from the switch
node (LL node) during the ON-state. An advantage of LL monitoring is that the loss in the high-side NFET is now
a part of the ON-time calculation, thereby making the frequency more stable with load.
Another consideration about frequency is jitter. Jitter may be caused by many reasons, but the constant on-time
D-CAP mode scheme has some amount of inherent jitter. Because the output voltage ripple height is in the
range of a couple of tens of millivolts. A millivolt order of noise on the feedback signal can affect the frequency by
a few to ten percent. This is normal operation and has little harm to the power supply performance.
7.3.2 Low-Side Driver
The low-side driver is designed to drive high-current, low RDS(on) N-channel MOSFETs. The drive capability is
represented by its internal resistance, which is 5 Ω for V5DRV to DRVL and 1.5 Ω for DRVL to PGND. A dead
time to prevent shoot-through is internally generated between high-side MOSFET off to low-side MOSFET on,
and low-side MOSFET off to high-side MOSFET on. A 5-V bias voltage is delivered from V5DRV supply. The
average drive current is calculated by the FET gate charge at Vgs = 5 V times the switching frequency. The
instantaneous drive current is supplied by an input capacitor connected between V5DRV and GND.
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Feature Description (continued)
7.3.3 High-Side Driver
The high-side driver is designed to drive high-current, low RDS(on) N-channel MOSFET(s). When configured as a
floating driver, 5-V bias voltage is delivered from V5DRV supply. An internal PN diode is connected between
V5DRV to VBST. The designer can add an external Schottky diode if forward drop is critical to drive the high-side
NFET or to achieve the last 1% efficiency improvement. The average drive current is also estimated by the gate
charge at Vgs = 5 V times the switching frequency. The instantaneous drive current is supplied by the flying
capacitor between the VBST pin and LL pin. The drive capability is represented by its internal resistance, which
is 5 Ω for VBST to DRVH and 1.5 Ω for DRVH to LL.
7.3.4 Soft-Start
The TPS51117 has an internal, 1.2-ms, voltage servo soft-start with overcurrent limit. When the EN_PSV pin
becomes high, an internal DAC begins ramping up the reference voltage to the error amplifier. Smooth control of
the output voltage is maintained during start-up.
7.3.5 Powergood
The TPS51117 has powergood output. PGOOD is an open-drain 7.5-mA pulldown output. This pin should be
typically connected to a 5-V power supply node through a 100-kΩ resistor. The powergood function is activated
after the soft start has finished. If the output voltage becomes within ±5% of the target value, internal
comparators detect the power good state and the powergood signal becomes high after a 64-μs internal delay. If
the output voltage goes outside ±10% of the target value, the powergood signal becomes low immediately.
7.3.6 Output Discharge Control (Soft-Stop)
The TPS51117 discharges output when EN_PSV is low or the converter is in a fault condition (UVP, OVP,
UVLO, or thermal shutdown). The TPS51117 discharges output using an internal 20-Ω MOSFET, which is
connected to VOUT and PGND. The discharge time-constant is a function of the output capacitance and
resistance of the discharge transistor.
7.3.7 Overcurrent Limit
The TPS51117 has cycle-by-cycle overcurrent limiting control. Inductor current is monitored during the OFF-state
and the controller keeps the OFF-state when inductor current is larger than the overcurrent trip level. To provide
both good accuracy and a cost-effective solution, the TPS51117 supports temperature compensated MOSFET
RDS(on) sensing. The TRIP pin should be connected to GND through the trip voltage setting resistor, RTRIP. The
TRIP terminal sources 10-μA ITRIP current, and the trip level is set to the OCL trip voltage, VTRIP as in the
following equation.
VTRIP (mV) = RTRIP (kW ) ´ 10 (mA )
(2)
Inductor current is monitored by the voltage between the PGND pin and the LL pin so the LL pin should be
connected to the drain terminal of the low-side MOSFET. ITRIP has 4500 ppm/°C temperature coefficient to
compensate the temperature dependency of the RDS(on). PGND is used as the positive current sensing node so
PGND should be connected to the source terminal of the bottom MOSFET.
As the comparison is done during the OFF-state, VTRIP sets the valley level of the inductor current. Thus, the load
current at overcurrent threshold, Iocp, can be calculated as follows;
(V - VOUT )´ VOUT
V
1
´ IN
Iocp = VTRIP / RDS(on) + IRIPPLE / 2 = TRIP +
RDS(on) 2 ´ L ´ ƒ
VIN
(3)
In an overcurrent condition, the current to the load exceeds the current to the output capacitor, thus the output
voltage tends to fall. Eventually, the output voltage crosses the undervoltage protection threshold and shutdown.
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Feature Description (continued)
7.3.8 Negative Overcurrent Limit (PWM-Only Mode)
The TPS51117 also supports cycle-by-cycle negative overcurrent limiting in PWM-only mode. The overcurrent
limit is set to be negative but is the same absolute value as the positive overcurrent limit. If output voltage
continues to rise, the bottom MOSFET stays on, thus inductor current is reduced and reverses direction after it
reaches zero. When there is too much negative current in the inductor, the bottom MOSFET is turned off and the
current flows to VIN through the body diode of the top MOSFET. Because this protection reduces current to
discharge the output capacitor, output voltage tends to rise, eventually hitting the overvoltage protection
threshold and shutdown. To prevent false OVP from triggering, the bottom MOSFET is turned on again 400 ns
after it is turned off. If the device hits the negative overcurrent threshold again before output voltage is
discharged to the target level, the bottom MOSFET is turned off and the process repeats, which is called NOCL
Buzz. The device ensures maximum allowable discharge capability when output voltage continues to rise. On the
other hand, if the output voltage is discharged to the target level before the NOCL threshold is reached, the
bottom MOSFET is turned off, the top MOSFET is then turned on, and the device resumes normal operation.
7.3.9 Overvoltage Protection
The TPS51117 monitors a resistor divided feedback voltage to detect overvoltage and undervoltage condition.
When the feedback voltage becomes higher than 115% of the target value, the top MOSFET is turned off and
the bottom MOSFET is turned on immediately. The output is also discharged by the internal 20-Ω transistor.
Also, the TPS51117 monitors VOUT terminal voltage directly and if it becomes greater than 5.75 V, it turns off
the top MOSFET driver.
7.3.10 Undervoltage Protection
When the feedback voltage becomes lower than 70% of the target value, the UVP comparator output goes high
and an internal UVP delay counter begins counting. After 32 μs, the TPS51117 latches off the high-side and lowside MOSFETs and discharges the output with the internal 20-Ω transistor. This function is enabled after 2 ms
from when EN_PSV is brought high, that is, UVP is disabled during start-up.
7.3.11 Start-Up Sequence
Referring to Figure 20 which shows the timing sequence, to ensure the proper start-up of the TPS51117, always
ensure that VEN_PSV is less or equal to that of VV5FILT prior to VV5FILT reaching VUVLO.
5V UVLO
V5DRV
V5FILT
EN_PSV
VOUT
PGOOD
t – Time
UDG-09142
Figure 20. Start-Up Timing Sequence
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Feature Description (continued)
7.3.12 UVLO Protection
The TPS51117 has V5FILT undervoltage lockout protection (UVLO). When the V5FILT voltage is lower than the
UVLO threshold voltage, the TPS51117 is shut off. This is a nonlatched protection.
7.3.13 Thermal Shutdown
The TPS51117 monitors the temperature of itself. If the temperature exceeds the threshold value (typically
160°C), the TPS51117 shuts itself off. Both top and bottom gate drivers are tied low with output discharged
through the VOUT terminal. This is also a nonlatched protection. The device recovers once the temperature has
decreased approximately 12°C.
7.4 Device Functional Modes
7.4.1 PWM Operation
The main control loop of the TPS51117 is designed as an adaptive on-time pulse width modulation (PWM)
controller. It supports proprietary D-CAP Mode that uses an internal compensation circuit and is suitable for
minimal external component count configuration when an appropriate amount of ESR at the output capacitor(s) is
allowed. Basic operation of D-CAP Mode can be described as follows.
At the beginning of each cycle, the synchronous high-side MOSFET is turned on, or becomes ON-state. This
MOSFET is turned off, or becomes OFF-state, after the internal one-shot timer expires. This one-shot is
determined by VIN and VOUT to keep the frequency fairly constant over the input voltage range at steady-state,
hence it is called adaptive on-time control or fixed frequency emulated on-time control (see PWM Frequency and
Adaptive On-Time Control). The MOSFET is turned on again when both feedback information, monitored at VFB
voltage, indicates insufficient output voltage and inductor current information indicates below the overcurrent
limit. Repeating the operation in this manner, the controller regulates the output voltage. The synchronous lowside or rectifying MOSFET is turned on each OFF-state to keep the conduction loss to a minimum.
The TPS51117 supports selectable PWM-only and auto-skip operation modes. If EN_PSV is grounded, the
switching regulator is disabled. If the EN_PSV pin is connected to 3.3 V or 5 V, the regulator is enabled with
auto-skip mode selected. The rectifying MOSFET is turned off when inductor current information detects zero
level. This enables a seamless transition to reduced frequency operation during a light-load condition so that high
efficiency is maintained over a broad range of load currents. If the EN_PSV pin is floated, it is internally pulled up
to 1.95 V, and the regulator is enabled with PWM-only mode selected. The rectifying MOSFET is not turned off
when inductor current reaches zero. The converter runs forced continuous conduction mode for the entire load
range. System designers may want to use this mode to avoid a certain frequency during a light-load condition but
with the cost of low efficiency. However, be aware the output has the capability to both source and sink current in
this mode. If the output terminal is connected to a voltage source higher than the target of the regulator, the
converter sinks current from the output and boosts the charge into the input capacitor. This may cause
unexpected high voltage at VIN and may damage the power FETs.
DC output voltage can be set by the external resistor divider as follows (refer to Figure 21, Figure 24, and
Figure 25).
æ
R ö
VOUT = ç 1 + 1 ÷ ´ 0.75 V
è R2 ø
(4)
7.4.2 Light-Load Condition With Auto-Skip Function
If auto-skip mode is selected, the TPS51117 automatically reduces the switching frequency during a light-load
condition to maintain high efficiency. This reduction of frequency is achieved smoothly and without an increase of
Vout ripple or load regulation. Detailed operation is described as follows. As the output current decreases from a
heavy load condition, the inductor current is also reduced and eventually comes to the point that its valley
touches zero current, which is the boundary between continuous conduction and discontinuous conduction
modes. The rectifying MOSFET is turned off when this zero inductor current is detected. Because the output
voltage is still higher than the reference at this moment, both high-side and low-side MOSFETs are turned off
and wait for the next cycle. As the load current decreases further, the converter runs in discontinuous conduction
14
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Device Functional Modes (continued)
mode, taking longer time to discharge the output capacitor below the reference voltage. The ON time is kept the
same as during the heavy load condition. In reverse, when the output current increases from a light load to a
heavy load, the switching frequency increases to the preset value as the inductor current reaches to the
continuous conduction. The transition load point to light-load operation, IOUT(LL) (that is, the threshold between
continuous and discontinuous conduction mode), can be calculated as follows:
(V - VOUT )´ VOUT
1
IOUT(LL) =
´ IN
2 ´ L ´ ƒSW
VIN
where
•
f
sw
is the PWM switching frequency
(5)
Switching frequency versus output current in the light-load condition is a function of L, f sw, VIN and VOUT, but it
decreases almost proportional to the output current from the IOUT(LL) given in Equation 5. For example, it is about
60 kHz at IOUT(LL)/5 if the PWM switching frequency is 300 kHz.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS51117 is a cost-effective, synchronous buck controller for POL voltage regulation in notebook PC
applications. The controller is dedicated for Adaptive On-Time D-CAP Mode operation. Use the following design
procedure to select component values for each device.
8.2 Typical Application
A buck converter system using D-CAP Mode can be simplified as shown in Figure 21.
VIN
R1
DRVH
PWM
-
VFB
Control
Logic
and
Driver
+
R2
+
Lx
Ic
IL
DRVL
0.75V
ESR
Vc
Voltage Divider
Io
RL
Switching Modulator
Co
Output Capacitor
Figure 21. Simplified Diagram of the Modulator
8.2.1 Design Requirements
For this design example, use Table 1 as the input parameters.
Table 1. Design Parameters
16
DESIGN PARAMETER
EXAMPLE VALUE
Input Voltage Range
1.8 V to 28 V
Output Voltage
1.05 V
Output Current Rating
10 A
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8.2.2 Detailed Design Procedure
8.2.2.1 D-CAP Mode Operation
The VFB voltage is compared with the internal reference voltage after the divider resistors. The PWM comparator
determines the timing to turn on the top MOSFET. The gain and speed of the comparator is high enough to keep
the voltage at the beginning of each on cycle (or the end of off cycle) substantially constant. The DC output
voltage may have line regulation due to ripple amplitude that slightly increases as the input voltage increases.
For loop stability, the 0 dB frequency, f 0, defined in Equation 6 must be lower than 1/4 of the switching
frequency.
ƒ
1
ƒO =
£ SW
2P ´ ESR ´ Co
4
(6)
As f 0 is determined solely by the output capacitor characteristics, loop stability of D-CAP Mode is determined by
capacitor chemistry. For example, specialty polymer capacitors (SP-CAP) have Co in the order of several 100 μF
and ESR in range of 10 mΩ. These values make f 0 in the order of 100 kHz or less and the loop is stable.
However, ceramic capacitors have f0 at more than 700 kHz, which is not suitable for this operational mode.
Although D-CAP Mode provides many advantages such as ease-of-use, minimum external component
configuration, and extremely short response time, due to not employing an error amplifier in the loop, a sufficient
feedback signal must be provided by an external circuit to reduce the jitter level. The required signal level is
approximately 15 mV at the comparing point. This generates Vripple = (VOUT/0.75) × 15 mV at the output node.
The output capacitor ESR should meet this requirement.
The external component selection is simple in D-CAP Mode:
1. Determine the value of R1 and R2
The recommended R2 value is 10 kΩ to 100 kΩ. Calculate R1 by Equation 7.
(V
-0.75
OUT
R1 =
0.75
2. Choose RTON
)´ R2
(7)
Switching frequency is usually determined by the overall view of the DC-DC converter design of: size,
efficiency or cost, and mostly dictated by external component constraints such as the size of inductor and/or
output capacitor. When an extremely low or high duty factor is expected, the minimum on-time or off-time
also must be considered to satisfy the required duty factor. Once the switching frequency is decided, RTON
can be determined by Equation 8 and Equation 9,
1 V
TON(max ) = ´ OUT
ƒ VIN(min)
(8)
RTON
(
)
TON(max)-50ns
VIN(min)
3
= ´
´
[W]
12
2
(VOUT + 150mV )
19 ´ 10
(9)
3. Choose inductor
A good starting point inductance value is where the ripple current is approximately 1/4 to 1/2 of the maximum
output current.
LIND =
1
IIND(ripple) ´ ƒ
´
(V
IN(max )
)
- VOUT ´ VOUT
VIN(max)
=
3
IOUT(max) ´ ƒ
´
(V
IN(max )
)
- VOUT ´ VOUT
VIN(max )
(10)
For applications that require fast transient response with minimum VOUT overshoot, consider a smaller
inductance than above. The cost of a small inductance value is higher steady-state ripple, larger line
regulation, and higher switching loss.
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The inductor also needs to have low DCR to achieve good efficiency, as well as enough room above peak
inductor current before saturation. The peak inductor current can be estimated in Equation 11.
IIND(peak ) =
(
)
VIN(max ) - VOUT ´ VOUT
VTRIP
1
+
´
RDS(on ) L ´ ƒ
VIN(max )
(11)
4. Choose output capacitor(s)
Organic semiconductor capacitor(s) or specialty polymer capacitor(s) are recommended. Determine ESR to
meet the required ripple voltage above. A quick approximation is shown in Equation 12.
VOUT ´ 0.015
VOUT
»
´ 60 [mW ]
ESR =
Iripple ´ 0.75
IOUT(max)
(12)
5. Choose MOSFETs
Loss-less current sensing and overcurrent protection of the TPS51117 is determined by RDS(on) of the lowside MOSFET. So, RDS(on) times the inductor current value at the overcurrent point should be in the range of
30 mV to 200 mV for the entire operational temperature range. Assuming a 20% guard band, RDS(on) in
Equation 13 should satisfy the full temperature range.
30mV
200mV
£ RDS(on) £
1.2 ´ IOUT(max) - 0.5 ´ Iripple
1.2 ´ IOUT(max) - 0.5 ´ Iripple
(13)
6. Choose Rtrip
Once the low-side FET is decided, select an appropriate Rtrip value that provides Vtrip equal to RDS(on) times
Ipeak.
7. LPF for V5FILT
To reject high-frequency noise and also secure safe start-up of the internal reference circuit, apply 1 μF of
MLCC closely at the V5FILT pin with a 300-Ω resistor to create a LPF between +5-V supply and the pin.
8. VBST capacitor, VBST diode
Apply 0.1-μF MLCC between VBST and the LL node as the flying capacitor for the high-side FET driver. The
TPS51117 has its own boost diode onboard between V5DRV and VBST. This is a PN junction diode and
strong enough for most typical applications. However, in case efficiency has priority over cost, the designer
may add a Schottky diode externally to improve gate drive voltage of the high-side FET. A Schottky diode
has a higher leakage current, especially at high temperature, than a PN junction diode. A low-leakage diode
should be selected in order to maintain VBST voltage during low-frequency operation in skip mode.
Table 2. Typical Application Circuit Components
18
SYMBOL
SPECIFICATION
MANUFACTURER
C1A, C1B
470 μF, 2.5 V, 12 mΩ
SANYO
PART NUMBER
2R5TPE470MC
C2
10 μF, 25 V, 2 pcs
Murata
GRM31CR61E106KA12B
L1
1.0 μH
Vishay, Toko
IHLP-5050, FDA1254-1R0M
Q1
30 V, 13 mΩ
International Rectifier
IRF7821
Q2
30 V, 5.8 mΩ
International Rectifier
IRF8113
R4
8.06 kΩ
—
Std
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8.2.3 Application Curves
EN_PSV (2 V/div)
EN_PSV (2 V/div)
VO (1 V/div)
VO (1 V/div)
PDOOD (5 V/div)
PGOOD (5 V/div)
DRVL (5 V/div)
t - Time - 10 ms/div
t - Time - 1 ms/div
Figure 22. 2.5-V Start-Up Waveforms
Figure 23. 2.5-V Shutdown Waveforms
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8.3 System Examples
+5V
+
TPS51117PW
EN_PSV
+ +VBAT
C4
0.1mF
1
EN_PSV
VBST
14
2
TON
DRVH
13
LL
12
TRIP
11
V5DRV
10
C2
2 0 mF
Q1
R3
249k W
R5
3
VOUT
4
V5FILT
5
VFB
6
PGOOD
DRVL
9
7
GND
PGND
8
300W
R6
100kW
+
R4
C3
1mF
L1
1.0mH
R1
8.5kW
C1A
GND
C1B
R2
22k W
Q2
PGOOD
VO
1.05V/10A
-
PGND
GND
Figure 24. 1.05-V/10-A Application from VBAT (PW Package)
+5V
+
+
TPS51117RGY
EN_PSV
R3
R6
100kW
TON
3
VOUT
300W
C3
1mF
PGOOD
14
VBST
2
249kW
R5
1
EN_PSV
4
V5FILT
5
VFB
C4
0.1mF
DRVH
13
LL
12
Q1
+
R4
TRIP
11
V5DRV
10
R1
8.5kW
GND
Q2
6
PGOOD
DRVL
GND
PGND
7
8
C2
20mF
L1
1.0mH
9
C1A
+VBAT
VO
1.05V/10A
C1B
R2
22kW
-
PGND
GND
Figure 25. 1.05-V/10-A Application from VBAT (RGY Package)
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9 Power Supply Recommendations
The devices are designed to operate at 5 V (4.5 V to 5.5 V) supply from another regulator such as the Limit
TPS51120 or TPS51020. The conversion input can be either VBAT or a 5-V rail, ranging from 1.8 V to 28 V. In
order to reject high-frequency noise and also secure safe start-up of the internal reference circuit, apply 1 μF of
MLCC closely at the V5FILT pin with a 300-Ω resistor to create a LPF between 5-V supply and the pin.
10 Layout
10.1 Layout Guidelines
Certain points must be considered before starting a layout work using the TPS51117.
• Connect the RC low-pass filter from 5-V supply to V5FILT, 300 Ω and 1 μF are recommended. Place the filter
capacitor close to the device, within 12 mm (0.5 inches) if possible.
• Connect the overcurrent setting resistors from TRIP to GND close to the device, right next to the device, if
possible. The trace from TRIP to resistor and resistor to GND should avoid coupling to a high-voltage
switching node.
• The discharge path (VOUT) should have a dedicated trace to the output capacitor(s); separate from the
output voltage sensing trace, and use a 1.5-mm (60 mils) or wider trace with no loops. Make sure the
feedback current setting resistor (the resistor between VFB to GND) is tied close to the device GND. The
trace from this resistor to the VFB pin should be short and thin. Place on the component side and avoid vias
between this resistor and the device.
• Connections from the drivers to the respective gate of the high-side or the low-side MOSFET should be as
short as possible to reduce stray inductance. Use a 0.65-mm (25 mils) or wider trace.
• All sensitive analog traces and components such as VOUT, VFB, GND, EN_PSV, PGOOD, TRIP, V5FILT,
and TON should be placed away from high-voltage switching nodes such as LL, DRVL, DRVH or VBST to
avoid coupling. Use internal layer(s) as ground plane(s) and shield feedback trace from power traces and
components.
• Gather the ground terminals of the VIN capacitor(s), VOUT capacitor(s), and the source of the low-side
MOSFETs as close as possible. GND (signal ground) and PGND (power ground) should be connected
strongly together near the device. The PCB trace defined as LL node, which connects to the source of the
high-side MOSFET, the drain of the low-side MOSFET, and the high-voltage side of the inductor, should be
as short and wide as possible.
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10.2 Layout Example
VBST
D
D
D
G
D
D
S
D
S
D
S
D
G
LL
SGND
V5FILT
TRIP
VFB
V5DRV
PGOOD
GND
PGND
SGND
S
VOUT
S
DRVH
S
TON
EN_PSV
VIN
INPUT
CAPACITOR
CONNECTED
TO POWER
GND ON
INTERNAL OR
BOTTOM
LAYER
DRVL
GND
OUTPUT
INDUCTOR
OUTPUT
CAPACITOR
VOUT
Figure 26. Layout Recommendation
10.3 Thermal Considerations
Power dissipation of the TPS51117 is mainly generated from the FET drivers. Average drive current can be
estimated by gate charge, Qg, times the switching frequency.
IG = Qg ´ ƒSW
(14)
Qg is the charge needed to charge gate capacitance up to the V5DRV voltage of 5 V. Actual values are shown
on MOSFET datasheets provided by the manufacturer. Total power dissipation, therefore, to drive the top and
bottom MOSFETs can be calculated by the following equation Equation 15.
(
WDRIVE = VV5DRV ´ Qg(top ) + Qg(btm )
) ´ƒ
SW
(15)
This power plus a small amount of dissipation (less than 5 mW) from controller circuitry needs to be effectively
dissipated from the package. Maximum power dissipation allowed for the package is calculated by:
TJ(max) - TA(max)
WPKG =
RQJA
where
•
•
•
TJ(max) is 125°C.
TA(max) is the maximum ambient temperature in the system.
RθJA is the thermal resistance from the silicon junction to the ambient.
(16)
This thermal resistance strongly depends on board layout. The TPS51117 is assembled in a standard TSSOP
package and the heat mainly moves to the board through its leads.
22
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.3 Trademarks
D-CAP, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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Copyright © 2005–2015, Texas Instruments Incorporated
Product Folder Links: TPS51117
23
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS51117PW
ACTIVE
TSSOP
PW
14
90
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
51117
Samples
TPS51117PWR
ACTIVE
TSSOP
PW
14
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
51117
Samples
TPS51117RGYR
ACTIVE
VQFN
RGY
14
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
51117
Samples
TPS51117RGYT
ACTIVE
VQFN
RGY
14
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
51117
Samples
TPS51117RGYTG4
ACTIVE
VQFN
RGY
14
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
51117
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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