TPS51317
SLUSAH9 – MARCH 2011
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3.3-V/5-V Input, 6-A, D-CAP+™ Mode Synchronous Step-Down Integrated FETs Converter
Check for Samples: TPS51317
FEATURES
DESCRIPTION
•
The TPS51317 is a fully integrated synchronous buck
regulator employing D-CAP+™ mode architechture. It
is used for 3.3-V and 5-V step-down systems where
space is a consideration, high-performance and
optimized component count are required. The
TPS51317 features four switching frequency settings
(up to 1.5 MHz), synchronous operation in SKIP,
droop support, external tracking support, pre-bias
startup, output soft discharge, integrated bootstrap
switch, power good function, enable function and
complete protection functions, and both output
ceramic and SP/POS capacitor support. It supports
supply and conversion voltages up to 6.0 V, and
output voltages adjustable from 0.6 V to 2.0V. The
TPS51317 is available in the 3.5 mm × 4 mm 20-pin
QFN package (Green RoHs compliant and Pb free)
and is specified from -40°C to 85°C.
1
2
•
•
•
•
•
•
•
•
Integrated FETs Converter w/TI Proprietary
D-CAP+™ Mode Architecture
Minimum External Components Count
Support all MLCC Output Capacitor and
SP/POSCAP
Auto-Skip Mode and Ripple Reduction Mode
Optimized Efficiency at Light and Heavy Loads
Selectable 800-kHz, 1-MHz, 1.2-MHz and
1.5-MHz Frequency
Up to 6.0-V Conversion Voltage Range
Adjustable Output Voltage Range From 0.6 V
to 2 V
Small 3.5 mm × 4 mm, 20-Pin QFN Package
APPLICATIONS
•
Low-Voltage Applications Stepping Down from
5-V or 3.3-V Rail
XXXX
XXXX
XXXX
XXXX
XXXX
TPS51317
VIN
VIN
17 EN
8
COMP
7
VREF
9
REFIN
BST 16
SW
VOUT
PGOOD 19
PGOOD
PGND
VOUT 10
PGND GND
MODE 18
GND GND
5VIN
20 V5IN
PGND
GND
6
GND
UDG-11041
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
D-CAP+ is a trademark of Texas Instruments.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2011, Texas Instruments Incorporated
TPS51317
SLUSAH9 – MARCH 2011
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Table 1. ORDERING INFORMATION (1)
TA
PACKAGE (2)
ORDERING NUMBER
PINS
OUTPUT SUPPLY
MINIMUM
QUANTITY
-40°C to 85°C
Plastic QFN
(RGB)
TPS51317RGBR
20
Tape and reel
3000
TPS51317RGBT
20
Mini reel
250
(1)
(2)
ECO PLAN
Green (RoHS and
no Pb/Br)
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or visit the TI
website at www.ti.com.
Package drawings, standard packing quantities, thermal data, symbolization, and PCB design guidelines are available at
www.ti.com/sc/package.
THERMAL INFORMATION
TPS51317
THERMAL METRIC (1)
RGB
UNITS
20 PINS
θJA
Junction-to-ambient thermal resistance (2)
35.5
θJCtop
Junction-to-case (top) thermal resistance (3)
39.6
θJB
Junction-to-board thermal resistance (4)
12.4
(5)
ψJT
Junction-to-top characterization parameter
ψJB
Junction-to-board characterization parameter (6)
12.5
θJCbot
Junction-to-case (bottom) thermal resistance (7)
3.7
(1)
(2)
(3)
(4)
(5)
(6)
(7)
2
0.5
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
The junction-to-ambient thermal resistance under natural convection is obtained in a simulation on a JEDEC-standard, high-K board, as
specified in JESD51-7, in an environment described in JESD51-2a.
The junction-to-case (top) thermal resistance is obtained by simulating a cold plate test on the package top. No specific
JEDEC-standard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
The junction-to-board thermal resistance is obtained by simulating in an environment with a ring cold plate fixture to control the PCB
temperature, as described in JESD51-8.
The junction-to-top characterization parameter, ψJT, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA, using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-board characterization parameter, ψJB, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA , using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-case (bottom) thermal resistance is obtained by simulating a cold plate test on the exposed (power) pad. No specific
JEDEC standard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
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ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
VALUE
MIN
UNIT
MAX
VIN, V5IN, BST (with respect to SW)
–0.3
7.0
BST
–0.3
14.0
SW
–2
7
EN
–0.3
7
MODE, REFIN
–0.3
3.6
–1
3.6
COMP, VREF
–0.3
3.6
PGOOD
–0.3
7.0
PGND
–0.3
0.3
Junction temperature
TJ
–40
150
Storage temperature
Tstg
–55
150
˚C
300
˚C
Input voltage range
VOUT
Output voltage range
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
(1)
V
V
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating
conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
RECOMMENDED OPERATING CONDITIONS
VALUE
MIN
–0.1
VIN
Input voltage range
Output voltage range
TYP
MAX
6.5
V5IN
4.5
6.5
BST
–0.1
13.5
SW
–1.0
6.5
EN
–0.7
6.5
VOUT, MODE, REFIN
–0.1
3.5
COMP, VREF
–0.1
3.5
PGOOD
–0.1
5.5
PGND
–0.1
0.1
-40
85
Operating temperature range, TA
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UNIT
V
V
°C
3
TPS51317
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ELECTRICAL CHARACTERISTICS
over recommended free-air temperature range, VV5IN = 5.0 V, PGND = GND (unless otherwise noted)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY: VOLTAGE, CURRENTS AND 5 V UVLO
IVINSD
VIN shutdown current
EN = 'LO'
V5VIN
5VIN supply voltage
V5IN voltage range
I5VIN
5VIN supply current
EN =’HI’, V5IN supply current
I5VINSD
5VIN shutdown current
EN = ‘LO’, V5IN shutdown current
VV5UVLO
V5IN UVLO
Ramp up; EN = 'HI'
VV5UVHYS
V5IN UVLO hysteresis
Falling hysteresis
VVREFUVLO
REF UVLO (1)
Rising edge of VREF, EN = 'HI'
VVREFUVHYS
REF UVLO hysteresis (1)
VPOR5VFILT
Reset
4.5
4.20
5
5.0
6.5
1.1
2
mA
0.2
7.0
µA
4.37
4.50
440
OVP latch is reset by V5IN falling below the reset threshold
1.5
µA
0.02
V
V
mV
1.8
V
100
mV
2.3
3.1
V
VOLTAGE FEEDBACK LOOP: VREF, VOUT, AND VOLTAGE GM AMPLIFIER
VOUTTOL
VOUT accuracy
VREFIN = 1 V, No droop
–1%
0%
1%
IVREF = 0 µA
1.98
2.00
2.02
IVREF = 50 µA
1.975
2.000
2.025
VVREF
VREF
IREFSNK
VREF sink current
GM
Transconductance
VCM
Common mode input voltage range (1)
0
2
V
VDM
Differential mode input voltage
0
80
mV
ICOMPSNK
COMP pin maximum sinking current
VCOMP = 2 V, (VREFIN - VOUT) = 80 mV
ICOMPSRC
COMP pin maximum sourcing current
VCOMP = 2 V
VOFFSET
Input offset voltage
TA = 25°C
RDSCH
Output voltage discharge resistance
f–3dbVL
–3dB Frequency (1)
VVREF = 2.05 V
2.5
V
mA
1.00
mS
µA
80
-80
µA
0
mV
Ω
42
4.5
6.0
7.5
MHz
43
53
57
mV/A
CURRENT SENSE: CURRENT SENSE AMPLIFIER, OVERCURRENT AND ZERO CROSSING
Gain from the current of the low-side FET to PWM comparator
when PWM = "OFF"
ACSINT
Internal current sense gain
IOCL
Positive overcurrent limit (valley)
7.6
IOCL(neg)
Negative overcurrent limit (valley)
–9.3
VZXOFF
Zero crossing comp internal offset
0
A
A
mV
DRIVERS: BOOT STRAP SWITCH
RDSONBST
Internal BST switch on-resistance
IBST = 10 mA, TA = 25°C
10
Ω
IBSTLK
Internal BST switch leakage current
VBST = 14 V, VSW = 7 V
1
µA
V
PROTECTION: OVP, UVP, PGOOD, and THERMAL SHUTDOWN
VPGDLL
PGOOD deassert to lower
(PGOOD → Low)
VPGHYSHL
PGOOD high hysteresis
VPGDLH
PGOOD de-assert to higher
(PGOOD → Low)
VPGHYSHH
PGOOD high hysteresis
VINMINPG
Minimum VIN voltage for valid
PGOOD
Measured at the VIN pin with a 2-mA sink current on PGOOD
pin
VOVP
OVP threshold
Measured at the VOUT pin wrt/ VREFIN
UVP threshold
Measured at the VOUT pin wrt/ VREFIN, device latches OFF,
begins soft-stop
VUVP
(1)
THSD
Thermal shutdown
THSD(hys)
Thermal Shutdown hysteresis (1)
(1)
4
Measured at the VOUT pin wrt/ VREFIN
84%
8%
Measured at the VOUT pin wrt/ VREFIN
116%
-8%
Latch off controller, attempt soft-stop.
Controller re-starts after temperature has dropped
0.9
1.3
1.5
117%
120%
123%
65%
68%
71%
145
°C
10
°C
Ensured by design, not production tested.
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ELECTRICAL CHARACTERISTICS (continued)
over recommended free-air temperature range, VV5IN = 5.0 V, PGND = GND (unless otherwise noted)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
TIMERS: ON-TIME, MINIMUM OFF-TIME, SS, AND I/O TIMINGS
tONESHOTC
PWM one-shot
(2)
VVIN = 5 V, VVOUT = 1.05 V, fSW = 860 KHz
240
VVIN = 5 V, VVOUT = 1.05 V, fSW = 1 MHz
210
VVIN = 5 V, VVOUT = 1.05 V, fSW = 1.2 MHz
175
VVIN = 5 V, VVOUT = 1.05 V, fSW = 1.5 MHz
140
360
ns
ms
ns
tMIN(off)
Minimum OFF time
VVIN = 5 V, VVOUT = 1.05 V, fSW = 1 MHz, DRVL on, SW =
PGND, VVOUT < VREFIN
tINT(SS)
Soft-start time
From EN = HI to VOUT =95%, default setting
1.6
tINT(SSDLY)
Internal soft-start delay time
From EN = HI to VOUT ramp starts
260
µs
tPGDDLY
PGOOD startup delay time
External tracking
8
ms
tPGDPDLYH
PGOOD high propagation delay time
50 mV over drive, rising edge
tPGDPDLYL
PGOOD low propagation delay time
50 mV over drive, falling edge
10
µs
tOVPDLY
OVP delay time
Time from the VOUT pin out of +20% of REFIN to OVP fault
10
µs
Time from EN_INT going high to undervoltage fault is ready
2
External tracking from VOUT ramp starts
8
tUVDLYEN
Undervoltage fault enable delay
tUVPDLY
UVP delay time
0.8
Time from the VOUT pin out of –30% of REFIN to UVP fault
1
1.2
ms
ms
µs
256
LOGIC PINS: I/O VOLTAGE AND CURRENT
VPGDPD
PGOOD pull-down voltage
PGOOD low impedance, ISINK = 4 mA, VV5IN = 4.5 V
IPGDLKG
PGOOD leakage current
PGOOD high impedance, forced to 5.5 V
VENH
EN logic high
EN, VCCP logic
VENL
EN logic low
EN, VCCP logic
IEN
EN input current
VMODETH
MODE threshold voltage (3)
IMODE
MODE current
(2)
(3)
–1
0
0.3
V
1
µA
2
V
0.5
V
1
µA
Threshold 1
80
130
180
Threshold 2
200
250
300
Threshold 3
370
420
470
Threshold 4
1.77
1.80
1.85
15
mV
V
µA
Ensured by design, not production tested.
See Table 4 for descriptions of MODE parameters.
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V5IN
PGOOD
MODE
EN
BST
TPS51317
RGB PACKAGE
(Top View)
20
19
18
17
16
PGND
2
14
SW
PGND
3
13
SW
VIN
4
12
SW
VIN
5
11
SW
7
8
9
10
VOUT
6
REFIN
SW
COMP
15
VREF
1
GND
PGND
Table 2. PIN FUNCTIONS
PIN
NO.
NAME
I/O
DESCRIPTION
16
BST
I
Power supply for internal high-side gate driver. Connect a 0.1-µF bootstrap capacitor between this pin and
the SW pin.
8
COMP
O
Connect series R-C filter between this pin and VREF for loop compensation.
17
EN
I
Enable of the SMPS (3.3-V logic compatible).
6
GND
–
Signal ground.
18
MODE
I
Allows selection of switching frequencies light-load modes. (See Table 4)
PGND
I
Power ground. Source terminal of the rectifying low-side power FET. Positive input for current sensing.
19
PGOOD
O
Power good output. Connect pull-up resistor.
9
REFIN
1
2
3
Target output voltageinput pin. Apply voltage between 0.6 V to 2.0 V.
11
12
13
SW
I/O
Switching node output. Connect to the external inductor. Also serve as current-sensing negative input.
V5IN
I
5-V power supply for analog circuits and gate drive.
VIN
I
Power supply input pin. Drain terminal of the switching high-side power FET.
14
15
20
4
5
6
10
VOUT
I
Output voltage monitor input pin.
7
VREF
O
2.0-V reference output. Connect a 0.22-µF ceramic capacitor to GND.
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BLOCK DIAGRAM
TPS51317
VREFIN –30%
VREFIN 16 %
+
UV
19 PGOOD
+
Delay
+
+
OV
VREFIN 16 %
VREFIN +20%
COMP
VS
Smplifier
REFIN
UVP
OVP
+
+
9
Ramp
Comp
SS
VREF
15 mA
8
+
PWM
Control Logic
On/Off Time
Minimum On/Off
SKIP/RR (OTP)
OCL/OVP/UVP
DIsharge
On-Time
Selection
18 MODE
16 BST
7
4
VIN
5
VIN
VBG
VOUT 10
Current Sense
Amplifier
+
11 SW
8R
12 SW
+
PGND
tON
OneShot
OC
R
GND
13 SW
14 SW
15 SW
SW
Current
Sense
XCON
18 V5IN
ZC
+
ZC Threshold
Modulation
EN 17
GND
6
1
PGND
2
PGND
3
PGND
Discharge
PGND
UDG-11058
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APPLICATION SCHEMATIC WITH TPS51317
EN
5VIN
2.2 mF
100 kW
PGOOD
20
19
18
V5IN PGOOD MODE
10 mF
10 mF
17
16
EN
BST
0.1 mF
1
PGND
SW 15
2
PGND
SW 14
3
PGND
4
VIN
SW 12
5
VIN
SW 11
0.42 mH
VOUT = 1.5 V
10 mF
0.1 mF
GND
SW 13
TPS51317
22 mF
22 mF
22 mF
22 mF
22 mF
22 mF
VREF COMP REFIN VOUT
6
7
0.22 mF
8
9
10
3.3 nF
100 kW
5 kW
300 kW
UDG-11059
Figure 1. Application Using Non-Droop Configuration
EN
5VIN
2.2 mF
100 kW
PGOOD
20
19
18
V5IN PGOOD MODE
10 mF
10 mF
17
16
EN
BST
0.1 mF
1
PGND
SW 15
2
PGND
SW 14
3
PGND
4
VIN
SW 12
5
VIN
SW 11
0.42 mH
VOUT = 1.5 V
10 mF
0.1 mF
GND
SW 13
TPS51317
22 mF
22 mF
22 mF
22 mF
22 mF
22 mF
VREF COMP REFIN VOUT
6
7
8
9
0.22 mF
5 kW
10
100 kW
300 kW
UDG-11060
Figure 2. Application Using Droop Configuration
8
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Application Circuit List of Materials
Recommended parts for key external components for the circuits in Figure 1 and Figure 2 are listed in Table 3.
Table 3. Key External Component Recommendations
(Figure 1 and Figure 2)
FUNCTION
MANUFACTURER
PART NUMBER
Output Inductor
Nec-Tokin
MPCG0740LR42C
Panasonic
ECJ2FB0J226M
Murata
GRM21BR60J226ME39L
Ceramic Output Capacitors
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APPLICATION INFORMATION
Functional Overview
The TPS51317 is a D-CAP+™ mode adaptive on-time converter. Integrated high-side and low-side FET supports
output current to a maximum of 6-ADC. The converter automatically runs in discontinuous conduction mode
(DCM) to optimize light-load efficiency. Multiple switching frequencies are provided to enable optimization of the
power chain for the cost, size and efficiency requirements of the design (see Table 4).
In adaptive on-time converters, the controller varies the on-time as a function of input and output voltage to
maintain a nearly constant frequency during steady-state conditions. In conventional constant on-time converters,
each cycle begins when the output voltage crosses to a fixed reference level. However, in the TPS51317, the
cycle begins when the current feedback reaches an error voltage level which is the amplified difference between
the reference voltage and the feedback voltage.
PWM Operation
Referring to Figure 3, in steady state, continuous conduction mode, the converter operates in the following way.
Starting with the condition that the top FET is off and the bottom FET is on, the current feedback (VCS) is higher
than the error amplifier output (VCOMP). VCS falls until it hits VCOMP, which contains a component of the output
ripple voltage. VCS is not directly accessible by measuring signals on pins of TPS51317. The PWM comparator
senses where the two waveforms cross and triggers the on-time generator.
Current
Feedback
Voltage (V)
VCS
VCOMP
VREF
tON
t
Time (ms)
UDG-10187
Figure 3. D-CAP+™ Mode Basic Waveforms
The current feedback is an amplified and filtered version of the voltage between PGND and SW during low-side
FET on-time. The TPS51317 also provides a single-ended differential voltage (VOUT) feedback to increase the
system accuracy and reduce the dependence of circuit performance on layout.
10
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PWM Frequency and Adaptive on Time Control
In general, the on-time (at the SW node) can be estimated by Equation 1.
V
1
tON = OUT ´
VIN
fSW
where
•
fSW is the frequency selected by the connection of the MODE pin
(1)
The on-time pulse is sent to the top FET. The inductor current and the current feedback rises to peak value.
Each ON pulse is latched to prevent double pulsing. Switching frequency settings are shown in .
Non-Droop Configuration
The TPS51317 can be configured as a non-droop solution. The benefit of a non-droop approach is that load
regulation is flat, therefore, in a system where tight DC tolerance is desired, the non-droop approach is
recommended. For the Intel system agent application, non-droop is recommended as the standard configuration.
The non-droop approach can be implemented by connecting a resistor and a capacitor between the COMP and
the VREF pins. The purpose of the type II compensation is to obtain high DC feedback gain while minimizing the
phase delay at unity gain cross over frequency of the converter.
The value of the resistor (RC) can be calculated using the desired unity gain bandwidth of the converter, and the
value of the capacitor (CC) can be calculated by knowing where the zero location is desired. An application tool
that calculates these values is available from your local TI Field Application Engineer.
Figure 4 shows the basic implementation of the non-droop mode using the TPS51317.
GMV = 1 mS
VSLEW
RC
CC
+
+
–
RDS(on)
LOUT
+
GMC= 1 mS
Driver
+
ESR
PWM
Comparator
ROUT
RLOAD
COUT
8 kW
+
–
VREF
UDG-10190
Figure 4. Non-Droop Mode Basic Implementation
Figure 5 shows shows the load regulation using non-droop configuration.
Figure 6 shows the transient response of TPS51317 using non-droop configuration, where COUT = 6 x 22 µF. The
applied step load is from 0 A to 3 A.
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1.650
VIN = 5 V
1.625
1.600
Output Voltage (V)
1.575
1.550
1.525
1.500
1.475
1.450
1.425
1.400
1.375
1.350
0
1
2
3
4
Output Current (A)
5
Figure 5. 1.5-V Load Regulation (VIN = 5 V)
Non-Droop Configuration
6
Figure 6. Non-Droop Configuration Transient
Response
Droop Configuration
The terminology for droop is the same as load line or voltage positioning as defined in the Intel CPU VCORE
specification. Based on the actual tolerance requirement of the application, load-line set points can be defined to
maximize either cost savings (by reducing output capacitors) or power reduction benefits.
Accurate droop voltage response is provided by the finite gain of the droop amplifier. The equation for droop
voltage is shown in Equation 2.
´ I(L)
A
VDROOP = CSINT
RDROOP ´ GM
where
•
•
•
•
•
low-side on-resistence is used as the current sensing element
ACSINT is a constant, which nominally is 53 mV/A.
I(L) is the DC current of the inductor, or the load current
RDROOP is the value of resistor from the COMP pin to the VREF pin
GM is the transconductance of the droop amplifier with nominal value of 1 mS
Equation 3 can be used to easily derive RDROOP for any load line slope/droop design target.
V
A CSINT
A CSINT
\ RDROOP =
RLOAD _ LINE = DROOP =
I(L)
RDROOP ´ GM
RLOAD _ LINE ´ GM
12
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(3)
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Figure 7 shows the basic implementation of the droop mode using the TPS51317.
GMV = 1 mS
VSLEW
RDROOP
+
+
–
RDS(on)
LOUT
+
GMC= 1 mS
Driver
+
ESR
PWM
Comparator
ROUT
RLOAD
COUT
8 kW
+
–
VREF
UDG-10188
Figure 7. DROOP Mode Basic Implementation
The droop (voltage positioning) method was originally recommended to reduce the number of external output
capacitors required. The effective transient voltage range is increased because of the active voltage positioning
(see Figure 8).
Load insertion
ILOAD
Load release
Droop
VOUT setpoint at 0 A
Maximum transient voltage
= (5%–1%) x 2 = 8% x VOUT
VOUT setpoint at 6 A
NonDroop
Maximum overshoot voltage =(5%–1%) x 1 = 4% x VOUT
VOUT setpoint at 0 A
Maximum undershoot voltage =(5%–1%) x 1 = 4% x VOUT
UDG-11080
Figure 8. DROOP vs Non-DROOP in Transient Voltage Window
In applications where the DC and the AC tolerances are not separated, which means there is not a strict DC
tolerance requirement, the droop method can be used.
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Table 4. Mode Definitions
MODE RESISTANCE
(kΩ)
LIGHT-LOAD POWER
SAVING MODE
SWITCHING
FREQUENCY
(fSW) (MHz)
1
0
SKIP
0.86
2
12
SKIP
1.2
3
22
SKIP
1.5
4
33
RR
(1)
5
47
RR (1)
0.86
6
68
PWM
1.2
7
100
PWM
1.5
8
OPEN
SKIP
1.0
MODE
(1)
1.0
Ripple reduction is a special light-load power saving feature. See (Light-Load Power Saving Features)
Figure 9 shows the load regulation of the 1.5-V rail using an RDROOP value of 5 kΩ.
Figure 10 shows the transient response of the TPS51317 using droop configuration and COUT = 6 × 22 µF. The
applied step load is from 0 A to 3 A.
1.650
VIN =3.3 V
1.625
1.600
Output Voltage (V)
1.575
1.550
1.525
1.500
1.475
1.450
1.425
1.400
1.375
1.350
0
1
2
3
4
Output Current (A)
5
Figure 9. 1.5-V Load Regulation (VIN = 5 V)
14
6
Figure 10. Droop Configuration Transient
Response, COUT = 6 x 22 µF and 0 A to 3 A
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Light-Load Power Saving Features
The TPS51317 has an automatic pulse-skipping mode to provide excellent efficiency over a wide load range.
The converter senses inductor current and prevents negative flow by shutting off the low-side gate driver. This
saves power by eliminating re-circulation of the inductor current. Further, when the bottom FET shuts off, the
converter enters discontinuous mode, and the switching frequency decreases, thus reducing switching losses as
well.
TPS51317 also provides a special light-load power saving feature, called ripple reduction. Essentially, it reduces
the on-time in SKIP mode to effectively reduce the output voltage ripple associated with using an all MLCC
capacitor output power stage design.
Power Sequences
Non-Tracking Startup
The TPS51317 can be configured for non-tracking application. When non-tracking is configured, output voltage is
regulated to the REFIN voltage which taps off the voltage dividers from the 2VREF. Either the EN pin or the V5IN
pin can be used to start up the device. The TPS51317 uses internal voltage servo DAC to provide a precise
1.6-ms soft-start time during soft-start initialization. (See Figure 11)
Tracking Startup
TPS51317 can also be configured for tracking application. When tracking configuration is desired, output voltage
is also regulated to the REFIN voltage which comes from external power source. In order for TPS51317 to
differentiate between a non-tracking configuration or a tracking configuration, there is a minimum delay time of
260 µs required between the time when the EN pin or the 5VIN pin is validated to the time when the REFIN pin
voltage can be applied, in order for the TPS51317 to track properly (see Figure 12). The valid REFIN voltage
range is between 0.6 V to 2 V.
Protection Features
The TPS51317 offers many features to protect the converter power chain as well as the system electronics.
5-V Undervoltage Protection (UVLO)
The TPS51317 continuously monitors the voltage on the V5IN pin to ensure that the voltage level is high enough
to bias the device properly and to provide sufficient gate drive potential to maintain high efficiency. The converter
starts with approximately 4.3 V and has a nominal of 440 mV of hysteresis. If the 5-V UVLO limit is reached, the
converter transitions the phase node into a off function. And the converter remains in the off state until the device
is reset by cycling 5 V until the 5-V POR is reached (2.3-V nominal). The power input does not have an UVLO
function
Power Good Signals
The TPS51317 has one open-drain power good (PGOOD) pin. During startup, there is a 1-ms power good high
propagation delay. The PGOOD pin de-asserts as soon as the EN pin is pulled low or an undervoltage condition
on V5IN or any other faults that require latch off action is detected.
Output Overvoltage Protection (OVP)
In addition to the power good function described above, the TPS51317 has additional OVP and UVP thresholds
and protection circuits.
An OVP condition is detected when the output voltage is approximately 120% × VREFIN. In this case, the
converter de-asserts the PGOOD signals and performs the overvoltage protection function. The converter
remains in this state until the device is reset by cycling 5 V until the 5-V POR threshold (2.3 V nominal) is
reached.
Output Undervoltage Protection (UVP)
Output undervoltage protection works in conjunction with the current protection described in the Overcurrent
Protection and Overcurrent Limit sections. If the output voltage drops below 70% of VREFIN, after an 8-µs delay,
the device latches OFF. Undervoltage protection can be reset only by EN or a 5-V POR.
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Overcurrent Protection
Both positive and negative overcurrent protection are provided in the TPS51317:
• Overcurrent Limit (OCL)
• Negative OCL (level same as positive OCL)
Overcurrent Limit
If the sensed current value is above the OCL setting, the converter delays the next ON pulse until the current
drops below the OCL limit. Current limiting occurs on a pulse-by-pulse basis. The TPS51317 uses a valley
current limiting scheme where the DC OCL trip point is the OCL limit plus half of the inductor ripple current. The
minimum valley OCL is 6 A over process and temperature.
During the overcurrent protection event, the output voltage likely droops until the UVP limit is reached. Then, the
converter de-asserts the PGOOD pin, and then latches OFF after an 8-µs delay. The converter remains in this
state until the device is reset.
1
IOCL(dc ) = IOCL(valley ) + ´ IP-P
2
(4)
Negative OCL
The negative OCL circuit acts when the converter is sinking current from the output capacitor(s). The converter
continues to act in a valley mode, the absolute value of the negative OCL set point is typically -6.5 A.
Thermal Protection
Thermal Shutdown
The TPS51317 has an internal temperature sensor. When the temperature reaches a nominal 145°C, the device
shuts down until the temperature cools by approximately 10°C. Then the converter restarts.
16
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Startup Timing Diagrams
Figure 11. Non-Tracking Start-Up
Figure 12. Tracking Start-Up
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TYPICAL CHARACTERISTICS
95
95
TA = 25°C
VIN = 3.3 V
TA = 25°C
VIN = 5 V
90
85
85
80
80
Efficiency (%)
75
70
65
60
Mode 1
Mode 3
Mode 4
Mode 7
Mode 8
55
50
45
40
0.01
0.1
1
Output Current (A)
75
70
65
60
Mode 1
Mode 3
Mode 4
Mode 7
Mode 8
55
50
45
40
0.01
10
0.1
1
Output Current (A)
Figure 13. Efficiency vs Output Current
Figure 14. Efficiency vs Output Current
1.50
1.50
Mode 1
Mode 3
Mode 4
Mode 7
Mode 8
1.00
0.75
0.50
0.25
1
Output Current (A)
1.00
0.75
0.50
0.25
TA = 25°C
VIN = 3.3 V
0.00
0.1
Mode 1
Mode 3
Mode 4
Mode 7
Mode 8
1.25
Power Loss (W)
Power Loss (W)
1.25
TA = 25°C
VIN = 5 V
0.00
0.1
10
1
Output Current (A)
Figure 15. Power Loss
350
30
310
40
Gain
30
260
250
20
210
150
-20
100
-30
-50
1000
10 k
10
160
0
Phase
-10
50
100 k
1M
0
10 M
110
-20
-30
25°C
-10°C
85°C
-40
Gain (dB)
200
Phase
-10
Phase (°)
10
Gain
300
20
Gain (dB)
360
60
50
40
-40
60
25°C
-10°C
85°C
-50
1000
10 k
Frequency (Hz)
10
100 k
1M
-40
10 M
Frequency (Hz)
Figure 17. Bode Plot (Non-Droop Mode) VIN = 5 V,
VOUT = 0.8 V, ILOAD = 5 A
18
10
Figure 16. Power Loss
400
50
0
10
Phase (°)
Efficiency (%)
90
Figure 18. Bode Plot (Droop Mode), VIN = 5 V,
VOUT = 0.8 V, ILOAD = 5 A
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DESIGN PROCEDURE
The simplified design procedure is done for a non-droop application using the TPS51317 converter.
Step One
Determine the specifications.
The Rail requirements provide the following key parameters:
1. VOUT = 1.5 V
2. ICC(max) = 6 A
3. IDYN(max) = 3 A
4. ICC(tdc) = 3 A
Step Two
Determine system parameters.
The input voltage range and operating frequency are of primary interest. For example:
1. VIN = 5 V
2. fSW = 1 MHz
Step Three
Determine inductor value and choose inductor.
Smaller values of inductor have better transient performance but higher ripple and lower efficiency. Higher values
have the opposite characteristics. It is common practice to limit the ripple current to 25% to 50% of the maximum
current. In this case, use 40%:
IP-P = 6 A ´ 0.4 = 2.4 A
(5)
At fSW = 1 MHz, with a 5-V input and a 1.5-V output:
ö
VOUT
÷÷
è (fSW ´ VIN ) ø
æ
L=
V ´ dT
=
IP-P
(VIN - VOUT )´ çç
IP-P
æ 1.5 ö
÷÷
è (1´ 5 ) ø
(5 - 1.5 )´ çç
=
1.5 A
= 0.43 mH
(6)
For this application, a 0.42-µH, 1.55-mΩ inductor from NEC-TOKIN with part number MPCG0740LR42C is
chosen.
Step Four
Set the output voltage.
VVREF
´ RLOWER
VOUT =
RUPPER + RLOWER
(7)
The output voltage is determined by the 2-V reference (VREF) and the resistor dividers (RUPPER and RLOWER).
The output voltage is regulated to the REFIN pin. Because the 2-V reference current capability is limited to less
than 50 µA, care should be taken when selecting the resistor dividers. For the current reference design of 1.5 V
(see application schematics shown in Figure 1 and Figure 2, RUPPER = 100 kΩ, RLOWER = 300 kΩ.
Step Five
Calculate OCL.
The DC OCL level of TPS51317 design is determined by Equation 8,
1
1
IOCL(dc ) = IOCL(valley ) + ´ IP-P = 6 A + ´ 1.5 A = 6.75 A
2
2
(8)
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The minimum valley OCL is 6 A over process and temperature, and IP-P = 1.5 A, the minimum DC OCL is
calculated to be 6.75A.
Step Six
Determine the output capacitance.
To determine COUT based on transient and stability requirement, first calculate the the minimum output
capacitance for a given transient.
Equation 10 and Equation 9 can be used to estimate the amount of capacitance needed for a given dynamic load
step/release. Please note that there are other factors that may impact the amount of output capacitance for a
specific design, such as ripple and stability. Equation 10 and Equation 9 are used only to estimate the transient
requirement, the result should be used in conjunction with other factors of the design to determine the necessary
output capacitance for the application.
æV
ö
´t
L ´ DILOAD(max )2 ´ ç VOUT SW + tMIN(off ) ÷
ç VIN(min )
÷
è
ø
COUT(min_ under ) =
ææ V
ö
ö
IN(min ) - VVOUT
÷
÷ ´ tSW - t
2 ´ DVLOAD(insert ) ´ ç ç
MIN(off ) ÷ ´ VVOUT
çç
÷
VIN(min )
ø
èè
ø
(9)
2
COUT(min_ over ) =
LOUT ´ DILOAD(max )
(
)
2 ´ DVLOAD(release ) ´ VVOUT
(10)
Equation 9 and Equation 10 calculate the minimum COUT for meeting the transient requirement, which is 84 µF
assuming the following:
• ±3% voltage allowance for load step and release
• MLCC capacitance derating of 60% due to DC and AC bias effect
In this reference design, 6, 22-µF capacitors are used in order to provide this amount of capacitance.
Step Seven
Determine the stability based on the output capacitance COUT.
In order to achieve stable operation. The 0-dB frequency, f0 should be kept less than 1/5 of the switching
frequency (1 MHz). (See Figure 4)
R
GM
1
´
´ C = 190kHz
f0 =
2p COUT RS
where
•
RS = RDS(on) × GMC × RLOAD
(11)
.
f ´ RS ´ 2p ´ COUT 190kHz ´ 53mW ´ 2p ´ 80 mF
=
» 5kW
RC = 0
GM
1mS
(12)
Using 6, 22-µF capacitors, the compensation resistance, RC can be calculated to be approximately 5 kΩ.
The purpose of the comparator capacitor (CC) is to reduce the DC component to obtain high DC feedback gain.
However, as it causes phase delay, another zero to cancel this effect at f0 is needed. This zero can be
determined by values of CC and the compensation resistor, RC.
f
1
= 0
fZ =
2p ´ RC ´ CC 10
(13)
And since RC has previously been derived, the value of CC is calculated to be 2.2 nF. In order to further boost
phase margin, a value of 3.3-nF is chosen for this reference design.
20
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Step Eight
Select decoupling and peripheral components.
For TPS51317 peripheral capacitors use the following minimum values of ceramic capacitance. X5R or better
temperature coefficient is recommended. Tighter tolerances and higher voltage ratings are always appropriate.
• V5IN decoupling ≥ 22 µF, ≥ 10 V
• VREF decoupling 0.22 µF to 1 µF, ≥ 4 V
• Bootstrap capacitors ≥ 0.1 µF, ≥ 10 V
• Pull-up resistors on PGOOD, 100 kΩ
Layout Considerations
Good layout is essential for stable power supply operation. Follow these guidelines for an efficient PCB layout.
• Connect PGND pins (or at least one of the pins) to the thermal PAD underneath the device. Also connect
GND pin to the thermal PAD underneath the device. Use four vias to connect the thermal pad to internal
ground planes.
• Place VIN, V5IN and 2VREF decoupling capacitors as close to the device as possible.
• Use wide traces for the VIN, VOUT, PGND and SW pins. These nodes carry high current and also serve as
heat sinks.
• Place feedback and compensation components as close to the device as possible.
• Keep COMP analog signal away from noisy signals (SW, BST).
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS51317RGBR
ACTIVE
VQFN
RGB
20
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
51317
TPS51317RGBT
ACTIVE
VQFN
RGB
20
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
51317
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
