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TPS54061-Q1
SLVSBM7A – MARCH 2013 – REVISED JANUARY 2016
TPS54061-Q1 Wide-Input, 60-V, 200-mA Synchronous Step-Down
DC–DC Converter With Low IQ
1 Features
3 Description
•
•
The TPS54061-Q1 device is a 60-V, 200-mA,
synchronous step-down DC-DC converter with
integrated high-side and low-side MOSFETs. Currentmode control provides simple external compensation
and flexible component selection. The non-switching
supply current is 90 µA. Using the enable pin reduces
the shutdown supply current to 1.4 µA.
1
•
•
•
•
•
•
•
•
•
•
•
•
Qualified for Automotive Applications
AEC-Q100 Qualified With the Following Results:
– Device Temperature Grade 1: –40°C to 125°C
Ambient Operating Temperature Range
– Device HBM ESD Classification Level H2
– Device CDM ESD Classification Level C3B
Integrated High-Side and Low-Side MOSFETs
Diode Emulation for Light-Load Efficiency
Peak-Current Mode Control
90-µA Operating Quiescent Current
1.4-µA Shutdown Supply Current
50-kHz to 1.1-MHz Adjustable Switching
Frequency
Synchronizes to External Clock
0.8 V ±1% Voltage Reference
Stable With Ceramic Output Capacitors or LowCost Aluminum Electrolytic
Cycle-by-Cycle Current Limit, Thermal, OVP, and
Frequency Foldback Protection
3-mm x 3-mm, 8-Pin SON Package With Thermal
Pad
–40°C to 150°C Operating Junction Temperature
The internal undervoltage lockout setting is 4.5 V, but
using two resistors on the enable pin can increase
the setting. The internal slow-start time controls the
output-voltage start-up ramp.
The adjustable switching-frequency range allows
optimization of efficiency and external component
size. Frequency foldback and thermal shutdown
protect the part during an overload condition.
The TPS54061-Q1 enables small designs by
integrating the MOSFETs and boot recharge diode,
and by minimizing the IC footprint with a small 3-mm
× 3-mm thermally-enhanced SON package.
The TPS54061-Q1 is supported in the WEBENCH™
Designer at www.ti.com.
Device Information(1)
2 Applications
•
•
To increase light-load efficiency, the low-side
MOSFET emulates a diode when the inductor current
reaches zero.
PART NUMBER
Low-Power Standby or Bias Voltage Supplies
High-Efficiency Replacement for High-Voltage
Linear Regulators
TPS54061-Q1
PACKAGE
BODY SIZE (NOM)
SON (8)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
Efficiency
100
VIN
BOOT
VIN
90
TPS54061-Q1
80
RT /CLK
VSNS
VOUT
Efficiency (%)
PH
EN
70
60
50
40
COMP
30
PowerPAD
GND
VOUT = 5 V, fSW = 50 kHz
VOUT = 5 V, fSW = 400 kHz
VOUT = 3.3 V, fSW = 400 kHz
20
10
0
0.001
0.010
Load Current (A)
0.100
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54061-Q1
SLVSBM7A – MARCH 2013 – REVISED JANUARY 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 10
7.1 Overview ................................................................. 10
7.2 Functional Block Diagram ....................................... 11
7.3 Feature Description................................................. 11
7.4 Device Functional Modes........................................ 15
8
Applications and Implementation ...................... 16
8.1 Application Information............................................ 16
8.2 Typical Applications ................................................ 16
9 Power Supply Recommendations...................... 30
10 Layout................................................................... 30
10.1 Layout Guidelines ................................................. 30
10.2 Layout Example .................................................... 30
11 Device and Documentation Support ................. 31
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
31
31
31
31
31
31
12 Mechanical, Packaging, and Orderable
Information ........................................................... 31
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Original (March 2013) to Revision A
•
2
Page
Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device Functional
Modes section, Application and Implementation section, Power Supply Recommendations section, Layout section,
Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information section .................. 1
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SLVSBM7A – MARCH 2013 – REVISED JANUARY 2016
5 Pin Configuration and Functions
DRB Package
8-Pin SON
Bottom View
PH 8
Thermal
Pad (9)
GND 7
COMP 6
VSENSE 5
1 BOOT
2 VIN
See appended
Mechanical
Data for
size and shape
3 EN
4 RT/CLK
Pin Functions
PIN
I/O
DESCRIPTION
NO.
NAME
1
BOOT
O
The device requires a bootstrap capacitor between BOOT and PH. If the voltage on this capacitor is
below the minimum required by the output device, the output switches off until refreshing of the
capacitor is complete.
2
VIN
I
Input supply voltage, 4.7 V to 60 V
3
EN
I
Enable pin with internal pullup current source. Pull below 1.18 V to disable. Float to enable. Adjust
the input undervoltage lockout (UVLO) with two resistors, see Enable and Adjusting Undervoltage
Lockout.
4
RT/CLK
I
Resistor timing and external clock. An internal amplifier holds this pin at a fixed voltage when using
an external resistor to ground to set the switching frequency. Pulling the pin above the PLL upper
threshold causes a mode change, and the pin becomes a synchronization input. The change
disables the internal amplifier, and the pin becomes a high-impedance clock input to the internal PLL.
Stoppage of the clocking edges re-enables the internal amplifier, and the mode returns to a resistor
frequency programming.
5
VSENSE
I
Inverting input of the transconductance (gm) error amplifier
6
COMP
O
Error amplifier output and input to the output switch current comparator. Connect frequency
compensation components to this pin.
7
GND
G
Ground
8
PH
O
The source of the internal high-side power MOSFET and drain of the internal low-side MOSFET
9
Thermal pad
–
Connect the GND pin electrically to the exposed pad on the printed circuit board for proper operation.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)
VIN
EN
Voltage
(2)
MAX
UNIT
62
V
–0.3
8
V
BOOT-PH
8
V
BOOT
70
V
VSENSE
–0.3
6
V
COMP
–0.3
3
V
PH
–0.6
62
V
–2
62
V
–0.3
6
V
100
mA
PH, 10-ns transient
RT/CLK
VIN
Current
MIN
–0.3
Internally limited
BOOT
PH
A
Internally limited
A
Operating junction temperature
–40
150
ºC
Storage temperature, Tstg
–65
150
ºC
(1)
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
See Enable and Adjusting Undervoltage Lockout.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human body model (HBM), per QSS 009-105 (JESD22-A114A) and
AEC-Q100 Classification Level H2 (1)
±2000
Charged-device model (CDM), per QSS 009-147 (JESD22-C101B.01)
and AEC-Q100 500V Classification Level C3B (2)
±750
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
Input voltage, VIN
4.7
Switching frequency synchronized to external clock
4
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UNIT
60
V
200
mA
50
1100
kHz
300
1100
kHz
Output current
Switching frequency set by RT/CLK resistor
MAX
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6.4 Thermal Information
TPS54061-Q1
THERMAL METRIC (1)
DRB (SON)
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance
42.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
46.0
°C/W
RθJB
Junction-to-board thermal resistance
18.1
°C/W
ψJT
Junction-to-top characterization parameter
0.5
°C/W
ψJB
Junction-to-board characterization parameter
18.3
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
3.0
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
6.5 Electrical Characteristics
TJ = –40°C to 150°C, VIN = 4.7 to 60 V (unless otherwise noted) (1)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
Operating input voltage
4.7
60
V
Shutdown supply current
EN = 0 V
1.4
µA
IQ operating; non-switching
VSENSE = 0.9 V, VIN = 12 V
90
110
µA
1.23
1.4
V
ENABLE AND UVLO (EN PIN)
Enable threshold
Input current
Rising
Falling
1
1.18
V
Enable threshold 50 mV
–4.7
µA
Enable threshold –50 mV
–1.2
µA
Hysteresis
–3.5
µA
Enable high to start switching time
450
µs
4.5
V
VIN
VIN start voltage
VIN rising
VOLTAGE REFERENCE
Voltage reference
TJ = 25°C, VIN = 12 V
0.792
0.8
0.808
1 mA < IOUT < Minimum current limit
0.784
0.8
0.816
BOOT-PH = 5.7 V
1.5
3
Ω
VIN = 12 V
0.8
1.5
Ω
V
HIGH-SIDE MOSFET
Switch resistance
LOW-SIDE MOSFET
Switch resistance
ERROR AMPLIFIER
Input current
VSENSE pin
Error-amplifier gm
–2 µA < I(COMP) < 2 µA, V(COMP) = 1 V
EA gm during slow-start
–2 µA < I(COMP) < 2 µA, V(COMP) = 1 V, VSENSE = 0.4 V
Error amplifier DC gain
VSENSE = 0.8 V
Minimum unity-gain bandwidth
Error amplifier source and sink
V(COMP) = 1 V, 100-mV overdrive
Start-switching threshold
COMP to Iswitch gm
(1)
20
nA
108
µMhos
27
µMhos
1000
V/V
0.5
MHz
±8
µA
0.57
V
1
A/V
The electrical ratings specified in this section apply to all specifications in this document unless otherwise noted. These specifications
are interpreted as conditions that do not degrade the parametric or functional specifications of the device for the life of the product
containing it.
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Electrical Characteristics (continued)
TJ = –40°C to 150°C, VIN = 4.7 to 60 V (unless otherwise noted)(1)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNIT
250
350
500
mA
CURRENT LIMIT
High-side sourcing current-limit
threshold
BOOT-PH = 5.7 V
Zero-cross detect current
–1.1
mA
176
C
THERMAL SHUTDOWN
Thermal shutdown
RT/CLK
Operating frequency using RT mode
Switching frequency
50
R(RT/CLK) = 120 kΩ
425
Minimum CLK pulse duration
RT/CLK voltage
472
1100
kHz
520
kHz
40
R(RT/CLK) = 120 kΩ
ns
0.53
RT/CLK high threshold
V
1.8
RT/CLK low threshold
0.5
RT/CLK falling-edge to PH risingedge delay
Measure at 500 kHz with RT resistor
PLL lock-in time
Measure at 500 kHz
PLL frequency range
V
V
67
ns
100
300
µs
1100
kHz
PH
Minimum ON-time
Measured at 50% to 50%, IOUT = 200 mA
120
ns
Dead time
VIN = 12 V, IOUT = 200 mA, one transition
30
ns
BOOT
BOOT-to-PH regulation voltage
VIN = 12 V
BOOT-PH UVLO
6
V
2.9
V
2.36
ms
INTERNAL SLOW-START TIME
Slow-start time
6
fSW = 472 kHz, RT = 120 kΩ, 10% to 90%
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3
1.4
2.5
1.2
On Resistance (Ω)
On Resistance (Ω)
6.6 Typical Characteristics
2
1.5
1
0.5
1
0.8
0.6
0.2
0
–50
–25
50
75
0
100
25
Junction Temperature (°C)
125
0
–50
150
VIN = 12 V
Figure 1. High-Side rDS(on) vs Temperature
150
Figure 2. Low-Side rDS(on) vs Temperature
% of Normal Switching Frequency
Voltage Reference (V)
125
120
0.801
0.799
0.797
0.795
0.793
0.791
0.789
–25
0
25
50
75
100
Junction Temperature (Deg)
125
100
Rising
80
60
40
20
0
150
Falling
0
100
200
300
400
500
600
700
800
Feedback Voltage (mV)
VIN = 12 V
Figure 3. VREF Voltage vs Temperature
RT = 120 kΩ
TJ = 25°C
Figure 4. Frequency vs VSENSE Voltage
540
1100
1000
520
Oscillator Frequency (kHz)
Oscillator Frequency (kHz)
50
75
0
25
100
Junction Temperature (°C)
VIN = 12 V
0.803
0.787
–50
–25
500
480
460
440
420
900
800
700
600
500
400
300
200
100
400
-50
-25
0
25
50
75
100
Junction Temperature (°C)
VIN = 12 V
125
150
RT = 120 kΩ
Figure 5. Frequency vs Temperature
0
25
100
Timing Resistance (kW)
VIN = 12 V
1K
25K
TJ = 25°C
Figure 6. Frequency vs RT/CLK Resistance
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140
1.26
120
1.24
Enable Voltage (V)
Transconductance (mS)
Typical Characteristics (continued)
100
80
60
40
VEN Rising
1.22
1.20
VEN Falling
1.18
1.16
1.14
20
1.12
0
–50
–25
50
75
0
25
100
Junction Temperature (°C)
125
1.10
–50
150
VIN = 12 V
0
25
50
75
100
Junction Temperature (°C)
125
150
VIN = 12 V
Figure 7. Error Amplifier Transconductance vs Temperature
Figure 8. Enable-Pin Voltage vs Temperature
-3.35
4.6
4.55
-3.40
4.5
-3.45
Input Voltage (V)
Enable Hysteresis Current (µA)
–25
-3.50
-3.55
-3.60
-3.65
4.45
4.4
4.35
4.3
UVLO Start
4.25
4.2
4.15
UVLO Stop
4.1
4.05
-3.70
-3.75
–50
–25
50
75
0
25
100
Junction Temperature (°C)
125
4
-50
150
-25
0
25
50
75
100
125
150
Junction Temperature (°C)
VIN = 12 V
Figure 9. Enable-Pin Hysteresis Current
vs Temperature
Figure 10. Input Voltage (UVLO) vs Temperature
3
-1
-1.05
2.5
Shutdown Current (µA)
Enable Current (µA)
-1.1
-1.15
-1.2
-1.25
-1.3
-1.35
-1.4
2
1.5
1
TJ = 150°C
TJ = −40°C
TJ = 25°C
0.5
-1.45
-1.5
0
0
5
10
15
20
25 30 35 40
Input Voltage (V)
VIN = 12 V
45
50
55
60
5
10
15
20
25 30 35 40
Input Voltage (V)
45
50
55
60
EN = 0 V
TJ = 25°C
Figure 11. Enable-Pin Pullup Current vs Input Voltage
8
0
Figure 12. Shutdown Supply Current (VIN Pin)
vs Input Voltage
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98
2
96
1.75
94
Supply Current (µA)
Supply Current (µA)
Typical Characteristics (continued)
92
90
88
86
84
TJ = 150°C
TJ = −40°C
TJ = 25°C
82
80
0
5
10
15
20
25 30 35 40
Input Voltage (V)
45
EN = Open
50
55
1.5
1.25
1
0.75
0.5
0.25
0
60
VSENSE = 0.83 V
1
2
3
Input Voltage (V)
4
5
Figure 14. Supply Current (VIN Pin)
vs Input Voltage (0 V to VSTART), EN Pin Low
160
2.48
TJ = 150°C
120
TJ = –40°C
100
TJ = 25°C
2.46
2.44
SS Time (ms)
140
Supply Current (µA)
0
EN = 0 V
Figure 13. Supply Current (VIN Pin) vs Input Voltage
80
60
2.42
2.40
2.38
40
2.36
20
2.34
0
TJ = 150°C
TJ = −40°C
TJ = 25°C
0
1
2
3
Input Voltage (V)
4
2.32
-50
5
EN = Open
-25
0
25
50
75
100
Junction Temperature (°C)
VIN = 12 V
Figure 15. Supply Current (VIN Pin) vs
Input Voltage (0 V to VSTART), EN Pin Open
125
150
fsw = 472 kHz
Figure 16. Slow-Start Time vs Temperature
Current Limit Threshold (A)
0.45
TJ = 25°C
TJ = –40°C
0.4
0.35
0.3
TJ = 150°C
0.25
0.2
0
5
10
15
20
25 30 35 40
Input Voltage (V)
45
50
55
60
Figure 17. Current Limit vs Input Voltage
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7 Detailed Description
7.1 Overview
The TPS54061-Q1 device is a 60-V, 200-mA, step-down (buck) regulator with integrated high-side and low-side
N-channel MOSFETs. To improve performance during line and load transients, the device implements a
constant-frequency, current-mode control which reduces output capacitance and simplifies external frequencycompensation design.
The switching frequency of 50 kHz to 1100 kHz allows for efficiency and size optimization when selecting the
output filter components. Adjustment of the switching frequency is by use of a resistor to ground on the RT/CLK
pin. The device has an internal phase-lock loop (PLL) on the RT/CLK pin that synchronizes the power-switch
turnon to a falling edge of an external system clock.
The TPS54061-Q1 has a default start-up voltage of approximately 4.5 V. The EN pin has an internal pullup
current source, a possible use of which is to adjust the input voltage undervoltage lockout (UVLO) threshold with
two external resistors. In addition, the pullup current provides a default condition. When the EN pin is floating, the
device operates. The operating current is 90 µA when not switching and under no load. When the device is
disabled, the supply current is 1.4 µA.
The integrated 1.5-Ω high-side MOSFET and 0.8-Ω low-side MOSFET allow for high-efficiency power-supply
designs capable of delivering 200 milliamperes of continuous current to a load.
The TPS54061-Q1 reduces the external component count by integrating the boot recharge diode. A capacitor
between the BOOT and PH pins supplies the bias voltage for the integrated high-side MOSFET. The boot
capacitor voltage is monitored by an UVLO circuit and turns the high-side MOSFET off when the boot voltage
falls below a preset threshold. The TPS54061-Q1 can operate at high duty cycles because of the boot UVLO.
The output voltage can be adjusted down to as low as the 0.8-V reference.
The TPS54061-Q1 has an internal output OV protection that disables the high-side MOSFET if the output voltage
is 109% of the nominal output voltage.
The TPS54061-Q1 reduces external component count by integrating the slow-start time using a reference DAC
system.
The TPS54061-Q1 resets the slow-start times during overload conditions with an overload recovery circuit. The
overload recovery circuit slow-starts the output from the fault voltage to the nominal regulation voltage once a
fault condition is removed. A frequency foldback circuit reduces the switching frequency during start-up and
overcurrent fault conditions to help control the inductor current.
10
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7.2 Functional Block Diagram
VIN
EN
Thermal
Shutdown
Enable
Comparator
UVLO
Shutdown
Shutdown
Logic
Enable
Threshold
VSENSE
Boot
Charge
Regulator
OV
ERROR
AMPLIFIER
Boot
UVLO
Minimum
Clamp
Current
Sense
PWM
Comparator
BOOT
Deadtime
Control Logic
Reference DAC
With
Slow Start
Shutdown
S
Slope
Compensation
PH
COMP
Frequency
Shift
DRV
REG
Maximum
Clamp
Oscillator
with PLL
ZX
detect
GND
RT /CLK
THERMAL PAD
7.3 Feature Description
7.3.1 Fixed Frequency PWM Control
The TPS54061-Q1 uses adjustable fixed frequency, peak current mode control. The output voltage is sensed
through external resistors on the VSENSE pin and compared to an internal voltage reference by an error
amplifier which drives the COMP pin. An internal oscillator initiates the turnon of the high-side power switch. The
error amplifier output is compared to the high-side power switch current. When the power switch current reaches
the level set by the COMP voltage, the power switch is turned off. The COMP pin voltage increases and
decreases as the output current increases and decreases. The device implements current limiting by clamping
the COMP pin voltage to a maximum level.
7.3.2 Slope Compensation Output Current
The TPS54061-Q1 adds a compensating ramp to the switch current signal. This slope compensation prevents
sub-harmonic oscillations.
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Feature Description (continued)
7.3.3 Error Amplifier
The TPS54061-Q1 uses a transconductance amplifier for the error amplifier. The error amplifier compares the
VSENSE voltage to the lower of the internal slow-start voltage or the internal 0.8-V voltage reference. The
transconductance (gm) of the error amplifier is 108 µA/V during normal operation. During the slow-start
operation, the transconductance is a fraction of the normal operating gm. The frequency compensation
components (capacitor, and the series resistor and capacitor) are added to the COMP pin to ground.
7.3.4 Voltage Reference
The voltage reference system produces a precise voltage reference over temperature by scaling the output of a
temperature stable band-gap circuit.
7.3.5 Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the VSENSE pin. TI recommends using
1% tolerance or better divider resistors. Start with 10 kΩ for the RLS resistor and use Equation 1 to calculate RHS.
- 0.8 V ö
æV
RHS = RLS ´ ç OUT
÷÷
ç
0.8 V
è
ø
(1)
7.3.6 Enable and Adjusting Undervoltage Lockout (UVLO)
The TPS54061-Q1 is enabled when the VIN pin voltage rises above 4.5 V and the EN pin voltage exceeds the
EN rising threshold of 1.23 V. The EN pin has an internal pull-up current source, I1, of 1.2 µA that provides the
default enabled condition when the EN pin floats.
If an application requires a higher input undervoltage lockout (UVLO) threshold, use the circuit shown in
Figure 18 to adjust the input voltage UVLO with two external resistors. When the EN pin voltage exceeds 1.23 V,
an additional 3.5 µA of hysteresis current, Ihys, is sourced out of the EN pin. When the EN pin is pulled below
1.18 V, the 3.5-µA Ihys current is removed. This additional current facilitates adjustable input voltage hysteresis.
Use Equation 2 to calculate RUVLO1 for the desired input start and stop voltages. Use Equation 3 to similarly
calculate RUVLO2.
In applications designed to start at relatively low input voltages (for example, from 4.7 V to 10 V) and withstand
high input voltages (for example, from 40 V to 60 V), the EN pin may experience a voltage greater than the
absolute maximum voltage of 8 V during the high input voltage condition. TI recommends to use a Zener diode to
clamp the pin voltage below the absolute maximum rating.
VIN
TPS54061
i1
ihys
Ruvlo1
EN
Optional
VEN
Ruvlo2
Figure 18. Adjustable Undervoltage Lockout
æV
ö
VSTART ç ENAFALLING ÷ - VSTOP
è VENARISING ø
RUVLO 1 =
æ V
ö
I1 × ç 1- ENAFALLING ÷ + IHYS
VENARISING ø
è
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Feature Description (continued)
RUVLO 2 =
RUVLO 1 ´ VENAFALLING
VSTOP - VENAFALLING + RUVLO 1 ´ (I1 + IHYS )
(3)
7.3.7 Internal Slow-Start
The TPS54061-Q1 has an internal digital slow-start that ramps the reference voltage from 0 V to its final value in
1114 switching cycles. The internal slow-start time is calculated by Equation 4:
1114
tss(ms) =
fSW (kHz)
(4)
If the EN pin is pulled below the stop threshold of 1.18 V, switching stops and the internal slow-start resets. The
slow-start also resets in thermal shutdown.
7.3.8 Constant Switching Frequency and Timing Resistor (RT/CLK Pin)
The switching frequency of the TPS54061-Q1 is adjustable over a wide range from 50 kHz to 1100 kHz by
varying the resistor on the RT/CLK pin. The RT/CLK pin voltage is typically 0.53 V and must have a resistor to
ground to set the switching frequency. To determine the timing resistance for a given switching frequency, use
Equation 5. To reduce the solution size, the switching frequency is typically set as high as possible, but tradeoffs
of the supply efficiency, maximum input voltage, and minimum controllable ON-time should be considered. The
minimum controllable ON-time is typically 120 ns and limits the operating frequency for high input voltages. The
maximum switching frequency is also limited by the frequency shift circuit. More discussion on the details of the
maximum switching frequency, refer to Selecting the Switching Frequency.
71657
RT (kW) =
fSW (kHz)1.039
(5)
7.3.9 Selecting the Switching Frequency
The TPS54061-Q1 implements current mode control which uses the COMP pin voltage to turn off the high-side
MOSFET on a cycle-by-cycle basis. Each cycle the switch current and COMP pin voltage are compared, when
the peak switch current intersects the COMP voltage, the high-side switch is turned off. During overcurrent
conditions that pull the output voltage low, the error amplifier responds by driving the COMP pin high, increasing
the switch current. The error amplifier output is clamped internally, which functions as a switch current limit.
To enable higher switching frequency at high input voltages, the TPS54061-Q1 implements a frequency shift.
The switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 V on VSENSE pin. The
device implements a digital frequency shift to enable synchronizing to an external clock during normal start-up
and fault conditions. Because the device can only divide the switching frequency by 8, there is a maximum input
voltage limit in which the device operates and still have frequency shift protection. During short-circuit events
(particularly with high input voltage applications), the control loop has a finite minimum controllable ON-time and
the output has a low voltage. During the switch ON-time, the inductor current ramps to the peak current limit
because of the high input voltage and minimum ON-time. During the switch OFF-time, the inductor would
normally not have enough OFF-time and output voltage for the inductor to ramp down by the ramp up amount.
The frequency shift effectively increases the OFF-time allowing the current to ramp down.
æ 1 ö
æ V OUT + R LS ´ I O + R DC ´ I O ö
fSW (maxskip) = ç
÷ ´ ç
÷
è t ON ø
è V IN - I O ´ R HS + I O ´ R LS ø
where
•
•
•
•
•
•
•
tON = Controllable ON-time
VOUT = Output Voltage
RLS = Low-side MOSFET resistance
IO = Output Current
RDC = Inductor resistance
VIN = Input Voltage
RHS = High-side MOSFET resistance
(6)
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Feature Description (continued)
f
SW (shift)
æ f div ö æ V OUTSC + R LS × I CL + R DC ´ I CL ö
= ç
÷ × ç
÷
è t ON ø è V IN - I CL ´ R HS + I CL ´ R LS ø
where
•
•
•
fdiv = Frequency divide (equals 1, 2, 4, or 8)
VOUTSC Output Voltage during short
ICL = Current Limit
(7)
7.3.10 Synchronization to RT/CLK Pin
The RT/CLK pin can be used to synchronize the regulator to an external system clock. To implement the
synchronization feature connect a square wave to the RT/CLK pin through one of the circuit networks shown in
Figure 19. The square wave amplitude must extend lower than 0.5 V and higher than 1.8 V on the RT/CLK pin
and have high and low states greater than 40 ns. The synchronization frequency range is 300 kHz to 1100 kHz.
The rising edge of the PH is synchronized to the falling edge of RT/CLK pin signal. The external synchronization
circuit should be designed in such a way that the device has the default frequency set resistor connected from
the RT/CLK pin to ground should the synchronization signal turn off. TI recommends using a frequency set
resistor connected as shown in Figure 19 through another resistor (for example, 50 Ω) to ground for clock signal
that are not Hi-Z or tristate during the OFF-state. The sum of the resistance should set the switching frequency
close to the external CLK frequency. TI recommends to AC couple the synchronization signal through a 10-pF
ceramic capacitor to RT/CLK pin. The first time the CLK is pulled above the CLK threshold the device switches
from the RT resistor frequency to PLL mode. The internal 0.5-V voltage source is removed and the CLK pin
becomes high impedance as the PLL starts to lock onto the external signal. The switching frequency can be
higher or lower than the frequency set with the RT/CLK resistor. The device transitions from the resistor mode to
the PLL mode and lock onto the CLK frequency within 100 microseconds. When the device transitions from the
PLL mode to the resistor mode, the switching frequency reduces from the external CLK frequency to 150 kHz,
then reapply the 0.5-V voltage source and the resistor then sets the switching frequency. The switching
frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 V on VSENSE pin. The device
implements a digital frequency shift to enable synchronizing to an external clock during normal start-up and fault
conditions.
TPS54061
TPS54061
RT/CLK
RT/CLK
PLL
PLL
RT
Hi -Z
Clock
Source
Clock
Source
RT
Figure 19. Synchronizing to a System Clock
7.3.11 Overvoltage Protection
The TPS54061-Q1 incorporates an output overvoltage transient protection (OVP) circuit to minimize voltage
overshoot when recovering from output fault conditions or strong unload transients on power supply designs with
low value output capacitance. For example, when the power supply output is overloaded the error amplifier
compares the actual output voltage to the internal reference voltage. If the VSENSE pin voltage is lower than the
internal reference voltage for a considerable time, the output of the error amplifier responds by clamping the error
amplifier output to a high voltage. Thus, requesting the maximum output current. Once the condition is removed,
the regulator output rises and the error amplifier output transitions to the steady-state duty cycle. In some
applications, the power supply output voltage can respond faster than the error amplifier output can respond, this
actuality leads to the possibility of an output overshoot.
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Feature Description (continued)
The OVP feature minimizes the output overshoot when using a low value output capacitor by comparing the
VSENSE pin voltage to OVP threshold which is 109% of the internal voltage reference. If the VSENSE pin
voltage is greater than the OVP threshold, the high-side MOSFET is disabled to minimize output overshoot.
When the VSENSE voltage drops lower than the OVP threshold, the high-side MOSFET resumes normal
operation.
7.3.12 Thermal Shutdown
The device implements an internal thermal shutdown until the junction temperature exceeds 176°C. The thermal
shutdown forces the device to stop switching until the junction temperature falls below the thermal trip threshold.
Once the die temperature decreases below 176°C, the device reinitiates the power up sequence by restarting the
internal slow-start.
7.4 Device Functional Modes
7.4.1 Operation Near Minimum Input Voltage
The TPS54061-Q1 is recommended to operate with input voltages above 4.7 V. The typical VIN UVLO threshold
is 4.5 V and the device may operate at input voltages down to the UVLO voltage. At input voltages below the
actual UVLO voltage the device does not switch. If EN is floating or externally pulled up to greater up than the
typical 1.23-V rising threshold, when V(VIN) passes the UVLO threshold the TPS54061-Q1 becomes active.
Switching is enabled and the slow-start sequence is initiated. The TPS54061-Q1 starts linearly ramping up the
internal reference DAC from 0 V to the reference voltage over the internal slow-start time period set by the
switching frequency.
7.4.2 Operation With Enable Control
The enable start threshold voltage is 1.23 V typical. With EN held below the 1.23-V typical rising threshold
voltage, the TPS54061-Q1 is disabled and switching is inhibited even if VIN is above its UVLO threshold. The
quiescent current is reduced in this state. If the EN voltage is increased above the rising threshold voltage while
V(VIN) is above the UVLO threshold, the device becomes active. Switching is enabled and the slow-start
sequence is initiated. The TPS54061-Q1 starts linearly ramping up the internal reference DAC from 0 V to the
reference voltage over the internal slow-start time period set by the switching frequency. If EN is pulled below the
1.18-V typical falling threshold, the TPS54061-Q1 enters the reduced quiescent current state again.
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8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS54061-Q1 is a 60-V, 200-mA step-down regulator with an integrated high-side and low-side MOSFET.
This device is typically used to convert a higher DC voltage to a lower DC voltage with a maximum available
output current of 200 mA. Example applications are: Low-Power Standby or Bias Voltage Supplies or an efficient
replacement for high-voltage linear regulator. Use the following design procedure to select component values for
the TPS54061-Q1. This procedure shows the design of a high-frequency switching regulator. These calculations
can be done with the aid of the TPS54x40/x60 design calculator (SLVC431). Alternatively, use the WEBENCH
software to generate a complete design. The WEBENCH software uses an iterative design procedure and
accesses a comprehensive database of components when generating a design.
8.2 Typical Applications
8.2.1 Continuous Conduction Mode Application
C BOOT
0.01 μ F
LO
U1
TPS54061
8 V to 60 V
R UVLO1
C IN
196 kΩ
2.2 μ F
R UVLO2
1
8
2
7
3
6
4
3.3 V 200 mA
2
1
R COMP
R HS
CO
5
RT
36.5 kΩ
100 μ H
31.6 kΩ
26.1 kΩ
C POLE
C COMP
33 pF
4700 pF
10 μ F
R LS
143 kΩ
10 kΩ
* See Enable and Adjusting Undervoltage Lockout.
Figure 20. CCM Application Schematic
8.2.1.1 Design Requirements
This example details the design of a continuous conduction mode (CCM) switching regulator design using
ceramic output capacitors. If a low output current design is needed, see Discontinuous Conduction Mode
Application. A few parameters must be known in order to start the design process. These parameters are
typically determined at the system level. For this example, the following known parameters are listed in Table 1.
Table 1. Design Parameters
PARAMETER
VALUE
Output voltage, VOUT
5.0 V
Transient response 50 to 150-mA load step
ΔVOUT = 4%
Maximum output current
200 mA
Input voltage, VIN
24 V nominal, 8 V to 60 V
Output voltage ripple
0.5% of VOUT
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Typical Applications (continued)
Table 1. Design Parameters (continued)
PARAMETER
VALUE
Start input voltage (rising VIN)
7.50 V
Stop input voltage (falling VIN)
6.50 V
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Selecting the Switching Frequency
The first step is to decide on a switching frequency for the regulator. Typically, the user wants to choose the
highest switching frequency possible because this produces the smallest solution size. The high switching
frequency allows for lower valued inductors and smaller output capacitors compared to a power supply that
switches at a lower frequency. The switching frequency is limited by the minimum on-time of the internal power
switch, the maximum input voltage, the output voltage and the frequency shift limitation.
Equation 6 and Equation 7 must be used to find the maximum switching frequency for the regulator, choose the
lower value of the two results. Switching frequencies higher than these values results in pulse skipping or a lack
of overcurrent protection during short circuit conditions. The typical minimum ON-time, tonmin, is 120 ns for the
TPS54061-Q1. To ensure overcurrent runaway does not occur during short circuits in your design, use
Equation 7 to determine the maximum switching frequency. With a maximum input voltage of 60 V, inductor
resistance of 0.77 Ω, high-side switch resistance of 3.0 Ω, low-side switch resistance of 1.5 Ω, a current limit
value of 350 mA, and a short circuit output voltage of 0.1 V, the maximum switching frequency is 524 kHz and
1003 kHz in each case respectively. A switching frequency of 400 kHz is used. To determine the timing
resistance for a given switching frequency, use Equation 5. The switching frequency is set by resistor RT shown
in Figure 20. RT is calculated to be 142 kΩ. A standard value of 143 kΩ is used.
8.2.1.2.2 Output Inductor Selection (LO)
To calculate the minimum value of the output inductor, use Equation 8. KIND is a coefficient that represents the
amount of inductor ripple current relative to the maximum output current. The inductor ripple current is filtered by
the output capacitor. Therefore, choosing high inductor ripple currents impacts the selection of the output
capacitor because the output capacitor must have a ripple current rating equal to or greater than the inductor
ripple current. In general, the inductor ripple value is at the discretion of the designer; however, the following
guidelines may be used. TI typically recommends using KIND values in the range of 0.2 to 0.4; however, for
designs using low ESR output capacitors such as ceramics and low output currents, a KIND value as high as 1
may be used. In a wide input voltage regulator, it is best to choose an inductor ripple current on the larger side.
This allows the inductor to still have a measurable ripple current with the input voltage at its minimum. For this
design example, use KIND of 0.4 and the minimum inductor value is calculated to be 97 µH. For this design, a
standard 100-µH value was chosen. It is important that the RMS current and saturation current ratings of the
inductor not be exceeded. The RMS and peak inductor current can be found from Equation 10 and Equation 11.
For this design, the RMS inductor current is 200 mA and the peak inductor current is 239 mA. The chosen
inductor is a Würth 74408943101. It has a saturation current rating of 680 mA and an RMS current rating of
520 mA. Equation 8 through Equation 11 show lower ripple currents reduce the output voltage ripple of the
regulator but require a larger value of inductance. Selecting higher ripple currents increases the output voltage
ripple of the regulator but allow for a lower inductance value. The current flowing through the inductor is the
inductor ripple current plus the average output current. During power up, faults or transient load conditions, the
inductor current can increase above the peak inductor current level calculated above. In transient conditions, the
inductor current can increase up to the switch current limit of the device. For this reason, the most conservative
approach is to specify an inductor with a saturation current rating equal to or greater than the switch current limit
rather than the calculated peak inductor current.
V max - VOUT
VOUT
´
LO min ³ IN
Kind ´ IO
VIN max ´ ¦ sw
(8)
IRIPPLE ³
VOUT ´
(VINmax
- VOUT )
VINmax ´ LO ´ fSW
(9)
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IL rms = IO
2
IL peak = IOUT
www.ti.com
æ VOUT ´ (VINmax - VOUT ) ö
1
+
´ ç
÷÷
ç
12
VINmax ´ LO ´ fSW
è
ø
2
I
+ RIPPLE
2
(10)
(11)
8.2.1.2.3 Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the most stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current until the regulator increases the inductor current. This situation occurs if there are
desired hold-up times for the regulator where the output capacitor must hold the output voltage above a certain
level for a specified amount of time after the input power is removed. The regulator also is temporarily not able to
supply sufficient output current if there is a large, fast increase in the current needs of the load such as
transitioning from no load to a full load. The regulator usually needs two or more clock cycles for the control loop
to see the change in load current and output voltage and adjust the duty cycle to react to the change. The output
capacitor must be sized to supply the extra current to the load until the control loop responds to the load change.
The output capacitance must be large enough to supply the difference in current for 2 clock cycles while only
allowing a tolerable amount of droop in the output voltage. Equation 15 shows the minimum output capacitance
necessary to accomplish this, where ΔIout is the change in output current, ƒsw is the regulators switching
frequency and ΔVOUT is the allowable change in the output voltage.
For this example, the transient load response is specified as a 4% change in VOUT for a load step from 50 mA to
150 mA. For this example, ΔIOUT = 0.150 – 0.05 = 0.10 and ΔVOUT = 0.04 × 3.3 = 0.132.
Using these values gives a minimum capacitance of 3.79 µF. This does not take the ESR of the output capacitor
into account in the output voltage change. For ceramic capacitors, the ESR is usually small enough to ignore in
this calculation. Aluminum electrolytic and tantalum capacitors have higher ESR that should be taken into
account.
The low-side FET of the regulator emulates a diode so it can not sink current so any stored energy in the
inductor produces an output voltage overshoot when the load current rapidly decreases, as in Figure 28. The
output capacitor must also be sized to absorb energy stored in the inductor when transitioning from a high load
current to a lower load current. The excess energy that gets stored in the output capacitor increases the voltage
on the capacitor. The capacitor must be sized to maintain the desired output voltage during these transient
periods. Equation 14 is used to calculate the minimum capacitance input the output voltage overshoot to a
desired value, where LO is the value of the inductor, IOH is the output current under heavy load, IOL is the output
under light load, VOUT + ΔVOUT is the final peak output voltage, and VI is the initial capacitor voltage. For this
example, the worst case load step is from 150 mA to 50 mA. The output voltage increases during this load
transition and must be limited to 4% of the output voltage to satisy the design goal. This makes VOUT+ΔVOUT =
1.04 × 3.3 = 3.432 V. VOUT is the initial capacitor voltage which is the nominal output voltage of 3.3 V. Using
these numbers in Equation 14 yields a minimum capacitance of 2.25 µF.
Equation 13 calculates the minimum output capacitance needed to meet the output voltage ripple specification,
where fSW is the switching frequency, VRIPPLE is the maximum allowable output voltage ripple, and IRIPPLE is the
inductor ripple current. Equation 13 yields 1.48 µF. Equation 16 calculates the maximum ESR an output
capacitor can have to meet the output voltage ripple specification. Equation 16 indicates the ESR should be less
than 0.160 Ω.
The most stringent criteria for the output capacitor is 3.79 µF of capacitance to maintain the output voltage
regulation during an load transient.
Additional capacitance de-ratings for aging, temperature, and DC bias increases this minimum value. For this
example, a 100-µF, 10-V X5R ceramic capacitor with 0.003 Ω of ESR if a 1206 package is used.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the Root Mean Square (RMS) value of the maximum ripple current.
Equation 12 can be used to calculate the RMS ripple current the output capacitor needs to support. For this
example, Equation 12 yields 10.23 mA.
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ICOrms =
CO 1 ³
æ VOUT ´ (VINmax - VOUT ) ö
´ ç
÷÷
ç
VINmax ´ LO ´ fSW
12
è
ø
1
(12)
æ
ö
IRIPPLE
1
´ ç
÷
VRIPPLE
è 8 ´ fSW ø
CO 2 ³ LO ´
IOH2
(VOUT
(13)
- IOL
2
+ DVOUT )
2
- VOUT 2
(14)
DIOUT
2
CO 3 ³
´
DVOUT f sw
RC
(15)
V
£ RIPPLE
IRIPPLE
(16)
8.2.1.2.4 Input Capacitor
The TPS54061-Q1 requires a high-quality ceramic, type X5R or X7R, input decoupling capacitor of at least
1-µF of effective capacitance. The effective capacitance includes any deration for DC bias effects. The voltage
rating of the input capacitor must be greater than the maximum input voltage. The capacitor must also have an
RMS current rating greater than the maximum RMS input current. The input RMS current can be calculated using
Equation 17. The value of a ceramic capacitor varies significantly over temperature and the dc bias applied to the
capacitor. The capacitance variations with temperature can be minimized by selecting a dielectric material that is
stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The effective value
of a capacitor decreases as the DC bias across a capacitor increases. For this example design, a ceramic
capacitor with at least a 60-V voltage rating is required to support the maximum input voltage. The input
capacitance value determines the input ripple voltage of the regulator. The input voltage ripple can be calculated
by rearranging Equation 18.
Using the design example values, IOUTmax = 200 mA, CIN = 2.2 µF, ƒSW = 400 kHz, yields an input voltage ripple
of 56.8 mV and an RMS input ripple current of 98.5 mA.
ICINrms = IOUT ´
CIN ³
VOUT
´
VINmin
(VINmin
- VOUT )
VINmin
(17)
æ 0.25 ö
IO
´ ç
÷
VINripple
è fSW ø
(18)
8.2.1.2.5 Bootstrap Capacitor Selection
A 0.01-µF ceramic capacitor must be connected between the BOOT and PH pins for proper operation. TI
recommends using a ceramic capacitor with X5R or better grade dielectric. The capacitor should have 10-V or
higher voltage rating.
8.2.1.2.6 Undervoltage Lockout Set Point
The Undervoltage Lockout (UVLO) can be adjusted using an external voltage divider on the EN pin of the
TPS54061-Q1. The UVLO has two thresholds: one for power up when the input voltage is rising and one for
power down or brown outs when the input voltage is falling. For the example design, the supply should turn on
and start switching once the input voltage increases above 7.50 V (enabled). After the regulator starts switching,
it should continue to do so until the input voltage falls below 6.50 V (UVLO stop). The programmable UVLO and
enable voltages are set by connecting resistor divider between VIN and ground to the EN pin. Equation 2 and
Equation 3 can be used to calculate the resistance values necessary. For example, a 196-kΩ resistor between
VIN and EN and a 36.5-kΩ resistor between EN and ground are required to produce the 7.50-V and 6.50-V start
and stop voltages. See Enable and Adjusting Undervoltage Lockout for additional considerations in high input
voltage applications.
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8.2.1.2.7 Output Voltage and Feedback Resistors Selection
For the example design, 10 kΩ was selected for RLS. Using Equation 1, RHS is calculated as 31.46 kΩ. The
nearest standard 1% resistor is 31.6 kΩ.
8.2.1.2.8 Closing the Loop
There are several methods used to compensate DC–DC regulators. The method presented here is easy to
calculate and ignores the effects of the slope compensation that is internal to the device. Because the slope
compensation is ignored, the actual cross over frequency is usually lower than the crossover frequency used in
the calculations. This method assume the crossover frequency is between the modulator pole and the ESR zero
and the ESR zero is at least 10 times greater the modulator pole.
To get started, the modulator pole, fpole, and the ESR zero, fzero must be calculated using Equation 19 and
Equation 20. For COUT, use a derated value of 6.0 µF. Use Equation 21 and Equation 22 to estimate a starting
point for the crossover frequency, fco, to design the compensation. For the example design, fpole is 1015 Hz and
fzero is 5584 kHz.
Equation 21 is the geometric mean of the modulator pole and the ESR zero and Equation 22 is the mean of
modulator pole and the switching frequency. Equation 21 yields 119.2 kHz and Equation 22 gives 17.9 kHz. Use
a frequency near the lower value of Equation 21 or Equation 22 for an initial crossover frequency.
For this example, fco of 17.9 kHz is used. Next, the compensation components are calculated. A resistor in
series with a capacitor is used to create a compensating zero. A capacitor in parallel to these two components
forms the compensating pole.
To determine the compensation resistor, RCOMP, use Equation 23. Assume the power stage transconductance,
gmps, is 1.00 A/V. The output voltage, VOUT, reference voltage, VREF, and amplifier transconductance, gmea, are
3.3 V, 0.8 V and 108 µA/V, respectively.
RCOMP is calculated to be 25.9 kΩ, use the nearest standard value of 26.1 kΩ. Use Equation 24 to set the
compensation zero equal to the modulator pole frequency. Equation 24 yields a 3790 pF for capacitor CCOMP and
a 4700 pF is chosen. Use the larger value of Equation 25 and Equation 26 to calculate the CPOLE value to set the
compensation pole. Equation 26 yields 30.5 pF so the nearest standard of 33 pF is selected.
1
f pole(Hz) =
VOUT
´ Co ´ 2 ´ p
Io
(19)
1
f zero(Hz) =
RC ´ CO ´ 2 ´ p
(20)
0.5
f co1(Hz) = ( f zero ´ f pole)
(21)
0.5
æ f sw
ö
f co2(Hz) = ç
´ f pole ÷
2
è
ø
RCOMP =
(22)
2 ´ p ´ ¦ CO ´ CO
VOUT
´
gmps
VREF ´ gmea
1
C5 =
2 ´ p ´ R4 ´ fPOLE
(24)
R ´ CO
C6 = C
R4
1
C6 =
R4 ´ fSW ´ p
20
(23)
(25)
(26)
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100
100
90
90
80
80
70
70
60
50
40
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
VIN = 60 V
30
20
10
0
0
0.025
0.05
0.075 0.1 0.125
Load Current (A)
0.15
0.175
Efficiency (%)
Efficiency (%)
8.2.1.3 Application Curves
60
50
40
20
10
0
0.001
0.2
fsw = 400 kHz
90
80
80
70
70
60
50
40
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
VIN = 60 V
30
20
10
0.075 0.1 0.125
Load Current (A)
0.15
VOUT = 5 V
0.175
Efficiency (%)
Efficiency (%)
100
90
0.05
50
40
10
0
0.001
0.2
1
VOUT = 5 V
20
60
0
0
–60
–120
–40
Gain
Phase
–180
100K
Output Voltage Normalized (%)
120
–20
fsw = 400 kHz
0.25
Phase (º)
Gain (dB)
40
0.2
0.15
0.1
0.05
0
–0.05
–0.1
–0.15
–0.2
–0.25
0
10
IOUT = 200 mA
Figure 25. Gain vs Phase
0.1
Figure 24. Efficiency vs Output Current
180
10K
0.01
Load Current (A)
60
1K
Frequency (Hz)
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
VIN = 60 V
30
20
fsw = 400 kHz
100
fsw = 400 kHz
60
Figure 23. Efficiency vs Output Current
–60
10
1
Figure 22. Efficiency vs Output Current
100
0.025
0.1
VOUT = 3.3 V
Figure 21. Efficiency vs Output Current
0
0.01
Load Current (A)
VOUT = 3.3 V
0
VIN = 8 V
VIN = 12 V
VIN = 24 V
VIN = 36 V
VIN = 60 V
30
30
20
40
Input Voltage (V)
50
60
fsw = 400 kHz
Figure 26. Output Voltage vs Input Voltage
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Output Voltage Normalized (%)
0.50
0.40
0.30
IOUT = 100 mA /div
0.20
0.10
0
–0.10
–0.20
–0.30
–0.40
–0.50
0
0.025
0.05
VIN = 24 V
0.075 0.1 0.125
Load Current (A)
0.15
VOUT = 3.3 V
0.175
VOUT = 50 mV /div ac coupled
0.2
fsw = 400 kHz
500 µs /div
Figure 27. Output Voltage vs Output Current
Figure 28. Load Transient
VIN = 10 V /div
VIN = 10 V /div
VEN = 5 V /div
VOUT = 20 mV /div ac coupled
VOUT = 2 V /div
1 ms /div
5 ms /div
Figure 30. Start-Up With ENA
Figure 29. Line Transient
PH = 20 V /div
VIN = 10 V /div
Inductor Current = 200 mA /div
VEN = 2 V /div
VOUT = 2 V /div
VIN = 10 mV /div ac coupled
2 ms /div
2 µs /div
Figure 31. Start-Up With VIN
22
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Figure 32. Input Ripple in DCM
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PH = 20 V /div
PH = 20 V /div
Inductor Current = 200 mA /div
Inductor Current = 200 mA /div
VIN = 50 mV /div ac coupled
VIN = 10 mV /div ac coupled
2 µs /div
2 µs /div
Figure 33. Input Ripple in CCM
Figure 34. Input Ripple Skip
PH = 20 V /div
PH = 20 V /div
Inductor Current = 200 mA /div
Inductor Current = 200 mA/div
VOUT = 10 mV /div
VOUT = 10 mV /div ac coupled
2 µs /div
2 µs /div
Figure 35. Output Ripple in DCM
Figure 36. Output Ripple in CCM
PH = 20 V /div
Inductor Current = 200 mA /div ac coupled
VOUT = 20 mV /div ac coupled
2 µs /div
Figure 37. Output Ripple Skip
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8.2.2 Discontinuous Conduction Mode Application
It is most desirable to have a power supply that is efficient and has a fixed switching frequency at low output
currents. A fixed-frequency power supply has a predictable output voltage ripple and noise. Using a traditional
continuous conduction mode (CCM) design method to calculate the output inductor yields a large inductance for
a low output current supply. Using a CCM inductor results in a large sized supply or affects efficiency from the
large DC resistance, an alternative is to operate in discontinuous conduction mode (DCM). Use the procedure
below to calculate the components values for designing a power supply operating in discontinuous conduction
mode. The advantage of operating a power supply in DCM for low output current is the fixed switching frequency,
lower output inductance, and lower DC resistance on the inductor. Use the frequency shift and skip equations to
estimate the maximum switching frequency.
C BOOT
0.01 µF
LO
U1
220 µH
TPS54061
8 V to 40 V
R UVLO1
C IN
2.2 µF
8
2
7
3
6
4
255 kΩ
1
5.0 V
2
744 053 221
R COMP
R HS
CO
5
52.3 kΩ
35.7 kΩ
RT
R UVLO2
45.3 kΩ
1
C POLE
C COMP
220 pF
0.33 µF
1240 kΩ
22 µF
R LS
10 kΩ
Figure 38. DCM Application Schematic
8.2.2.1 Design Requirements
This example details the design of a low output current, fixed switching regulator design using ceramic output
capacitors. A few parameters must be known in order to start the design process. These parameters are typically
determined at the system level. For this example, follow the design parameters listed in Table 2.
Table 2. Design Parameters
PARAMETER
VALUE
Output voltage
5.0 V
Transient response 37.5 to 75-mA load step
ΔVOUT = 4%
Maximum output current
75 mA
Minimum output currert
1 mA
Input voltage
24 V nominal, 8 V to 40 V
Output voltage ripple
1 % of VOUT
Switching frequency
50 kHz
Start input voltage (rising VIN)
8V
Stop input voltage (falling VIN)
6.8 V
8.2.2.2 Detailed Design Procedure
The TPS54061-Q1 is designed for applications which require a fixed operating frequency and low output voltage
ripple at low output currents, thus, the TPS54061-Q1 does not have a pulse skip mode at light loads. Because
the device has a minimum controllable ON-time, there is an output current at which the power supply pulse skips.
To ensure that the supply does not pulse skip at output current of the application the inductor value needs to be
selected greater than a minimum value. The minimum inductance needed to maintain a fixed switching frequency
24
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at the minimum load is calculated to be 227 µH using Equation 27. Because the equation is ideal and was
derived without losses, assume the minimum controllable light load ON-time, tonminll, is 180 ns. To maintain DCM
operation, the inductor value and output current need to stay below a maximum value. The maximum inductance
is calculated to be 250 µH using Equation 28. A 744053221 inductor from Würth Elektronik is selected. If CCM
operation is necessary, use the previous design procedure.
Use Equation 29, to make sure the minimum current limit on the high-side power switch is not exceeded at the
maximum output current. The peak current is calculated as 244 mA and is lower than the 350-mA current limit.
To determine the RMS current for the inductor and output capacitor, it is necessary to calculate the duty cycle.
The duty cycle, D1, for a step down regulator in DCM is calculated in Equation 30. D1 is the portion of the
switching cycle the high-side power switch is on, and is calculated to be 0.1345. D2 is the portion of the
switching cycle the low-side power switch is on, and is calculated to be 0.5111.
Using the Equation 32 and Equation 33, the RMS current of the inductor and output capacitor are calculated, to
be 0.1078 A and 0.0774 A, respectively. Select components that ratings exceed the calculated RMS values.
Calculate the output capacitance using the Equation 34 to Equation 36 and use the largest value. VRIPPLE is the
steady-state voltage ripple and ΔV is voltage change during a transient. A minimum of 7.5-µF capacitance is
calculated. Additional capacitance de-ratings for aging, temperature, and DC bias should be factored in, which
increases this minimum value. For this example, a 22-µF, 10-V X7R ceramic capacitor with 5-mΩ ESR is used.
To have a low output ripple power supply use a low ESR capacitor. Use Equation 37 to estimate the maximum
ESR for the output capacitor. Equation 38 and Equation 39 estimate the RMS current and capacitance for the
input capacitor. An RMS current of 38.7 mA and capacitance of 1.56 µF is calculated. A 2.2-µF, 100-V X7R
ceramic is used for this example.
æ V max - VOUT ö
t Onmin2
æ VINmax ö
x f sw
Lomin ³ ç IN
´
÷ ´ ç
÷
VOUT
2
IOmin
è
ø
è
ø
(27)
æ VOUT ö
1
æ V min - VOUT ö
LOmax £ ç IN
÷ ´
÷ ´ ç
f sw ´ IO
2
è
ø
è VINmin ø
(28)
æ 2 ´ VOUT ´ Iomax ´ (VINmax - VOUT ) ö
IL peak = ç
÷÷
ç
VINmax ´ LO ´ f sw
è
ø
æ 2 ´ VOUT ´ IO ´ LO ´ f sw
D1 = ç
ç
VIN ´ (VIN - VOUT )
è
ö
÷÷
ø
0.5
(29)
0.5
(30)
æ V - VOUT ö
D2 = ç IN
÷ ´ D1
VOUT
è
ø
(31)
æ D1 + D2 ö
IL rms = IL peak ´ ç
÷
3
è
ø
0.5
(32)
æ æ D1 + D2 ö
ICOrms = IL peak ´ ç ç
÷ çè
3
ø
è
(D1 + D2 )2 ö÷
4
0.5
÷
ø
(33)
æ D1 + D2 ö
I peak
CO 1 £ L
´ ç
÷
VRIPPLE
è 8 ´ fSW ø
CO 2 ³ LO ´
Co3 ³
RC
Io
(VOUT
2
- 0
2
+ DV )
(34)
2
- VOUT 2
(35)
IOUT
1
´
DVOUT fco
(36)
V
£ RIPPLE
IL peak
(37)
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æ
æ (D1)2
æ D1 ö
ç
ICINrms = IL peak ´ ç ç
ç è 3 ÷ø
ç 4
è
è
CIN ³
www.ti.com
öö
÷÷
÷÷
øø
0.5
(38)
æ 0.25 ö
IO
´ ç
÷
VINRIPPLE
è fSW ø
(39)
8.2.2.2.1 Closing the Feedback Loop
The method presented here is easy to calculate and includes the effect of the slope compensation that is internal
to the TPS54061-Q1. This method assumes the crossover frequency is between the modulator pole and the ESR
zero and the ESR zero is at least 10 times greater than the modulator pole. Once the output components are
determined, use the equations below to close the feedback loop. A current mode controlled power supply
operating in DCM has a transfer function which includes an ESR zero and pole as shown in Equation 40. To
calculate the current mode power stage gain, first calculate, Kdcm, the DCM gain, and Fm, the modulator gain,
using Equation 41 and Equation 42. Kdcm and Fm are 32.4 and 0.475, respectively. The location of the pole and
ESR zero are calculated using Equation 43 and Equation 44 . The pole and zero are 491 Hz and 2.8 MHz,
respectively. Use the lower value of Equation 45 and Equation 46 as a starting point for the crossover frequency.
Equation 45 is the geometric mean of the power stage pole and the ESR zero and Equation 46 is the mean of
power stage pole and the switching frequency. The crossover frequency is chosen as 5 kHz from Equation 46.
To determine the compensation resistor, RCOMP, use Equation 47. Assume the power stage transconductance,
gmps, is 1.0 A/V. The output voltage, VOUT, reference voltage, VREF, and amplifier transconductance, gmea, are
5.0 V, 0.8 V and 108 µA/V, respectively. RCOMP is calculated to be 38.3 kΩ; use the nearest standard value of
35.7 kΩ. Use Equation 48 to set the compensation zero to equal the modulator pole frequency. Equation 48
yields 290 nF for compensating capacitor CCOMP, and a 330 nF is used. Use the larger value of Equation 49 or
Equation 50 to calculate the CPOLE, which sets the compensation pole. Equation 50 yields 178-pF standard value
of 220 pF is selected.
s
1+
2 ´ p ´ f ZERO
Gdcm(s) » Fm ´ Kdcm ´
s
1+
2 ´ p ´ fPOLE
(40)
Kdcm =
(VIN
VOUT ´
2
´
D1
- VOUT )
æ
ö
ç
Rdc ÷÷
VIN ´ ç 2 +
- VOUT
VOUT ÷
ç
ç
IO ÷ø
è
gmps
Fm =
æ VIN - VOUT ö
ç
÷ + 0.380
è LO ´ f sw ø
V
æ
2 - OUT
ç
VIN
1
fPOLE (Hz) =
´ ç
VOUT
VOUT
ç
´ CO ´ 2 ´ p
ç1 - V
IO
IN
è
1
f ZERO (Hz) =
RC ´ CO ´ 2 ´ p
fCO1(Hz) =
fCO2 (Hz) =
RCOMP =
26
( f ZERO
( fSW
(41)
(42)
ö
÷
÷
÷
÷
ø
(43)
(44)
0.5
´ fPOLE )
(45)
0.5
´ fPOLE )
¦ co
Kdcm ´ Fm ´ ¦POLE
(46)
x
VOUT
VREF ´ gmea
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1
CCOMP =
CPOLE1
2 ´ p ´ RCOMP ´ Kdcm ´ Fm
(48)
R ´ CO
= C
RCOMP
(49)
1
RCOMP ´ fSW ´ p
CPOLE2 =
(50)
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
8.2.2.3 Application Curves
60
50
40
30
VIN
VIN
VIN
VIN
20
10
0
0
0.025
0.05
Load Current (A)
VOUT = 5 V
0.075
=
=
=
=
60
50
40
30
8V
12 V
24 V
36 V
10
0
0.001
0.1
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
100
60
50
40
V IN = 8 V
V IN = 12 V
V IN = 24 V
V IN = 36 V
10
0
0
0.025
0.05
Load Current (A)
VOUT = 3.3 V
0.075
0.1
fsw = 50 kHz
Figure 40. Efficiency vs Load Current
Figure 39. Efficiency vs Load Current
20
0.01
Load Current (A)
VOUT = 5 V
fsw = 50 kHz
30
V IN = 8 V
V IN = 12 V
V IN = 24 V
V IN = 36 V
20
0.1
fsw = 50 kHz
60
50
40
30
VIN
VIN
VIN
VIN
20
10
0
0.001
0.01
Load Current (A)
VOUT = 3.3 V
Figure 41. Efficiency vs Load Current
=8V
= 12 V
= 24 V
= 36 V
0.1
fsw = 50 kHz
Figure 42. Efficiency vs Load Current
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180
Gain
Phase 135
20
90
10
45
0
0
0.50
Phase (º)
Gain (dB)
30
–10
–45
–20
–90
–30
–135
–40
–180
100K
10
100
1K
Frequency (Hz)
10K
Output Voltage Normalized (%)
40
0.40
0.30
0.20
0.10
0.00
–0.10
–0.20
–0.30
–0.40
–0.50
0
0.025
VIN = 24 V
Figure 43. Frequency Response
0.05
Load Current (A)
0.075
VOUT = 5 V
0.1
fsw = 50 kHz
Figure 44. Output Voltage Normalized vs Load Current
Output Voltage Normalized (%)
0.25
VOUT = 100
100mV
mV/div
/divacac
coupled
coupled
0.2
0.15
0.1
0.05
0
–0.05
–0.1
–0.15
–0.2
–0.25
20mA/div
mA /div
IOUT = 20
0
10
20
50
40
30
Input Voltage (V)
IOUT = 37.5 mA
60
fsw = 50 kHz
2 ms /div
Figure 46. Load Transient
Figure 45. Output Voltage Normalized vs Input Voltage
VOUT = 100 mV /div ac coupled
VIN = 10 V /div
VOUT = 2 V /div
EN= 5 V /div
I OUT = 20 mA /div
10 ms /div
4 ms /div
Figure 47. Unload Transient
28
I OUT = 50 mA /div
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Figure 48. Start-Up With ENA
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VIN = 10 V /div
VIN = 10 V /div
VOUT = 2 V /div
VOUT = 2 V /div
EN = 5 V /div
EN = 5 V /div
IOUT = 50 mA /div
I OUT = 50 mA /div
10 ms /div
10 ms /div
Figure 49. Start-Up With VIN
Figure 50. Prebias Start-Up With ENA
PH = 20 V /div
VIN = 10 V /div
VIN = 100 nV /div ac coupled
VOUT = 2 V /div
VOUT = 50 mV /div ac coupled
EN = 5 V /div
I OUT = 50 mA /div
Inductor current = 100 mA /div
4 µ s /div
10 ms /div
Figure 52. Input and Output Ripple in DCM
spacer
Figure 51. Prebias Start-Up With VIN
spacer
PH = 20 V /div
VIN = 20 mV /div ac coupled
VOUT = 20 mV /div ac coupled
Inductor current = 20 mA /div
4 µs /div
Figure 53. Input and Output Ripple in PSM
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9 Power Supply Recommendations
The TPS54061-Q1 is designed to operate from an input voltage supply range between 4.7 V and 60 V. This input
supply should remain within the input voltage supply range. If the input supply is located more than a few inches
from the TPS54061-Q1 converter, bulk capacitance may be required in addition to the ceramic bypass
capacitors.
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. There are several signals paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade the power supplies performance. To help eliminate these problems, the VIN pin should be bypassed
to ground with a low ESR ceramic bypass capacitor with X5R or X7R dielectric. Take care to minimize the loop
area formed by the bypass capacitor connections, the VIN pin, and the GND pin. See Figure 54 for a PCB layout
example. Because the PH connection is the switching node and the output inductor must be located close to the
PH pin, the area of the PCB conductor is minimized to prevent excessive capacitive coupling. The RT/CLK pin is
sensitive to noise, so the RT resistor should be located as close as possible to the IC and routed with minimal
lengths of trace. The additional external components can be placed approximately as shown. It may be possible
to obtain acceptable performance with alternate PCB layouts; however, this layout has been shown to produce
good results and is meant as a guideline.
10.2 Layout Example
VOUT
Route Boot Capacitor
Trace on another layer to
provide wide path for
topside ground
GND
Input
Capacitor
Output
Inductor
Boot
Capacitor
BOOT
VIN
UVLO
Adjust
Resistor
Output
Capacitor
·
PH
VIN
GND
EN
COMP
RT/CLK
Frequency Set
Resistor
Compensation
Network
Feedback
Resistors
·
VSENSE
·
Signal VIA
Figure 54. PCB Layout Example
30
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
• For TPS54061-Q1 Design Excel Spreadsheet Tool, see SLVC431.
• For the WEBENCH Design Center, go to www.ti.com/WEBENCH.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation, see the following:
• Evaluation Module for TPS54061 Synchronous Step-Down SWIFT Converter (SLVU721)
• Using the TPS54061 to Create an Inverting Power Supply (SLVA524)
• Linear Versus Switching regulators in Industrial Applications With a 24-V Bus (SLYT527)
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
WEBENCH, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54061QDRBRQ1
ACTIVE
SON
DRB
8
2500
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 150
61Q1
TPS54061QDRBTQ1
ACTIVE
SON
DRB
8
250
RoHS & Green
NIPDAU
Level-3-260C-168 HR
-40 to 150
61Q1
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
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RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of