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TPS54260
SLVSA86D – MARCH 2010 – REVISED OCTOBER 2018
TPS54260 3.5-V to 60-V Input, 2.5-A, Step-Down Converter With Eco-mode™
1 Features
3 Description
•
•
•
The TPS54260 device is a 60-V, 2.5-A, step-down
regulator with an integrated high-side MOSFET.
Current mode control provides simple external
compensation and flexible component selection. A
low-ripple, pulse skip mode reduces the no load,
regulated output supply current to 138 μA. Using the
enable pin, shutdown supply current is reduced to
1.3 μA, when the enable pin is low.
1
•
•
•
•
•
•
•
•
•
•
3.5-V to 60-V Input Voltage Range
200-mΩ High-Side MOSFET
High Efficiency at Light Loads With a PulseSkipping Eco-mode™
138-μA Operating Quiescent Current
1.3-μA Shutdown Current
100-kHz to 2.5-MHz Switching Frequency
Synchronizes to External Clock
Adjustable Slow Start and Sequencing
UV and OV Powergood Output
Adjustable UVLO Voltage and Hysteresis
0.8-V Internal Voltage Reference
10-Pin MSOP and 10-Pin 3-mm × 3-mm VSON
With PowerPAD™ Packages
Create a Custom Design Using the TPS54260
With WEBENCH® Power Designer
2 Applications
•
•
12-V, 24-V and 48-V Industrial and Commercial
Low-Power Systems
GSM, GPRS Modules in Fleet Management, EMeters, and Security Systems
Undervoltage lockout (UVLO) is internally set at
2.5 V, but can be increased using the enable pin. The
output voltage startup ramp is controlled by the slowstart pin that can also be configured for sequencing
and tracking. An open-drain powergood signal
indicates the output is within 94% to 107% of its
nominal voltage.
A wide switching frequency range allows efficiency
and external component size to be optimized.
Frequency foldback and thermal shutdown protects
the part during an overload condition.
The TPS54260 is available in 10-pin thermally
enhanced MSOP and 10-pin 3-mm × 3-mm SON
PowerPAD packages.
Device Information(1)
PART NUMBER
TPS54260
PACKAGE
BODY SIZE (NOM)
HVSSOP (10)
3.00 mm × 3.00 mm
VSON (10)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
Efficiency vs Load Current
100
VIN
PWRGD
VIN
90
80
TPS54260
70
SS /TR
BOOT
PH
V OUT
RT /CLK
Efficiency - %
EN
60
50
40
30
COMP
VSENSE
VIN=12V
VOUT=3.3V
fsw=300kHz
20
10
GND
0
0
0.5
1.0
1.5
2.0
IO - Output Current - A
2.5
3.0
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54260
SLVSA86D – MARCH 2010 – REVISED OCTOBER 2018
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 11
7.1 Overview ................................................................. 11
7.2 Functional Block Diagram ....................................... 12
7.3 Feature Description................................................. 12
7.4 Device Functional Modes........................................ 28
8
Application and Implementation ........................ 29
8.1 Application Information............................................ 29
8.2 Typical Applications ............................................... 29
9 Power Supply Recommendations...................... 41
10 Layout................................................................... 41
10.1 Layout Guidelines ................................................. 41
10.2 Layout Example .................................................... 41
11 Device and Documentation Support ................. 42
11.1
11.2
11.3
11.4
11.5
11.6
Device Support ....................................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
42
42
42
42
42
43
12 Mechanical, Packaging, and Orderable
Information ........................................................... 43
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (May 2016) to Revision D
Page
•
Added top navigator icon for TI reference design .................................................................................................................. 1
•
Added links for WEBENCH ................................................................................................................................................... 1
•
Changed the minimum value for PH, 10-ns transient from –2 V to –5 V ............................................................................... 4
Changes from Revision B (December 2014) to Revision C
Page
•
Changed unit "A/V" to "S" throughout the document.............................................................................................................. 1
•
Changed the DGQ and DRC package images in Pin Configuration and Functions ............................................................. 3
•
Deleted 25°C from the Test conditions of Enable threshold voltage in the Electrical Characteristics ................................... 5
•
Changed Figure 9................................................................................................................................................................... 7
•
Changed TPS54360 To: TPS54260 in Figure 27................................................................................................................. 15
•
Changed TPS54360 To: TPS54260 in Figure 28 ............................................................................................................... 15
•
Added text "Do not place a low-impedance..." and Figure 29 to the Enable and Adjusting Undervoltage Lockout section 16
•
Changed TPS54360 To: TPS54260 in Figure 42 ............................................................................................................... 23
•
Changed 350 µA/V To: 310 µS for gmea, and 10.5 A/V To: 10.5 S for gmps in Figure 46 ................................................... 25
•
Changed text From: "yields an input voltage ripple of 206 mV" To: "yields an input voltage ripple of 473 mV" in the
Input Capacitor section......................................................................................................................................................... 33
Changes from Revision A (December 2010) to Revision B
Page
•
Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device Functional
Modes, Application and Implementation section, Power Supply Recommendations section, Layout section, Device
and Documentation Support section, and Mechanical, Packaging, and Orderable Information section .............................. 1
•
Simplified front page schematic for clarity .............................................................................................................................. 1
•
Deleted Junction-to-ambient thermal resistance (custom board) from Thermal Information ................................................. 5
•
Added "the level set" to Fixed Frequency PWM Control description ................................................................................... 12
2
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Changes from Original (March 2010) to Revision A
Page
•
Added 10-Pin 3mm × 3mm SON to Features and Description .............................................................................................. 1
•
Added SON package ............................................................................................................................................................. 3
•
Added DRC thermal data to thermal table and deleted dissipation rating table .................................................................... 5
5 Pin Configuration and Functions
DGQ Package
10-Pin HVSSOP
Top View
DRC Package
10-Pin VSON
Top View
BOOT
1
10
VIN
2
9
GND
EN
3
8
Thermal
PH
BOOT
1
COMP
VIN
2
Pad
10
Thermal
Pad
PH
9
GND
8
COMP
SS/TR
4
7
VSENSE
EN
3
RT/CLK
5
6
PWRGD
SS/TR
4
7
VSENSE
RT/CLK
5
6
PWRGD
Pin Functions
PIN
NAME
NO.
I/O
DESCRIPTION
BOOT
1
O
A bootstrap capacitor is required between BOOT and PH. If the voltage on this capacitor is below the
minimum required by the output device, the output is forced to switch off until the capacitor is refreshed.
VIN
2
I
Input supply voltage, 3.5 V to 60 V.
EN
3
I
Enable pin, internal pull-up current source. Pull below 1.2V to disable. Float to enable. Adjust the input
undervoltage lockout with two resistors.
SS/TR
4
I
Slow-start and Tracking. An external capacitor connected to this pin sets the output rise time. Since the
voltage on this pin overrides the internal reference, it can be used for tracking and sequencing.
RT/CLK
5
I
Resistor Timing and External Clock. An internal amplifier holds this pin at a fixed voltage when using an
external resistor to ground to set the switching frequency. If the pin is pulled above the PLL upper threshold,
a mode change occurs and the pin becomes a synchronization input. The internal amplifier is disabled and
the pin is a high impedance clock input to the internal PLL. If clocking edges stop, the internal amplifier is reenabled and the mode returns to a resistor set function.
PWRGD
6
O
An open drain output, asserts low if output voltage is low due to thermal shutdown, dropout, over-voltage or
EN shut down.
VSENSE
7
I
Inverting node of the transconductance (gm) error amplifier.
COMP
8
O
Error amplifier output, and input to the output switch current comparator. Connect frequency compensation
components to this pin.
GND
9
—
Ground
PH
10
O
The source of the internal high-side power MOSFET.
Thermal Pad
--
—
GND pin must be electrically connected to the exposed pad on the printed circuit board for proper operation.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating temperature range (unless otherwise noted) (1)
Input voltage
Output voltage
MIN
MAX
VIN
–0.3
65
EN
–0.3
5
VSENSE
–0.3
3
COMP
–0.3
3
PWRGD
–0.3
6
SS/TR
–0.3
3
RT/CLK
–0.3
3.6
BOOT-PH
–0.3
8
PH
–0.6
65
–5
65
–200
200
PH, 10-ns transient
Voltage difference
Source current
PAD to GND
V
V
mV
EN
100
μA
BOOT
100
mA
10
μA
100
μA
VSENSE
PH
Current Limit
A
RT/CLK
VIN
Sink current
UNIT
Current limit
A
COMP
100
μA
10
mA
PWRGD
200
μA
Operating junction temperature, TJ
SS/TR
–40
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001
(1)
UNIT
±2000
Charged-device model (CDM), per JEDEC specification JESD22-C101 (2)
V
±500
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VI
Input supply voltage
IO
Output current
TJ
Operating junction temperature
4
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MIN
MAX
3.5
60
UNIT
2.5
A
–40
150
°C
V
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SLVSA86D – MARCH 2010 – REVISED OCTOBER 2018
6.4 Thermal Information
TPS54260
THERMAL METRIC (1) (2)
DGQ (HVSOP)
DRC (VSON)
10 PINS
10 PINS
UNIT
62.5
40
°C/W
RθJA
Junction-to-ambient thermal resistance (standard board)
RθJC(top)
Junction-to-case (top) thermal resistance
83
65
°C/W
RθJB
Junction-to-board thermal resistance
28
8
°C/W
ψJT
Junction-to-top characterization parameter
1.7
0.6
°C/W
ψJB
Junction-to-board characterization parameter
20.1
7.5
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
21
7.8
°C/W
(1)
(2)
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
Power rating at a specific ambient temperature TA should be determined with a junction temperature of 150°C. This is the point where
distortion starts to substantially increase. See power dissipation estimate in application section of this data sheet for more information.
6.5 Electrical Characteristics
TJ = –40°C to 150°C, VIN = 3.5 to 60 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
Operating input voltage
3.5
Internal undervoltage lockout threshold No voltage hysteresis, rising and falling
60
2.5
V
V
Shutdown supply current
EN = 0 V, 25°C, 3.5 V ≤ VIN ≤ 60 V
1.3
4
Operating: nonswitching supply
current
VSENSE = 0.83 V, VIN = 12 V, 25°C
138
200
1.25
1.36
μA
ENABLE AND UVLO (EN PIN)
Enable threshold voltage
Input current
No voltage hysteresis, rising and falling
1.15
Enable threshold +50 mV
–3.8
Enable threshold –50 mV
–0.9
Hysteresis current
V
μA
–2.9
μA
VOLTAGE REFERENCE
Voltage reference
TJ = 25°C
0.792
0.8
0.808
0.784
0.8
0.816
V
HIGH-SIDE MOSFET
VIN = 3.5 V, BOOT-PH = 3 V
300
VIN = 12 V, BOOT-PH = 6 V
200
50
nA
Error amplifier transconductance (gM)
–2 μA < ICOMP < 2 μA, VCOMP = 1 V
310
μS
Error amplifier transconductance (gM)
during slow-start
–2 μA < ICOMP < 2 μA, VCOMP = 1 V,
VVSENSE = 0.4 V
70
μS
Error amplifier dc gain
VVSENSE = 0.8 V
10,000
V/V
2700
kHz
±27
μA
10.5
S
6.1
A
182
°C
On-resistance
410
mΩ
ERROR AMPLIFIER
Input current
Error amplifier bandwidth
Error amplifier source/sink
V(COMP) = 1 V, 100 mV overdrive
COMP to switch current
transconductance
CURRENT LIMIT
Current limit threshold
VIN = 12 V, TJ = 25°C
3.5
THERMAL SHUTDOWN
Thermal shutdown
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Electrical Characteristics (continued)
TJ = –40°C to 150°C, VIN = 3.5 to 60 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
2500
kHz
720
kHz
2200
kHz
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
Switching frequency range using RT
mode
fSW
Switching frequency
100
RT = 200 kΩ
450
Switching frequency range using CLK
mode
581
300
Minimum CLK input pulse width
40
RT/CLK high threshold
1.9
RT/CLK low threshold
0.5
RT/CLK falling edge to PH rising edge
delay
Measured at 500 kHz with RT resistor in series
PLL lock in time
Measured at 500 kHz
ns
2.2
V
0.7
V
60
ns
100
μs
2
μA
45
mV
SLOW-START AND TRACKING (SS/TR)
Charge current
VSS/TR = 0.4 V
SS/TR-to-VSENSE matching
VSS/TR = 0.4 V
SS/TR-to-reference crossover
98% nominal
1.15
V
SS/TR discharge current (overload)
VSENSE = 0 V, V(SS/TR) = 0.4 V
382
μA
SS/TR discharge voltage
VSENSE = 0 V
54
mV
VSENSE falling
92%
VSENSE rising
94%
VSENSE rising
109%
VSENSE falling
107%
Hysteresis
VSENSE falling
2%
Output high leakage
VSENSE = VREF, V(PWRGD) = 5.5 V, 25°C
10
On resistance
I(PWRGD) = 3 mA, VSENSE < 0.79 V
50
Minimum VIN for defined output
V(PWRGD) < 0.5 V, II(PWRGD) = 100 μA
POWERGOOD (PWRGD PIN)
VVSENSE
6
VSENSE threshold
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0.95
nA
Ω
1.5
V
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0.816
500
375
BOOT-PH = 3 V
Voltage Reference (V)
Static Drain-Source On-State Resistance (mW)
6.6 Typical Characteristics
250
BOOT-PH = 6 V
125
0
–50
–25
0
75
25
50
100
Junction Temperature (°C)
125
0.808
0.800
0.792
0.784
–50
150
–25
0
C001
VI = 12 V
75
25
50
100
Junction Temperature (°C)
125
150
C002
VI = 12 V
Figure 1. On Resistance vs Junction Temperature
Figure 2. Voltage Reference vs Junction Temperature
610
7.0
VI = 12 V
Switching Frequency (kHz)
600
Switch Current - A
6.5
6.0
5.5
590
580
570
560
5.0
-50
-25
0
25
50
75
100
125
550
–50
150
–25
0
TJ - Junction Temperature - °C
VI = 12 V
Figure 3. Switch Current Limit vs Junction Temperature
150
C004
Figure 4. Switching Frequency vs Junction Temperature
500
2000
400
Switching Frequency (kHz)
Switching Frequency (kHz)
125
VI = 12 V
RT = 200 kΩ
2500
1500
1000
500
0
0
75
25
50
100
Junction Temperature (°C)
25
50
75
100
125
150
RT/CLK Resistance (kW)
175
200
300
200
100
0
200
300
C005
VI = 12 V
TJ = 25°C
400
500 600 700 800 900 1000 1100 1200
RT/CLK Resistance (kW)
C006
VI = 12 V
TJ = 25°C
Figure 5. Switching Frequency vs RT/CLK Resistance HighFrequency Range
Figure 6. Switching Frequency vs RT/CLK Resistance LowFrequency Range
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Typical Characteristics (continued)
500
120
VI = 12 V
VI = 12 V
450
100
400
gm - mS
gm - mS
80
350
60
300
40
250
20
-50
-25
0
25
50
75
100
TJ - Junction Temperature - °C
125
200
-50
150
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
Figure 7. EA Transconductance During Slow-Start vs
Junction Temperature
Figure 8. EA Transconductance vs Junction Temperature
1.40
–3.25
I(EN) (μA)
EN Threshold (V)
–3.5
1.30
–3.75
1.20
–4
1.10
-50
-25
0
25
50
75
100
Junction Temperature (qC)
125
–4.25
–50
150
D001
–25
0
VI = 12 V
75
25
50
100
Junction Temperature (°C)
125
150
C010
VI = 12 V
VI(EN) = Theshold + 50 mV
Figure 9. EN Pin Voltage vs Junction Temperature
–1
–0.85
–1.5
I(SS/TR) (μA)
I(EN) (μA)
Figure 10. EN Pin Current vs Junction Temperature
–0.8
–0.9
–0.95
–2.5
–1
–50
–25
0
75
25
50
100
Junction Temperature (°C)
125
150
–3
–50
C011
VI = 12 V
VI(EN) = Theshold – 50 mV
–25
0
75
25
50
100
Junction Temperature (°C)
125
150
C012
VI = 12 V
Figure 11. EN Pin Current vs Junction Temperature
8
–2
Figure 12. SS/TR Charge Current vs Junction Temperature
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Typical Characteristics (continued)
100
575
VI = 12 V
80
% of Nominal fsw
II(SS/TR) - mA
500
425
350
60
40
20
275
0
200
-50
0
50
100
TJ - Junction Temperature - °C
0
150
0.2
0.4
VSENSE (V)
0.6
0.8
C014
VI = 12 V
TJ = 25°C
Figure 14. Switching Frequency vs VSENSE
Figure 13. SS/TR Discharge Current vs Junction
Temperature
1.5
1.5
I(VIN) (μA)
2
I(VIN) (μA)
2
1
0.5
1
0.5
0
0
–50
–25
0
75
25
50
100
Junction Temperature (°C)
125
0
150
10
20
C015
VI = 12 V
30
40
Input Voltage (V)
50
60
C016
TJ = 25°C
Figure 15. Shutdown Supply Current vs Junction
Temperature
Figure 16. Shutdown Supply Current vs Input Voltage (Vin)
210
190
170
VI = 12 V,
VI(VSENSE) = 0.83 V
o
TJ = 25 C,
VI(VSENSE) = 0.83 V
170
150
I(VIN) - mA
I(VIN) - mA
150
130
130
110
90
70
-50
110
0
50
100
TJ - Junction Temperature - °C
150
Figure 17. VIN Supply Current vs Junction Temperature
0
20
40
VI - Input Voltage - V
60
Figure 18. VIN Supply Current vs Input Voltage
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Typical Characteristics (continued)
80
110
PWRGD Threshold (% of Vref)
115
RDSON (W)
100
60
40
20
0
–50
–25
0
75
25
50
100
Junction Temperature (°C)
125
VSENSE Rising
105
VSENSE Falling
100
VSENSE Rising
95
90
VSENSE Falling
85
–50
150
–25
VI = 12 V
75
25
50
100
Junction Temperature (°C)
125
150
C020
VI = 12 V
3
2.25
2.75
VI(VIN) (V)
Figure 20. PWRGD Threshold vs Junction Temperature
2.5
VI(BOOT-PH) (V)
Figure 19. PWRGD On Resistance vs Junction Temperature
2
1.75
2.50
2.25
1.5
–50
2
–25
0
75
25
50
100
Junction Temperature (°C)
125
150
-50
VI = 12 V,
o
TJ = 25 C
125
150
C022
V(SS/TR) = 0.4 V
VI = 12 V
50
40
Offset - mV
Offset - mV
75
25
50
100
Junction Temperature (°C)
60
400
300
30
200
20
100
10
0
0
0
Figure 22. Input Voltage (UVLO) vs Junction Temperature
600
500
-25
C021
Figure 21. BOOT-PH UVLO vs Junction Temperature
100
200
300
400
500
600
700
800
0
-50
-25
0
25
50
75
100
125
150
TJ - Junction Temperature - °C
VSENSE - mV
Figure 23. SS/TR to VSENSE Offset vs VSENSE
10
0
C019
Figure 24. SS/TR to VSENSE Offset vs Temperature
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7 Detailed Description
7.1 Overview
The TPS54260 device is a 60-V, 2.5-A, step-down (buck) regulator with an integrated high-side N-channel
MOSFET. To improve performance during line and load transients the device implements a constant frequency,
current mode control which reduces output capacitance and simplifies external frequency compensation design.
The wide switching frequency of 100kHz to 2500kHz allows for efficiency and size optimization when selecting
the output filter components. The switching frequency is adjusted using a resistor to ground on the RT/CLK pin.
The device has an internal phase lock loop (PLL) on the RT/CLK pin that is used to synchronize the power
switch turn on to a falling edge of an external system clock.
The TPS54260 has a default start up voltage of approximately 2.5V. The EN pin has an internal pull-up current
source that can be used to adjust the input voltage under voltage lockout (UVLO) threshold with two external
resistors. In addition, the pull up current provides a default condition. When the EN pin is floating the device will
operate. The operating current is 138μA when not switching and under no load. When the device is disabled, the
supply current is 1.3μA.
The integrated 200mΩ high side MOSFET allows for high efficiency power supply designs capable of delivering
2.5 amperes of continuous current to a load. The TPS54260 reduces the external component count by
integrating the boot recharge diode. The bias voltage for the integrated high side MOSFET is supplied by a
capacitor on the BOOT to PH pin. The boot capacitor voltage is monitored by an UVLO circuit and will turn the
high side MOSFET off when the boot voltage falls below a preset threshold. The TPS54260 can operate at high
duty cycles because of the boot UVLO. The output voltage can be stepped down to as low as the 0.8V
reference.
The TPS54260 has a powergood comparator (PWRGD) which asserts when the regulated output voltage is less
than 92% or greater than 109% of the nominal output voltage. The PWRGD pin is an open drain output which
deasserts when the VSENSE pin voltage is between 94% and 107% of the nominal output voltage allowing the
pin to transition high when a pull-up resistor is used.
The TPS54260 minimizes excessive output overvoltage (OV) transients by taking advantage of the OV
powergood comparator. When the OV comparator is activated, the high-side MOSFET is turned off and masked
from turning on until the output voltage is lower than 107%.
The SS/TR (slow-start / tracking) pin is used to minimize inrush currents or provide power supply sequencing
during power-up. A small value capacitor should be coupled to the pin to adjust the slow-start time. A resistor
divider can be coupled to the pin for critical power supply sequencing requirements. The SS/TR pin is discharged
before the output powers up. This discharging ensures a repeatable restart after an over-temperature fault,
UVLO fault or a disabled condition.
The TPS54260, also, discharges the slow-start capacitor during overload conditions with an overload recovery
circuit. The overload recovery circuit will slow start the output from the fault voltage to the nominal regulation
voltage once a fault condition is removed. A frequency foldback circuit reduces the switching frequency during
start-up and overcurrent fault conditions to help control the inductor current.
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7.2 Functional Block Diagram
PWRGD
6
EN
3
VIN
2
Shutdown
UV
Thermal
Shutdown
Enable
Comparator
Logic
UVLO
Shutdown
Shutdown
Logic
OV
Enable
Threshold
Boot
Charge
Voltage
Reference
Boot
UVLO
Minimum
Clamp
Pulse
Skip
ERROR
AMPLIFIER
PWM
Comparator
VSENSE 7
Current
Sense
1 BOOT
Logic
And
PWM Latch
SS/TR 4
Shutdown
Slope
Compensation
10 PH
COMP 8
11 POWERPAD
Frequency
Shift
Overload
Recovery
Maximum
Clamp
Oscillator
with PLL
TPS54260 Block Diagram
9 GND
5
RT/CLK
Copyright © 2016, Texas Instruments Incorporated
7.3 Feature Description
7.3.1 Fixed Frequency PWM Control
The TPS54260 uses an adjustable fixed frequency, peak current mode control. The output voltage is compared
through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives
the COMP pin. An internal oscillator initiates the turn on of the high-side power switch. The error amplifier output
is compared to the high-side power switch current. When the power switch current reaches the level set by the
COMP voltage, the power switch is turned off. The COMP pin voltage will increase and decrease as the output
current increases and decreases. The device implements a current limit by clamping the COMP pin voltage to a
maximum level. The Eco-Mode is implemented with a minimum clamp on the COMP pin.
7.3.2 Slope Compensation Output Current
The TPS54260 adds a compensating ramp to the switch current signal. This slope compensation prevents subharmonic oscillations. The available peak inductor current remains constant over the full duty cycle range.
12
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Feature Description (continued)
7.3.3 Pulse-Skip Eco-Mode
The TPS54260 operates in a pulse-skip Eco-Mode at light-load currents to improve efficiency by reducing
switching and gate drive losses. The TPS54260 is designed so that if the output voltage is within regulation and
the peak switch current at the end of any switching cycle is below the pulse-skipping current threshold, the
device enters Eco-Mode. This current threshold is the current level corresponding to a nominal COMP voltage or
500mV.
When in Eco-Mode, the COMP pin voltage is clamped at 500 mV and the high-side MOSFET is inhibited. Further
decreases in load current or in output voltage can not drive the COMP pin below this clamp voltage level.
Since the device is not switching, the output voltage begins to decay. As the voltage control loop compensates
for the falling output voltage, the COMP pin voltage begins to rise. At this time, the high-side MOSFET is enabled
and a switching pulse initiates on the next switching cycle. The peak current is set by the COMP pin voltage. The
output voltage re-charges the regulated value, then the peak switch current starts to decrease, and eventually
falls below the Eco-Mode threshold at which time the device again enters Eco-Mode.
For Eco-Mode operation, the TPS54260 senses peak current, not average or load current, so the load current
where the device enters Eco-Mode is dependent on the output inductor value. For example, the circuit in
Figure 50 enters Eco-Mode at about 5 mA of output current. When the load current is low and the output voltage
is within regulation, the device enters a sleep mode and draws only 138-μA input quiescent current. The internal
PLL remains operating when in sleep mode. When operating at light-load currents in the pulse-skip mode, the
switching transitions occur synchronously with the external clock signal.
7.3.4 Low-Dropout Operation and Bootstrap Voltage (BOOT)
The TPS54260 has an integrated boot regulator, and requires a small ceramic capacitor between the BOOT and
PH pins to provide the gate drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when the
high-side MOSFET is off and the low-side diode conducts. The value of this ceramic capacitor should be 0.1 μF.
A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10 V or higher is recommended
because of the stable characteristics overtemperature and voltage.
To improve dropout, the TPS54260 is designed to operate at 100% duty cycle as long as the BOOT to PH pin
voltage is greater than 2.1 V. When the voltage from BOOT to PH drops below 2.1 V, the high-side MOSFET is
turned off using an UVLO circuit which allows the low-side diode to conduct and refresh the charge on the BOOT
capacitor. Since the supply current sourced from the BOOT capacitor is low, the high-side MOSFET can remain
on for more switching cycles than are required to refresh the capacitor, thus the effective duty cycle of the
switching regulator is high.
The effective duty cycle during dropout of the regulator is mainly influenced by the voltage drops across the
power MOSFET, inductor resistance, low-side diode and printed circuit board resistance. During operating
conditions in which the input voltage drops and the regulator is operating in continuous conduction mode, the
high-side MOSFET can remain on for 100% of the duty cycle to maintain output regulation, until the BOOT to PH
voltage falls below 2.1 V.
Pay attention in maximum duty cycle applications which experience extended time periods with light loads or no
load. When the voltage across the BOOT capacitor falls below the 2.1V UVLO threshold, the high-side MOSFET
is turned off, but there may not be enough inductor current to pull the PH pin down to recharge the BOOT
capacitor. The high-side MOSFET of the regulator stops switching because the voltage across the BOOT
capacitor is less than 2.1 V. The output capacitor then decays until the difference in the input voltage and output
voltage is greater than 2.1 V, at which point the BOOT UVLO threshold is exceeded, and the device starts
switching again until the desired output voltage is reached. This operating condition persists until the input
voltage and/or the load current increases. TI recommends adjusting the VIN stop voltage greater than the BOOT
UVLO trigger condition at the minimum load of the application using the adjustable VIN UVLO feature with
resistors on the EN pin.
The start and stop voltages for typical 3.3-V and 5-V output applications are shown in Figure 25 and Figure 26.
The voltages are plotted versus load current. The start voltage is defined as the input voltage needed to regulate
the output within 1%. The stop voltage is defined as the input voltage at which the output drops by 5% or stops
switching.
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Feature Description (continued)
4
5.6
3.8
5.4
3.6
Input Voltage (V)
Input Voltage (V)
During high duty cycle conditions, the inductor current ripple increases while the BOOT capacitor is being
recharged resulting in an increase in ripple voltage on the output. This is due to the recharge time of the boot
capacitor being longer than the typical high-side off-time when switching occurs every cycle.
Start
3.4
Stop
5.2
Start
5
Stop
3.2
4.8
3
4.6
0
0.05
0.10
Output Current (A)
0.15
0.20
0
C025
Figure 25. 3.3V Start / Stop Voltage
0.05
0.10
Output Current (A)
0.15
0.20
C026
Figure 26. 5.0V Start / Stop Voltage
7.3.5 Error Amplifier
The TPS54260 has a transconductance amplifier for the error amplifier. The error amplifier compares the
VSENSE voltage to the lower of the SS/TR pin voltage or the internal 0.8-V voltage reference. The
transconductance (gm) of the error amplifier is 310 μS during normal operation. During the slow-start operation,
the transconductance is a fraction of the normal operating gm. When the voltage of the VSENSE pin is below 0.8
V and the device is regulating using the SS/TR voltage, the gm is 70 μS.
The frequency compensation components (capacitor, series resistor and capacitor) are added to the COMP pin
to ground.
7.3.6 Voltage Reference
The voltage reference system produces a precise ±2% voltage reference over temperature by scaling the output
of a temperature stable bandgap circuit.
7.3.7 Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the VSENSE pin. TI recommends using
1% tolerance or better divider resistors. Start with a 10-kΩ for the R2 resistor and use the Equation 1 to calculate
R1. To improve efficiency at light loads consider using larger value resistors. If the values are too high, the
regulator will be more susceptible to noise and voltage errors from the VSENSE input current will be noticeable.
æ Vout - 0.8V ö
R1 = R2 ´ ç
÷
0.8 V
è
ø
(1)
7.3.8 Enable and Adjusting Undervoltage Lockout
The TPS54260 is disabled when the VIN pin voltage falls below 2.5 V. If an application requires a higher
undervoltage lockout (UVLO), use the EN pin as shown in Figure 27 to adjust the input voltage UVLO by using
the two external resistors. Though it is not necessary to use the UVLO adjust registers, for operation it is highly
recommended to provide consistent power-up behavior. The EN pin has an internal pullup current source, I1, of
0.9μA that provides the default condition of the TPS54260 operating when the EN pin floats. Once the EN pin
voltage exceeds 1.25 V, an additional 2.9 μA of hysteresis, Ihys, is added. This additional current facilitates input
voltage hysteresis. Use Equation 2 to set the external hysteresis for the input voltage. Use Equation 3 to set the
input start voltage.
14
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Feature Description (continued)
TPS54260
VIN
Ihys
I1
1.18 mA
R1
3.35 mA
+
R2
EN
-
1.25 V
Copyright © 2016, Texas Instruments Incorporated
Figure 27. Adjustable Undervoltage Lockout (UVLO)
V
- VSTOP
R1 = START
IHYS
R2 =
(2)
VENA
VSTART - VENA
+ I1
R1
(3)
Another technique to add input voltage hysteresis is shown in Figure 28. This method may be used, if the
resistance values are high from the previous method and a wider voltage hysteresis is needed. The resistor R3
sources additional hysteresis current into the EN pin.
TPS54260
VIN
Ihys
R1
I1
1.18 mA
3.35 mA
+
R2
EN
1.25 V
R3
-
VOUT
Copyright © 2016, Texas Instruments Incorporated
Figure 28. Adding Additional Hysteresis
R1 =
R2 =
VSTART - VSTOP
V
IHYS + OUT
R3
(4)
VENA
VSTART - VENA
V
+ I1 - ENA
R1
R3
(5)
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Feature Description (continued)
Do not place a low-impedance voltage source with greater than 5 V directly on the EN pin. Do not place a
capacitor directly on the EN pin if VEN > 5 V when using a voltage divider to adjust the start and stop voltage.
The node voltage, (see Figure 29) must remain equal to or less than 5.8 V. The zener diode can sink up to
100 µA. The EN pin voltage can be greater than 5 V if the VIN voltage source has a high impedance and does
not source more than 100 µA into the EN pin.
VIN
R1
Node
ENA
10 kW
R2
5.8V
Figure 29. Node Voltage
7.3.9 Slow-Start / Tracking Pin (SS/TR)
The TPS54260 effectively uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as
the power-supply's reference voltage and regulates the output accordingly. A capacitor on the SS/TR pin-toground implements a slow-start time. The TPS54260 has an internal pullup current source of 2 μA that charges
the external slow-start capacitor. The calculations for the slow-start time (10% to 90%) are shown in Equation 6.
The voltage reference (VREF) is 0.8 V and the slow-start current (ISS) is 2 μA. The slow-start capacitor should
remain lower than 0.47 μF and greater than 0.47 nF.
Tss(ms) ´ Iss(m A)
Css(nF) =
Vref (V) ´ 0.8
(6)
At power-up, the TPS54260 will not start switching until the slow-start pin is discharged to less than 40 mV to
ensure a proper power-up, see Figure 30.
Also, during normal operation, the TPS54260 will stop switching and the SS/TR must be discharged to 40 mV,
when the VIN UVLO is exceeded, EN pin pulled below 1.25 V, or a thermal shutdown event occurs.
The VSENSE voltage will follow the SS/TR pin voltage with a 45-mV offset up to 85% of the internal voltage
reference. When the SS/TR voltage is greater than 85% on the internal reference voltage the offset increases as
the effective system reference transitions from the SS/TR voltage to the internal voltage reference (see
Figure 23). The SS/TR voltage will ramp linearly until clamped at 1.7 V.
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Feature Description (continued)
EN
SS/TR
VSENSE
VOUT
Figure 30. Operation of SS/TR Pin when Starting
7.3.10 Overload Recovery Circuit
The TPS54260 has an overload recovery (OLR) circuit. The OLR circuit will slow start the output from the
overload voltage to the nominal regulation voltage once the fault condition is removed. The OLR circuit will
discharge the SS/TR pin to a voltage slightly greater than the VSENSE pin voltage using an internal pulldown of
382 μA when the error amplifier is changed to a high voltage from a fault condition. When the fault condition is
removed, the output will slow start from the fault voltage to nominal output voltage.
7.3.11 Sequencing
Many of the common power supply sequencing methods can be implemented using the SS/TR, EN and PWRGD
pins. The sequential method can be implemented using an open-drain output of a power-on reset pin of another
device. The sequential method is illustrated in Figure 31 using two TPS54260 devices. The powergood is
coupled to the EN pin on the TPS54260 which will enable the second power supply once the primary supply
reaches regulation. If needed, a 1-nF ceramic capacitor on the EN pin of the second power supply will provide a
1-ms start-up delay. Figure 32 shows the results of Figure 31.
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Feature Description (continued)
TPS54260
EN
PWRGD
EN
EN1
SS /TR
SS /TR
PWRGD1
PWRGD
VOUT1
VOUT2
Figure 31. Schematic for Sequential Start-Up Sequence
Figure 32. Sequential Start-Up using EN and PWRGD
TPS54160
TPS54260
3
EN
4
SS/TR
6
PWRGD
EN1, EN2
VOUT1
TPS54260
TPS54160
VOUT2
3
EN
4
SS/TR
6
PWRGD
Figure 33. Schematic for Ratiometric Start-Up Sequence
18
Figure 34. Ratiometric Start-Up using Coupled SS/TR
Pins
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Figure 33 shows a method for ratio-metric start-up sequence by connecting the SS/TR pins together. The
regulator outputs will ramp up and reach regulation at the same time. When calculating the slow-start time, the
pullup current source must be doubled in Equation 6. Figure 34 shows the results of Figure 33.
TPS54260
EN
VOUT 1
SS/TR
PWRGD
TPS54260
VOUT 2
EN
R1
SS/ TR
R2
PWRGD
R3
R4
Copyright © 2016, Texas Instruments Incorporated
Figure 35. Schematic for Ratio-Metric and Simultaneous Start-Up Sequence
Ratio-metric and simultaneous power supply sequencing can be implemented by connecting the resistor network
of R1 and R2 shown in Figure 35 to the output of the power supply that needs to be tracked or another voltage
reference source. Using Equation 7 and Equation 8, the tracking resistors can be calculated to initiate the Vout2
slightly before, after or at the same time as Vout1. Equation 9 is the voltage difference between Vout1 and Vout2
at the 95% of nominal output regulation.
The deltaV variable is zero volts for simultaneous sequencing. To minimize the effect of the inherent SS/TR to
VSENSE offset (Vssoffset) in the slow-start circuit and the offset created by the pullup current source (Iss) and
tracking resistors, the Vssoffset and Iss are included as variables in the equations.
To design a ratio-metric start-up in which the Vout2 voltage is slightly greater than the Vout1 voltage when Vout2
reaches regulation, use a negative number in Equation 7 through Equation 9 for deltaV. Equation 9 will result in a
positive number for applications which the Vout2 is slightly lower than Vout1 when Vout2 regulation is achieved.
Since the SS/TR pin must be pulled below 40 mV before starting after an EN, UVLO or thermal shutdown fault,
careful selection of the tracking resistors is needed to ensure the device will restart after a fault. Make sure the
calculated R1 value from Equation 7 is greater than the value calculated in Equation 10 to ensure the device can
recover from a fault.
As the SS/TR voltage becomes more than 85% of the nominal reference voltage the Vssoffset becomes larger
as the slow-start circuits gradually handoff the regulation reference to the internal voltage reference. The SS/TR
pin voltage needs to be greater than 1.3 V for a complete handoff to the internal voltage reference as shown in
Figure 23.
Vout2 + deltaV
Vssoffset
R1 =
´
VREF
Iss
(7)
VREF ´ R1
R2 =
Vout2 + deltaV - VREF
(8)
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deltaV = Vout1 - Vout2
R1 > 2800 ´ Vout1 - 180 ´ deltaV
(9)
(10)
EN
EN
VOUT1
VOUT1
VOUT2
Figure 36. Ratiometric Start-Up With Tracking Resistors
VOUT2
Figure 37. Ratiometric Start-Up With Tracking Resistors
EN
VOUT1
VOUT2
Figure 38. Simultaneous Start-Up With Tracking Resistor
7.3.12 Constant Switching Frequency and Timing Resistor (RT/CLK Pin)
The switching frequency of the TPS54260 is adjustable over a wide range from approximately 100 kHz to 2500
kHz by placing a resistor on the RT/CLK pin. The RT/CLK pin voltage is typically 0.5 V and must have a resistorto-ground to set the switching frequency. To determine the timing resistance for a given switching frequency, use
Equation 11 or the curves in Figure 39 or Figure 40. To reduce the solution size one would typically set the
switching frequency as high as possible, but tradeoffs of the supply efficiency, maximum input voltage and
minimum controllable on time should be considered.
The minimum controllable on time is typically 135 ns and limits the maximum operating input voltage.
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The maximum switching frequency is also limited by the frequency shift circuit. More discussion on the details of
the maximum switching frequency is located below.
206033
RT (k W) =
¦ sw (kHz )1.0888
(11)
500
2500
2000
Switching Frequency (kHz)
fs - Switching Frequency - kHz
VI = 12 V,
TJ = 25°C
1500
1000
500
0
0
25
50
75
100
125
150
RT/CLK - Clock Resistance - kW
175
200
Figure 39. Switching Frequency vs RT/CLK Resistance
High-Frequency Range
400
300
200
100
0
200
300
400
500 600 700 800 900 1000 1100 1200
RT/CLK Resistance (kW)
C006
Figure 40. Switching Frequency vs RT/CLK Resistance
Low-Frequency Range
7.3.13 Overcurrent Protection and Frequency Shift
The TPS54260 implements current mode control which uses the COMP pin voltage to turn off the high-side
MOSFET on a cycle-by-cycle basis. Each cycle the switch current and COMP pin voltage are compared, when
the peak switch current intersects the COMP voltage, the high-side switch is turned off. During overcurrent
conditions that pull the output voltage low, the error amplifier will respond by driving the COMP pin high,
increasing the switch current. The error amplifier output is clamped internally, which functions as a switch current
limit.
To increase the maximum operating switching frequency at high input voltages the TPS54260 implements a
frequency shift. The switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 volts on
VSENSE pin.
The device implements a digital frequency shift to enable synchronizing to an external clock during normal
startup and fault conditions. Since the device can only divide the switching frequency by 8, there is a maximum
input voltage limit in which the device operates and still have frequency shift protection.
During short-circuit events (particularly with high input voltage applications), the control loop has a finite minimum
controllable on time and the output has a low voltage. During the switch on time, the inductor current ramps to
the peak current limit because of the high input voltage and minimum on time. During the switch off time, the
inductor would normally not have enough off time and output voltage for the inductor to ramp down by the ramp
up amount. The frequency shift effectively increases the off time allowing the current to ramp down.
7.3.14 Selecting the Switching Frequency
The switching frequency that is selected should be the lower value of the two equations, Equation 12 and
Equation 13. Equation 12 is the maximum switching frequency limitation set by the minimum controllable on time.
Setting the switching frequency above this value will cause the regulator to skip switching pulses.
Equation 13 is the maximum switching frequency limit set by the frequency shift protection. To have adequate
output short circuit protection at high input voltages, the switching frequency should be set to be less than the
fsw(maxshift) frequency. In Equation 13, to calculate the maximum switching frequency one must take into
account that the output voltage decreases from the nominal voltage to 0 V, the fdiv integer increases from 1 to 8
corresponding to the frequency shift.
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In Figure 41, the solid line illustrates a typical safe operating area regarding frequency shift and assumes the
output voltage is 0 V, and the resistance of the inductor is 0.130 Ω, FET on resistance of 0.2 Ω and the diode
voltage drop is 0.5 V. The dashed line is the maximum switching frequency to avoid pulse skipping. Enter these
equations in a spreadsheet or other software or use the SwitcherPro design software to determine the switching
frequency.
æ 1 ö æ (IL ´ Rdc + VOUT + Vd) ö
fSW (max skip ) = ç
÷
÷ ´ çç
÷
è tON ø è (VIN - IL ´ Rhs + Vd) ø
(12)
fSW (shift ) =
fdiv æ (IL ´ Rdc + VOUTSC + Vd) ö
´ç
÷
tON çè (VIN - IL x Rhs + Vd) ÷ø
where
•
•
•
•
•
•
•
•
•
IL = inductor current
Rdc = inductor resistance
VIN = maximum input voltage
VOUT = output voltage
VOUTSC =output voltage during short
Vd = diode voltage drop
RDS(on) = switch on resistance
tON = controllable on time
ƒDIV = frequency divide equals (1, 2, 4, or 8)
(13)
Switching Frequency (kHz)
2500
2000
Shift
1500
Skip
1000
500
0
10
20
30
40
Input Voltage (V)
50
60
C027
Figure 41. Maximum Switching Frequency vs Input Voltage
7.3.15 How to Interface to RT/CLK Pin
The RT/CLK pin can be used to synchronize the regulator to an external system clock. To implement the
synchronization feature connect a square wave to the RT/CLK pin through the circuit network shown in
Figure 42. The square wave amplitude must transition lower than 0.5 V and higher than 2.2 V on the RT/CLK pin
and have an on time greater than 40 ns and an off-time greater than 40 ns. The synchronization frequency range
is 300 kHz to 2200 kHz. The rising edge of the PH will be synchronized to the falling edge of RT/CLK pin signal.
The external synchronization circuit should be designed in such a way that the device will have the default
frequency set resistor connected from the RT/CLK pin to ground should the synchronization signal turn off. It is
recommended to use a frequency set resistor connected as shown in Figure 42 through a 50-Ω resistor-toground. The resistor should set the switching frequency close to the external CLK frequency. TI recommends to
AC couple the synchronization signal through a 10-pF ceramic capacitor to RT/CLK pin and a 4-kΩ series
resistor. The series resistor reduces PH jitter in heavy-load applications when synchronizing to an external clock
and in applications which transition from synchronizing to RT mode. The first time the CLK is pulled above the
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CLK threshold the device switches from the RT resistor frequency to PLL mode. The internal 0.5-V voltage
source is removed and the CLK pin becomes high impedance as the PLL starts to lock onto the external signal.
Since there is a PLL on the regulator the switching frequency can be higher or lower than the frequency set with
the external resistor. The device transitions from the resistor mode to the PLL mode and then will increase or
decrease the switching frequency until the PLL locks onto the CLK frequency within 100 µs.
When the device transitions from the PLL to resistor mode the switching frequency will slow down from the CLK
frequency to 150 kHz, then reapply the 0.5-V voltage and the resistor will then set the switching frequency. The
switching frequency is divided by 8, 4, 2, and 1 as the voltage ramps from 0 to 0.8 V on VSENSE pin. The device
implements a digital frequency shift to enable synchronizing to an external clock during normal start-up and fault
conditions. Figure 43, Figure 44, and Figure 45 show the device synchronized to an external system clock in
continuous conduction mode (CCM) discontinuous conduction (DCM) and pulse-skip mode (PSM).
TPS54260
10 pF
4 kW
PLL
Rfset
EXT
Clock
Source
50 W
RT/CLK
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Figure 42. Synchronizing to a System Clock
PH
PH
EXT
EXT
IL
IL
Figure 43. Plot of Synchronizing in CCM
Figure 44. Plot of Synchronizing in DCM
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PH
EXT
IL
Figure 45. Plot of Synchronizing in PSM
7.3.16 Powergood (PWRGD Pin)
The PWRGD pin is an open-drain output. Once the VSENSE pin is between 94% and 107% of the internal
voltage reference the PWRGD pin is de-asserted and the pin floats. TI recommends using a pullup resistor
between the values of 10 and 100 kΩ to a voltage source that is 5.5 V or less. The PWRGD is in a defined state
once the VIN input voltage is greater than 1.5 V but with reduced current sinking capability. The PWRGD will
achieve full current sinking capability as VIN input voltage approaches 3 V.
The PWRGD pin is pulled low when the VSENSE is lower than 92% or greater than 109% of the nominal internal
reference voltage. Also, the PWRGD is pulled low, if the UVLO or thermal shutdown are asserted or the EN pin
pulled low.
7.3.17 Overvoltage Transient Protection
The TPS54260 incorporates an overvoltage transient protection (OVTP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients on power supply designs with low value
output capacitance. For example, when the power supply output is overloaded the error amplifier compares the
actual output voltage to the internal reference voltage. If the VSENSE pin voltage is lower than the internal
reference voltage for a considerable time, the output of the error amplifier will respond by clamping the error
amplifier output to a high voltage. Thus, requesting the maximum output current. Once the condition is removed,
the regulator output rises and the error amplifier output transitions to the steady state duty cycle. In some
applications, the power supply output voltage can respond faster than the error amplifier output can respond, this
actuality leads to the possibility of an output overshoot. The OVTP feature minimizes the output overshoot, when
using a low-value output capacitor, by implementing a circuit to compare the VSENSE pin voltage to OVTP
threshold which is 109% of the internal voltage reference. If the VSENSE pin voltage is greater than the OVTP
threshold, the high-side MOSFET is disabled preventing current from flowing to the output and minimizing output
overshoot. When the VSENSE voltage drops lower than the OVTP threshold, the high-side MOSFET is allowed
to turn on at the next clock cycle.
7.3.18 Thermal Shutdown
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 182°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal
trip threshold. Once the die temperature decreases below 182°C, the device reinitiates the power-up sequence
by discharging the SS/TR pin.
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7.3.19 Small Signal Model for Loop Response
Figure 46 shows an equivalent model for the TPS54260 control loop which can be modeled in a circuit simulation
program to check frequency response and dynamic load response. The error amplifier is a transconductance
amplifier with a gmEA of 310 μS. The error amplifier can be modeled using an ideal voltage controlled current
source. The resistor Ro and capacitor Co model the open loop gain and frequency response of the amplifier. The
1-mV AC voltage source between the nodes a and b effectively breaks the control loop for the frequency
response measurements. Plotting C/A shows the small signal response of the frequency compensation. Plotting
a/b shows the small signal response of the overall loop. The dynamic loop response can be checked by
replacing RL with a current source with the appropriate load step amplitude and step rate in a time domain
analysis. This equivalent model is only valid for continuous conduction mode designs.
PH
VO
Power Stage
gmps 10.5 S
a
b
RESR
R1
RL
COMP
c
0.8 V
R3
RO
C0
C2
COUT
VSENSE
gmea
R2
310 mS
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 46. Small Signal Model for Loop Response
7.3.20 Simple Small Signal Model for Peak Current Mode Control
Figure 47 describes a simple small signal model that can be used to understand how to design the frequency
compensation. The TPS54260 power stage can be approximated to a voltage-controlled current source (duty
cycle modulator) supplying current to the output capacitor and load resistor. The control to output transfer
function is shown in Equation 14 and consists of a DC gain, one dominant pole, and one ESR zero. The quotient
of the change in switch current and the change in COMP pin voltage (node C in Figure 46) is the power stage
transconductance. The gmPS for the TPS54260 is 10.5 S. The low-frequency gain of the power stage frequency
response is the product of the transconductance and the load resistance as shown in Equation 15.
As the load current increases and decreases, the low-frequency gain decreases and increases, respectively. This
variation with the load may seem problematic at first glance, but fortunately the dominant pole moves with the
load current (see Equation 16). The combined effect is highlighted by the dashed line in the right half of
Figure 47. As the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB
crossover frequency the same for the varying load conditions which makes it easier to design the frequency
compensation. The type of output capacitor chosen determines whether the ESR zero has a profound effect on
the frequency compensation design. Using high-ESR aluminum electrolytic capacitors may reduce the number
frequency compensation components needed to stabilize the overall loop because the phase margin increases
from the ESR zero at the lower frequencies (see Equation 17).
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VO
Adc
VC
RESR
fp
RL
gmps
COUT
fz
Figure 47. Simple Small Signal Model and Frequency Response for Peak Current Mode Control
æ
s
ç1 +
2p ´ fZ
VOUT
= Adc ´ è
VC
æ
s
ç1 +
2p ´ fP
è
Adc = gmps ´ RL
ö
÷
ø
ö
÷
ø
(14)
(15)
1
fP =
COUT ´ RL ´ 2p
(16)
1
fZ =
COUT ´ RESR ´ 2p
(17)
7.3.21 Small Signal Model for Frequency Compensation
The TPS54260 uses a transconductance amplifier for the error amplifier and readily supports three of the
commonly-used frequency compensation circuits. Compensation circuits Type 2A, Type 2B, and Type 1 are
shown in Figure 48. Type 2 circuits most likely implemented in high-bandwidth power-supply designs using lowESR output capacitors. The Type 1 circuit is used with power-supply designs with high-ESR aluminum
electrolytic or tantalum capacitors.. Equation 18 and Equation 19 show how to relate the frequency response of
the amplifier to the small signal model in Figure 48. The open-loop gain and bandwidth are modeled using the RO
and CO shown in Figure 48. See the application section for a design example using a Type 2A network with a
low-ESR output capacitor.
Equation 18 through Equation 27 are provided as a reference for those who prefer to compensate using the
preferred methods. Those who prefer to use prescribed method use the method outlined in the application
section or use switched information.
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VO
R1
VSENSE
gmea
Type 2A
COMP
Type 2B
Type 1
Vref
R2
RO
R3
CO
C2
R3
C2
C1
C1
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Figure 48. Types of Frequency Compensation
Aol
A0
P1
Z1
P2
A1
BW
Figure 49. Frequency Response of the Type 2A and Type 2B Frequency Compensation
Aol(V/V)
gmea
gmea
=
2p ´ BW (Hz)
Ro =
CO
(18)
(19)
æ
ö
s
ç1 +
÷
2p ´ fZ1 ø
è
EA = A0 ´
æ
ö æ
ö
s
s
ç1 +
÷ ´ ç1 +
÷
2p ´ fP1 ø è
2p ´ fP2 ø
è
A0 = gmea
A1 = gmea
P1 =
(20)
R2
´ Ro ´
R1 + R2
R2
´ Ro| | R3 ´
R1 + R2
(21)
(22)
1
2p ´ Ro ´ C1
(23)
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Z1 =
P2 =
P2 =
P2 =
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1
2p ´ R3 ´ C1
(24)
1
2p ´ R3 | | RO ´ (C2 + CO )
type 2a
(25)
1
type 2b
2p ´ R3 | | RO ´ CO
2p ´ R O
(26)
1
type 1
´ (C2 + C O )
(27)
7.4 Device Functional Modes
7.4.1 Operation Near Minimum Input Voltage
The TPS54260 is recommended to operate with input voltages above 3.5 V. The typical VIN UVLO threshold is
2.5 V and the device may operate at input voltages down to the UVLO voltage. At input voltages below the actual
UVLO voltage, the device will not switch. If EN is floating or externally pulled up to greater than the typical 1.25-V
threshold, when V(VIN) passes the UVLO threshold the TPS54260 will become active. Switching is enabled and
the slow-start sequence is initiated. The TPS54260 ramps up the output voltage at the slow-start time determined
by the capacitance on the SS/TR pin.
7.4.2 Operation With Enable Control
The enable start threshold voltage is 1.25 V typical. With EN held below the 1.25-V typical threshold voltage the
TPS54260 is disabled and switching is inhibited even if VIN is above its UVLO threshold. The input current is
reduced in this state. If the EN voltage is increased above the rising threshold voltage while V(VIN) is above the
UVLO threshold, the device becomes active. Switching is enabled and the slow-start sequence is initiated. The
TPS54260 ramps up the output voltage at the slow-start time determined by the capacitance on the SS/TR pin. If
EN is pulled below the 1.25-V typical threshold the TPS54260 will enter the reduced input current state again.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS54260 is a 60-V, 2.5-A, step-down regulator with an integrated high-side MOSFET. This device is
typically used to convert a higher DC voltage to a lower DC voltage with a maximum available output current of
2.5 A. Example applications are 12-V, 24-V and 48-V Industrial, Automotive and Commercial power systems.
Use the following design procedure to select component values for the TPS54260. This procedure illustrates the
design of a high-frequency switching regulator. The Excel® spreadsheet (SLVC432) located on the product page
can help on all calculations. Alternatively, use the WEBENCH software to generate a complete design. The
WEBENCH software uses an iterative design procedure and accesses a comprehensive database of
components when generating a design.
8.2 Typical Applications
8.2.1 3.3-V Output Application
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A.
The estimated printed circuit board area for the components used in this design is 0.55 inch2. This area does not
include test points or connectors.
Figure 50. 3.3-V Output Design Example
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Typical Applications (continued)
8.2.1.1 Design Requirements
Table 1. Design Parameters
PARAMETER
VALUE
Output Voltage
3.3 V
Transient Response 0 to 1.5-A
load step
ΔVout = 3 %
Maximum Output Current
2.5 A
Input Voltage
12 V nom. 10.8 V to 13.2 V
Output Voltage Ripple
1% of Vout
Start Input Voltage (rising VIN)
6.0 V
Stop Input Voltage (falling VIN)
5.5 V
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS54260 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.1.2.2 Selecting the Switching Frequency
The first step is to decide on a switching frequency for the regulator. Typically, the user will want to choose the
highest switching frequency possible since this will produce the smallest solution size. The high-switching
frequency allows for lower valued inductors and smaller output capacitors compared to a power supply that
switches at a lower frequency. The switching frequency that can be selected is limited by the minimum on-time of
the internal power switch, the input voltage and the output voltage and the frequency shift limitation.
Equation 12 and Equation 13 must be used to find the maximum switching frequency for the regulator, choose
the lower value of the two equations. Switching frequencies higher than these values will result in pulse skipping
or the lack of overcurrent protection during a short circuit.
The typical minimum on time, tonmin, is 135 ns for the TPS54260. For this example, the output voltage is 3.3 V
and the maximum input voltage is 13.2 V, which allows for a maximum switch frequency up to 2247 kHz when
including the inductor resistance, on resistance output current and diode voltage in Equation 12. To ensure
overcurrent runaway is not a concern during short circuits in your design use Equation 13 or the solid curve in
Figure 41 to determine the maximum switching frequency. With a maximum input voltage of 13.2 V, assuming a
diode voltage of 0.7 V, inductor resistance of 26 mΩ, switch resistance of 200 mΩ, a current limit value of 3.5 A
and a short circuit output voltage of 0.2 V. The maximum switching frequency is approximately 4449 kHz.
For this design, a much lower switching frequency of 300 kHz is used. To determine the timing resistance for a
given switching frequency, use Equation 11 or the curve in Figure 40.
The switching frequency is set by resistor R3 shown in Figure 50 For 300 kHz operation a 412 kΩ resistor is
required.
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8.2.1.2.3 Output Inductor Selection (LO)
To calculate the minimum value of the output inductor, use Equation 28.
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current.
The inductor ripple current will be filtered by the output capacitor. Therefore, choosing high inductor ripple
currents will impact the selection of the output capacitor since the output capacitor must have a ripple current
rating equal to or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion
of the designer; however, the following guidelines may be used.
For designs using low-ESR output capacitors such as ceramics, a value as high as KIND = 0.3 may be used.
When using higher ESR output capacitors, KIND = 0.2 yields better results. Since the inductor ripple current is
part of the PWM control system, the inductor ripple current should always be greater than 150 mA for
dependable operation. In a wide input voltage regulator, it is best to choose an inductor ripple current on the
larger side. This allows the inductor to still have a measurable ripple current with the input voltage at its
minimum.
For this design example, use KIND = 0.3 and the minimum inductor value is calculated to be 11 μH. For this
design, a nearest standard value was chosen: 10 μH. For the output filter inductor, it is important that the RMS
current and saturation current ratings not be exceeded. The RMS and peak inductor current can be found from
Equation 30 and Equation 31.
For this design, the RMS inductor current is 2.51 A and the peak inductor current is 2.913 A. The chosen
inductor is a Coilcraft MSS1038-103NLB . It has a saturation current rating of 4.52 A and an RMS current rating
of 4.05 A.
As the equation set demonstrates, lower ripple currents will reduce the output voltage ripple of the regulator but
will require a larger value of inductance. Selecting higher ripple currents will increase the output voltage ripple of
the regulator but allow for a lower inductance value.
The current flowing through the inductor is the inductor ripple current plus the output current. During power-up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
Vinmax - Vout
Vout
Lo min =
´
Io ´ KIND
Vinmax ´ ƒsw
(28)
IRIPPLE =
IL(rms) =
VOUT ´
(Vin max
- VOUT )
Vin max ´ L O ´ fSW
1
- VOUT ) ö
÷
÷
Vinmax ´ LO ´ fSW
ø
æ VOUT ´
(IO )2 + 12 ´ çç
è
(29)
(Vinmax
2
(30)
Iripple
ILpeak = Iout +
2
(31)
8.2.1.2.4 Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor will
determine the modulator pole, the output voltage ripple, and how the regulators responds to a large change in
load current. The output capacitance needs to be selected based on the more stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator can not. This situation would occur if there are desired hold-up
times for the regulator where the output capacitor must hold the output voltage above a certain level for a
specified amount of time after the input power is removed. The regulator also will temporarily not be able to
supply sufficient output current if there is a large, fast increase in the current needs of the load such as
transitioning from no load to a full load. The regulator usually needs two or more clock cycles for the control loop
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to see the change in load current and output voltage and adjust the duty cycle to react to the change. The output
capacitor must be sized to supply the extra current to the load until the control loop responds to the load change.
The output capacitance must be large enough to supply the difference in current for 2 clock cycles while only
allowing a tolerable amount of droop in the output voltage. Equation 32 shows the minimum output capacitance
necessary to accomplish this.
Where ΔIout is the change in output current, ƒsw is the regulators switching frequency and ΔVout is the
allowable change in the output voltage. For this example, the transient load response is specified as a 3%
change in Vout for a load step from 1.5 A to 2.5 A (full load). For this example, ΔIout = 2.5-1.5 = 1.0 A and
ΔVout = 0.03 × 3.3 = 0.099 V. Using these numbers gives a minimum capacitance of 67 μF. This value does not
take the ESR of the output capacitor into account in the output voltage change. For ceramic capacitors, the ESR
is usually small enough to ignore in this calculation. Aluminum electrolytic and tantalum capacitors have higher
ESR that should be taken into account.
The catch diode of the regulator can not sink current so any stored energy in the inductor will produce an output
voltage overshoot when the load current rapidly decreases, see Figure 51. The output capacitor must also be
sized to absorb energy stored in the inductor when transitioning from a high load current to a lower load current.
The excess energy that gets stored in the output capacitor will increase the voltage on the capacitor. The
capacitor must be sized to maintain the desired output voltage during these transient periods. Equation 33 is
used to calculate the minimum capacitance to keep the output voltage overshoot to a desired value. Where L is
the value of the inductor, IOH is the output current under heavy load, IOL is the output under light load, Vf is the
final peak output voltage, and Vi is the initial capacitor voltage. For this example, the worst case load step will be
from 2.5 A to 1.5 A. The output voltage will increase during this load transition and the stated maximum in our
specification is 3 % of the output voltage. This will make Vf = 1.03 × 3.3 = 3.399. Vi is the initial capacitor voltage
which is the nominal output voltage of 3.3 V. Using these numbers in Equation 33 yields a minimum capacitance
of 60 μF.
Equation 34 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
Where fsw is the switching frequency, Voripple is the maximum allowable output voltage ripple, and Iripple is the
inductor ripple current. Equation 34 yields 12 μF.
Equation 35 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 35 indicates the ESR should be less than 36 mΩ.
The most stringent criteria for the output capacitor is 67 μF of capacitance to keep the output voltage in
regulation during an load transient.
Additional capacitance de-ratings for aging, temperature and dc bias should be factored in which will increase
this minimum value. For this example, 2 x 47 μF, 10 V ceramic capacitors with 3 mΩ of ESR will be used. The
derated capacitance is 72.4 µF, above the minimum required capacitance of 67 µF.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the Root Mean Square (RMS) value of the maximum ripple current. Equation 36 can be used
to calculate the RMS ripple current the output capacitor needs to support. For this application, Equation 36 yields
238 mA.
2 ´ DIout
Cout >
¦ sw ´ DVout
(32)
(Ioh
(V ¦
2
Cout > Lo ´
1
Cout >
8 ´ ¦ sw
´
)
- Vi )
- Iol2
2
2
(33)
1
VORIPPLE
IRIPPLE
(34)
V
RESR < ORIPPLE
IRIPPLE
Icorms =
32
(35)
Vout ´ (Vin max - Vout)
12 ´ Vin max ´ Lo ´ ¦ sw
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8.2.1.2.5 Catch Diode
The TPS54260 requires an external catch diode between the PH pin and GND. The selected diode must have a
reverse voltage rating equal to or greater than Vinmax. The peak current rating of the diode must be greater than
the maximum inductor current. The diode should also have a low forward voltage. Schottky diodes are typically a
good choice for the catch diode due to their low forward voltage. The lower the forward voltage of the diode, the
higher the efficiency of the regulator.
Typically, the higher the voltage and current ratings the diode has, the higher the forward voltage will be.
Although the design example has an input voltage up to 13.2 V, a diode with a minimum of 60-V reverse voltage
is selected to allow input voltage transients up to the rated voltage of the TPS54260.
For the example design, the B360B-13-F Schottky diode is selected for its lower forward voltage and it comes in
a larger package size which has good thermal characteristics over small devices. The typical forward voltage of
the B360B-13-F is 0.70 volts.
The diode must also be selected with an appropriate power rating. The diode conducts the output current during
the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input
voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by
the forward voltage of the diode which equals the conduction losses of the diode. At higher switch frequencies,
the AC losses of the diode need to be taken into account. The AC losses of the diode are due to the charging
and discharging of the junction capacitance and reverse recovery. Equation 37 is used to calculate the total
power dissipation, conduction losses plus ac losses, of the diode.
The B360B-13-F has a junction capacitance of 200 pF. Using Equation 37, the selected diode will dissipate 1.32
Watts.
If the power supply spends a significant amount of time at light-load currents or in sleep mode consider using a
diode which has a low leakage current and slightly higher forward voltage drop.
2
Pd =
(Vin max - Vout) ´ Iout ´ Vƒd Cj ´ ƒsw ´ (Vin + Vƒd)
+
2
Vin max
(37)
8.2.1.2.6 Input Capacitor
The TPS54260 requires a high-quality ceramic, type X5R or X7R, input decoupling capacitor of at least 3 μF of
effective capacitance and in some applications a bulk capacitance. The effective capacitance includes any dc
bias effects. The voltage rating of the input capacitor must be greater than the maximum input voltage. The
capacitor must also have a ripple current rating greater than the maximum input current ripple of the TPS54260.
The input ripple current can be calculated using Equation 38.
The value of a ceramic capacitor varies significantly over temperature and the amount of dc bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the dc bias taken into account. The capacitance value of a capacitor
decreases as the dc bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 60-V voltage rating is required to support the
maximum input voltage. Common standard ceramic capacitor voltage ratings include 4 V, 6.3 V, 10 V, 16 V, 25
V, 50 V or 100 V so a 100-V capacitor should be selected. For this example, two 2.2-μF, 100-V capacitors in
parallel have been selected. Table 2 shows a selection of high-voltage capacitors. The input capacitance value
determines the input ripple voltage of the regulator. The input voltage ripple can be calculated using Equation 39.
Using the design example values, Ioutmax = 2.5 A, Cin = 4.4 μF, ƒsw = 300 kHz, yields an input voltage ripple of
473 mV and a RMS input ripple current of 1.15 A.
Icirms = Iout ´
Vout
´
Vin min
(Vin min
- Vout )
Vin min
(38)
Iout max ´ 0.25
ΔVin =
Cin ´ ¦ sw
(39)
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Table 2. Capacitor Types
VENDOR
VALUE (μF)
1.0 to 2.2
Murata
1.0 to 4.7
1.0
1.0 to 2.2
1.0 10 1.8
Vishay
1.0 to 1.2
1.0 to 3.9
1.0 to 1.8
1.0 to 2.2
TDK
1.5 to 6.8
1.0. to 2.2
1.0 to 3.3
1.0 to 4.7
AVX
1.0
1.0 to 4.7
1.0 to 2.2
EIA Size
1210
1206
2220
2225
1812
1210
1210
1812
VOLTAGE
DIALECTRIC
100 V
COMMENTS
GRM32 series
50 V
100 V
GRM31 series
50 V
50 V
100 V
VJ X7R series
50 V
100 V
100 V
50 V
100 V
50 V
X7R
C series C4532
C series C3225
50 V
100 V
50 V
X7R dielectric series
100 V
8.2.1.2.7 Slow-Start Capacitor
The slow-start capacitor determines the minimum amount of time it will take for the output voltage to reach its
nominal programmed value during power-up. This is useful if a load requires a controlled voltage slew rate. This
is also used if the output capacitance is large and would require large amounts of current to quickly charge the
capacitor to the output voltage level. The large currents necessary to charge the capacitor may make the
TPS54260 reach the current limit or excessive current draw from the input power supply may cause the input
voltage rail to sag. Limiting the output voltage slew rate solves both of these problems.
The slow-start time must be long enough to allow the regulator to charge the output capacitor up to the output
voltage without drawing excessive current. Equation 40 can be used to find the minimum slow-start time, tss,
necessary to charge the output capacitor, Cout, from 10% to 90% of the output voltage, Vout, with an average
slow-start current of Issavg. In the example, to charge the effective output capacitance of 72.4 µF up to 3.3 V
while only allowing the average output current to be 1 A would require a 0.19-ms slow-start time.
Once the slow-start time is known, the slow-start capacitor value can be calculated using Equation 6. For the
example circuit, the slow-start time is not too critical since the output capacitor value is 2 x 47 μF which does not
require much current to charge to 3.3 V. The example circuit has the slow-start time set to an arbitrary value of
3.5 ms which requires a 8.75-nF slow-start capacitor. For this design, the next larger standard value of 10 nF is
used.
Cout ´ Vout ´ 0.8
tss >
Issavg
(40)
8.2.1.2.8 Bootstrap Capacitor Selection
A 0.1-μF ceramic capacitor must be connected between the BOOT and PH pins for proper operation. It is
recommended to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have a 10-V
or higher voltage rating.
8.2.1.2.9 Undervoltage Lock Out Set Point
The Under Voltage Lock Out (UVLO) can be adjusted using an external voltage divider on the EN pin of the
TPS54260. The UVLO has two thresholds, one for power-up when the input voltage is rising and one for power
down or brown outs when the input voltage is falling. For the example design, the supply should turn on and start
switching once the input voltage increases above 6.0 V (enabled). After the regulator starts switching, it should
continue to do so until the input voltage falls below 5.5 V (UVLO stop).
34
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The programmable UVLO and enable voltages are set using the resistor divider of R1 and R2 between Vin and
ground to the EN pin. Equation 2 through Equation 3 can be used to calculate the resistance values necessary.
For the example application, a 124 kΩ between Vin and EN (R1) and a 30.1 kΩ between EN and ground (R2)
are required to produce the 6.0 and 5.5 volt start and stop voltages.
8.2.1.2.10 Output Voltage and Feedback Resistors Selection
The voltage divider of R5 and R6 is used to set the output voltage. For the example design, 10.0 kΩ was
selected for R6. Using Equation 1, R5 is calculated as 31.25 kΩ. The nearest standard 1% resistor is 31.6 kΩ.
Due to current leakage of the VSENSE pin, the current flowing through the feedback network should be greater
than 1 μA in order to maintain the output voltage accuracy. This requirement makes the maximum value of R2
equal to 800 kΩ. Choosing higher resistor values will decrease quiescent current and improve efficiency at low
output currents but may introduce noise immunity problems.
8.2.1.2.11 Compensation
There are several methods used to compensate DC - DC regulators. The method presented here is easy to
calculate and ignores the effects of the slope compensation that is internal to the device. Since the slope
compensation is ignored, the actual crossover frequency will usually be lower than the crossover frequency used
in the calculations. This method assumes the crossover frequency is between the modulator pole and the esr
zero and the esr zero is at least 10 times greater the modulator pole. Use SwitcherPro software for a more
accurate design.
To get started, the modulator pole, fpmod, and the ESR zero, fz1 must be calculated using Equation 41 and
Equation 42. For Cout, use a derated value of 40 μF. Use equations Equation 43 and Equation 44, to estimate a
starting point for the crossover frequency, fco, to design the compensation. For the example design, fpmod is
1206 Hz and fzmod is 530.5 kHz. Equation 43 is the geometric mean of the modulator pole and the esr zero and
Equation 44 is the mean of modulator pole and the switching frequency. Equation 43 yields 25.3 kHz and
Equation 44 gives 13.4 kHz. Use the lower value of Equation 43 or Equation 44 for an initial crossover frequency.
For this example, a higher fco is desired to improve transient response. the target fco is 35.0 kHz. Next, the
compensation components are calculated. A resistor in series with a capacitor is used to create a compensating
zero. A capacitor in parallel to these two components forms the compensating pole.
Ioutmax
¦p mod =
2 × p × Vout × Cout
(41)
1
¦ z mod =
2 ´ p ´ Resr × Cout
(42)
fco =
f p mod ´ f z mod
fco =
f p mod ´
(43)
f sw
2
(44)
To determine the compensation resistor, R4, use Equation 45. Assume the power stage transconductance,
gmps, is 10.5S. The output voltage, Vo, reference voltage, VREF, and amplifier transconductance, gmea, are
3.3V, 0.8V and 310 μS, respectively. R4 is calculated to be 20.2 kΩ, use the nearest standard value of 20.0 kΩ.
Use Equation 46 to set the compensation zero to the modulator pole frequency. Equation 46 yields 4740 pF for
compensating capacitor C5, a 4700-pF is used for this design.
ö
æ 2 ´ p ´ fco ´ Cout ö æ
Vout
R4 = ç
÷
÷´ç
gmps
è
ø è Vref ´ gmea ø
1
C5 =
2 ´ p ´ R4 ´ fpmod
(45)
(46)
A compensation pole can be implemented if desired using an additional capacitor C8 in parallel with the series
combination of R4 and C5. Use the larger value of Equation 47 and Equation 48 to calculate the C8, to set the
compensation pole. C8 is not used for this design example.
C ´ Re sr
C8 = o
R4
(47)
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C8 =
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1
R4 ´ f sw ´ p
(48)
8.2.1.2.12 Discontinuous Mode and Eco-Mode Boundary
With an input voltage of 12 V, the power supply enters discontinuous mode when the output current is less than
337 mA. The power supply enters Eco-Mode when the output current is lower than 5 mA.
The input current draw at no load is 392 μA.
8.2.1.2.13 Power Dissipation Estimate
The following formulas show how to estimate the IC power dissipation under continuous conduction mode (CCM)
operation. These equations should not be used if the device is working in discontinuous conduction mode (DCM).
The power dissipation of the IC includes conduction loss (Pcon), switching loss (PSW), gate drive loss (PGD)
and supply current (Pq).
Vout
Pcon = Io2 ´ RDS(on) ´
Vin
(49)
Psw = Vin 2 ´ ¦ sw ´ lo ´ 0.25 ´ 10-9
Pgd = Vin ´ 3 ´ 10
Pq = 116 ´ 10
-6
-9
´ ¦ sw
(50)
(51)
´ Vin
where
•
•
•
•
•
IOUT is the output current (A)
RDS(on) is the on-resistance of the high-side MOSFET (Ω)
VOUT is the output voltage (V)
VIN is the input voltage (V)
fsw is the switching frequency (Hz)
(52)
So
Ptot = Pcon + Psw + Pgd + Pq
(53)
For given TA,
TJ = TA + Rth ´ Ptot
(54)
For given TJMAX = 150°C
TAmax = TJmax - Rth ´ Ptot
where
•
•
•
•
•
•
Ptot is the total device power dissipation (W)
TA is the ambient temperature (°C)
TJ is the junction temperature (°C)
Rth is the thermal resistance of the package (°C/W)
TJMAX is maximum junction temperature (°C)
TAMAX is maximum ambient temperature (°C).
(55)
There will be additional power losses in the regulator circuit due to the inductor ac and dc losses, the catch diode
and trace resistance that will impact the overall efficiency of the regulator.
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8.2.1.3 Application Curves
Vout = 50 mv / div (ac coupled)
Vin = 10 V / div
Vout = 2 V / div
Output Current = 1 A / div (Load Step 1.5 A to 2.5 A)
EN = 2 V / div
SS/TR = 2 V / div
Time = 200 usec / div
Time = 5 msec / div
Figure 51. Load Transient
Figure 52. Start-Up With VIN
Vout = 20 mV / div (ac coupled)
Vout = 20 mV / div (ac coupled)
PH = 5 V / div
PH = 5 V / div
Time = 2 usec / div
Time = 2 usec / div
Figure 53. Output Ripple CCM
Figure 54. Output Ripple, DCM
Vin = 200 mV / div (ac coupled)
Vout = 20 mV / div (ac coupled)
PH = 5 V / div
PH = 5 V / div
Time = 2 usec / div
Time = 10 usec / div
Figure 55. Output Ripple, PSM
Figure 56. Input Ripple, CCM
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100
90
Vin = 50 mV / div (ac coupled)
80
Efficiency - %
70
PH = 5 V / div
60
50
40
30
VIN=12V
VOUT=3.3V
fsw=300kHz
20
10
0
0
0.5
Time = 2 usec / div
Figure 57. Input Ripple, DCM
2.5
1.0
1.5
2.0
IO - Output Current - A
3.0
Figure 58. Efficiency vs Load Current
100
60
180
90
40
80
120
Phase
70
60
Gain
50
40
0
0
-20
30
VIN=12V
VOUT=3.3V
fsw=300kHz
20
-60
VIN=12 V
VOUT=3.3V
IOUT=2.5A
-40
10
0
0.001
0.1
0.01
IO - Output Current - A
-60
10
3.4
3.4
3.38
3.38
VO - Output Voltage - V
VO - Output Voltage - V
-120
1-104
1-103
f - Frequency - Hz
100
3.36
3.34
3.32
3.36
3.34
VIN=12V
VOUT=3.3V
fsw=300kHz
IOUT=1.5A
3.32
VIN=12V
VOUT=3.3V
fsw=300kHz
3.3
0.5
1.5
1.0
2.0
IO - Output Current - A
2.5
3.0
3.3
10.8
Figure 61. Regulation vs Load Current
38
-180
1-106
1-105
Figure 60. Overall Loop Frequency Response
Figure 59. Light-Load Efficiency
0
Phase - o
Gain - dB
Efficiency - %
20
60
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11.2
11.6
12.4
12
IO - Output Current - A
12.8
13.2
Figure 62. Regulation vs Input Voltage
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8.2.2 Inverting Power Supply
This application circuit shows how to use the TPS54260 to convert a positive-input voltage to a negative-output
voltage. Ideal applications are amplifiers requiring a negative power supply. For a more-detailed example, see
Create an Inverting Power Supply From a Step-Down Regulator, application report SLVA317.
VIN
+
Cin
Cboot
Lo
VIN
Cd
PH
BOOT
GND
R1
+
GND
Co
R2
TPS54260
VOUT
VSENSE
EN
COMP
SS/TR
Rcomp
RT/CLK
Css
Czero
RT
Cpole
Copyright © 2016, Texas Instruments Incorporated
Figure 63. TPS54260 Inverting Power Supply from SLVA317 Application Note
8.2.3 Split-Rail Power Supply
This application circuit shows how to use the TPS54260 to convert a positive-input voltage to a split-rail positive
and negative-output voltage by using a coupled inductor. Ideal applications are amplifiers requiring a split-rail
positive- and negative-voltage power supply. For a more-detailed example, see Creating a Split-Rail Power
Supply With a Wide Input Voltage Buck Regulator, application report SLVA369.
VOPOS
+
VIN
Copos
+
Cin
Cboot
BOOT
VIN
GND
PH
Lo
Cd
R1
GND
+
Coneg
R2
TPS54260
VONEG
VSENSE
EN
COMP
SS/TR
Rcomp
RT/CLK
Css
RT
Czero
Cpole
Copyright © 2016, Texas Instruments Incorporated
Figure 64. TPS54260 Split-Rail Power Supply based on SLVA369 Application Note
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8.2.4 12-V to 3.8-V GSM Power Supply
This application circuit is designed with TPS54260 device to power GSM-GPRS modules. GSM-GPRS modules
typically require a power supply that can support large output current transients. For a more-detailed example,
see Creating GSM-GPRS Power Supply, application report SLVA412.
12V nom. 8V to 40V
3.8V, 2.0A
L1: MSS1260-103
C4, C5: 47mF 10V X5R
Copyright © 2016, Texas Instruments Incorporated
Figure 65. 12 V to 3.8 V GSM Power Supply
8.2.5 24-V to 4.2-V GSM Power Supply
This application circuit is also designed to power GSM-GPRS modules. For a more-detailed example, see
Creating GSM-GPRS Power Supply, application report SLVA412.
4.2V, 2.0A
24V nom. 18V to 40V
L1: MSS1260-103
C4: 100mF 10V X5R
Copyright © 2016, Texas Instruments Incorporated
Figure 66. 24-V to 4.2-V GSM Power Supply
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9 Power Supply Recommendations
The design of the device is for operation from an input voltage supply range between 3.5 V and 60 V. This input
supply should remain within the input voltage supply range. If the input supply is more distant than a few inches
from the TPS54260 converter, the circuit may require additional bulk capacitance in addition to the ceramic
bypass capacitors. An electrolytic capacitor with a value of 100 µF is a typical choice.
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. There are several signals paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade the power supplies performance. To help eliminate these problems, the VIN pin should be bypassed
to ground with a low-ESR ceramic bypass capacitor with X5R or X7R dielectric. Take care to minimize the loop
area formed by the bypass capacitor connections, the VIN pin, and the anode of the catch diode. See Figure 67
for a PCB layout example. The GND pin must be tied directly to the power pad under the IC and the power pad.
The power pad should be connected to any internal PCB ground planes using multiple vias directly under the IC.
The PH pin should be routed to the cathode of the catch diode and to the output inductor. Since the PH
connection is the switching node, the catch diode and output inductor should be located close to the PH pins,
and the area of the PCB conductor minimized to prevent excessive capacitive coupling. For operation at full rated
load, the top side ground area must provide adequate heat dissipating area. The RT/CLK pin is sensitive to noise
so the RT resistor should be located as close as possible to the IC and routed with minimal lengths of trace. The
additional external components can be placed approximately as shown. It may be possible to obtain acceptable
performance with alternate PCB layouts, however this layout has been shown to produce good results and is
meant as a guideline.
10.2 Layout Example
Vout
Output
Capacitor
Topside
Ground
Area
Route Boot Capacitor
Trace on another layer to
provide wide path for
topside ground
Input
Bypass
Capacitor
BOOT
Vin
UVLO
Adjust
Resistors
Slow Start
Capacitor
Output
Inductor
Catch
Diode
PH
VIN
GND
EN
COMP
SS/TR
VSENSE
RT/CLK
PWRGD
Frequency
Set Resistor
Compensation
Network
Resistor
Divider
Thermal VIA
Signal VIA
Figure 67. PCB Layout Example
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
11.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS54260 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
Eco-mode, PowerPAD, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
Excel is a registered trademark of Microsoft Corporation.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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18-Sep-2018
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
TPS54260DGQ
ACTIVE
MSOPPowerPAD
DGQ
10
80
Green (RoHS
& no Sb/Br)
CU NIPDAU |
CU NIPDAUAG
Level-1-260C-UNLIM
-40 to 125
54260
TPS54260DGQR
ACTIVE
MSOPPowerPAD
DGQ
10
2500
Green (RoHS
& no Sb/Br)
CU NIPDAU |
CU NIPDAUAG
Level-1-260C-UNLIM
-40 to 125
54260
TPS54260DRCR
ACTIVE
VSON
DRC
10
3000
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
-40 to 125
54260
TPS54260DRCT
ACTIVE
VSON
DRC
10
250
Green (RoHS
& no Sb/Br)
CU NIPDAU
Level-1-260C-UNLIM
-40 to 125
54260
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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