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TPS54334
ZHCSDR7 – MAY 2015
TPS54334 4.2V 至 28V 输入电压、3A 输出电流同步 SWIFT™
降压控制器
1 特性
•
1
•
•
•
•
•
•
•
•
•
3 说明
两个用于提供 3A 持续输出电流的 128mΩ/84mΩ
金属氧化物半导体场效应晶体管 (MOSFET)
电源正常
关断时静态电流低至 2µA
0.8V 内部参考电压，整个温度范围内的精度为
±1.5%
定频电流模式控制
脉冲跳跃提升了轻负载时的效率
对于严重故障条件，用断续模式来对两个 MOSFET
进行过流保护
过热和过压瞬态保护
采用易于使用的 8 引脚小外形尺寸集成电路
(SOIC) PowerPAD™ 和 10 引脚小外形尺寸无引线
(SON) 封装
针对预偏置输出的单调性启动
TPS54334 是一款带有 MOSFET 的 28V、3A、低静
态电源电流 (IQ) 同步单片降压转换器。
TPS54334 通过集成 MOSFET 并实施电流模式控制来
减少外部元件数，从而实现小型设计。
该器件凭借集成的 128mΩ/84mΩ MOSFET、低静态电
源电流以及轻负载条件下的脉冲跳跃实现了效率的最大
化。 器件可利用使能引脚进入关断模式，关断电源电
流可降至
2µA。
TPS54334 的参考电压在整个温度范围内保持 1.5% 的
精度，可为各类负载提供精确稳压。
高侧场效应晶体管 (FET) 的逐周期电流限制可在过载
情况下保护 TPS54334，并通过低侧电源限流防止电
流失控，从而实现功能增强。 此外，还提供关闭低侧
MOSFET 的低侧吸收电流限值，以防止过多的反向电
流。 在过流情况持续时间超过预设时间时，将触发断
续保护功能。 当裸片温度超过热关断温度时，过热断
续保护功能将禁用该部件，而在内置热断续时间之后，
则可重新启用该部件。
2 应用
•
•
•
•
消费类应用，例如数字电视 (DTV)、机顶盒
（STB、数字化视频光盘 (DVD)/蓝光播放器）、液
晶显示器、客户端设备 (CPE)（电缆调制解调器、
WiFi 路由器）、数字光处理 (DLP) 投影仪、智能
电表
电池充电器
工业用和车载音频电源
5V、12V 和 24V 分布式电源总线供电
器件信息(1)
器件型号
封装
TPS54334
封装尺寸（标称值）
SO（8 引脚）
4.89mm x 3.90mm
VSON（10 引脚）
3.0mm × 3.0mm
(1) 要了解所有可用封装，请见数据表末尾的可订购产品附录。
4 简化电路原理图
U1
TPS54334DRC
VIN
VIN=4.2 V to 24 V
C1
10µF
C2
0.1µF
1
R1
221 kΩ
VIN
PGOOD
VDD
10
C3
BOOT 9
8 EN
SW
7
R2
84.5 kΩ
C5
150 pF
R7 100 kΩ
R3
2.05 kΩ
C4
0.015 µF
COMP
2
0.1 µF
VSENSE 6
GND
L1 6.8 µH
VOUT= 3.3 V
C6
22 µF
C7
22 µF
VOUT
R4
51.1 Ω
C8
200 pF
R5
31.6 kΩ
3,4,5
R6
10 kΩ
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
English Data Sheet: SLUSC26
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
目录
1
2
3
4
5
6
7
8
特性 ..........................................................................
应用 ..........................................................................
说明 ..........................................................................
简化电路原理图 ........................................................
修订历史记录 ...........................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
1
2
3
4
7.1
7.2
7.3
7.4
7.5
7.6
4
4
4
5
5
7
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information .................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
8.4 Device Functional Modes........................................ 13
9
Application and Implementation ........................ 16
9.1 Application Information............................................ 16
9.2 Typical Applications ................................................ 16
10 Power Supply Recommendations ..................... 24
11 Layout................................................................... 24
11.1 Layout Guidelines ................................................. 24
11.2 Layout Example .................................................... 25
12 器件和文档支持 ..................................................... 26
12.1
12.2
12.3
12.4
12.5
12.6
Detailed Description .............................................. 9
8.1 Overview ................................................................... 9
8.2 Functional Block Diagram ....................................... 10
8.3 Feature Description................................................. 10
器件支持................................................................
文档支持................................................................
社区资源................................................................
商标 .......................................................................
静电放电警告.........................................................
术语表 ...................................................................
26
26
26
26
26
26
13 机械、封装和可订购信息 ....................................... 26
5 修订历史记录
2
日期
修订版本
注释
2015 年 5 月
*
首次发布。
Copyright © 2015, Texas Instruments Incorporated
TPS54334
www.ti.com.cn
ZHCSDR7 – MAY 2015
6 Pin Configuration and Functions
DRC Package
10 Pin VSON
Top View
DDA Package
8 Pin SO
Top View
VIN
1
10
PGOOD
SW
2
9
BOOT
GND
3
8
EN
Thermal Pad
BOOT
1
VIN
2
8
PGOOD
7
EN
PowerPAD
GND
4
7
COMP
GND
5
6
VSENSE
SW
3
6
COMP
GND
4
5
VSENSE
Pin Functions
PIN
NAME
NUMBER
I/O
DESCRIPTION
DDA
DRC
BOOT
1
9
O
A bootstrap capacitor is required between BOOT and SW. If the voltage on this capacitor is below the
minimum required by the output device, the output is forced to switch off until the capacitor is
refreshed.
COMP
6
7
O
Error amplifier output, and input to the output switch current comparator. Connect frequency
compensation components to this pin.
EN
7
8
I
Enable pin. Float to enable.
GND
4
3, 4, 5
–
Ground.
PGOOD
8
10
O
PGOOD open drain output. Connect a pull-up resistor with a value of 100kΩ to this pin.
SW
3
2
O
The source of the internal high side power MOSFET.
Vin
2
1
–
Input supply voltage, 4.2 V to 28 V.
VSENSE
5
6
I
Inverting node of the gm error amplifier.
–
GND pin should be connected to the exposed thermal pad for proper operation. This thermal pad
should be connected to any internal PCB ground plane using multiple vias for good thermal
performance.
PowerPad (SO only)
Thermal pad (VSON only)
Copyright © 2015, Texas Instruments Incorporated
3
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
Input Voltage
(1) (2)
MIN
MAX
UNIT
VIN
–0.3
30
V
EN
–0.3
6
V
BOOT
–0.3
(SW+7.5)
V
VSENSE
–0.3
3
V
COMP
–0.3
3
V
PGOOD
–0.3
6
V
0
7.5
V
BOOT-SW
Output Voltage
SW
SW 10ns Transient
Vdiff(GND to Exposed Thermal Pad)
–1
30
V
–3.5
30
V
–0.2
0.2
V
100
µA
EN
Source Current
Sink Current
SW
Current Limit
SW
Current Limit
COMP
200
PGOOD
A
A
200
µA
mA
–0.1
5
Operating Junction Temperature
–40
150
Storage temperature, Tstg
–65
150
(1)
(2)
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
The human body model is a 100-pF capacitor discharged through a 1.5-kΩ resistor into each pin. The machine model is a 200-pF
capacitor discharged directly into each pin.
7.2 ESD Ratings
V(ESD)
(1)
(2)
Electrostatic
discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins
(1)
Charged device model (CDM), per JEDEC specification JESD22-C101, all
pins (2)
VALUE
UNIT
±2000
V
±500
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VSS
Supply input voltage
VOUT Output voltage
IOUT
TJ
(1)
4
Output current
Operating junction temperature
(1)
MIN
MAX
UNIT
4.2
28
V
0.8
24
V
0
3
A
–40
150
°C
The device must operate within 150°C to ensure continuous function and operation of the device.
Copyright © 2015, Texas Instruments Incorporated
TPS54334
www.ti.com.cn
ZHCSDR7 – MAY 2015
7.4 Thermal Information
TPS54334
THERMAL METRIC (1)
DDA (8 PINS)
DRC (10 Pins)
UNIT
RθJA
Junction-to-ambient thermal resistance
42.1
43.9
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
50.9
55.4
°C/W
RθJB
Junction-to-board thermal resistance
31.8
18.9
°C/W
ψJT
Junction-to-top characterization parameter
5
0.7
°C/W
ψJB
Junction-to-board characterization parameter
13.5
19.1
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
7.1
5.3
°C/W
(1)
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
7.5 Electrical Characteristics
The Electrical Ratings specified in this section will apply to all specifications in this document unless otherwise noted. These
specifications will be interpreted as conditions that will not degrade the device’s parametric or functional specifications for the
life of the product containing it. TJ = –40°C to 150°C, VIN =4.2 to 28V, (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
3.9
4.2
V
180
400
mV
2
10
µA
SUPPLY VOLTAGE AND UVLO (VIN PIN)
Operating input voltage
Input UVLO threshold
4.2
Rising Vin
Input UVLO hysteresis
28
V
VIN Shutdown Supply Current
EN = 0V
VIN Operating– non switching supply current
VSENSE=810mV
310
800
µA
Enable threshold
Rising
1.21
1.28
V
Enable threshold
Falling
1.17
V
Input current
EN=1.1V
1.15
µA
Hysteresis current
EN=1.3V
3.3
µA
ENABLE (EN PIN)
1.1
VOLTAGE REFERENCE
Reference
TJ = 25°C
0.792
0.8
0.808
V
0.788
0.8
0.812
V
BOOT-SW = 3V
160
290
mΩ
BOOT-SW = 6V
128
240
mΩ
84
170
mΩ
MOSFET
High side switch resistance (1)
Low side switch resistance (1)
ERROR AMPLIFIER
Error amplifier transconductance (gm)
–2 µA < ICOMP < 2 µA V(COMP)=1V
Error amplifier source/sink
V(COMP)=1V, 100 mV Overdrive
Start switching peak current threshold
1300
µmhos
100
µA
0.5
COMP to Iswitch gm
A
8
A/V
CURRENT LIMIT
High side switch current limit threshold
Low side switch sourcing current limit
Low side switch sinking current limit
Hiccup wait time
Hiccup time before re-start
(1)
4
5.2
6.5
A
3.5
4.7
6.1
A
0
A
512
Cycles
16384
Cycles
Measured at pins.
Copyright © 2015, Texas Instruments Incorporated
5
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
Electrical Characteristics (continued)
The Electrical Ratings specified in this section will apply to all specifications in this document unless otherwise noted. These
specifications will be interpreted as conditions that will not degrade the device’s parametric or functional specifications for the
life of the product containing it. TJ = –40°C to 150°C, VIN =4.2 to 28V, (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
THERMAL SHUTDOWN
Thermal shutdown
165
Thermal shutdown hysterisis
°C
10
Thermal shutdown hiccup time
°C
32768
Cycles
SW (SW PIN)
Minimum on time
Measured at 90% to 90% of VIN, ISW = 2A
Minimum off time
BOOT-SW ≥ 3V
94
145
ns
3
V
0%
BOOT (BOOT PIN)
BOOT-SW UVLO
2.2
SLOW START
Internal slow start time
2
ms
SWITCHING FREQUENCY
Internal switching frequency
456
570
684
kHz
POWER GOOD (PGOOD PIN)
VSENSE falling (Fault)
84
% Vref
VSENSE rising (Good)
90
% Vref
VSENSE rising (Fault)
116
% Vref
VSENSE falling (Good)
110
% Vref
Output high leakage
VSENSE = Vref, V(PGOOD) = 5.5 V
Output low
I(PGOOD) = 0.35 mA
Minimum VIN for valid output (2)
V(PGOOD) < 0.5V at 100 µA
30
0.6
500
nA
0.3
V
1
V
MAXIMUM OUTPUT VOLTAGE UNDER MINIMUM VIN (2)
Maximum output voltage
(2)
6
VIN = 4.2V, Iout = 3A
2.9
VIN = 4.2V, Iout = 2.5A
3.2
VIN = 4.2V, Iout = 2A
3.4
VIN = 4.2V, Iout = 1.5A
3.5
V
Not tested for mass production.
Copyright © 2015, Texas Instruments Incorporated
TPS54334
www.ti.com.cn
ZHCSDR7 – MAY 2015
7.6 Typical Characteristics
VIN = 12 V, unless otherwise specified.
140
210
130
On Resistance (m
On Resistance (m
190
170
150
130
110
120
110
100
90
80
70
90
60
50
70
±50
±25
0
25
50
75
100
125
Junction Temperature (C)
±50
150
±25
0
25
50
75
100
125
Junction Temperature (C)
C001
150
C002
Figure 2. Low-Side Resistance vs Junction Temperature
Figure 1. High-Side Resistance vs Junction Temperature
575
0.808
Switching Frequnecy (kHz)
Voltage Reference (V)
570
0.804
0.800
0.796
560
555
550
545
540
535
530
0.792
±50
±25
0
25
50
75
100
125
Junction Temperature (C)
150
525
-50
C003
Figure 3. Reference Voltage vs Junction Temperature
0
25
50
75
100
Junction Temperature (qC)
125
150
D004
2
3
Soft Start Time (ms)
1.98
2.5
2
1.5
1.96
1.94
1.92
1
0.5
-50
-25
Figure 4. Switching Frequency vs Junction Temperature
3.5
Shut Down Supply Current (PA)
565
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
D005
Figure 5. Shutdown Current vs Junction Temperature
Copyright © 2015, Texas Instruments Incorporated
1.9
-50
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
D006
Figure 6. Softstart Time vs Junction Temperature
7
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
Typical Characteristics (continued)
VIN = 12 V, unless otherwise specified.
108
380
106
Minimum On Time (ns)
Supply Current (PA)
360
340
320
300
-25
0
25
50
75
100
Junction Temperature (qC)
125
100
96
-50
150
-25
0
D007
Figure 7. Non-Switching Operating Quiescent Current vs
Junction Temperature
25
50
75
100
Junction Temperature (qC)
125
150
D008
Figure 8. Minimum On Time vs Junction Temperature
5.4
2.5
LoH
HoL
5.3
High Side Current Limit (A)
2.45
BOOT UVLO Threshold (V)
102
98
280
-50
2.4
2.35
2.3
2.25
2.2
5.2
5.1
5
4.9
4.8
2.15
2.1
-50
-25
0
25
50
75
100
Junction Temperature (qC)
125
4.7
-50
150
-25
0
D009
Figure 9. BOOT-SW UVLO Threshold vs Junction
Temperature
25
50
75
100
Junction Temperature (qC)
125
150
D010
Figure 10. High-Side Current Limit Threshold vs Junction
Temperature
1.3
4
LoH
HoL
3.95
1.26
UVLO Threshold (V)
EN-UVLO Threshold (V)
104
1.22
1.18
3.9
3.85
3.8
3.75
1.14
3.7
1.1
-50
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
D011
Figure 11. EN Threshold vs Junction Temperature
8
3.65
-50
LoH
HoL
-25
0
25
50
75
100
Junction Temperature (qC)
125
150
D012
Figure 12. VIN UVLO Threshold vs Junction Temperature
Copyright © 2015, Texas Instruments Incorporated
TPS54334
www.ti.com.cn
ZHCSDR7 – MAY 2015
8 Detailed Description
8.1 Overview
The device is a 28-V, 3-A, synchronous step-down (buck) converter with two integrated n-channel MOSFETs. To
improve performance during line and load transients the device implements a constant frequency, peak current
mode control which reduces output capacitance and simplifies external frequency compensation design.
The device is designed for safe monotonic startup into pre-biased loads. It has a typical default start up voltage
of 3.9 V. The EN pin has an internal pull-up current source that can provide a default condition when the EN pin
is floating for the device to operate. The total operating current for the device is typically 310µA when not
switching and under no load. When the device is disabled, the supply current is less than 5μA.
The integrated 128mΩ/84mΩ MOSFETs allow for high efficiency power supply designs with continuous output
currents up to 3 amperes.
The device reduces the external component count by integrating the boot recharge diode. The bias voltage for
the integrated high-side MOSFET is supplied by a capacitor between the BOOT and SW pins. The boot capacitor
voltage is monitored by an UVLO circuit and turns off the high-side MOSFET when the voltage falls below a
preset threshold. The output voltage can be stepped down to as low as the 0.8 V reference.
The device has a power good comparator (PGOOD) with hysteresis which monitors the output voltage through
the VSENSE pin. The PGOOD pin is an open drain MOSFET which is pulled low when the VSENSE pin voltage
is less than 84% or greater than 116% of the reference voltage Vref and asserts high when the VSENSE pin
voltage is 90% to 110% of the Vref.
The device minimizes excessive output over-voltage transients by taking advantage of the over-voltage power
good comparator. When the regulated output voltage is greater than 116% of the nominal voltage, the overvoltage comparator is activated, and the high-side MOSFET is turned off and masked from turning on until the
output voltage is lower than 110%.
The TPS54334 operating frequency is fixed at 570 kHz and at 2 ms slow start time.
Copyright © 2015, Texas Instruments Incorporated
9
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
8.2 Functional Block Diagram
PGOOD
VIN
EN
Shutdown
Ip
UV
Ih
Enable
Comparator
Thermal
Hiccup
UVLO
Shutdown
Logic
Logic
Hiccup
Shutdown
Enable
Threshold
OV
Boot
Charge
Current
Sense
Minimum Clamp
Pulse Skip
ERROR
AMPLIFIER
VSENSE
BOOT
Boot
UVLO
HS MOSFET
Current
Comparator
Voltage
Reference
Power Stage
& Deadtime
Control
Logic
SW
Slope
Compensation
VIN
Regulator
Hiccup
Shutdown
Overload
Recovery
Maximum
Clamp
Oscillator
LS MOSFET
Current Limit
Current
Sense
GND
COMP
EXPOSED THERMAL PAD
8.3 Feature Description
8.3.1 Fixed Frequency PWM Control
The device uses a fixed frequency, peak current mode control. The output voltage is compared through external
resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives the COMP pin.
An internal oscillator initiates the turn on of the high side power switch. The error amplifier output is compared to
the high side power switch current. When the power switch current reaches the COMP voltage level the high side
power switch is turned off and the low side power switch is turned on. The COMP pin voltage increases and
decreases as the output current increases and decreases. The device implements a current limit by clamping the
COMP pin voltage to a maximum level and also implements a minimum clamp for improved transient response
performance.
8.3.2 Light Load Operation
The device monitors the peak switch current of the high-side MOSFET. Once the peak switch current is lower
than typically 0.5A, the device stops switching to boost the efficiency until the peak switch current is again higher
than typically 0.5A.
10
Copyright © 2015, Texas Instruments Incorporated
TPS54334
www.ti.com.cn
ZHCSDR7 – MAY 2015
Feature Description (continued)
8.3.3 Slope Compensation and Output Current
The device adds a compensating ramp to the switch current signal. This slope compensation prevents subharmonic oscillations as duty cycle increases. The available peak inductor current remains constant over the full
duty cycle range.
8.3.4 Bootstrap Voltage (BOOT) and Low Dropout Operation
The device has an integrated boot regulator and requires a small ceramic capacitor between the BOOT and SW
pin to provide the gate drive voltage for the high-side MOSFET. The value of the ceramic capacitor should be
0.1μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10 V or higher is
recommended because of the stable characteristics over temperature and voltage.
When the voltage between BOOT and SW pins drops below the BOOT-SW UVLO threshold, which is 2.2 V
(typical), the high-side MOSFET turns off and the low-side MOSFET turns on, allowing the boot capacitor to
recharge.
The device may work at 100% duty ratio as long as the BOOT-SW voltage is higher than the BOOT-SW UVLO
threshold; but, do not operate the device at 100% duty ratio with no load.
8.3.5 Error Amplifier
The device has a transconductance amplifier. The error amplifier compares the VSENSE voltage to the lower of
the internal slow start voltage or the internal 0.8 V voltage reference. The transconductance of the error amplifier
is 1300μA/V typically. The frequency compensation components are placed between the COMP pin and ground.
8.3.6 Voltage Reference
The voltage reference system produces a precise ±1.5% voltage reference over temperature by scaling the
output of a temperature stable bandgap circuit. The bandgap and scaling circuits produce 0.8 V at the noninverting input of the error amplifier.
8.3.7 Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the VSENSE pin. It is recommended to
use divider resistors with 1% tolerance or better. Start with a 10 kΩ for the R1 resistor and use the Equation 1 to
calculate R2. To improve efficiency at light loads consider using larger value resistors. If the values are too high
the regulator is more susceptible to noise and voltage errors from the VSENSE input current are noticeable.
VREF
R2 =
´ R1
VOUT - VREF
(1)
8.3.8 Enable and Undervoltage Lockout
The EN pin provides electrical on/off control of the device. Once the EN pin voltage exceeds the threshold
voltage, the device starts operation. If the EN pin voltage is pulled below the threshold voltage, the regulator
stops switching and enters the low-quiescent (IQ) state.
The EN pin has an internal pull-up current source, allowing the user to float the EN pin for enabling the device. If
an application requires controlling the EN pin, use open drain or open collector output logic to interface with the
pin
The device implements internal UVLO circuitry on the VIN pin. The device is disabled when the VIN pin voltage
falls below the internal VIN UVLO threshold. The internal VIN UVLO threshold has a hysteresis of 180mV.
If an application requires a higher UVLO threshold on the VIN pin, then the EN pin can be configured as shown
in Figure 13. When using the external UVLO function it is recommended to set the hysteresis to be greater than
500mV.
The EN pin has a small pull-up current Ip which sets the default state of the pin to enable when no external
components are connected. The pull-up current is also used to control the voltage hysteresis for the UVLO
function since it increases by Ih once the EN pin crosses the enable threshold. The UVLO thresholds can be
calculated using Equation 2, and Equation 3.
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Feature Description (continued)
TPS54334
VIN
Ip
Ih
R1
R2
EN
Figure 13. Adjustable VIN Undervoltage Lockout
æ VENfalling ö
VSTART ç
÷ - VSTOP
ç VENri sin g ÷
è
ø
R1 =
æ
VENfalling ö
Ip ç 1 ÷ +I
ç
VENri sin g ÷ø h
è
(2)
space
R1´ VENfalling
R2 =
VSTOP - VENfalling + R1(Ip + Ih )
where
•
•
•
•
IP = 1.15 μA
IH = 3.3 μA
VENfalling = 1.17 V
VENrising = 1.21 V
(3)
8.3.9 Slow Start
The internal 2-ms soft-start time is implemented to minimize inrush currents. If during normal operation, the VIN
goes below the UVLO, EN pin pulled below 1.21 V, or a thermal shutdown event occurs, the device stops
switching and the internal slow start voltage is discharged to 0 volts before reinitiating a powering up sequence.
8.3.10 Safe Start-up into Pre-Biased Outputs
The device is designed to prevent the low-side MOSFET from discharging a pre-biased output. During monotonic
pre-biased startup, both high-side and low-side MOSFETs are not allowed to be turned on until the internal softstart voltage is higher than VSENSE pin voltage.
8.3.11 Power Good (PGOOD)
The PGOOD pin is an open drain output. Once the VSENSE pin is between 90% and 110% of the internal
voltage reference the PGOOD pin pull-down is de-asserted and the pin floats. It is recommended to use a pull up
resistor between the values of 10kΩ and 100kΩ to a voltage source that is 5.5V or less. The PGOOD is in a
defined state once the VIN input voltage is greater than 1V but with reduced current sinking capability. The
PGOOD achieves full current sinking capability once the VIN input voltage is above 4.2V.
The PGOOD pin is pulled low when VSENSE is lower than 84% or greater than 116% of the nominal internal
reference voltage. Also, the PGOOD is pulled low, if the input UVLO or thermal shutdown is asserted, the EN pin
is pulled low.
12
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8.4 Device Functional Modes
8.4.1 Overcurrent/Overvoltage/Thermal Protection
The device is protected from output overvoltage, overload and thermal fault conditions. The device minimizes
excessive output overvoltage transients by taking advantage of the overvoltage circuit power good comparator.
When the overvoltage comparator is activated, the high-side MOSFET is turned off and prevented from turning
on until the VSENSE pin voltage is lower than 106% of the Vref. The device implements both high-side MOSFET
overload protection and bidirectional low-side MOSFET overload protections which help control the inductor
current and avoid current runaway. If the overcurrent condition has lasted for more than the hiccup wait time, the
device will shut down and re-start after the hiccup time. The device also shuts down if the junction temperature is
higher than thermal shutdown trip point. When the junction temperature drops 10°C typically below the thermal
shutdown trip point, the built-in thermal shutdown hiccup timer is triggered. The device will be restarted under
control of the slow start circuit automatically after the thermal shutdown hiccup time is over.
Furthermore, if the overcurrent condition has lasted for more than the hiccup wait time which is programmed for
512 switching cycles, the device will shut down itself and re-start after the hiccup time which is set for 16384
cycles. The hiccup mode helps to reduce the device power dissipation under severe overcurrent conditions.
8.4.2 Thermal Shutdown
The internal thermal shutdown circuitry forces the device to stop switching if the junction temperature exceeds
165°C typically. Once the junction temperature drops below 155°C typically, the internal thermal hiccup timer will
start to count. The device reinitiates the power up sequence after the built-in thermal shutdown hiccup time
(32768 cycles) is over.
8.4.3 Small Signal Model for Loop Response
Figure 14 shows an equivalent model for the device control loop which can be modeled in a circuit simulation
program to check frequency response and transient responses. The error amplifier is a trans conductance
amplifier with a gm of 1300μA/V. The error amplifier can be modeled using an ideal voltage controlled current
source. The resistor Roea (3.07 MΩ) and capacitor Coea (20.7 pF) model the open loop gain and frequency
response of the error amplifier. The 1-mV ac voltage source between the nodes a and b effectively breaks the
control loop for the frequency response measurements. Plotting a/c and c/b show the small signal responses of
the power stage and frequency compensation respectively. Plotting a/b shows the small signal response of the
overall loop. The dynamic loop response can be checked by replacing the RL with a current source with the
appropriate load step amplitude and step rate in a time domain analysis.
SW
VOUT
Power Stage
8 A/V
a
b
c
R3 Coea
C2
R1
RESR
VSENSE
CO
COMP
C1
0.8 V
Roea
gm
1300 mA/V
RL
R2
Figure 14. Small Signal Model for Loop Response
8.4.4 Small Signal Model for Peak Current Mode Control
Figure 15 is a simple small signal model that can be used to understand how to design the frequency
compensation. The device power stage can be approximated to a voltage controlled current source (duty cycle
modulator) supplying current to the output capacitor and load resistor. The control to output transfer function is
shown in Equation 4 and consists of a dc gain, one dominant pole and one ESR zero. The quotient of the
change in switch current and the change in COMP pin voltage (node c in Figure 14) is the power stage trans
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Device Functional Modes (continued)
conductance (gmps) which is 8 A/V for the device. The DC gain of the power stage is the product of gmps and the
load resistance, RL, as shown in Equation 6 with resistive loads. As the load current increases, the DC gain
decreases. This variation with load may seem problematic at first glance, but fortunately the dominant pole
moves with load current (see Equation 6). The combined effect is highlighted by the dashed line in Figure 16. As
the load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB crossover
frequency the same for the varying load conditions which makes it easier to design the frequency compensation.
VOUT
VC
RESR
RL
gm ps
CO
Figure 15. Small Signal Model for Peak Current Mode Control
VOUT
Adc
VC
RESR
fp
RL
gmps
CO
fz
Figure 16. Simplified Frequency Response for Peak Current Mode Control
æ
s
ç1 +
2p ´ ¦ z
VOUT
= Adc ´ è
VC
æ
s
ç1 +
ç
2p ´ ¦ p
è
Adc = gmPS ´ RL
ö
÷
ø
ö
÷
÷
ø
1
¦p =
Co ´ RL ´ 2p
¦z =
1
Co ´ RESR ´ 2p
(4)
(5)
(6)
(7)
Where
gmea is the GM amplifier gain (1300µA/V)
gmPS is the power stage gain (8A/V)
14
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Device Functional Modes (continued)
RL is the load resistance
CO is the output capacitance
RESR is the equivalent series resistance of the output capacitors
8.4.5 Small Signal Model for Frequency Compensation
The device uses a transconductance amplifier for the error amplifier and readily supports two of the commonly
used Type II compensation circuits and a Type III frequency compensation circuit, as shown in Figure 17. In
Type 2A, one additional high frequency pole, C6, is added to attenuate high frequency noise. In Type III, one
additional capacitor, C11, is added to provide a phase boost at the crossover frequency. See Designing Type III
Compensation for Current Mode Step-Down Converters (SLVA352) for a complete explanation of Type III
compensation.
The design guidelines below are provided for advanced users who prefer to compensate using the general
method. The below equations only apply to designs whose ESR zero is above the bandwidth of the control loop.
This is usually true with ceramic output capacitors.
VOUT
C11
R8
Type 3
VSENSE
COMP Type 2A
Vref
R9
gmea
Roea
R4
Coea
C6
Type 2B
R4
C4
C4
Figure 17. Types of Frequency Compensation
The general design guidelines for device loop compensation are as follows:
1. Determine the crossover frequency, fc. A good starting point is 1/10th of the switching frequency, fsw.
2. R4 can be determined by:
2p ´ ¦ c ´ VOUT ´ Co
R4 =
gmea ´ Vref ´ gmPS
where
•
•
•
gmea is the GM amplifier gain (1300µA/V)
gmPS is the power stage gain (8A/V)
Vref is the reference voltage (0.8V)
3. Place a compensation zero at the dominant pole:
1
¦p =
Co ´ RL ´ 2p
(8)
(9)
4. C4 can be determined by:
R ´ CO
C4 = L
R4
(10)
5. C6 is optional. It can be used to cancel the zero from the ESR of the output capacitor Co
´ CO
R
C6 = ESR
R4
(11)
6. Type III compensation can be implemented with the addition of one capacitor, C11. This allows for slightly
higher loop bandwidths and higher phase margins. If used, C11 can be estimated from Equation 12.
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Device Functional Modes (continued)
C11 =
1
2p ´ R8 ´ ¦
(12)
C
9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The following design procedure can be used to select component values for the TPS54334. Alternately, the
WEBENCH® software may be used to generate a complete design. The WEBENCH® software uses an iterative
design procedure and accesses a comprehensive database of components when generating a design. This
section presents a simplified discussion of the design process using the TPS54334.
9.2 Typical Applications
9.2.1 TPS54334 Application
U1
TPS54334DRC
VIN
VIN=4.2 V to 24 V
C1
10µF
1
C2
0.1µF
R1
221 kΩ
VIN
PGOOD
VDD
10
C3
BOOT 9
8 EN
SW
7
R2
84.5 kΩ
C5
150 pF
R7 100 kΩ
R3
2.05 kΩ
C4
0.015 µF
COMP
2
L1 6.8 µH
0.1 µF
VOUT= 3.3 V
VSENSE 6
GND
C7
22 µF
VOUT
R4
51.1 Ω
C8
200 pF
R5
31.6 kΩ
C6
22 µF
3,4,5
R6
10 kΩ
Figure 18. Typical Application Schematic, TPS54334
9.2.1.1 Design Requirements
For this design example, use the parameters listed in Table 1.
Table 1. Design Parameters
16
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage range
4.2 to 24 V
Output voltage
3.3 V
Transient response, 1.5-A load step
ΔVO = ±5 %
Input ripple voltage
400 mV
Output ripple voltage
30 mV
Output current rating
3A
Operating Frequency
570 kHz
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9.2.1.2 Detailed Design Procedure
The following design procedure can be used to select component values for the TPS54334. Alternately, the
WEBENCH® software may be used to generate a complete design. The WEBENCH software uses an iterative
design procedure and accesses a comprehensive database of components when generating a design. This
section presents a simplified discussion of the design process using the TPS54334 device.
For this design example, use the input parameters listed in Table 1.
9.2.1.2.1 Switching Frequency
The switching frequency of the TPS54334 device is set at 570 kHz to match the internally set frequency of the
TPS54334 device for this design.
9.2.1.2.2 Output Voltage Set Point
The output voltage of the TPS54334 device is externally adjustable using a resistor divider network. In the
application circuit of Figure 18, this divider network is comprised of R5 and R6. Use Equation 13 and Equation 14
to calculate the relationship of the output voltage to the resistor divider.
R5 ´ Vref
R6 =
VOUT - Vref
(13)
é R5 ù
VOUT = Vref ´ ê
+1ú
ë R6 û
(14)
Select a value of R5 to be approximately 31.6 kΩ. Slightly increasing or decreasing R5 can result in closer
output-voltage matching when using standard value resistors. In this design, R5 = 31.6 kΩ and R6 = 10 kΩ which
results in a 3.328-V output voltage. The 51.1-Ω resistor, R4, is provided as a convenient location to break the
control loop for stability testing.
9.2.1.2.3 Undervoltage Lockout Set Point
The undervoltage lockout (UVLO) set point can be adjusted using the external-voltage divider network of R1 and
R2. R1 is connected between the VIN and EN pins of the TPS54334 device. R2 is connected between the EN
and GND pins. The UVLO has two thresholds, one for power up when the input voltage is rising and one for
power down or brown outs when the input voltage is falling. For the example design, the minimum input voltage
is 4.2V, so the start-voltage threshold is set to 4.1 V and the stop-voltage threshold is set to VIN UVLO (3.7V).
Use Equation 2 and Equation 3 to calculate the values for the upper and lower resistor values of R1 and R2.
9.2.1.2.4 Input Capacitors
The TPS54334 device requires an input decoupling capacitor and, depending on the application, a bulk input
capacitor. The typical recommended value for the decoupling capacitor is 10 μF. A high-quality ceramic type X5R
or X7R is recommended. The voltage rating should be greater than the maximum input voltage. A smaller value
can be used as long as all other requirements are met; however a 10-μF capacitor has been shown to work well
in a wide variety of circuits. Additionally, some bulk capacitance may be needed, especially if the TPS54334
circuit is not located within about 2 inches from the input voltage source. The value for this capacitor is not critical
but should be rated to handle the maximum input voltage including ripple voltage, and should filter the output so
that input ripple voltage is acceptable. For this design, a 10-μF, X7R dielectric capacitor rated for 35 V is used for
the input decoupling capacitor. The ESR is approximately 2 mΩ, and the current rating is 3 A. Additionally, a
small 0.1-μF capacitor is included for high frequency filtering.
Use Equation 15 to calculate the input ripple voltage (ΔVIN).
IOUT(MAX) ´ 0.25
DVIN =
+ IOUT(MAX) ´ ESRMAX
CBULK ´ ƒSW
(
)
where
•
•
•
•
CBULK is the bulk capacitor value
ƒSW is the switching frequency
IOUT(MAX) is the maximum load current
ESRMAX is the maximum series resistance of the bulk capacitor
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The maximum RMS (root mean square) ripple current must also be checked. For worst case conditions, use
Equation 16 to calculate ICIN(RMS).
IO(MAX)
ICIN(RMS) =
(16)
2
In this case, the input ripple voltage is 138 mV and the RMS ripple current is 1.5 A.
NOTE
The actual input-voltage ripple is greatly affected by parasitics associated with the layout
and the output impedance of the voltage source.
Design Requirements shows the actual input voltage ripple for this circuit which is larger than the calculated
value. This measured value is still below the specified input limit of 400 mV. The maximum voltage across the
input capacitors is VIN(MAX) + ΔVIN / 2. The selected bypass capacitor is rated for 35 V and the ripple current
capacity is greater than 3 A. Both values provide ample margin. The maximum ratings for voltage and current
must not be exceeded under any circumstance.
9.2.1.2.5 Output Filter Components
Two components must be selected for the output filter, the output inductor (LO) and CO. Because the TPS54334
device is an externally compensated device, a wide range of filter component types and values can be
supported.
9.2.1.2.5.1 Inductor Selection
Use Equation 17 to calculate the minimum value of the output inductor (LMIN).
LMIN =
VOUT ´
(VIN(MAX) - VOUT )
VIN(MAX) ´ KIND ´ IOUT ´ ƒSW
where
•
KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output
current
(17)
In general, the value of KIND is at the discretion of the designer; however, the following guidelines may be used.
For designs using low-ESR output capacitors, such as ceramics, a value as high as KIND = 0.3 can be used.
When using higher ESR output capacitors, KIND = 0.2 yields better results.
For this design example, use KIND = 0.3. The minimum inductor value is calculated as 5.6 μH. For this design, a
standard value of 6.8 µH was selected for LMIN.
For the output filter inductor, the RMS current and saturation current ratings must not be exceeded. Use
Equation 18 to calculate the RMS inductor current (IL(RMS)).
IL(RMS) =
18
2
IOUT(MAX)
(
)
æ VOUT ´ VIN(MAX) - VOUT ö
1
÷
´ ç
+
ç VIN(MAX) ´ LOUT ´ ƒSW ´ 0.8 ÷
12
è
ø
2
(18)
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Use Equation 19 to calculate the peak inductor current (IL(PK)).
IL(PK) = IOUT(MAX) +
VOUT ´
(VIN(MAX)
- VOUT
)
1.6 ´ VIN(MAX) ´ LOUT ´ ƒSW
(19)
For this design, the RMS inductor current is 3.01 A and the peak inductor current is 3.459 A. The selected
inductor is a Vishay 6.8 μH, IHLP-4040DZ-01. Smaller or larger inductor values can be used depending on the
amount of ripple current the designer wants to allow so long as the other design requirements are met. Larger
value inductors have lower AC current and result in lower output voltage ripple. Smaller inductor values increase
AC current and output voltage ripple. In general, for the TPS54334 device, use inductors with values in the range
of 0.68 μH to 100 μH.
9.2.1.2.5.2 Capacitor Selection
Consider three primary factors when selecting the value of the output capacitor. The output capacitor determines
the modulator pole, the output voltage ripple, and how the regulator responds to a large change in load current.
The output capacitance must be selected based on the more stringent of these three criteria.
The desired response to a large change in the load current is the first criterion. The output capacitor must supply
the load with current when the regulator cannot. This situation occurs if the desired hold-up times are present for
the regulator. In this case, the output capacitor must hold the output voltage above a certain level for a specified
amount of time after the input power is removed. The regulator is also temporarily unable to supply sufficient
output current if a large, fast increase occurs affecting the current requirements of the load, such as a transition
from no load to full load. The regulator usually requires two or more clock cycles for the control loop to notice the
change in load current and output voltage and to adjust the duty cycle to react to the change. The output
capacitor must be sized to supply the extra current to the load until the control loop responds to the load change.
The output capacitance must be large enough to supply the difference in current for 2 clock cycles while only
allowing a tolerable amount of drop in the output voltage. Use Equation 20 to calculate the minimum required
output capacitance.
2 ´ DIOUT
CO >
ƒSW ´ DVOUT
where
•
•
•
ΔIOUT is the change in output current
ƒSW is the switching frequency of the regulator
ΔVOUT is the allowable change in the output voltage
(20)
For this example, the transient load response is specified as a 5% change in the output voltage, VOUT, for a load
step of 1.5 A. For this example, ΔIOUT = 1.5 A and ΔVOUT = 0.05 × 3.3 = 0.165 V. Using these values results in a
minimum capacitance of 31.9 μF. This value does not consider the ESR of the output capacitor in the output
voltage change. For ceramic capacitors, the ESR is usually small enough to ignore in this calculation.
Equation 21 calculates the minimum output capacitance required to meet the output voltage ripple specification.
In this case, the maximum output voltage ripple is 30 mV. Under this requirement, Equation 21 yields 3.65 µF.
1
1
CO >
´
8 ´ ƒSW VOUTripple
Iripple
where
•
•
•
ƒSW is the switching frequency
VOUTripple is the maximum allowable output voltage ripple
Iripple is the inductor ripple current
(21)
Use Equation 22 to calculate the maximum ESR an output capacitor can have to meet the output-voltage ripple
specification. Equation 22 indicates the ESR should be less than 40.9 mΩ. In this case, the ESR of the ceramic
capacitor is much smaller than 40.9 mΩ.
VOUTripple
RESR <
Iripple
(22)
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Additional capacitance deratings for aging, temperature, and DC bias should be considered which increases this
minimum value. For this example, two 22-μF 25-V X7R ceramic capacitors with 3 mΩ of ESR are used.
Capacitors generally have limits to the amount of ripple current they can handle without failing or producing
excess heat. An output capacitor that can support the inductor ripple current must be specified. Some capacitor
data sheets specify the RMS value of the maximum ripple current. Use Equation 23 to calculate the RMS ripple
current that the output capacitor must support. For this application, Equation 23 yields 106 mA for each
capacitor.
ICOUT(RMS) =
(
)
æ VOUT ´ VIN(MAX) - VOUT
´ ç
ç VIN(MAX) ´ LOUT ´ ƒSW ´ NC
12
è
1
ö
÷
÷
ø
(23)
9.2.1.2.6 Compensation Components
Several possible methods exist to design closed loop compensation for DC-DC converters. For the ideal currentmode control, the design equations can be easily simplified. The power stage gain is constant at low frequencies,
and rolls off at –20 dB/decade above the modulator pole frequency. The power stage phase is 0 degrees at low
frequencies and begins to fall one decade below the modulator pole frequency reaching a minimum of –90
degrees which is one decade above the modulator pole frequency. Use Equation 24 to calculate the simple
modulator pole (ƒp_mod).
IOUT max
ƒp_mod =
2p ´ VOUT ´ COUT
(24)
For the TPS54334 device, most circuits have relatively high amounts of slope compensation. As more slope
compensation is applied, the power stage characteristics deviate from the ideal approximations. The phase loss
of the power stage will now approach –180 degrees, making compensation more difficult. The power stage
transfer function can be solved but it requires a tedious calculation. Use the PSpice model to accurately model
the power-stage gain and phase so that a reliable compensation circuit can be designed. Alternately, a direct
measurement of the power stage characteristics can be used which is the technique used in this design
procedure. For this design, the calculated values are as follows:
L1 = 6.8 µH
C6 and C7 = 22 µF
ESR = 3 mΩ
Figure 19 shows the power stage characteristics.
180
Gain = –2.088 dB
@ ƒ = 54.26 kHz
Gain (dB)
40
120
20
60
0
0
-20
Phase (q)
60
-60
-40
-120
Gain
Phase
-60
10
100
1k
Frequency (Hz)
10k
-180
100k
D016
Figure 19. Power Stage Gain and Phase Characteristics
For this design, the intended crossover frequency is 54.26 kHz (an actual measured data point exists for that
frequency). From the power stage gain and phase plots, the gain at 54.26 kHz is –2.088 dB and the phase is
about –121 degrees. For 60 degrees of phase margin, additional phase boost from a feed-forward capacitor in
parallel with the upper resistor of the voltage set point divider is needed. R3 sets the gain of the compensated
error amplifier to be equal and opposite the power stage gain at crossover. Use Equation 25 to calculate the
required value of R3.
20
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ZHCSDR7 – MAY 2015
10
R3 =
-GPWRSTG
20
gmea
´
VOUT
VREF
(25)
To maximize phase gain, the compensator zero is placed one decade below the crossover frequency of 54.26
kHz. Use Equation 26 to calculate the required value for C4.
1
C4 =
ƒ
2 ´ p ´ R3 ´ CO
10
(26)
To maximize phase gain the high frequency pole is placed one decade above the crossover frequency of 54.26
kHz. The pole can also be useful to offset the ESR of aluminum electrolytic output capacitors. Use Equation 27
to calculate the value of C5.
1
C5 =
2 ´ p ´ R3 ´ 10 ´ ƒCO
(27)
To Maximize Phase margin, use Type-lll compensation to provide a zero around the desired crossover frequency
(ƒco) with R5, VOUT and VREF.
C8 =
1
´
2p ´ R5 ´ ¦ CO
VOUT
VREF
(28)
For this design the calculated values for the compensation components are as follows:
R3 = 2.05 kΩ
C4 = 0.015 µF
C5 = 150 pF
C8 = 200 pF
9.2.1.2.7 Bootstrap Capacitor
Every TPS54334 design requires a bootstrap capacitor, C3. The bootstrap capacitor value must 0.1 μF. The
bootstrap capacitor is located between the SW and BOOT pins. The bootstrap capacitor should be a high-quality
ceramic type with X7R or X5R grade dielectric for temperature stability.
9.2.1.2.8 Power Dissipation Estimate
The following formulas show how to estimate the device power dissipation under continuous-conduction mode
operations. These formulas should not be used if the device is working in the discontinuous conduction mode
(DCM) or pulse-skipping Eco-mode™.
The device power dissipation includes:
1. Conduction loss:
PCON = IOUT2 × rDS(on) × VOUT / VIN
where
•
•
•
•
IOUT is the output current (A)
rDS(on) is the on-resistance of the high-side MOSFET (Ω)
VOUT is the output voltage (V)
VIN is the input voltage (V)
(29)
2. Switching loss:
E = 0.5 × 10–9 × VIN 2 × IOUT × ƒSW
where
•
ƒSW is the switching frequency (Hz)
(30)
3. Gate charge loss:
PG = 22.8 × 10–9 × ƒSW
(31)
4. Quiescent current loss:
PQ = 0.31 × 10-3 × VIN
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(32)
21
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
Therefore:
Ptot = PCON + E + PG + PQ
where
•
Ptot is the total device power dissipation (W)
(33)
For given TA :
TJ = TA + Rth × Ptot
where
•
•
•
TA is the ambient temperature (°C)
TJ is the junction temperature (°C)
Rth is the thermal resistance of the package (°C/W)
(34)
For given TJmax = 150°C:
TAmax = TJmax – Rth × Ptot
where
•
•
TAmax is the maximum ambient temperature (°C)
TJmax is the maximum junction temperature (°C)
(35)
100
100
90
90
80
80
70
70
Efficiency %
Efficiency %
9.2.1.3 Application Curves
60
50
40
60
50
40
30
30
20
20
VIN = 12 V
VIN = 24 V
10
0
1m
0
0
0.5
1
1.5
2
Output Current (A)
2.5
VIN = 12 V
VIN = 24 V
10
3
Figure 20. TPS54334 Efficiency
1
3
D018
0.1
VIN = 12 V
VIN = 24 V
0.4
0.08
0.3
0.06
0.2
0.04
Line Regulation %
Load Regulation %
100m
Output Current (A)
Figure 21. TPS54334 Low-Current Efficiency
0.5
0.1
0
-0.1
-0.2
0.02
0
-0.02
-0.04
-0.3
-0.06
-0.4
-0.08
-0.1
-0.5
0
0.5
1
1.5
2
Output Current (A)
2.5
Figure 22. TPS54334 Load Regulation
22
10m
D017
3
D019
4
6
8
10
12
14
16
Input Voltage (V)
18
20
22
24
D020
Figure 23. TPS54334 Line Regulation
Copyright © 2015, Texas Instruments Incorporated
TPS54334
ZHCSDR7 – MAY 2015
Gain (dB)
VOUT = 200 mV/div(AC coupled)
60
180
40
120
20
60
0
0
-20
Phase (q)
www.ti.com.cn
-60
IOUT = 1 A/div
-40
-120
Gain
Phase
Time = 200 µs/div
0.75- to 2.25-A load step
Slew rate = 500 mA/µs
-60
100
1k
10k
Frequency (Hz)
VOUT = 20 mV/div (AC coupled)
VOUT = 20 mV/div (AC coupled)
SW = 5 V/div
SW = 5 V/div
Time = 1 µs/div
Figure 27. TPS54334 200-mA Output Ripple
VIN = 200 mV/div (AC coupled)
SW = 5 V/div
SW = 5 V/div
Figure 28. TPS54334 No-Load Output Ripple
Copyright © 2015, Texas Instruments Incorporated
D021
Time = 2 µs/div
VOUT = 20 mV/div (AC coupled)
Time = 2 ms/div
-180
1M
Figure 25. TPS54334 Loop Response
Figure 24. TPS54334 Transient Response
Figure 26. TPS54334 Full-Load Output Ripple
100k
Time = 1 µs/div
Figure 29. TPS54334 Full-Load Input Ripple
23
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
VIN = 5 V/div
VIN = 5 V/div
EN = 2 V/div
EN = 2 V/div
PGOOD = 2 V/div
PGOOD = 2 V/div
VOUT = 2 V/div
VOUT = 2 V/div
Time = 2 ms/div
Time = 2 ms/div
Figure 30. TPS54334 Startup Relative To VIN
Figure 31. TPS54334 Startup Relative To Enable
VIN = 5 V/div
VIN = 5 V/div
EN = 2 V/div
EN = 2 V/div
PGOOD = 2 V/div
PGOOD = 2 V/div
VOUT = 2 V/div
VOUT = 2 V/div
Time = 2 ms/div
Figure 32. TPS54334 Shutdown Relative To VIN
Time = 2 ms/div
Figure 33. TPS54334 Shutdown Relative To EN
10 Power Supply Recommendations
The devices are designed to operate from an input supply ranging from 4.2 V to 28 V. The input supply should
be well regulated. If the input supply is located more than a few inches from the converter an additional bulk
capacitance, typically 100 μF, may be required in addition to the ceramic bypass capacitors.
11 Layout
11.1 Layout Guidelines
The VIN pin should be bypassed to ground with a low ESR ceramic bypass capacitor. Care should be taken to
minimize the loop area formed by the bypass capacitor connection. the VIN pin, and the GND pin of the IC. The
typical recommended bypass capacitance is 10-μF ceramic with a X5R or X7R dielectric and the optimum
placement is closest to the VIN and GND pins of the device. See Figure 34 for a PCB layout example. The GND
pin should be tied to the PCB ground plane at the pin of the IC. To facilitate close placement of the input bypass
capacitors, The SW pin should be routed to a small copper area directly adjacent to the pin. Use vias to rout the
SW signal to the bottom side or an inner layer. If necessary you can allow the top side copper area to extend
slightly under the body of the closest input bypass capacitor. Make the copper trace on the bottom or internal
layer short and wide as practical to reduce EMI issues. Connect the trace with vias back to the top side to
connect with the output inductor as shown after the GND pin. In the same way use a bottom or internal layer
trace to rout the SW signal across the VIN pin to connect to the BOOT capacitor as shown. Make the circulating
loop from SW to the output inductor, output capacitors and back to GND as tight as possible while preserving
adequate etch width to reduce conduction losses in the copper.
24
Copyright © 2015, Texas Instruments Incorporated
TPS54334
www.ti.com.cn
ZHCSDR7 – MAY 2015
Layout Guidelines (continued)
For operation at full rated load, the ground area near the IC must provide adequate heat dissipating area.
Connect the exposed thermal pad to bottom or internal layer ground plane using vias as shown. Additional vias
may be used adjacent to the IC to tie top side copper to the internal or bottom layer copper. The additional
external components can be placed approximately as shown. Use a separate ground trace to connect the
feedback, compensation, UVLO and RT returns. Connect this ground trace to the main power ground at a single
point to minimize circulating currents. It may be possible to obtain acceptable performance with alternate layout
schemes, however this layout has been shown to produce good results and is intended as a guideline.
11.2 Layout Example
VIA to Power Ground Plane
VIA to SW Copper Pour on Bottom
or Internal Layer
VIN
VIN
INPUT
BYPASS
CAPACITOR
Connect to VIN on
internal or bottom
layer
VIN
HIGH FREQUENCY
BOOT
BYPASS
CAPACITOR
CAPACITOR
ANALOG
GROUND
TRACE
PGOOD
PULLUP RESISTOR
VDD
BOOT
PGOOD
VIN
EN
SW
COMP
GND
VSENSE
UVLO
RESISTORS
COMPENSATION
NETWORK
EXPOSED
THERMAL PAD
AREA
POWER
GROUND
FEEDBACK
RESISTORS
OUTPUT
INDUCTOR
SW node copper pour
area on internal or
bottom layer
POWER
GROUND
VOUT
OUTPUT
FILTER
CAPACITOR
Figure 34. TPS54334DDA Board Layout
版权 © 2015, Texas Instruments Incorporated
25
TPS54334
ZHCSDR7 – MAY 2015
www.ti.com.cn
12 器件和文档支持
12.1 器件支持
12.1.1 开发支持
如需 WEBENCH 电路设计和选择仿真服务，请访问：www.ti.com/WEBENCH。
12.2 文档支持
12.2.1 相关文档
相关文档如下：
《设计电流模式降压转换器的第 III 类补偿》（文献编号：SLVA352）
12.3 社区资源
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 商标
SWIFT, PowerPAD, Eco-mode, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 静电放电警告
这些装置包含有限的内置 ESD 保护。 存储或装卸时，应将导线一起截短或将装置放置于导电泡棉中，以防止 MOS 门极遭受静电损
伤。
12.6 术语表
SLYZ022 — TI 术语表。
这份术语表列出并解释术语、首字母缩略词和定义。
13 机械、封装和可订购信息
以下页中包括机械、封装和可订购信息。 这些信息是针对指定器件可提供的最新数据。 这些数据会在无通知且不
对本文档进行修订的情况下发生改变。 欲获得该数据表的浏览器版本，请查阅左侧的导航栏。
26
版权 © 2015, Texas Instruments Incorporated
PACKAGE OPTION ADDENDUM
www.ti.com
11-Aug-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS54334DDA
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 85
54334
Samples
TPS54334DDAR
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
SN
Level-2-260C-1 YEAR
-40 to 85
54334
Samples
TPS54334DRCR
ACTIVE
VSON
DRC
10
3000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
54334
Samples
TPS54334DRCT
ACTIVE
VSON
DRC
10
250
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
54334
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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