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TPS54360B-Q1
SLVSDV1 – FEBRUARY 2017
TPS54360B-Q1 60 V Input, 3.5 A, Step Down DC-DC Converter with Eco-mode™
1 Features
3 Description
•
•
The TPS54360B-Q1 is a 60-V 3.5-A step-down
regulator with an integrated high-side MOSFET. The
device survives load dump pulses up to 65 V per ISO
7637. Current mode control provides simple external
compensation and flexible component selection. A
low-ripple pulse-skip mode reduces the no-load
supply current to 146 μA. Shutdown supply current is
reduced to 2 μA when the enable pin is pulled low.
1
•
•
•
•
•
•
•
•
•
•
Qualified for Automotive Applications
AEC-Q100 Qualified With the Following Results:
– Device Temperature Grade 1: –40°C to 125°C
Ambient Operating Temperature Range
– Device HBM ESD Classification Level H1C
– Device CDM ESD Classification Level C3B
High Efficiency at Light Loads With Pulse Skipping
Eco-Mode™
92-mΩ High-Side MOSFET
146-μA Operating Quiescent Current and 2 µA
Shutdown Current
100-kHz to 2.5-MHz Adjustable Switching
Frequency
Synchronizes to External Clock
Low Dropout at Light Loads With Integrated
BOOT Recharge FET
Adjustable UVLO Voltage and Hysteresis
0.8-V 1% Internal Voltage Reference
8-Pin HSOP With PowerPAD™ Package
–40°C to 150°C TJ Operating Range
Undervoltage lockout is internally set at 4.3 V but can
be increased using an external resistor divider at the
enable pin. The output voltage start-up ramp is
internally controlled to provide a controlled start up
and eliminate overshoot.
A wide adjustable frequency range allows either
efficiency or external component size to be optimized.
Frequency foldback and thermal shutdown protects
internal and external components during an overload
condition.
The TPS54360B-Q1 is available in an 8-pin thermally
enhanced HSOP PowerPAD package.
Device Information(1)
PART NUMBER
TPS54360B-Q1
2 Applications
•
•
•
•
Vehicle Accessories: GPS (See SLVA412),
Entertainment, ADAS, eCall
USB Dedicated Charging Ports and Battery
Chargers (See SLVA464)
Industrial Automation and Motor Controls
12-V, 24-V, and 48-V Industrial, Automotive, and
Communications Power Systems
PACKAGE
BODY SIZE (NOM)
HSOP (8)
4.89 mm x 3.90 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
Simplified Schematic
Efficiency vs Load Current
100
VIN
VIN
90
80
TPS54360B-Q1
BOOT
VOUT
RT/CLK
SW
Efficiency - %
70
EN
5V
3.3 V
60
50
40
VIN = 12 V
30
20
COMP
FB
VOUT = 5 V, fsw = 600 kHz
10
VOUT = 3.3 V, fsw = 300 kHz
0
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
GND
IO - Output Current - A
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54360B-Q1
SLVSDV1 – FEBRUARY 2017
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
6.7
4
4
4
4
5
6
7
Absolute Maximum Ratings ......................................
ESD Ratings..............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Typical Characteristics ..............................................
8
8.1 Application Information............................................ 23
8.2 Typical Application .................................................. 23
9 Power Supply Recommendations...................... 35
10 Layout................................................................... 35
10.1 Layout Guidelines ................................................. 35
10.2 Layout Example .................................................... 35
10.3 Estimated Circuit Area .......................................... 36
11 Device and Documentation Support ................. 36
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Detailed Description ............................................ 11
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
Application and Implementation ........................ 23
11
12
12
22
Device Support ....................................................
Documentation Support .......................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
36
36
36
36
36
36
36
12 Mechanical, Packaging, and Orderable
Information ........................................................... 37
4 Revision History
2
DATE
REVISION
NOTES
February 2017
*
Initial release.
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SLVSDV1 – FEBRUARY 2017
5 Pin Configuration and Functions
DDA Package
HSOP (8 Pin)
(Top View)
BOOT
1
VIN
2
8
SW
7
GND
6
COMP
5
FB
Thermal
EN
3
RT/CLK
4
Pad
Not to scale
Pin Functions
PIN
NAME
NO.
I/O
DESCRIPTION
BOOT
1
O
A bootstrap capacitor is required between BOOT and SW. If the voltage on this capacitor is below the
minimum required to operate the high-side MOSFET, the output is switched off until the capacitor is
refreshed.
VIN
2
I
Input supply voltage with 4.5-V to 60-V operating range.
EN
3
I
Enable pin, with internal pullup current source. Pull below 1.2 V to disable. Float to enable. Adjust the input
undervoltage lockout with two resistors. See the Enable and Adjusting Undervoltage Lockout section.
RT/CLK
4
I
Resistor Timing and External Clock. An internal amplifier holds this pin at a fixed voltage when using an
external resistor to ground to set the switching frequency. If the pin is pulled above the PLL upper
threshold, a mode change occurs and the pin becomes a synchronization input. The internal amplifier is
disabled and the pin is a high impedance clock input to the internal PLL. If clocking edges stop, the internal
amplifier is re-enabled and the operating mode returns to resistor frequency programming.
FB
5
I
Inverting input of the transconductance (gm) error amplifier.
COMP
6
O
Error amplifier output and input to the output switch current (PWM) comparator. Connect frequency
compensation components to this pin.
GND
7
–
Ground
SW
8
I
The source of the internal high-side power MOSFET and switching node of the converter.
Thermal
Pad
–
–
GND pin must be electrically connected to the exposed pad on the printed circuit board for proper
operation.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
Input voltage
Output voltage
(1)
MIN
MAX
VIN
–0.3
65
EN
–0.3
8.4
FB
–0.3
3
COMP
–0.3
3
RT/CLK
–0.3
3.6
BOOT-SW
–0.3
8
SW
–0.6
65
SW, 10-ns Transient
UNIT
V
V
–2
65
Storage temperature range, Tstg
–65
150
°C
Operating junction temperature, TJ
-40
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic discharge
Human body model (HBM), per AEC Q100-002 (1)
±2000
Charged device model (CDM), per AEC Q100-011
±500
UNIT
V
AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VI Input voltage range
(1)
TJ Operating junction temperature
(1)
MAX
UNIT
VO + Vdo
60
V
–40
150
°C
See Equation 1 in the Feature Description section
6.4 Thermal Information
TPS54360B-Q1
THERMAL METRIC (1)
DDA (HSOP)
UNIT
8 PINS
θJA
Junction-to-ambient thermal resistance (standard board)
42
ψJT
Junction-to-top characterization parameter
5.9
ψJB
Junction-to-board characterization parameter
23.4
θθJC(top)
Junction-to-case (top) thermal resistance
45.8
θθJC(bot)
Junction-to-case (bottom) thermal resistance
3.6
θJB
Junction-to-board thermal resistance
23.4
(1)
4
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SLVSDV1 – FEBRUARY 2017
6.5 Electrical Characteristics
TJ = –40°C to 150°C, VIN = 4.5 to 60 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
Operating input voltage
Internal undervoltage lockout
threshold
4.5
Rising
4.1
Internal undervoltage lockout
threshold hysteresis
4.3
60
V
4.48
V
325
mV
Shutdown supply current
EN = 0 V, 25°C, 4.5 V ≤ VIN ≤ 60 V
2.25
4.5
Operating: nonswitching supply
current
FB = 0.9 V, TA = 25°C
146
175
1.2
1.3
μA
ENABLE AND UVLO (EN PIN)
Enable threshold voltage
Input current
No voltage hysteresis, rising and falling
–4.6
Enable threshold –50 mV
Hysteresis current
Enable to COMP active
1.1
Enable threshold +50 mV
–0.58
–1.2
–1.8
–2.2
–3.4
-4.5
VIN = 12 V, TA = 25°C
V
μA
μA
346
µs
INTERNAL SOFT-START TIME
Soft-Start Time
fSW = 500 kHz, 10% to 90%
2.1
ms
Soft-Start Time
fSW = 2.5 MHz, 10% to 90%
0.42
ms
VOLTAGE REFERENCE
Voltage reference
0.792
0.8
0.808
92
190
V
HIGH-SIDE MOSFET
On-resistance
VIN = 12 V, BOOT-SW = 6 V
mΩ
ERROR AMPLIFIER
Input current
Error amplifier transconductance
(gM)
–2 μA < ICOMP < 2 μA, VCOMP = 1 V
Error amplifier transconductance
(gM) during soft-start
–2 μA < ICOMP < 2 μA, VCOMP = 1 V, VFB = 0.4 V
Error amplifier DC gain
VFB = 0.8 V
Min unity gain bandwidth
Error amplifier source/sink
V(COMP) = 1 V, 100 mV overdrive
COMP to SW current
transconductance
50
nA
350
μS
77
μS
10,000
V/V
2500
kHz
±30
μA
12
A/V
CURRENT LIMIT
Current limit threshold
All VIN and temperatures, Open-loop (1)
4.5
5.5
6.8
All temperatures, VIN = 12 V, Open-loop (1)
4.5
5.5
6.25
5.2
5.5
5.85
VIN = 12 V, TA = 25°C, Open-loop
(1)
Current limit threshold delay
A
60
ns
176
°C
12
°C
THERMAL SHUTDOWN
Thermal shutdown
Thermal shutdown hysteresis
(1)
Open-loop current limit measured directly at the SW pin and is independent of the inductor value and slope compensation.
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6.6 Timing Requirements
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
ƒSW
MIN
Switching frequency range using RT mode
100
Switching frequency
450
RT = 200 kΩ
Switching frequency range using CLK mode
500
160
Minimum CLK input pulse width
MAX
UNIT
2500
kHz
550
kHz
2300
kHz
15
RT/CLK high threshold
1.55
RT/CLK low threshold
6
TYP
0.5
ns
2
V
1.2
V
RT/CLK falling edge to SW rising edge
delay
Measured at 500 kHz with RT resistor in
series
55
ns
PLL lock in time
Measured at 500 kHz
78
μs
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6.7 Typical Characteristics
0.814
VIN = 12 V
BOOT-SW = 3 V
BOOT-SW = 6 V
VFB − Voltage Reference (V)
RDS(ON) − On-State Resistance (Ω)
0.25
0.2
0.15
0.1
0.05
0
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
0.809
0.804
0.799
0.794
0.789
0.784
−50
Figure 1. On-Resistance vs Junction Temperature
High-Side Switch Current (A)
High-Side Switch Current (A)
150
G002
TJ = −40°C
TJ = 25°C
TJ = 150°C
6.3
6.1
5.9
5.7
5.5
5.3
5.1
4.9
4.7
6.1
5.9
5.7
5.5
5.3
5.1
4.9
4.7
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
4.5
150
0
10
G003
Figure 3. High-side Switch Current Limit vs Junction
Temperature
500
ƒSW − Switching Frequency (kHz)
RT = 200 kΩ, VIN = 12 V
540
530
520
510
500
490
480
470
460
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
50
60
G004
ƒSW (kHz) = 92417 × RT (kΩ)−0.991
RT (kΩ) = 101756 × fSW (kHz)−1.008
450
400
350
300
250
200
150
100
50
0
200
300
G005
Figure 5. Switching Frequency vs Junction Temperature
20
30
40
VIN − Input Voltage (V)
Figure 4. High-side Switch Current Limit vs Input Voltage
550
ƒSW − Switching Frequency (kHz)
125
6.5
VIN = 12 V
6.3
450
−50
0
25
50
75
100
TJ − Junction Temperature (°C)
Figure 2. Voltage Reference vs Junction Temperature
6.5
4.5
−50
−25
G001
400
500
600
700
800
RT/CLK − Resistance (kΩ)
900
1000
G006
Figure 6. Switching Frequency vs RT/CLK Resistance Low
Frequency Range
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Typical Characteristics (continued)
500
VIN = 12 V
450
2000
400
1500
gm (uS)
ƒSW − Switching Frequency (kHz)
2500
1000
350
300
500
0
250
0
50
100
150
RT/CLK − Resistance (kΩ)
200
−50
200
Figure 7. Switching Frequency vs RT/CLK Resistance High
Frequency Range
VIN = 12 V
EN − Threshold (V)
100
gm (uS)
90
80
70
60
50
40
30
20
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
150
G008
VIN = 12 V
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
G010
−4
VIN = 5 V, I EN = Threshold-50mV
−0.7
−0.9
−4.2
−1.1
−4.3
−1.3
−4.4
−1.5
−1.7
−4.5
−4.6
−1.9
−4.7
−2.1
−4.8
−2.3
−4.9
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
VIN = 12 V, I EN = Threshold+50mV
−4.1
IEN (µA)
IEN (µA)
125
Figure 10. EN Pin Voltage vs Junction Temperature
−0.5
150
−5
−50
−25
G011
Figure 11. EN Pin Current vs Junction Temperature
8
1.3
1.29
1.28
1.27
1.26
1.25
1.24
1.23
1.22
1.21
1.2
1.19
1.18
1.17
1.16
1.15
−50
G009
Figure 9. EA Transconductance During Soft-Start vs
Junction Temperature
−2.5
−50
0
25
50
75
100
TJ − Junction Temperature (°C)
Figure 8. EA Transconductance vs Junction Temperature
120
110
−25
G007
0
25
50
75
100
Tj − Junction Temperature (°C)
125
150
G012
Figure 12. EN Pin Current vs Junction Temperature
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Typical Characteristics (continued)
100
−2.7
−2.9
−3.1
−3.3
−3.5
−3.7
−3.9
−4.1
−4.3
VIN = 12 V
−4.5
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
VFB Falling
VFB Rising
% of Nominal Switching Frequency
EN PIN Current Hysteresis (µA)
−2.5
75
50
25
0
150
0
0.1
0.2
G112
Figure 13. EN Pin Current Hysteresis vs Junction
Temperature
0.3
0.4
VFB (V)
0.5
G013
TJ = 25°C
Shutdown Supply Current (µA)
Shutdown Supply Current (µA)
0.8
3
VIN = 12 V
2.5
2
1.5
1
0.5
0
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
2.5
2
1.5
1
0.5
0
150
0
10
G014
Figure 15. Shutdown Supply Current vs Junction
Temperature
20
30
40
VIN − Input Voltage (V)
50
60
G015
Figure 16. Shutdown Supply Current vs Input Voltage (VIN)
210
210
VIN = 12 V
TJ = 25°C
190
190
VIN − Supply Current (µA)
VIN − Supply Current (uA)
0.7
Figure 14. Switching Frequency vs FB
3
170
150
130
110
90
70
−50
0.6
170
150
130
110
90
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
70
0
G016
Figure 17. VIN Supply Current vs Junction Temperature
10
20
30
40
VIN − Input Voltage (V)
50
60
G017
Figure 18. VIN Supply Current vs Input Voltage
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Typical Characteristics (continued)
2.6
4.4
2.4
4.3
VIN UVLO (V)
BOOT-SW UVLO (V)
2.5
4.5
BOOT-SW UVLO Falling
BOOT-SW UVLO Rising
2.3
2.2
2.1
4.2
4.1
4
2
3.9
1.9
3.8
1.8
−50
−25
0
25
50
75
100
TJ − Junction Temperature (°C)
125
150
3.7
−50
UVLO Start Switching
UVLO Stop Switching
−25
G018
Figure 19. BOOT-SW UVLO vs Junction Temperature
0
25
50
75
100
Tj − Junction Temperature (°C)
125
150
G019
Figure 20. Input Voltage UVLO vs Junction Temperature
10
VIN = 12V,
o
TJ = 25 C
9
Soft-Start Time (ms)
8
7
6
5
4
3
2
1
0
100 300 500 700 900 11001300 1500 17001900 2100 2300 2500
Switching Frequency (kHz)
G021
Figure 21. Soft-Start Time vs Switching Frequency
10
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7 Detailed Description
7.1 Overview
The TPS54360B-Q1 is a 60-V, 3.5-A, step-down (buck) regulator with an integrated high-side n-channel
MOSFET. The device implements constant frequency, current mode control which reduces output capacitance
and simplifies external frequency compensation. The wide switching frequency range of 100 kHz to 2500 kHz
allows either efficiency or size optimization when selecting the output filter components. The switching frequency
is adjusted using a resistor to ground connected to the RT/CLK pin. The device has an internal phase-locked
loop (PLL) connected to the RT/CLK pin that synchronizes the power switch turn on to a falling edge of an
external clock signal.
The TPS54360B-Q1 has a default input start-up voltage of approximately 4.3 V. The EN pin can be used to
adjust the input voltage undervoltage lockout (UVLO) threshold with two external resistors. An internal pull up
current source enables operation when the EN pin is floating. The operating current is 146 μA under no load
condition (not switching). When the device is disabled, the supply current is 2 μA.
The integrated 92-mΩ high-side MOSFET supports high efficiency power supply designs capable of delivering
3.5 A of continuous current to a load. The gate drive bias voltage for the integrated high-side MOSFET is
supplied by a bootstrap capacitor connected from the BOOT to SW pins. The TPS54360B-Q1 reduces the
external component count by integrating the bootstrap recharge diode. The BOOT pin capacitor voltage is
monitored by a UVLO circuit which turns off the high-side MOSFET when the BOOT to SW voltage falls below a
preset threshold. An automatic BOOT capacitor recharge circuit allows the TPS54360B-Q1 to operate at high
duty cycles approaching 100%. Therefore, the maximum output voltage is near the minimum input supply voltage
of the application. The minimum output voltage is the internal 0.8 V feedback reference.
Output overvoltage transients are minimized by an Overvoltage Transient Protection (OVP) comparator. When
the OVP comparator is activated, the high-side MOSFET is turned off and remains off until the output voltage is
less than 106% of the desired output voltage.
The TPS54360B-Q1 includes an internal soft-start circuit that slows the output rise time during start-up to reduce
in-rush current and output voltage overshoot. Output overload conditions reset the soft-start timer. When the
overload condition is removed, the soft-start circuit controls the recovery from the fault output level to the nominal
regulation voltage. A frequency foldback circuit reduces the switching frequency during start-up and overcurrent
fault conditions to help maintain control of the inductor current.
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7.2 Functional Block Diagram
EN
VIN
Shutdown
OV
Thermal
Shutdown
Enable
Comparator
Logic
UVLO
Shutdown
Shutdown
Logic
Enable
Threshold
Boot
Charge
Voltage
Reference
Error
Amplifier
FB
Boot
UVLO
Maximum
Clamp
Pulse
Skip
PMW
Comparator
±
Current
Sense
BOOT
+
+
Logic
Shutdown
6
Slope
Compensation
SW
COMP
Frequency
Foldback
Reference
DAC for
Soft-Start
Maximum
Clamp
GND
Oscillator
with PLL
POWERPAD
RT/CLK
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7.3 Feature Description
7.3.1 Fixed Frequency PWM Control
The TPS54360B-Q1 uses fixed frequency, peak current mode control with adjustable switching frequency. The
output voltage is compared through external resistors connected to the FB pin to an internal voltage reference by
an error amplifier. An internal oscillator initiates the turn on of the high-side power switch. The error amplifier
output at the COMP pin controls the high-side power switch current. When the high-side MOSFET switch current
reaches the threshold level set by the COMP voltage, the power switch is turned off. The COMP-pin voltage
increases and decreases as the output current increases and decreases. The device implements current limiting
by clamping the COMP-pin voltage to a maximum level. The pulse skipping Eco-Mode is implemented with a
minimum voltage clamp on the COMP pin.
7.3.2 Slope Compensation Output Current
The TPS54360B-Q1 adds a compensating ramp to the MOSFET switch current sense signal. This slope
compensation prevents sub-harmonic oscillations at duty cycles greater than 50%. The peak current limit of the
high-side switch is not affected by the slope compensation and remains constant over the full duty cycle range.
12
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Feature Description (continued)
7.3.3 Pulse Skip Eco-Mode™
The TPS54360B-Q1 operates in a pulse skipping Eco-mode at light load currents to improve efficiency by
reducing switching and gate drive losses. If the output voltage is within regulation and the peak switch current at
the end of any switching cycle is below the pulse skipping current threshold, the device enters Eco-Mode. The
pulse skipping current threshold is the peak switch current level corresponding to a nominal COMP voltage of
600 mV.
When in Eco-Mode, the COMP-pin voltage is clamped at 600 mV and the high-side MOSFET is inhibited.
Because the device is not switching, the output voltage begins to decay. The voltage control loop responds to the
falling output voltage by increasing the COMP-pin voltage. The high-side MOSFET is enabled and switching
resumes when the error amplifier lifts COMP above the pulse skipping threshold. The output voltage recovers to
the regulated value, and COMP eventually falls below the Eco-Mode pulse skipping threshold at which time the
device again enters Eco-Mode. The internal PLL remains operational when in Eco-Mode. When operating at light
load currents in Eco-Mode, the switching transitions occur synchronously with the external clock signal.
During Eco-Mode operation, the TPS54360B-Q1 senses and controls peak switch current, not the average load
current. Therefore the load current at which the device enters Eco-Mode is dependent on the output inductor
value. The circuit in Figure 33 enters Eco-Mode at about 24 mA output current. As the load current approaches
zero, the device enters a pulse skip mode during which it draws only 146 μA input quiescent current.
7.3.4 Low Dropout Operation and Bootstrap Voltage (BOOT)
The TPS54360B-Q1 provides an integrated bootstrap voltage regulator. A small capacitor between the BOOT
and SW pins provides the gate drive voltage for the high-side MOSFET. The BOOT capacitor is refreshed when
the high-side MOSFET is off and the external low-side diode conducts. The recommended value of the BOOT
capacitor is 0.1 μF. A ceramic capacitor with an X7R or X5R grade dielectric with a voltage rating of 10 V or
higher is recommended for stable performance over temperature and voltage.
When operating with a low voltage difference from input to output, the high-side MOSFET of the TPS54360B-Q1
operates at 100% duty cycle as long as the BOOT to SW-pin voltage is greater than 2.1 V. When the voltage
from BOOT to SW drops below 2.1V, the high-side MOSFET is turned off and an integrated low-side MOSFET
pulls SW low to recharge the BOOT capacitor. To reduce the losses of the small low-side MOSFET at high
output voltages, it is disabled at 24 V output and re-enabled when the output reaches 21.5 V.
Because the gate drive current sourced from the BOOT capacitor is small, the high-side MOSFET can remain on
for many switching cycles before the MOSFET is turned off to refresh the capacitor. Thus the effective duty cycle
of the switching regulator can be high, approaching 100%. The effective duty cycle of the converter during
dropout is mainly influenced by the voltage drops across the power MOSFET, the inductor resistance, the lowside diode voltage and the printed circuit board resistance.
Equation 1 calculates the minimum input voltage required to regulate the output voltage and ensure proper
operation of the device. This calculation must include tolerance of the component specifications and the variation
of these specifications at their maximum operating temperature in the application.
+ VF + Rdc ´ IOUT
V
+ RDS (on ) ´ IOUT - VF
VIN (min ) = OUT
D
where
•
•
•
•
VF = Schottky diode forward voltage
RDC = DC resistance of inductor
RDS(on) = High-side MOSFET resistance
D = Effective duty cycle of 99%.
(1)
During high duty cycle (low dropout) conditions, inductor current ripple increases when the BOOT capacitor is
being recharged resulting in an increase in output voltage ripple. Increased ripple occurs when the off time
required to recharge the BOOT capacitor is longer than the high-side off time associated with cycle by cycle
PWM control.
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Feature Description (continued)
7.3.5 Error Amplifier
The TPS54360B-Q1 voltage regulation loop is controlled by a transconductance error amplifier. The error
amplifier compares the FB-pin voltage to the lower of the internal soft-start voltage or the internal 0.8 V voltage
reference. The transconductance (gm) of the error amplifier is 350 μS during normal operation. During soft-start
operation, the transconductance is reduced to 78 μS and the error amplifier is referenced to the internal soft-start
voltage.
The frequency compensation components (capacitor, series resistor and capacitor) are connected between the
error amplifier output COMP pin and GND pin.
7.3.6 Adjusting the Output Voltage
The internal voltage reference produces a precise 0.8 V ±1% voltage reference over the operating temperature
and voltage range by scaling the output of a bandgap reference circuit. The output voltage is set by a resistor
divider from the output node to the FB pin. TI recommends to use 1% tolerance or better divider resistors. Select
the low-side resistor RLS for the desired divider current and use Equation 2 to calculate RHS. To improve
efficiency at light loads consider using larger value resistors. However, if the values are too high, the regulator is
more susceptible to noise and voltage errors from the FB input current can become noticeable.
æ Vout - 0.8V ö
RHS = RLS ´ ç
÷
0.8 V
è
ø
(2)
7.3.7 Enable and Adjusting Undervoltage Lockout
The TPS54360B-Q1 is enabled when the VIN-pin voltage rises above 4.3 V and the EN-pin voltage exceeds the
enable threshold of 1.2 V. The TPS54360B-Q1 is disabled when the VIN-pin voltage falls below 4 V or when the
EN-pin voltage is below 1.2 V. The EN pin has an internal pullup current source, I1, of 1.2 μA that enables
operation of the TPS54360B-Q1 when the EN pin floats.
If an application requires a higher undervoltage lockout (UVLO) threshold, use the circuit shown in Figure 22 to
adjust the input voltage UVLO with two external resistors. When the EN-pin voltage exceeds 1.2 V, an additional
3.4 μA of hysteresis current, Ihys, is sourced out of the EN pin. When the EN pin is pulled below 1.2 V, the 3.4
μA Ihys current is removed. This addional current facilitates adjustable input voltage UVLO hysteresis. Use
Equation 3 to calculate RUVLO1 for the desired UVLO hysteresis voltage. Use Equation 4 to calculate RUVLO2 for
the desired VIN start voltage.
In applications designed to start at relatively low input voltages (such as, from 4.5 V to 9 V) and withstand high
input voltages (such as, from 40 V to 60 V), the EN pin experiences a voltage greater than the absolute
maximum voltage of 8.4 V during the high input voltage condition. When using an external EN resistor divider the
EN pin voltage is clamped internally with a 5.8 V zener diode. The zener diode will sink up to 150 µA.
VIN
TPS54360B-Q1
i1
VIN
ihys
RUVLO1
RUVLO1
EN
EN
10 k
Node
Optional
VEN
RUVLO2
RUVLO2
Copyright © 2016, Texas Instruments Incorporated
Figure 22. Adjustable Undervoltage Lockout
(UVLO)
14
5.8 V
Copyright © 2016, Texas Instruments Incorporated
Figure 23. Internal EN Clamp
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Feature Description (continued)
- VSTOP
V
RUVLO1 = START
IHYS
(3)
VENA
RUVLO2 =
VSTART - VENA
+ I1
RUVLO1
(4)
7.3.8 Internal Soft-Start
The TPS54360B-Q1 has an internal digital soft-start that ramps the reference voltage from 0 V to the final value
in 1024 switching cycles. The internal soft-start time (10% to 90%) is calculated using Equation 5
1024
tSS (ms) =
fSW (kHz)
(5)
If the EN pin is pulled below the stop threshold of 1.2 V, switching stops and the internal soft-start resets. The
soft-start also resets in thermal shutdown.
7.3.9 Constant Switching Frequency and Timing Resistor (RT/CLK) Pin)
The switching frequency of the TPS54360B-Q1 is adjustable over a wide range from 100 kHz to 2500 kHz by
placing a resistor between the RT/CLK pin and GND pin. The RT/CLK-pin voltage is typically 0.5 V and must
have a resistor to ground to set the switching frequency. To determine the timing resistance for a given switching
frequency, use Equation 6 or Equation 7 or the curves in Figure 5 and Figure 6. To reduce the solution size one
would typically set the switching frequency as high as possible, but tradeoffs of the conversion efficiency,
maximum input voltage and minimum controllable on time must be considered. The minimum controllable on
time is typically 135 ns which limits the maximum operating frequency in applications with high input to output
step down ratios. The maximum switching frequency is also limited by the frequency foldback circuit. A more
detailed discussion of the maximum switching frequency is provided in the next section.
101756
RT (kW) =
f sw (kHz)1.008
(6)
f sw (kHz) =
92417
RT (kW)0.991
(7)
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Feature Description (continued)
7.3.10 Accurate Current Limit Operation and Maximum Switching Frequency
The TPS54360B-Q1 implements peak current mode control in which the COMP-pin voltage controls the peak
current of the high-side MOSFET. A signal proportional to the high-side switch current and the COMP-pin voltage
are compared each cycle. When the peak switch current intersects the COMP control voltage, the high-side
switch is turned off. During overcurrent conditions that pull the output voltage low, the error amplifier increases
switch current by driving the COMP pin high. The error amplifier output is clamped internally at a level which sets
the peak switch current limit. The TPS54360B-Q1 provides an accurate current limit threshold with a typical
current limit delay of 60 ns. With smaller inductor values, the delay results in a higher peak inductor current. The
relationship between the inductor value and the peak inductor current is shown in Figure 24.
Inductor Current (A)
Peak Inductor Current
ΔCLPeak
Open Loop Current Limit
ΔCLPeak = VIN/L x tCLdelay
tCLdelay
tON
Figure 24. Current Limit Delay
To protect the converter in overload conditions at higher switching frequencies and input voltages, the
TPS54360B-Q1 implements a frequency foldback. The oscillator frequency is divided by 1, 2, 4, and 8 as the FBpin voltage falls from 0.8 V to 0 V. The TPS54360B-Q1 uses a digital frequency foldback to enable
synchronization to an external clock during normal start-up and fault conditions. During short-circuit events, the
inductor current can exceed the peak current limit because of the high input voltage and the minimum
controllable on time. When the output voltage is forced low by the shorted load, the inductor current decreases
slowly during the switch-off time. The frequency foldback effectively increases the off time by increasing the
period of the switching cycle providing more time for the inductor current to ramp down.
With a maximum frequency foldback ratio of 8, there is a maximum frequency at which the inductor current is
controlled by frequency foldback protection. Equation 9 calculates the maximum switching frequency at which the
inductor current remains under control when VOUT is forced to VOUT(SC). The selected operating frequency must
not exceed the calculated value.
Equation 8 calculates the maximum switching frequency limitation set by the minimum controllable on time and
the input to output step down ratio. Setting the switching frequency above this value causes the regulator to skip
switching pulses to achieve the low duty cycle required at maximum input voltage.
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Feature Description (continued)
æ I ´R + V
dc
OUT + Vd
´ç O
ç VIN - IO ´ RDS(on ) + Vd
è
ö
÷
÷
ø
fDIV æç ICL ´ Rdc + VOUT(sc ) + Vd
´
tON ç VIN - ICL ´ RDS(on ) + Vd
è
ö
÷
÷
ø
fSW (max skip ) =
fSW(shift) =
1
tON
(8)
where
•
•
•
•
•
•
•
•
•
•
IO is Output current
ICL is Current limit
Rdc is inductor resistance
VIN is maximum input voltage
VOUT is output voltage
VOUTSC is output voltage during short
Vd is diode voltage drop
RDS(on) is switch on resistance
tON is minimum controllable on time
ƒDIV is frequency divide equals (1, 2, 4, or 8)
(9)
7.3.11 Synchronization to RT/CLK Pin
The RT/CLK pin can receive a frequency synchronization signal from an external system clock. To implement
this synchronization feature connect a square wave to the RT/CLK pin through either circuit network shown in
Figure 25. The square wave applied to the RT/CLK pin must switch lower than 0.5 V and higher than 1.7 V and
have a pulse-width greater than 15 ns. The synchronization frequency range is 160 kHz to 2300 kHz. The rising
edge of the SW synchronizes to the falling edge of RT/CLK pin signal. The external synchronization circuit must
be designed such that the default frequency set resistor is connected from the RT/CLK pin to ground when the
synchronization signal is off. When using a low impedance signal source, the frequency set resistor is connected
in parallel with an AC-coupling capacitor to a termination resistor (for example: 50 Ω) as shown in Figure 25. The
two resistors in series provide the default frequency setting resistance when the signal source is turned off. The
sum of the resistance should set the switching frequency close to the external CLK frequency. TI recommends to
AC couple the synchronization signal through a 10 pF ceramic capacitor to RT/CLK pin.
The first time the RT/CLK is pulled above the PLL threshold the TPS54360B-Q1 switches from the RT resistor
free-running frequency mode to the PLL synchronized mode. The internal 0.5 V voltage source is removed and
the RT/CLK pin becomes high impedance as the PLL starts to lock onto the external signal. The switching
frequency can be higher or lower than the frequency set with the RT/CLK resistor. The device transitions from
the resistor mode to the PLL mode and locks onto the external clock frequency within 78 µs. During the transition
from the PLL mode to the resistor programmed mode, the switching frequency falls to 150 kHz and then
increases or decreases to the resistor programmed frequency when the 0.5-V bias voltage is reapplied to the
RT/CLK resistor.
The switching frequency is divided by 8, 4, 2, and 1 as the FB-pin voltage ramps from 0 to 0.8 V. The device
implements a digital frequency foldback to enable synchronizing to an external clock during normal start-up and
fault conditions. Figure 26, Figure 27 and Figure 28 show the device synchronized to an external system clock in
continuous conduction mode (CCM), discontinuous conduction (DCM), and pulse skip mode (Eco-Mode).
SPACER
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Feature Description (continued)
TPS54360B-Q1
TPS54360B-Q1
RT/CLK
RT/CLK
PLL
PLL
RT
Hi-Z
Clock
Source
Clock
Source
RT
Copyright © 2016, Texas Instruments Incorporated
Figure 25. Synchronizing to a System Clock
SW
SW
EXT
EXT
IL
IL
Figure 26. Plot of Synchronizing in CCM
Figure 27. Plot of Synchronizing in DCM
SW
EXT
IL
Figure 28. Plot of Synchronizing in Eco-Mode™
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Feature Description (continued)
7.3.12 Overvoltage Protection
The TPS54360B-Q1 incorporates an output overvoltage protection (OVP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients in designs with low output capacitance.
For example, when the power supply output is overloaded the error amplifier compares the actual output voltage
to the internal reference voltage. If the FB-pin voltage is lower than the internal reference voltage for a
considerable time, the output of the error amplifier increases to a maximum voltage corresponding to the peak
current limit threshold. When the overload condition is removed, the regulator output rises and the error amplifier
output transitions to the normal operating level. In some applications, the power supply output voltage increases
faster than the response of the error amplifier output resulting in an output overshoot.
The OVP feature minimizes output overshoot when using a low value output capacitor by comparing the FB-pin
voltage to the rising OVP threshold which is nominally 109% of the internal voltage reference. If the FB-pin
voltage is greater than the rising OVP threshold, the high-side MOSFET is immediately disabled to minimize
output overshoot. When the FB voltage drops below the falling OVP threshold which is nominally 106% of the
internal voltage reference, the high-side MOSFET resumes normal operation.
7.3.13 Thermal Shutdown
The TPS54360B-Q1 provides an internal thermal shutdown to protect the device when the junction temperature
exceeds 176°C. The high-side MOSFET stops switching when the junction temperature exceeds the thermal trip
threshold. Once the die temperature falls below 164°C, the device reinitiates the power up sequence controlled
by the internal soft-start circuitry.
7.3.14 Small Signal Model for Loop Response
Figure 29 shows an equivalent model for the TPS54360B-Q1 control loop which can be simulated to check the
frequency response and dynamic load response. The error amplifier is a transconductance amplifier with a gmEA
of 350 μS. The error amplifier can be modeled using an ideal voltage controlled current source. The resistor Ro
and capacitor Co model, the open-loop gain, and frequency response of the amplifier. The 1-mV AC voltage
source between the nodes a and b effectively breaks the control loop for the frequency response measurements.
Plotting c/a provides the small signal response of the frequency compensation. Plotting a/b provides the small
signal response of the overall loop. The dynamic loop response can be evaluated by replacing RL with a current
source with the appropriate load step amplitude and step rate in a time domain analysis. This equivalent model is
only valid for continuous conduction mode (CCM) operation.
SW
VO
Power Stage
gmps 12 A/V
a
b
RESR
R1
RL
COMP
c
0.8 V
R3
CO
C2
RO
FB
COUT
gmea
R2
350 mA/V
C1
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Figure 29. Small Signal Model For Loop Response
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Feature Description (continued)
7.3.15 Simple Small Signal Model for Peak Current Mode Control
Figure 30 describes a simple small signal model that can be used to design the frequency compensation. The
TPS54360B-Q1 power stage can be approximated by a voltage-controlled current source (duty cycle modulator)
supplying current to the output capacitor and load resistor. The control to output transfer function is shown in
Equation 10 and consists of a DC gain, one dominant pole, and one ESR zero. The quotient of the change in
switch current and the change in COMP-pin voltage (node c in Figure 29) is the power stage transconductance,
gmPS. The gmPS for the TPS54360B-Q1 is 12 A/V. The low-frequency gain of the power stage is the product of
the transconductance and the load resistance as shown in Equation 11.
As the load current increases and decreases, the low-frequency gain decreases and increases, respectively. This
variation with the load is problematic at first glance, but fortunately the dominant pole moves with the load current
(see Equation 12). The combined effect is highlighted by the dashed line in the right half of Figure 30. As the
load current decreases, the gain increases and the pole frequency lowers, keeping the 0-dB crossover frequency
the same with varying load conditions. The type of output capacitor chosen determines whether the ESR zero
has a profound effect on the frequency compensation design. Using high-ESR aluminum-electrolytic capacitors
can reduce the number frequency compensation components required to stabilize the overall loop because the
phase margin is increased by the ESR zero of the output capacitor (see Equation 13).
VO
Adc
VC
RESR
fp
RL
gmps
COUT
fz
Figure 30. Simple Small Signal Model and Frequency Response for Peak Current Mode Control
æ
s ö
ç1 +
÷
2p ´ fZ ø
VOUT
è
= Adc ´
VC
æ
s ö
ç1 +
÷
2
p
´ fP ø
è
Adc = gmps ´ RL
(10)
(11)
1
fP =
COUT ´ RL ´ 2p
(12)
1
fZ =
COUT ´ RESR ´ 2p
(13)
7.3.16 Small Signal Model for Frequency Compensation
The TPS54360B-Q1 uses a transconductance amplifier for the error amplifier and supports three of the
commonly-used frequency compensation circuits. Compensation circuits Type 2A, Type 2B, and Type 1 are
shown in Figure 31. Type 2 circuits are typically implemented in high bandwidth power-supply designs using lowESR output capacitors. The Type 1 circuit is used with power-supply designs with high-ESR aluminumelectrolytic or tantalum capacitors. Equation 14 and Equation 15 relate the frequency response of the amplifier to
the small signal model in Figure 31. The open-loop gain and bandwidth are modeled using the RO and CO shown
in Figure 31. See the application section for a design example using a Type 2A network with a low ESR output
capacitor.
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Feature Description (continued)
Equation 14 through Equation 23 are provided as a reference. An alternative is to use WEBENCH software tools
to create a design based on the power supply requirements.
VO
R1
FB
gmea
Type 2A
COMP
Type 2B
Type 1
Vref
R2
RO
R3
CO
C2
R3
C2
C1
C1
Copyright © 2016, Texas Instruments Incorporated
Figure 31. Types of Frequency Compensation
Aol
A0
P1
Z1
P2
A1
BW
Figure 32. Frequency Response of the Type 2A and Type 2B Frequency Compensation
Aol(V/V)
gmea
gmea
=
2p ´ BW (Hz)
Ro =
CO
(14)
(15)
æ
ö
s
ç1 +
÷
2p ´ fZ1 ø
è
EA = A0 ´
æ
ö æ
ö
s
s
ç1 +
÷ ´ ç1 +
÷
2
2
p
´
p
´
f
f
P1 ø è
P2 ø
è
(16)
R2
R1 + R2
R2
´ Ro| | R3 ´
R1 + R2
A0 = gmea ´ Ro ´
A1 = gmea
(17)
(18)
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Feature Description (continued)
P1 =
Z1 =
P2 =
1
2p ´ Ro ´ C1
(19)
1
2p ´ R3 ´ C1
(20)
1
type 2a
2p ´ R3 | | RO ´ (C2 + CO )
(21)
1
P2 =
type 2b
2p ´ R3 | | RO ´ CO
P2 =
2p ´ R O
1
type 1
´ (C2 + C O )
(22)
(23)
7.4 Device Functional Modes
7.4.1 Operation near Minimum VIN (VVIN = < 4.5 V)
The TPS54360B-Q1 is designed to operate with input voltage above 4.5 V. The typical VIN UVLO threshold is
4.3 V and the device may operate at input voltages down to the UVLO voltage. At input voltages below the
UVLO voltage the device does not switch. If an external resistor divider pulls the EN pin up to VIN or the EN pin
is floating, when VIN passes the UVLO threshold the device becomes active. When the device is active switching
begins and the soft-start sequence initiates. The TPS54360B-Q1 ramps up the output voltage at a rate based on
the internal digital soft-start.
7.4.2 Operation with EN Control
The enabled threshold voltage is 1.2 V typical. With EN held below the threshold voltage the device is shut down
and switching is inhibited even if the VIN voltage is above its UVLO threshold. The IC quiescent current
decreases to a minimum in this state. If the EN pin voltage is increased above its threshold while the VIN voltage
is also above its UVLO threshold, the device becomes active. When the device is active switching begins and the
soft-start sequence initiates. The TPS54360B-Q1 ramps up the output voltage at a rate based on the internal
digital soft-start
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The TPS54360B-Q1 device is a 60-V, 3.5-A, step down regulator with an integrated high side MOSFET. This
device is typically used to convert a higher DC voltage to a lower DC voltage with a maximum available output
current of 3.5 A. Example applications are: 12 V, 24 V and 48 V Industrial, Automotive and Communications
Power Systems. Use the following design procedure to select component values for the TPS54360B-Q1 device.
This procedure illustrates the design of a high frequency switching regulator using ceramic output capacitors.
Calculations can be done with the excel spreadsheet (SLVC452) located on the product page. Alternately, use
the WEBENCH software to generate a complete design. The WEBENCH software uses an iterative design
procedure and accesses a comprehensive database of components when generating a design.
8.2 Typical Application
8.2.1 5-V Output TPS54360B-Q1 Design Example
L1
8.2 H
C4
0.1 F
5.0V, 3.5A
U1
TPS54360B-Q1 (DDA)
VIN
2
3
C1
2.2 F
C2
2.2 F
R1
523 k
4
BOOT
SW
VIN
GND
EN
COMP
RT/CLK
PWRPD
1
8.5V to 60V
9
GND
R2
84.5 k
R3
162 k
GND
FB
D1
C6
C7
B560C
47 F
47 F
8
VOUT
R5
53.6 k
7
6
5
GND
FB
R4
13.0 k
C5
C8
FB
R6
10.2 k
39 pF
6800 pF
GND
GND
GND
Copyright © 2016, Texas Instruments Incorporated
Figure 33. 5 V Output TPS54360B-Q1 Design Example
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Typical Application (continued)
8.2.1.1 Design Requirements
A few parameters must be known in order to start the design process. These requirements are typically
determined at the system level. This example is designed to the following known parameters:
DESIGN PARAMETER
EXAMPLE VALUE
Output Voltage
5V
Transient Response 0.875 A to 2.625 A load step
ΔVOUT = 4 %
Maximum Output Current
3.5 A
Input Voltage
12 V nom. 8.5 V to 60 V
Output Voltage Ripple
0.5% of VOUT
Start Input Voltage (rising VIN)
8V
Stop Input Voltage (falling VIN)
6.25 V
8.2.1.2 Detailed Design Procedure
8.2.1.2.1 Selecting the Switching Frequency
The first step is to choose a switching frequency for the regulator. Typically, the designer uses the highest
switching frequency possible because this produces the smallest solution size. High switching frequency allows
for lower value inductors and smaller output capacitors compared to a power supply that switches at a lower
frequency. The switching frequency that can be selected is limited by the minimum on-time of the internal power
switch, the input voltage, the output voltage and the frequency foldback protection.
Equation 8 and Equation 9 must be used to calculate the upper limit of the switching frequency for the regulator.
Choose the lower value result from the two equations. Switching frequencies higher than these values results in
pulse skipping or the lack of overcurrent protection during a short circuit.
The typical minimum on time, tonmin, is 135 ns for the TPS54360B-Q1. For this example, the output voltage is 5 V
and the maximum input voltage is 60 V, which allows for a maximum switch frequency up to 710 kHz to avoid
pulse skipping from Equation 8. To ensure overcurrent runaway is not a concern during short circuits use
Equation 9 to determine the maximum switching frequency for frequency foldback protection. With a maximum
input voltage of 60 V, assuming a diode voltage of 0.7 V, inductor resistance of 25 mΩ, switch resistance of 92
mΩ, a current limit value of 4.7 A and short circuit output voltage of 0.1 V, the maximum switching frequency is
902 kHz.
For this design, a lower switching frequency of 600 kHz is chosen to operate comfortably below the calculated
maximums. To determine the timing resistance for a given switching frequency, use Equation 6 or the curve in
Figure 6. The switching frequency is set by resistor R3 shown in Figure 33. For 600 kHz operation, the closest
standard value resistor is 162 kΩ.
1
æ 3.5 A x 25 mW + 5 V + 0.7 V ö
fSW(max skip) =
´ ç
÷ = 710 kHz
135ns
è 60 V - 3.5 A x 92 mW + 0.7 V ø
(24)
8
æ 4.7 A x 25 mW + 0.1 V + 0.7 V ö
´ ç
÷ = 902 kHz
135 ns
è 60 V - 4.7 A x 92 mW + 0.7 V ø
101756
RT (kW) =
= 161 kW
600 (kHz)1.008
fSW(shift) =
(25)
(26)
8.2.1.2.2 Output Inductor Selection (LO)
To calculate the minimum value of the output inductor, use Equation 27.
KIND is a ratio that represents the amount of inductor ripple current relative to the maximum output current. The
inductor ripple current is filtered by the output capacitor. Therefore, choosing high inductor ripple currents
impacts the selection of the output capacitor because the output capacitor must have a ripple current rating equal
to or greater than the inductor ripple current. In general, the inductor ripple value is at the discretion of the
designer, however, the following guidelines are used.
24
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For designs using low-ESR output capacitors such as ceramics, a value as high as KIND = 0.3 is desirable. When
using higher ESR output capacitors, KIND = 0.2 yields better results. Because the inductor ripple current is part of
the current mode PWM control system, the inductor ripple current must always be greater than 150 mA for stable
PWM operation. In a wide input voltage regulator, it is best to choose relatively large inductor ripple current. This
provides sufficienct ripple current with the input voltage at the minimum.
For this design example, KIND = 0.3 and the minimum inductor value is calculated to be 7.3 μH. The nearest
standard value is 8.2 μH. It is important that the RMS current and saturation current ratings of the inductor not be
exceeded. The RMS and peak inductor current can be found from Equation 29 and Equation 30. For this design,
the RMS inductor current is 3.5 A and the peak inductor current is 3.97 A. The chosen inductor is a WE
7447797820, which has a saturation current rating of 5.8 A and an RMS current rating of 5.05 A.
As the equation set demonstrates, lower ripple currents reduce the output voltage ripple of the regulator but
require a larger value of inductance. Selecting higher ripple currents increases the output voltage ripple of the
regulator but allow for a lower inductance value.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the peak inductor current level
calculated above. In transient conditions, the inductor current can increase up to the switch current limit of the
device. For this reason, the most conservative design approach is to choose an inductor with a saturation current
rating equal to or greater than the switch current limit of the TPS54360B-Q1 which is nominally 5.5 A.
LO(min ) =
VIN(max ) - VOUT
IOUT ´ KIND
´
VOUT
60 V - 5 V
5V
=
´
= 7.3 mH
VIN(max ) ´ fSW
3.5 A x 0.3
60 V ´ 600 kHz
(27)
spacer
IRIPPLE =
VOUT ´ (VIN(max ) - VOUT )
VIN(max ) ´ LO ´ fSW
=
5 V x (60 V - 5 V)
= 0.932 A
60 V x 8.2 mH x 600 kHz
(28)
spacer
(
æ
1 ç VOUT ´ VIN(max ) - VOUT
2
´ç
IL(rms ) = (IOUT ) +
12 ç
VIN(max ) ´ LO ´ fSW
è
)ö÷
2
÷ =
÷
ø
2
(3.5 A )2 +
æ
5 V ´ (60 V - 5 V ) ö
1
´ ç
÷ = 3.5 A
ç 60 V ´ 8.2 mH ´ 600 kHz ÷
12
è
ø
(29)
spacer
IL(peak ) = IOUT +
IRIPPLE
0.932 A
= 3.5 A +
= 3.97 A
2
2
(30)
8.2.1.2.3 Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. The output capacitor
determines the modulator pole, the output voltage ripple, and how the regulator responds to a large change in
load current. The output capacitance must be selected based on the most stringent of these three criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor must supply
the increased load current until the regulator responds to the load step. The regulator does not respond
immediately to a large, fast increase in the load current such as transitioning from no load to a full load. The
regulator usually requires two or more clock cycles for the control loop to sense the change in output voltage and
adjust the peak switch current in response to the higher load. The output capacitance must be large enough to
supply the difference in current for 2 clock cycles to maintain the output voltage within the specified range.
Equation 31 shows the minimum output capacitance necessary, where ΔIOUT is the change in output current, ƒSW
is the regulators switching frequency and ΔVOUT is the allowable change in the output voltage. For this example,
the transient load response is specified as a 4% change in VOUT for a load step from 0.875 A to 2.625 A.
Therefore, ΔIOUT is 2.625 A - 0.875 A = 1.75 A and ΔVOUT = 0.04 × 5 = 0.2 V. Using these numbers gives a
minimum capacitance of 29.2 μF. This value does not take the ESR of the output capacitor into account in the
output voltage change. For ceramic capacitors, the ESR is usually small enough to be ignored. Aluminum
electrolytic and tantalum capacitors have higher ESR that must be included in load step calculations.
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The output capacitor must also be sized to absorb energy stored in the inductor when transitioning from a high to
low load current. The catch diode of the regulator can not sink current so energy stored in the inductor can
produce an output voltage overshoot when the load current rapidly decreases. A typical load step response is
shown in Figure 34. The excess energy absorbed in the output capacitor increases the voltage on the capacitor.
The capacitor must be sized to maintain the desired output voltage during these transient periods. Equation 32
calculates the minimum capacitance required to keep the output voltage overshoot to a desired value, where LO
is the value of the inductor, IOH is the output current under heavy load, IOL is the output under light load, Vf is the
peak output voltage, and Vi is the initial voltage. For this example, the worst case load step is from 2.625 A to
0.875 A. The output voltage increases during this load transition and the stated maximum in our specification is
4 % of the output voltage. This makes Vf = 1.04 × 5 = 5.2. Vi is the initial capacitor voltage which is the nominal
output voltage of 5 V. Using these numbers in Equation 32 yields a minimum capacitance of
24.6 μF.
Equation 33 calculates the minimum output capacitance required to meet the output voltage ripple specification,
where ƒSW is the switching frequency, VORIPPLE is the maximum allowable output voltage ripple, and IRIPPLE is the
inductor ripple current. Equation 33 yields 7.8 μF.
Equation 34 calculates the maximum ESR an output capacitor can have to meet the output voltage ripple
specification. Equation 34 indicates the ESR must be less than 27 mΩ.
The most stringent criteria for the output capacitor is 29.2 μF required to maintain the output voltage within
regulation tolerance during a load transient.
Capacitance de-ratings for aging, temperature and DC bias increases this minimum value. For this example, two
47-μF 10-V ceramic capacitors with 5 mΩ of ESR are used. The derated capacitance is 58.3 µF, well above the
minimum required capacitance of 29.2 µF.
Capacitors are generally rated for a maximum ripple current that can be filtered without degrading capacitor
reliability. Some capacitor data sheets specify the Root Mean Square (RMS) value of the maximum ripple
current. Equation 35 can be used to calculate the RMS ripple current that the output capacitor must support. For
this example, Equation 35 yields 269 mA.
2 ´ DIOUT
2 ´ 1.75 A
=
= 29.2 mF
COUT >
fSW ´ DVOUT 600 kHz x 0.2 V
(31)
((I ) - (I ) ) = 8.2 mH x (2.625 A - 0.875 A ) = 24.6 mF
x
(5.2 V - 5 V )
((V ) - (V ) )
2
OH
COUT > LO
2
2
2
OL
2
f
2
2
2
I
1
1
1
1
´
=
= 7.8 mF
COUT >
x
8 ´ fSW æ VORIPPLE ö 8 x 600 kHz
æ 25 mV ö
ç 0.932 A ÷
ç
÷
è
ø
è IRIPPLE ø
V
25 mV
= 27 mW
RESR < ORIPPLE =
IRIPPLE
0.932 A
ICOUT(rms) =
(
VOUT ´ VIN(max ) - VOUT
)=
12 ´ VIN(max ) ´ LO ´ fSW
5V ´
(60 V
(32)
(33)
(34)
- 5 V)
12 ´ 60 V ´ 8.2 mH ´ 600 kHz
= 269 mA
(35)
8.2.1.2.4 Catch Diode
The TPS54360B-Q1 requires an external catch diode between the SW pin and GND. The selected diode must
have a reverse voltage rating equal to or greater than VIN(max). The peak current rating of the diode must be
greater than the maximum inductor current. Schottky diodes are typically a good choice for the catch diode due
to their low forward voltage. The lower the forward voltage of the diode, the higher the efficiency of the regulator.
Typically, diodes with higher voltage and current ratings have higher forward voltages. A diode with a minimum of
60 V reverse voltage is preferred to allow input voltage transients up to the rated voltage of the TPS54360B-Q1.
For the example design, the B560C-13-F Schottky diode is selected for its lower forward voltage and good
thermal characteristics compared to smaller devices. The typical forward voltage of the B560C-13-F is 0.70 V at
5 A.
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The diode must also be selected with an appropriate power rating. The diode conducts the output current during
the off-time of the internal power switch. The off-time of the internal switch is a function of the maximum input
voltage, the output voltage, and the switching frequency. The output current during the off-time is multiplied by
the forward voltage of the diode to calculate the instantaneous conduction losses of the diode. At higher
switching frequencies, the AC losses of the diode must be taken into account. The AC losses of the diode are
due to the charging and discharging of the junction capacitance and reverse recovery charge. Equation 36 is
used to calculate the total power dissipation, including conduction losses and AC losses of the diode.
The B560C-13-F diode has a junction capacitance of 300 pF. Using Equation 36, the worst case total loss in the
diode using the maximum input voltage is 2.58 Watts.
If the power supply spends a significant amount of time at light load currents or in sleep mode, consider using a
diode which has a low leakage current and slightly higher forward voltage drop.
PD =
(V
IN(max ) - VOUT
)´ I
OUT
+
VIN(max )
(60 V
2
´ Vf d
- 5 V ) ´ 3.5 A x 0.7 V
60 V
+
C j ´ fSW ´ (VIN + Vf d)
=
2
300 pF x 600 kHz x (60 V + 0.7 V)2
= 2.58 W
2
(36)
8.2.1.2.5 Input Capacitor
The TPS54360B-Q1 requires a high quality ceramic type X5R or X7R input decoupling capacitor with at least 3
μF of effective capacitance. Some applications benefit from additional bulk capacitance. The effective
capacitance includes any loss of capacitance due to DC bias effects. The voltage rating of the input capacitor
must be greater than the maximum input voltage. The capacitor must also have a ripple current rating greater
than the maximum input current ripple of the TPS54360B-Q1. The input ripple current can be calculated using
Equation 37.
The value of a ceramic capacitor varies significantly with temperature and the DC bias applied to the capacitor.
The capacitance variations due to temperature can be minimized by selecting a dielectric material that is more
stable over temperature. X5R and X7R ceramic dielectrics are usually selected for switching regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The input capacitor
must also be selected with consideration for the DC bias. The effective value of a capacitor decreases as the DC
bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 60-V voltage rating is required to support the
maximum input voltage. Common standard ceramic capacitor voltage ratings include 4 V, 6.3 V, 10 V, 16 V, 25
V, 50 V or 100 V. For this example, two 2.2-μF 100-V capacitors in parallel are used. Table 1 shows several
choices of high voltage capacitors.
The input capacitance value determines the input ripple voltage of the regulator. The input voltage ripple can be
calculated using Equation 38. Using the design example values, IOUT = 3.5 A, CIN = 4.4 μF, ƒsw = 600 kHz,
yields an input voltage ripple of 331 mV and a RMS input ripple current of 1.72 A.
ICI(rms ) = IOUT x
VOUT
x
VIN(min )
(V
IN(min ) - VOUT
VIN(min )
) = 3.5 A
5V
´
8.5 V
(8.5 V
- 5 V)
8.5 V
= 1.72 A
(37)
´ 0.25
I
3.5 A ´ 0.25
DVIN = OUT
=
= 331 mV
CIN ´ fSW
4.4 mF ´ 600 kHz
(38)
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Table 1. Capacitor Types
VENDOR
VALUE (μF)
1 to 2.2
Murata
1 to 4.7
1
1 to 2.2
1 to 1.8
Vishay
1 to 1.2
1 to 3.9
1 to 1.8
1 to 2.2
TDK
1.5 to 6.8
1 to 2.2
1 to 3.3
1 to 4.7
AVX
1
1 to 4.7
1 to 2.2
EIA Size
1210
1206
2220
2225
1812
1210
1210
1812
VOLTAGE (V)
DIALECTRIC
100
COMMENTS
GRM32 series
50
100
GRM31 series
50
50
100
VJ X7R series
50
100
X7R
100
C series C4532
50
100
C series C3225
50
50
100
X7R dielectric series
50
100
8.2.1.2.6 Bootstrap Capacitor Selection
A 0.1-μF ceramic capacitor must be connected between the BOOT and SW pins for proper operation. A ceramic
capacitor with X5R or better grade dielectric is recommended. The capacitor should have a 10 V or higher
voltage rating.
8.2.1.2.7 Undervoltage Lockout Set Point
The Undervoltage Lockout (UVLO) is adjusted using an external voltage divider on the EN pin of the
TPS54360B-Q1. The UVLO has two thresholds, one for power up when the input voltage is rising and one for
power down or brown outs when the input voltage is falling. For the example design, the supply should turn on
and start switching once the input voltage increases above 8 V (UVLO start). After the regulator starts switching,
it should continue to do so until the input voltage falls below 6.25 V (UVLO stop).
Programmable UVLO threshold voltages are set using the resistor divider of RUVLO1 and RUVLO2 between Vin and
ground connected to the EN pin. Equation 3 and Equation 4 calculate the resistance values necessary. For the
example application, a 523 kΩ between Vin and EN (RUVLO1) and a 84.5 kΩ between EN and ground (RUVLO2)
are required to produce the 8 V and 6.25 V start and stop voltages.
V
- VSTOP
8 V - 6.25 V
=
= 515 kW
RUVLO1 = START
IHYS
3.4 mA
(39)
RUVLO2 =
VENA
1.2 V
=
= 84.5 kW
VSTART - VENA
8 V - 1.2 V
+
1.2
m
A
+ I1
523 kW
RUVLO1
(40)
8.2.1.2.8 Output Voltage and Feedback Resistors Selection
The voltage divider of R5 and R6 sets the output voltage. For the example design, 10.2 kΩ was selected for R6.
Using Equation 2, R5 is calculated as 53.5 kΩ. The nearest standard 1% resistor is 53.6 kΩ. Due to the input
current of the FB pin, the current flowing through the feedback network must be greater than 1 μA to maintain the
output voltage accuracy. This requirement is satisfied if the value of R6 is less than 800 kΩ. Choosing higher
resistor values decreases quiescent current and improves efficiency at low output currents but can also introduce
noise immunity problems.
V
- 0.8 V
æ 5 V - 0.8 V ö
= 10.2 kW x ç
RHS = RLS x OUT
÷ = 53.5 kW
0.8 V
0.8 V
è
ø
(41)
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8.2.1.2.9 Minimum VIN
To ensure proper operation of the device and to keep the output voltage in regulation, the input voltage at the
device must be above the value calculated with . Using the typical values for the RDS(on), RDC and VF in this
application example, the minimum input voltage is 5.56 V. The BOOT-SW = 3 V curve in Figure 1 was used for
RDS(on) = 0.12 Ω because the device will be operating with low drop out. When operating with low dropout, the
BOOT-SW voltage is regulated at a lower voltage because the BOOT-SW capacitor is not refreshed every
switching cycle. In the final application, the values of RDS(on), RDC and VF used in this equation must include
tolerance of the component specifications and the variation of these specifications at their maximum operating
temperature in the application.
V
+ VF + Rdc ´ IOUT
VIN (min ) = OUT
+ RDS (on )´ IOUT - VF
0.99
5V + 0.5V + 0.0253W ´ 3.5A
+ 0.12W ´ 3.5A - 0.5V = 5.56V
VIN (min ) =
0.99
(42)
8.2.1.2.10 Compensation
There are several methods to design compensation for DC-DC regulators. The method presented here is easy to
calculate and ignores the effects of the slope compensation that is internal to the device. Because the slope
compensation is ignored, the actual crossover frequency is lower than the crossover frequency used in the
calculations. This method assumes the crossover frequency is between the modulator pole and the ESR zero
and the ESR zero is at least 10 times greater the modulator pole.
To get started, the modulator pole, ƒp(mod), and the ESR zero, ƒz1 must be calculated using Equation 43 and
Equation 44. For COUT, use a derated value of 58.3 μF. Use equations Equation 45 and Equation 46 to estimate
a starting point for the crossover frequency, ƒco. For the example design, ƒp(mod) is 1912 Hz and ƒz(mod) is 1092
kHz. Equation 44 is the geometric mean of the modulator pole and the ESR zero and Equation 46 is the mean of
modulator pole and the switching frequency. Equation 45 yields 45.7 kHz and Equation 46 gives 23.9 kHz. Use
the lower value of Equation 45 or Equation 46 for an initial crossover frequency. For this example, the target ƒco
is 23.9 kHz.
Next, the compensation components are calculated. A resistor in series with a capacitor is used to create a
compensating zero. A capacitor in parallel to these two components forms the compensating pole.
IOUT(max )
3.5 A
fP(mod) =
=
= 1912 Hz
2 ´ p ´ VOUT ´ COUT 2 ´ p ´ 5 V ´ 58.3 mF
(43)
f Z(mod) =
1
2 ´ p ´ RESR ´ COUT
fco =
fp(mod) x f z(mod) =
fco =
fp(mod) x
fSW
2
=
=
1
= 1092 kHz
2 ´ p ´ 2.5 mW ´ 58.3 mF
1912 Hz x 1092 kHz
1912 Hz x
600 kHz
2
= 45.7 kHz
(44)
(45)
= 23.9 kHz
(46)
To determine the compensation resistor, R4, use Equation 47. Assume the power stage transconductance,
gmps, is 12 A/V. The output voltage, VO, reference voltage, VREF, and amplifier transconductance, gmea, are 5
V, 0.8 V and 350 μA/V, respectively. R4 is calculated to be 13 kΩ which is a standard value. Use Equation 48 to
set the compensation zero to the modulator pole frequency. Equation 48 yields 6404 pF for compensating
capacitor C5. 6800 pF is used for this design.
ö
VOUT
æ 2 ´ p ´ fco ´ COUT ö æ
ö
5V
æ 2 ´ p ´ 23.9 kHz ´ 58.3 mF ö æ
R4 = ç
xç
÷ = ç
÷ x ç
÷ = 13 kW
÷
gmps
12 A / V
è
ø è 0.8 V x 350 mA / V ø
è
ø è VREF x gmea ø
(47)
C5 =
1
1
=
= 6404 pF
2 ´ p ´ R4 x fp(mod)
2 ´ p ´ 13 kW x 1912 Hz
(48)
A compensation pole can be implemented if desired by adding capacitor C8 in parallel with the series
combination of R4 and C5. Use the larger value calculated from Equation 49 and Equation 50 for C8 to set the
compensation pole. The selected value of C8 is 39 pF for this design example.
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COUT x RESR
58.3 mF x 2.5 mW
=
= 11.2 pF
R4
13 kW
1
1
=
= 40.8 pF
C8 =
R4 x f sw x p
13 kW x 600 kHz x p
C8 =
(49)
(50)
8.2.1.2.11 Discontinuous Conduction Mode and Eco-Mode™ Boundary
With an input voltage of 12 V, the power supply enters discontinuous conduction mode when the output current
is less than 300 mA. The power supply enters Eco-Mode when the output current is lower than 24 mA. The input
current draw is 270 μA with no load.
8.2.1.2.12 Power Dissipation Estimate
The following formulas show how to estimate the TPS54360B-Q1 power dissipation under continuous conduction
mode (CCM) operation. These equations should not be used if the device is operating in discontinuous
conduction mode (DCM).
The power dissipation of the IC includes conduction loss (PCOND), switching loss (PSW), gate drive loss (PGD) and
supply current (PQ). Example calculations are shown with the 12 V typical input voltage of the design example.
PCOND =
(IOUT )2
æV
ö
5V
´ RDS(on ) ´ ç OUT ÷ = 3.5 A 2 ´ 92 mW ´
= 0.47 W
12 V
è VIN ø
(51)
spacer
PSW = VIN ´ fSW ´ IOUT ´ trise = 12 V ´ 600 kHz ´ 3.5 A ´ 4.9 ns = 0.123 W
(52)
spacer
PGD = VIN ´ QG ´ fSW = 12 V ´ 3nC ´ 600 kHz = 0.022 W
(53)
spacer
PQ = VIN ´ IQ = 12 V ´ 146 mA = 0.0018 W
(54)
Where:
• IOUT is the output current (A)
• RDS(on) is the on-resistance of the high-side MOSFET (Ω)
• VOUT is the output voltage (V)
• VIN is the input voltage (V)
• ƒSW is the switching frequency (Hz)
• trise is the SW-pin voltage rise time and is estimated by trise = VIN × 0.16 ns/V + 3 ns
• QG is the total gate charge of the internal MOSFET
• IQ is the operating nonswitching supply current
Therefore,
PTOT = PCOND + PSW + PGD + PQ = 0.47 W + 0.123 W + 0.022 W + 0.0018 W = 0.616 W
(55)
For given TA,
TJ = TA + RTH ´ PTOT
(56)
For given TJMAX = 150°C
TA (max ) = TJ(max ) - RTH ´ PTOT
(57)
Where:
• PTOT is the total device power dissipation (W)
• TA is the ambient temperature (°C)
• TJ is the junction temperature (°C)
• RTH is the thermal resistance of the package (°C/W)
• TJ(max) is maximum junction temperature (°C)
• TA(max) is maximum ambient temperature (°C)
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There are additional power losses in the regulator circuit due to the inductor AC and DC losses, the catch diode
and PCB trace resistance impacting the overall efficiency of the regulator.
10 V/div
1 A/div
8.2.1.3 Application Curves
C4: IOUT
VIN
C3
C3: VOUT ac coupled
20 mV/div
100 mV/div
C4
VOUT -5 V offset
Time = 5 ms/div
Figure 35. Line Transient (8 V To 40 V)
5 V/div
5 V/div
Time = 100 ms/div
Figure 34. Load Transient
C1: VIN
C1: VIN
2 V/div
1 V/div
C1
C2: EN
C2
C1
C2: EN
C3: VOUT
2 V/div
2 V/div
C2
C3
C3: VOUT
C3
Time = 2 ms/div
Figure 36. Startup With VIN
20 mV/div
10 V/div
IOUT = 3.5 A
C3: VOUT ac coupled
C3
C4
C1
C4: IL
500 mA/div
C4: IL
C1: SW
C4
20 mV/div
C1
1 A/div
10 V/div
C1: SW
Time = 2 ms/div
Figure 37. Startup With EN
C3
IOUT = 100 mA
C3: VOUT ac coupled
Time = 2 ms/div
Figure 38. Output Ripple CCM
Time = 2 ms/div
Figure 39. Output Ripple DCM
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C1: SW
C1
C1: SW
C1
1 A/div
C4: IL
C4: IL
C4
IOUT = 3.5 A
C3: VOUT ac coupled
C3: VIN ac coupled
C3
200 mV/div
20 mV/div
200 mA/div
10 V/div
SLVSDV1 – FEBRUARY 2017
No Load
C3
C4
Time = 2 ms/div
Figure 41. Input Ripple CCM
C1: SW
2 V/div
C1: SW
C1
200 mA/div
C4: IL
C4
IOUT = 100 mA
C3: VIN ac coupled
20 mV/div
20 mV/div
500 mA/div
10 V/div
Time = 2 ms/div
Figure 40. Output Ripple PSM
C3
C4
C4: IL
C3
C3: VOUT ac coupled
VIN = 5.5 V
VOUT = 5 V
Time = 20 ms/div
Figure 43. Low Dropout Operation
100
100
90
90
80
80
70
70
Efficiency - %
Efficiency - %
Time = 2 ms/div
Figure 42. Input Ripple DCM
60
50
40
VOUT = 5V, fsw = 600 kHz
30
20
VOUT = 5V, fsw = 600 kHz
60
50
40
30
20
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
0
0.001
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
32
No Load
EN Floating
0.01
0.1
IO - Output Current - A
IO - Output Current - A
Figure 44. Efficiency vs Load Current
Figure 45. Light Load Efficiency
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100
100
90
90
80
80
70
70
Efficiency - %
Efficiency - %
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60
50
40
VOUT = 3.3V, fsw = 300 kHz
30
VOUT = 3.3V, fsw = 300 kHz
60
50
40
30
20
20
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
0
0
0.5
1.0
1.5
2.5
2.0
3.0
3.5
0
0.001
4.0
Figure 47. Light Load Efficiency
180
60
Output Voltage Deviation - %
60
Gain
0
0
-60
-20
VIN = 12V,
VOUT = 5V,
IOUT = 3.5A
100
Phase - degree
120
10
-120
0.6
0.4
0.2
0
-0.2
0.4
-0.6
-0.8
-180
1000
VIN = 12V, VOUT = 5V,
fsw = 600 kHz
0.8
20
Gain - dB
1
Phase
-60
10000
1
IO - Output Current - A
Figure 46. Efficiency vs Load Current
-40
0.1
0.01
IO - Output Current - A
40
36Vin
48Vin
60Vin
8Vin
12Vin
24Vin
10
100000
-1
0
1000000
0.5
1.0
1.5
2.0
2.5
3.0
3.5
IO - Output Current - A
Frequency - Hz
Figure 48. Overall Loop Frequency Response
Figure 49. Regulation vs Load Current
0.5
VOUT = 5V,
fsw = 600 kHz, IOUT = 3.5A
Output Voltage Deviation - %
0.4
0.3
0.2
0.1
0
-0.1
0.2
-0.3
-0.4
-0.5
0
5
10
15
20
25
30
35
40
45
50
55
60
VIN - Input Voltage - V
Figure 50. Regulation vs Input Voltage
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8.2.2 TPS54360B-Q1 Inverting Power Supply
The TPS54360B-Q1 can be used to convert a positive input voltage to a negative output voltage. Example
applications are amplifiers requiring a negative power supply.
VIN
+
CIN
CBOOT
Lo
BOOT
VIN
CD
GND
SW
R1
+
GND
CO
R2
TPS54360B-Q1 FB
VOUT
EN
COMP
RCOMP
RT/CLK
RT
CZERO
CPOLE
Copyright © 2016, Texas Instruments Incorporated
Figure 51. TPS54360B-Q1 Inverting Power Supply
8.2.3 TPS54360B-Q1 Split Rail Power Supply
The TPS54360B-Q1 device can be used to convert a positive input voltage to a split rail positive and negative
output voltage by using a coupled inductor. Example applications are amplifiers requiring a split rail positive and
negative voltage power supply.
VOPOS
+
VIN
COPOS
+
CIN
CBOOT
GND
BOOT
VIN
SW
LO
CD
R1
GND
+
CONEG
R2
TPS54360B-Q1
VONEG
FB
EN
COMP
RCOMP
RT/CLK
RT
CZERO
CPOLE
Copyright © 2016, Texas Instruments Incorporated
Figure 52. TPS54360B-Q1 Split Rail Power Supply
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9 Power Supply Recommendations
The TPS54360B-Q1 is designed to operate from an input voltage supply range between 4.5 V and 60 V. This
input supply should be well regulated. If the input supply is located more than a few inches from the TPS54360BQ1 converter, in addition to the ceramic bypass capacitors, bulk capacitance may be required. An electrolytic
capacitor with a value of 100 μF is a typical choice.
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. There are several signal paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade performance. To reduce parasitic effects, the VIN pin must be bypassed to ground with a low ESR
ceramic bypass capacitor with X5R or X7R dielectric. Care must be taken to minimize the loop area formed by
the bypass capacitor connections, the VIN pin, and the anode of the catch diode. See Figure 53 for a PCB layout
example. The GND pin must be tied directly to the power pad under the IC and the power pad.
The power pad must be connected to internal PCB ground planes using multiple vias directly under the IC. The
SW pin must be routed to the cathode of the catch diode and to the output inductor. Because the SW connection
is the switching node, the catch diode and output inductor must be located close to the SW pins, and the area of
the PCB conductor minimized to prevent excessive capacitive coupling. For operation at full rated load, the top
side ground area must provide adequate heat dissipating area. The RT/CLK pin is sensitive to noise so the RT
resistor must be located as close as possible to the IC and routed with minimal lengths of trace. The additional
external components can be placed approximately as shown. Obtaining acceptable performance with alternate
PCB layouts is possible, however this layout has been shown to produce good results and is meant as a
guideline.
10.2 Layout Example
Vout
Output
Capacitor
Topside
Ground
Area
Output
Inductor
Route Boot Capacitor
Trace on another layer to
provide wide path for
topside ground
Input
Bypass
Capacitor
Vin
UVLO
Adjust
Resistors
BOOT
Catch
Diode
SW
VIN
GND
EN
COMP
RT/CLK
FB
Frequency
Set Resistor
Compensation
Network
Resistor
Divider
Thermal VIA
Signal VIA
Figure 53. PCB Layout Example
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10.3 Estimated Circuit Area
Boxing in the components in the design of Figure 33 the estimated printed circuit board area is 1.025 in2 (661
mm2). This area does not include test points or connectors.
11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation see the following:
• Create an Inverting Power Supply from a Step-Down Regulator, SLVA317
• Create a Split-Rail Power Supply With a Wide Input oltage Buck Regulator, SLVA369
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.5 Trademarks
Eco-Mode, PowerPAD, E2E are trademarks of Texas Instruments.
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
36
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12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54360BQDDAQ1
ACTIVE SO PowerPAD
DDA
8
75
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
54360B
TPS54360BQDDARQ1
ACTIVE SO PowerPAD
DDA
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 125
54360B
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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