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TPS54362-Q1
SLVS845G – MARCH 2009 – REVISED AUGUST 2014
TPS54362-Q1 3-A, 60-V Step-Down DC-DC Converter With Low I(q)
1 Features
2 Applications
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Qualified for Automotive Applications
AEC-Q100 Qualified With the Following Results:
– Device Temperature Grade 1: –40°C to 125°C
Ambient Operating Temperature
– Device HBM ESD Classification Level 2
– Device CDM ESD Classification Level C5
Withstands Transients up to 60 V With an
Operating Range of 3.6 V to 48 V
Asynchronous Switch-Mode Regulator With
External Components (L and C), Load Current up
to 3 A (max.)
0.8 V ± 1.5% Voltage Reference
200-kHz to 2.2-MHz Switching Frequency
High-Voltage-Tolerant Enable Input for ON/OFF
State
Soft Start on Enable Cycle
Slew-Rate Control on Internal Power Switch
External Clock Input for Synchronization
Pulse-Skip Mode (PFM) During Light Output
Loads With Quiescent Current = 65 μA Typical
(LPM Operation)
External Compensation for Wide-Bandwidth Error
Amplifier
Internal Undervoltage Lockout, UVLO
Programmable Reset Power-On Delay
Reset-Function Filter Time for Fast Negative
Transients
Programmable Overvoltage Output Monitoring
Programmable Undervoltage Output Monitoring,
Issuance of Reset if Output Falls Below Threshold
Thermal Shutdown During Excessive Power
Dissipation
Switch Current-Limit Protection
Short-Circuit and Overcurrent Protection of FET
Junction Temperature Range: –40°C to 150°C
20-Pin HTSSOP PowerPAD™ Package
Qualified for Automotive Applications
Automotive Telematics
Navigation Systems
In-Dash Instrumentation
Battery-Powered Applications
3 Description
The TPS54362-Q1 device is a step-down switchmode power supply with a low-power mode and a
programmable voltage supervisor. Integrated input
voltage-line feed-forward topology improves line
transient regulation of the voltage-mode buck
regulator. The regulator has a cycle-by-cycle current
limit. Pulse-skip mode operation under no load
reduces the supply current to 65 μA. Using the
enable pin reduces the supply shutdown current to
1 μA.
An open-drain reset signal indicates when the
nominal output drops below the threshold set by an
external resistor-divider network. A soft-start
capacitor controls the output voltage start-up ramp.
The device activates an internal undervoltage
shutdown when the input supply ramps down to
2.6 V.
Frequency foldback operation protects the device
during an overload conditions on the output. The
device also has thermal shutdown protection due to
excessive power dissipation.
The B-revision has an improved leakage current
parameter, and improved discharged function while in
disable mode.
Device Information(1)
PART NUMBER
TPS54362-Q1
PACKAGE
HTSSOP (20)
BODY SIZE (NOM)
6.50 mm × 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
Simplified Schematic
LPM
V(VIN)
VIN
EN
VIN
Rslew
SS
SYNC
RST
VReg
V(Vreg)
BOOT
RT
PH
TPS54362x-Q1
Cdly
VSENSE
COMP
RST_TH
GND
OV_TH
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54362-Q1
SLVS845G – MARCH 2009 – REVISED AUGUST 2014
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
4
5
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
5
5
6
6
7
8
8
9
Absolute Maximum Ratings ......................................
Handling Ratings.......................................................
Recommended Operating Conditions.......................
Thermal Information .................................................
Electrical Characteristics...........................................
Timing Requirements ................................................
Switching Characteristics ..........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 12
7.1 Overview ................................................................. 12
7.2 Functional Block Diagram ....................................... 12
7.3 Feature Description................................................. 13
7.4 Device Functional Modes........................................ 20
8
Application and Implementation ........................ 22
8.1 Application Information............................................ 22
8.2 Typical Application .................................................. 22
9 Power Supply Recommendations...................... 30
10 Layout................................................................... 30
10.1 Layout Guidelines ................................................. 30
10.2 Layout Example .................................................... 31
11 Device and Documentation Support ................. 32
11.1 Trademarks ........................................................... 32
11.2 Electrostatic Discharge Caution ............................ 32
11.3 Glossary ................................................................ 32
12 Mechanical, Packaging, and Orderable
Information ........................................................... 32
4 Revision History
Changes from Revision F (May 2014) to Revision G
Page
•
Added AEC-Q100 qualification features to the Features section .......................................................................................... 1
•
Added product improvements for B-revision silicon under the Product Description section.................................................. 1
•
Increased the max output voltage value for the SS pin in the Absolute Maximum Ratings table ......................................... 5
•
Changed column in Thermal Table from TPS54362A, to TPS54362x................................................................................... 6
•
Changed the IIKG row in Elec Chara Table from TPS54362A-Q1 to A-revision and TPS54362B-Q1 to B-revision .............. 7
•
Increased the maximum Ilkg value for both B-revision test conditions in the Electrical Characteristics table ....................... 7
•
Changed in CAUTION: TPS54362A-Q1 to TPS54362Ax-Q1 and deleted TPS54362B-Q1................................................ 16
•
Changed in Soft-Start Capacitor paragraph TPS54362A-Q1 to TPS54362Ax-Q1 and deleted TPS54362B-Q1 ................ 26
•
Changed Power Supply Recommendations section: TPS54362A-Q1 to TPS54362Ax-Q1 and deleted TPS54362B-Q1 .. 30
Changes from Revision E (May 2013) to Revision F
Page
•
Changed all text, tables and graphics to the new data sheet template. ................................................................................ 1
•
Changed pinout drawing......................................................................................................................................................... 4
•
Changed parameter symbols for JEDEC compliance throughout the data sheet ................................................................. 4
•
Added a row for the Rslew pin to the Absolute Maximum Ratings table ............................................................................... 5
•
Changed symbol for thermal resistance from θ to Rθ in the Thermal Information table ....................................................... 6
•
Added Ilkg parameters for EN pin on TPS54362B-Q1 device................................................................................................. 7
•
Revised Figure 22 ............................................................................................................................................................... 22
•
Changed value of R4 in Output Voltage and Feedback Resistor Selection section ............................................................ 25
•
Changed several values in the Overvoltage Resistor Selection section .............................................................................. 25
•
Changed several values in the Reset-Threshold Resistor Selection section....................................................................... 25
•
Changed the voltage value in the Undervoltage Threshold for Low-Power Mode and Load-Transient Operation section . 26
•
Added the TPS54362B-Q1 part number to the text of the Soft-Start Capacitor section ...................................................... 26
•
Changed calculated values for loop compensation components ......................................................................................... 27
2
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SLVS845G – MARCH 2009 – REVISED AUGUST 2014
Changes from Revision D (October 2011) to Revision E
Page
•
Removed TPS54362-Q1 and TPS54362 from data sheet; added -Q1 to TPS54362A part numbers. .................................. 1
•
Removed Ordering Information table; see Package Option Addendum for ordering information. ......................................... 1
•
Removed items 3 and 4 from the Soft Start (SS) section, also removed the sentence: Item 3 and item 4 are not
applicable for TPS54362A-Q1. ............................................................................................................................................. 15
•
Removed the following sentence from the Soft-Start Capacitor section: Equation 4 has to be satisfied in addition to
the other conditions stated in the soft start section of this document (not applicable for TPS54362A-Q1). ........................ 26
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5 Pin Configuration and Functions
PWP 20-Pin Package
20-Pin HTSSOP With Thermal Pad
Top View
NU
NU
SYNC
LPM
EN
RT
Rslew
RST
Cdly
GND
1
20
2
19
3
18
4
17
5
16
6
15
7
14
8
13
9
12
10
11
BOOT
VIN
VIN
PH
VReg
COMP
VSENSE
RST_TH
OV_TH
SS
NU – Make no external connection
Pin Functions
PIN
I/O
DESCRIPTION
NAME
NO.
BOOT
20
O
External bootstrap capacitor to PH to drive the gate of the internal switching FET
Cdly
9
I/O
External capacitor to ground to program power-on-reset delay.
COMP
15
I/O
Error-amplifier output to connect external compensation components
EN
5
I
Enable pin, internally pulled up. This pin requires an external pullup or pulldown to enable or disable the
device.
GND
10
O
Ground pin
LPM
4
I
Low-power mode control using digital input signal. An internal pulldown resistor of 62 kΩ (typical)
connects to ground.
NU
1
—
OV_TH
12
I
Sense input for overvoltage detection on regulated output. This pin monitors the V(Vreg) output voltage as
divided by the external resistor network connecting between the VReg pin and ground. The resistor
network programs the threshold voltage.
PH
17
O
Source of the internal switching FET
Rslew
7
O
External resistor to ground to control the slew rate of the internal switching FET
RST
8
O
Active-low, open-drain reset output connected to external bias voltage through a resistor, asserted high
after the device starts regulating
RST_TH
13
I
Sense input for undervoltage and reset voltage detection on regulated output to initiate a reset-output
signal. This pin monitors the V(Vreg) output voltage as divided by the external resistor network connecting
between the VReg pin and ground. The resistor network programs the threshold voltage.
Connect to ground
2
RT
6
O
External resistor to ground to program the internal oscillator frequency
SS
11
I/O
External capacitor to ground to program soft-start time
SYNC
3
I
External synchronization clock input to override the internal oscillator clock. An internal pulldown resistor
of 62 kΩ (typical) connects to ground.
VIN
18
I
Unregulated input voltage. Connect pin 18 and pin 19 together externally.
I
Internal low-side FET to load output during start-up or limit overshoot
I
Inverting node of error amplifier for voltage-mode control
19
VReg
16
VSENSE
14
Thermal pad
4
—
The thermal pad connects electrically to exposed ground pad on PCB for proper thermal performance.
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature (unless otherwise noted)
Input voltage
MIN
MAX
EN
–0.3
60
VIN
–0.3
60
VReg
–0.3
20
LPM
–0.3
5.5
OV_TH
–0.3
5.5
RST_TH
–0.3
5.5
SYNC
–0.3
5.5
VSENSE
–0.3
5.5
BOOT
–0.3
65
–0.3
60
30 ns
–2
60
200 ns
–1
60
TJ = –40
–0.85
60
TJ = 125
–0.5
60
RT
–0.3
5.5
RST
–0.3
5.5
Rslew
–0.3
5.5
Cdly
–0.3
8
SS
–0.3
8
COMP
–0.3
7
Operating virtual junction temperature range
–40
150
PH
Output voltage
TJ
(1)
(1)
UNIT
V
V
V
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 Handling Ratings
Tstg
Storage temperature range
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per AEC Q100-002
(1)
Charged-device model (CDM), per AEC Q100-011
MIN
MAX
UNIT
–55
165
°C
–2
2
kV
–750
750
V
AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
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6.3 Recommended Operating Conditions
over operating free-air temperature (unless otherwise noted)
MIN
MAX
UNIT
VI
Unregulated buck supply input voltage (VIN, EN)
3.6
48
V
V(VReg)
Regulator voltage range
0.9
18
V
Power up in low-power mode (LPM) or discontinuous mode (DCM)
0.9
5.5
0
5.25
Logic level inputs (RST, VSENSE, OV_TH, RST_TH, SYNC, RT)
Logic level inputs (SS, Cdly, COMP)
Operating junction temperature range (1)
TJ
(1)
V
0
6.5
V
–40
150
°C
This assumes TA = TJ – Power dissipation × RθJA (junction-to-ambient).
6.4 Thermal Information
TPS54362-Q1
THERMAL METRIC (1)
PWP
UNIT
20 PINS
RθJA
Junction-to-ambient thermal resistance
43.8
RθJC(top)
Junction-to-case (top) thermal resistance
21.4
RθJB
Junction-to-board thermal resistance
18.5
ψJT
Junction-to-top characterization parameter
0.5
ψJB
Junction-to-board characterization parameter
18.3
RθJC(bot)
Junction-to-case (bottom) thermal resistance
1.2
(1)
6
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
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6.5 Electrical Characteristics
V(VIN) = 7 V to 48 V, V(EN) = V(VIN), TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
Normal mode–buck mode after initial start-up
3.6
TYP
MAX
UNIT
TEST (1)
INPUT POWER SUPPLY
Falling threshold (LPM disabled)
V(VIN)
Supply voltage on VIN pin
Low-power mode
Rising threshold (LPM activated)
High-voltage threshold (LPM
disabled)
I(q-Normal)
I(q-LPM)
I(SD)
Quiescent current, normal mode
Quiescent current; low-power
mode
Shutdown
48
V
8
8.5
25
V
27
30
Open-loop test – maximum duty cycle
V(VIN) = 7 V to 48 V
5
10
I(VReg) < 1 mA, V(VIN) = 12 V, TA = 25°C
65
75
I(VReg) < 1 mA, V(VIN) = 12 V, –40 < TJ < 150°C
75
I(VReg) < 1 mA, V(VIN) = 24 V, TA = 25°C
85
I(VReg) < 1 mA, V(VIN) = 24 V, –40 < TJ < 150°C
85
V(EN) = 0 V, device is OFF, TA = –40°C to 125°C,
V(VIN) = 24 V
10
V(EN) = 0 V, device is OFF, TA = 25°C, V(VIN) = 12 V
1
Info
mA
PT
μA
PT
μA
PT
4
TRANSITION TIMES (LOW-POWER AND NORMAL MODES)
td(1)
Transition delay from normal
mode to low-power mode
V(VIN) = 12 V, V(VReg) = 5 V, I(VReg) = 1 A to 1 mA
100
μs
CT
td(2)
Transition delay from low-power
mode to normal mode
V(VIN) = 12 V, V(VReg) = 5 V, I(VReg) = 1 mA to 1 A
5
μs
CT
18
V
Info
0.812
V
CT
mΩ
PT
SWITCH-MODE SUPPLY (VReg)
V(VReg)
Regulator output
V(VSENSE) = 0.8-V reference
V(VSENSE)
Feedback voltage
V(VReg) = 0.9 V to 18 V, V(VIN) = 7 V to 48 V
rDS(on)
Internal switch resistance
Measured across VIN and PH, I(VReg) = 500 mA
I(CL)
Switch current limit, cycle-bycycle
V(VIN) = 12 V
t(ON-Min)
t(OFF-Min)
f(SW)
Switch-mode frequency
0.9
0.788
0.8
500
4
6
8
A
Info
Duty-cycle pulse duration (ON)
50
100
150
ns
Info
Duty-cycle pulse duration (OFF)
100
200
250
ns
Info
0.2
2.2
MHz
PT
–10%
10%
Set using external resistor on RT pin
Accuracy of f(SW)
I(Sink)
Sink current in start-up
condition
V(OV_TH) = 0 V, V(VReg) = 10 V
I(Limit)
Sink-current limit
0 V < V(OV_TH) < 0.8 V, V(VReg) = 10 V
PT
1
mA
80
mA
Info
0.7
V
PT
V
PT
μA
PT
μA
PT
V
PT
V
PT
V
PT
95
μA
PT
0.832
V
PT
60
μA
PT
Info
ENABLE (EN)
VIL
Low input threshold
VIH
High input threshold
Ilkg
Leakage into EN pin
1.7
A-revision, V(EN) = 60 V
100
135
A-revision,, V(EN) = 12 V
8
15
B-revision, V(EN) = 60 V
10
B-revision, V(EN) = 12 V
2
RESET DELAY (Cdly)
IO
External capacitor charge
current
V(EN) = high
VThreshold
Switching threshold
Output voltage in regulation
1.4
2
2.6
2
LOW-POWER MODE (LPM)
VIL
Low input threshold
V(VIN) = 12 V
VIH
High input threshold
V(VIN) = 12 V
Ilkg
Leakage into LPM pin
V(LPM) = 5 V
0.7
1.7
65
RESET OUTPUT (RST)
V(RST_TH)
Reset threshold for RST_TH pin
0.768
SOFT START (SS)
I(SS)
(1)
Soft-start source current
40
50
PT = Production tested; CT = Characterization tested only, not production tested; Info = User information only, not production tested
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Electrical Characteristics (continued)
V(VIN) = 7 V to 48 V, V(EN) = V(VIN), TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
SYNCHRONIZATION (SYNC)
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST (1)
V
PT
V
PT
(2)
VIL(SYNC)
Low input threshold
VIH(SYNC)
High input threshold
Ilkg
Leakage
Duty(min)
Minimum duty cycle
Duty(miax)
Maximum duty cycle
0.7
1.7
SYNC = 5 V
65
μA
95
30%
PT
CT
70%
CT
Rslew
I(Rslew)
Output current
Rslew = 50 kΩ
20
Rslew = 10 kΩ
100
μA
CT
OVERVOLTAGE SUPERVISORS (OV_TH)
V(OV_TH)
Threshold for OV_TH pin during
OV
Internal switch is OFF.
Internal pulldown current on
OV_TH pin
OV_TH = 1 V, V(VReg) = 5 V
0.768
0.832
V
PT
70
mA
175
°C
CT
30
°C
CT
THERMAL SHUTDOWN
T(SD)
Thermal shutdown junction
temperature
T(HYS)
Temperature hysteresis
(2)
The SYNC input clock can have a maximum frequency of 2× the programmed clock frequency up to a maximum value of 1.1 MHz.
6.6 Timing Requirements
V(VIN) = 7 V to 48 V, V(EN) = V(VIN), TJ = –40°C to 150°C (unless otherwise noted)
MIN
SYNCHRONIZATION (SYNC)
NOM
UNIT
TEST (1)
μs
Info
2200
kHz
CT
7
ms
PT
35
μs
PT
MAX
(2)
t(trans-ItoE)
Internal clock to external clock
External clock = 1 MHz, V(VIN) = 12 V,
V(VReg) = 5 V
f(SYNC)
Input clock
V(VIN) = 12 V, V(VReg) = 5 V,
f(sw) < f(ext) < 2 × f(sw)
180
C2 = 4.7 nF
3.6
2.5
RESET OUTPUT (RST)
td(POR)
POR delay timer
td(RSTdly)
Filter time
(1)
(2)
10
20
PT = Production tested; CT = Characterization tested only, not production tested; Info = User information only, not production tested
The SYNC input clock can have a maximum frequency of 2× the programmed clock frequency up to a maximum value of 1.1 MHz.
6.7 Switching Characteristics
V(VIN) = 7 V to 48 V, V(EN) = V(VIN), TJ = –40°C to 150°C (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TEST (1)
μs
Info
SYNCHRONIZATION (SYNC) (2)
t(trans-EtoI)
(1)
(2)
8
External clock to internal clock
Remove external clock, V(VIN) = 12 V,
V(VReg) = 5 V
32
PT = Production tested; CT = Characterization tested only, not production tested; Info = User information only, not production tested
The SYNC input clock can have a maximum frequency of 2× the programmed clock frequency up to a maximum value of 1.1 MHz.
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6.8 Typical Characteristics
90
95
85
90
85
80
Efficiency (%)
Efficiency (%)
6.8.1 Efficiency Data of Power Supply
75
70
80
75
70
65
Rslew = 35 kW
60
0
0.5
1
1.5
2
2.5
V(VIN) = 6 V
65
Rslew = 5 kW
V(VIN) = 20 V
60
0
3
0.5
1
Load Current (A)
V(VIN) = 12 V
L = 22 µH
1.5
2
2.5
3
Load Current (A)
V(VReg) = 5 V
C4 (output) = 100 µF
f(SW) = 500 kHz
TA = 25ºC
V(VReg) = 5 V
C4 (output) = 100 µF
Figure 1. FET Switching (Slow Slew Rate)
f(SW) = 500 kHz
Rslew = 5 kΩ
L = 22 µH
TA = 25ºC
Figure 2. Fast Slew Rate on Switching FET
6.8.2 Output Voltage Dropout
7
7
6.5
Intput Voltage (V)
Intput Voltage (V)
6
5.5
5
4.5
4
3.5
200 mA
3
536 mA
6
5
4
No Load
10 mA
3
50 mA
1.4 A
2.5
3A
2
0
1
2
3
4
5
6
2
2.5
100 mA
3
4
4.5
5
5.5
Output Voltage (V)
Output Voltage (V)
V(VReg) = 5 V
3.5
V(VReg) = 5 V
f(SW) = 500 kHz
Figure 3. Load Current > 100 mA
f(SW) = 500 kHz
Figure 4. Load Current < 100 mA
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Output Voltage Dropout (continued)
8
Start (Power Up)
Tracking (Power Down)
7
Input Voltage (V)
6
5
4
3
2
1
0
0
0.05
0.1
0.15
0.2
Load Current (A)
V(VReg) = 5 V
TA = 25ºC
Figure 5. Output-Voltage Tracking
NOTE
Tracking: The input voltage at which the output voltage drops approximately –0.7 V of the
regulated voltage or for low input voltages (tracking function) over the load range.
Start: The input voltage required to achieve 5-V regulation on power up with the stated
load currents.
6.8.3 Quiescent and Standby Current
77
68
76
Quiescent Current (μA)
Quiescent Current (μA)
67
66
65
64
63
62
61
75
74
73
72
71
70
69
68
60
59
–40
–20
0
20
40
60
80
100
120
140
67
–40
0
–20
40
60
80
100
120
140
V(VIN) = 24 V
V(VIN) = 12 V
Figure 6. LPM, Quiescent Current Variation With
Temperature
10
20
Free-Air Temperature (°C)
Free-Air Temperature (°C)
Figure 7. LPM, Quiescent Current Variation with
Temperature
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4.5
10
4
9
3.5
8
Shutdown Current (μA)
Shutdown Current (μA)
Quiescent and Standby Current (continued)
3
2.5
2
1.5
1
7
6
5
4
3
2
0.5
0
–40
1
–20
0
20
40
60
80
100
120
0
–40
140
0
–20
20
40
60
80
100
120
140
Free-Air Temperature (°C)
Free-Air Temperature (°C)
V(VIN) = 12 V
V(VIN) = 24 V
Figure 8. Shutdown Current
Figure 9. Shutdown Current
Current Consumption (mA)
5.6
5.55
5.5
5.45
5.4
5.35
5.3
5.25
–50
–30 –10
10
30
50
70
90
110
130 150
Free-Air Temperature (°C)
EN = High
V(VIN) = 12 V
Figure 10. Current Consumption vs Temperature
798.5
1006
798
1004
Voltage Drop on Rslew (mV)
Internal Reference Voltage (mV)
6.8.4 Reference Voltages
797.5
797
796.5
796
795.5
–40
–20
0
20
40
60
80
100
120
140
1002
1000
998
996
994
–40
–20
0
20
40
60
80
100
120
140
Free-Air Temperature (°C)
Free-Air Temperature (°C)
V(VIN) = 12 V
Figure 11. Internal Reference Voltage
Figure 12. Voltage Drop on Rslew for Current Reference
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7 Detailed Description
7.1 Overview
The TPS54362-Q1 device is a 60-V, 3-A DC-DC step-down (buck) converter using a voltage-control-mode
scheme. The device features a supervisory function for power-on-reset during system power on. When the
output voltage has exceeded the threshold set by the resistor network connected to the RST_TH pin, a delay of 1
ms per nF (based on the capacitor value on the RSTDLY pin) occurs before the release to high of the RST pin.
Conversely on power down, once the output voltage falls below the same set threshold, the device pulls RST low
only after a de-glitch filter of approximately 20 μs (typical) expires. The implementation of this function prevents
the triggering of RST due to fast transient line noise on the output supply.
An overvoltage monitor function limits output voltage to the threshold set by OV_TH. The external resistor
network sets both the RST_TH and OV_TH monitoring voltages to be a pre-scale of the output voltage, and the
internal bias voltages of the voltage comparators (0.8 V typical) are the basis of the thresholds.
The RST_TH setting is the basis for detection of undervoltage on the output, which invokes low assertion of the
RST pin. The OV_TH setting is the basis for detection of overvoltage on the output, which does not invoke low
assertion of the RST pin. However, the device commands the internal switch to turn OFF.
In systems where power consumption is critical, implementation of low-power mode reduces the non-switching
quiescent current during light load conditions. The system entering discontinuous current mode (DCM) for at
least 100 μs determines PFM operation. When the device enters discontinuous mode depends on the selection
of external components.
If excessive power dissipation causes invocation of thermal shutdown, the device disables the internal switch,
and the regulated output voltage starts to decrease. Depending on the load line, the regulated voltage could
decay and the RST_TH threshold may assert the RST output low.
7.2 Functional Block Diagram
BOOT
20
Band-Gap
Reference
LPM 4
D1
VIN
V(supply)
18
C1
R11
7
R7
Internal
supply
Internal
Voltage
Rail
19
16
Gate Drive with
Overcurrent Limit
for Internal Switch
5
R10
Rslew
0.8-V Vref(1)
0.2-V Vref(2)
EN
VIN
VReg
C3
L
PH
V(VReg)
17
6
RT
R8
Selectable
Oscillator
Thermal
Sensor
14
+
0.8-V Vref(1)
9
11
15
+ 0.8-V V
ref(1)
-
8
RST
Reset with
Delay Timer
GND
10
Voltage
comp
+
-
0.82-V Vref(3)
13
-
12
+ 0.8-V V
ref(1)
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R4
VSENSE
SS
C6
V(Vreg)
R12
12
R9
-
Cdly
C4
C7
Error
amp
SYNC 3
C2
D2
R5
C8
COMP
RST_TH
OV_TH
C5
R6
R1
R2
R3
C10
C9
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7.3 Feature Description
The TPS54362-Q1 device is a DC-DC converter using a voltage-control-mode scheme with an input voltage
feed-forward technique. The device is programmable for a range of output voltages with a wide input-voltage
range. The following are details with regard to the pin functionality.
7.3.1 Input Voltage
The VIN pin is the input power source for the TPS54362-Q1 device. This pin requires external protection against
voltage levels greater than 60 V and reverse battery. In buck mode, the device draws pulsed input current from
this pin, with fast rise and fall times. Therefore, this input line requires a filter capacitor to minimize noise.
Additionally, for EMI considerations, the line may also require an input-filter inductor.
7.3.2
Function Mode
FUNCTION
Buck
OPERATING VOLTAGE
RANGE
OUTPUT CURRENT CAPABILITY
COMMENTS
3.6 V to 48 V
V(VReg) = 0.9 V to 18 V and I(VReg) up to 3
A; however, at higher output power the
device requires derating for maximum
temperature rating.
Optimum performance: always set V(VIN)-toV(VReg) ratios such that the minimum required
duty cycle pulse (t(ON-Min)) > 150 ns. The
minimum off-time is 250 ns for all conditions.
7.3.3 Output Voltage V(VReg)
The converter supplied from battery voltage V(VIN) and the external components (L, C) generate the output
voltage, V(VReg). The VSENSE pin senses the output through an external resistor divider and compares it with an
internal reference voltage.
Selecting the external resistors according to the relationship in Equation 1 selects the value of the adjustable
output voltage between 0.9 V and 18 V in buck mode. Reference designators for the resistors in the following
equations refer to the Functional Block Diagram.
V(VReg) = Vref (1 + R4 / R5)
where
•
•
R5 and R4 are feedback resistors
Vref = 0.8 V (typical)
(1)
The internal reference voltage has a ±1.5% tolerance. The overall output voltage tolerance depends on the
external feedback resistors. To determine the overall output voltage tolerance, use the following relationship:
V(VReg-tol) = V(VRef-tol) + (R4 / (R4 + R5)) ´ (R4-tol+ R5-tol)
where
•
•
R4 and R5 are feedback resistors
Vref = 0.8 V (typical)
(2)
The VReg pin also connects internally to a load of 100 Ω, which turns ON in the following conditions:
• During startup conditions, when the device is powered up with no load, or whenever EN is toggled, the
internal load connected to the VReg pin turns ON for about 100 µs to charge the bootstrap capacitor to
provide gate drive voltage to the switching transistor.
• During normal operating conditions, when the regulated output voltage exceeds the overvoltage threshold
(preset by external resisitors R1, R2, and R3), the internal load turns ON, pulling this pin down to bring the
regulated output voltage down.
Typically, the output uses a capacitor within the range of 10 μF to 400 μF. This pin must have a filter capacitor
with low ESR characteristics in order to minimize output ripple voltage.
7.3.4 Oscillator Frequency (RT)
Oscillator frequency is selectable by means of a resistor placed at the RT pin. The switching frequency (f(SW)) can
be set in the range of 200 kHz–2200 kHz. In addition, a clock signal (f(ext)) at the SYNC pin with f(SW) < f(ext) < 2 ×
f(SW) can externally impose the switching frequency. In this case, the external clock overrides the switching
frequency determined by the RT pin, and the external synchronization signal clocks the internal oscillator.
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Switching Frequency (kHz)
2000
8V
1800
14 V
1600
24 V
1400
40 V
1200
1000
800
600
400
200
0
100
200
300
400
500
600
Resistor on RT (kW)
Figure 13. Switching Frequency vs Resistor Value
7.3.5 Synchronization (SYNC)
The SYNC pin is an external input signal to synchronize the switching frequency using an external clock signal.
The synchronization input overrides the internally fixed oscillator signal. The synchronization signal must be valid
for approximately 2 clock cycles (pulses) before the transition is made for synchronization with the external
frequency input. If the external clock input does NOT transition low or high for 32 μS (typical), the system
defaults to the internal clock set by the RT pin. The SYNC input clock can have a maximum frequency of 2× the
programmed clock frequency up to a maximum value of 2.2 MHz.
7.3.6 Enable or Shutdown (EN)
The enable pin provides electrical on-off control of the regulator. Once the enable pin voltage exceeds the
threshold voltage, the regulator starts operation, and the internal soft start begins to ramp. Pulling the enable pin
voltage below the threshold voltage stops the regulator from switching, and the internal soft start resets.
Connecting the pin to ground or to any voltage less than 0.7 V disables the regulator and activates the shutdown
mode. This pin must have an external pullup or pulldown to change the state of the device.
7.3.7 Reset Delay (Cdly)
The reset-delay pin sets the desired delay time to assert the RESET pin high after the supply has exceeded the
programmed VReg_RST voltage. One can program the delay in the range of 2.2 ms to 200 ms using capacitors
in the range of 2.2 nF to 200 nF. Use Equation 3 to calculate the delay time.
t d(POR) = 1 ms / nF ´ C
where
•
C = capacitance on the Cdly pin
(3)
7.3.8 Reset Pin (RST)
The RST pin is an open-drain output. The device asserts the power-on-reset output low until the output voltage
exceeds the programmed VReg_RST voltage threshold and the reset delay timer has expired. Additionally,
whenever the EN pin is low or open, the device immediately asserts RST low regardless of the output voltage. A
reset filter timer prevents reset being invoked because of short negative transients on the output line. If a thermal
shutdown occurs due to excessive thermal conditions, the device asserts this pin low when the switching FET is
OFF and output falls below the reset threshold.
14
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VIN
VIN
SS
SS
VReg
0.92 ´ V(VReg)
Set by RST_TH
Terminal
VReg
Cdly
Cdly
td(POR)
RST
RST
20 ms (Typical
Deglitch Time)
Figure 14. Power-On Condition, Reset Line
Figure 15. Power-Down Condition, Reset Line
7.3.9 Boost Capacitor (BOOT)
This capacitor provides the gate-drive voltage for the internal MOSFET switch. TI recommends X7R or X5R
grade dielectrics because of the stable values over temperature of these dielectrics. Decreasing the value of the
boost capacitor for low-Vreg applications, high-frequency applications, or combination applications may be
necessary. Increasing the value of the boost capacitor for high-Vreg applications, low-frequency applications, or
combination applications may be necessary (for example, 100 nF for 500 kHz at 5 V or 220 nF for 500 kHz at 8
V).
7.3.10 Soft Start (SS)
On power up or after a short-circuit event , TI recommends the following conditions:
• V(VIN) – V(VReg) > 2.5 V
• Load current < 1 A, until RST goes high
C(SS) is 1 nF to 220 nF. If the buck converter starts up with output shorted to ground, the value of C(SS) must be a
minimum of 150 nF.
7.3.11 Short-Circuit Protection
The TPS54362-Q1 device features output short-circuit protection. Detection of short-circuit conditions is by
monitoring RST_TH, and when the voltage on this node drops below 0.2 V, the switching frequency decreases
and the current limit folds back to protect the device. The switching frequency folds back to approximately 25 kHz
and the current limit reduces to 30% of the current-limit typical value.
7.3.12 Overcurrent Protection
Implementation of overcurrent protection is by sensing the current through the NMOS switch FET and a
comparison of the sensed current to a current reference level representing the overcurrent threshold limit.
Sensed current exceeding the overcurrent threshold limit sets the overcurrent indicator to true. The system
ignores the overcurrent indicator for the leading edge blanking time at the beginning of each cycle to avoid any
turnon-noise glitches.
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Setting the overcurrent indicator to true triggers overcurrent protection. The MOSFET turns off for the rest of the
cycle after a propagation delay. The name of the overcurrent protection scheme is cycle-by-cycle current limiting.
If the sensed current continues to increase during cycle-by-cycle current limiting, the temperature of the device
starts rising. At the temperature limit, thermal shutdown (TSD) kicks in and shuts down switching until the device
cools sufficiently.
CAUTION
In certain conditions, device damage may occur under a shorted load condition,
depending on the supply voltage. The design of the TPS54362-Q1 devices is for
protection from damage due to a shorted load condition using a cycle-by-cycle current
limit, the short-circuit protection function, and/or the thermal shutdown function.
Short-circuit detection protects the device from damage when encountering a 0-Ω
short-circuit condition. However, damage to a device may occur when the shorted load
has some resistivity and the output level stays higher than the short-circuit detection
level of 0.2 V. In this case, the inductor current increases until the junction temperature
of the device hits the thermal shutdown threshold, but damage to the switching FET
may occur before thermal shutdown.
This failure only occurs during an output short circuit with some resistivity when the
supply voltage is above 18 V.
7.3.13 Slew Rate Control (Rslew)
The Rslew pin controls the switching slew rate of the internal power NMOS. An external resistor with a slew-rate
range for rise and fall times as shown in Figure 16 and Figure 17 sets the slew rate. The range of rise time tr =
15 ns to 35 ns, and fall time tf = 15 ns to 200 ns, with Rslew range of 10 kΩ to 50 kΩ (see Figure 16 and
Figure 17).
35
350
30
300
25
250
8V
14 V
Fall Time (ns)
Rise Time (ns)
24 V
20
15
8V
10
40 V
200
150
100
14 V
5
50
24 V
40 V
0
10
20
30
40
50
60
0
70
10
Slew Resistor (kW)
20
30
40
50
60
70
Slew Resistor (kW)
Figure 16. FET Rise Time
Figure 17. FET Fall Time
7.3.14 Thermal Shutdown
The TPS54362-Q1 device protects itself from overheating with an internal thermal-shutdown circuit. If the
junction temperature exceeds the thermal-shutdown trip point, the MOSFET turns off. The device restarts
automatically under control of the slow-start circuit when the junction temperature drops below the thermalshutdown hysteretic trip point. Operating in low-power mode disables the thermal-shutdown sensing circuitry for
low current consumption. Asserting RST or V(VReg_UV) low activates thermal-shutdown monitoring.
7.3.15 Regulation Voltage (VSENSE)
Use of the VSENSE pin is for programming the regulated output voltage based on a resistor feedback network
monitoring the V(VReg) output voltage. The selected ratio of R4 to R5 sets the VReg voltage.
16
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7.3.16 RESET Threshold (RST_TH)
This pin is programmable for setting the undervoltage threshold level (V(VReg_UV)) for proper regulation in lowpower mode and the reset threshold level (V(VReg_RST)) to initiate a reset-output signal. The resistor combination
of R1 to R3 programs the threshold for detection of undervoltage. Voltage bias on R2 + R3 sets the reset
threshold.
Undervoltage for transient and low-power-mode operation:
V(VReg_UV) = 0.82 V ´ (R1 + R2 + R3 / (R2 + R3)
(4)
Reset threshold = V(VReg_RST) = 0.8 V ´ (R1 + R2 + R3 / (R2 + R3)
(5)
Recommended range: 70% to 92% of the regulation voltage
7.3.17 Overvoltage Supervisor for V(VReg) (OV_TH)
This pin is programmable to set the overvoltage monitoring of the regulated output voltage. The resistor
combination of R1 to R3 programs the threshold for detection of overvoltage. The bias voltage of R3 sets the OV
threshold and the output voltage accuracy in hysteretic mode during transient events.
Overvoltage reference = V(VReg _OV) = 0.8 V ´ (R1 + R2 + R3) / (R3)
(6)
Recommended range: 106% to 110% of the regulation voltage
7.3.18 Noise Filter on RST_TH and OV_TH Pins
Some noise sensitivity exists on the RST_TH and OV_TH pins, and added capacitance filters this noise. The
noise is more pronounced with fast falling edges on the PH pin. So a smaller Rslew resistor (minimum
recommended value is 10 kΩ) may require more capacitance on RST_TH and OV_TH. Users should use the
smallest capacitance necessary, because larger values increase the loop response time and degrade shortcircuit protection and transient response. The 2-μs maximum time constant seen on OV_TH and RST_TH when
V(VReg) = 0 V (that is, [R2 + R3] × [C9 + C10] < 2 μs) determines the upper limit. The noise in the RST_TH and
OV_TH resistor chain may change with PCB layout or application setup, so there may not be a requirement for
the RST_TH and/or OV_TH capacitor in all applications. Users can place the footprint and then populate it only if
necessary.
Example:
R1 = 36 kΩ
R2 = 600 Ω
R3 = 6.6 kΩ
V(VReg _RST) = 0.8 ´ (43.2 kW) / 7.2 kW = 4.8 V
(7)
V(VReg _OV) = 0.8 ´ (43.2 kW) / 6.6 kW) = 5.24 V
(8)
VReg
RST _TH
C4
R1
R2
C10
OV_TH
R3
C9
Figure 18. Resistor Network and Noise Filters
Typical values for the RST_TH and OV_TH capacitors are in the 10-pF to 100-pF range for total resistance on
the RST_TH-OV_TH divider of < 200 kΩ.
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7.3.19 Output Tolerances Based on Modes of Operation
V(VReg_OV)
V(VReg_UL)
VReg Output
V(VReg_LL)
V(VReg_UV) = 0.82 V ´ (R1 + R2 + R3) / (R2 + R3)
V(VReg_RST) = 0.8 V ´ (R1 + R2 + R3) / (R2 + R3)
Load Reg. and
Line Reg. in
Hysteretic Mode
Low-Power
Mode (LPM)
Active
(Normal)
Figure 19. Modes of Operation
empty paragraph for space between the illustration and table
Table 1. Values for Threshold Voltages in Each Mode of Operation
MODE OF OPERATION
V(VReg) - LOWER LIMIT
V(VReg) – UPPER LIMIT
COMMENTS
Hysteretic mode
0.82 V × (R1 + R2 + R3) / (R2 + R3)
0.8 V × (R1 + R2 + R3) / (R3)
Minimum to maximum ripple on
output
Low-power mode
0.82 V × (R1 + R2 + R3) / (R2 + R3)
V(VReg) + V(VReg-tol)
Minimum to maximum ripple on
output
Active (normal)
V(VReg) – V(VReg-tol)
V(VReg) + V(VReg-tol)
Minimum to maximum ripple on
output
empty paragraph for space between the two tables
Table 2. Values for Threshold Voltages of Voltage Supervisors
SUPERVISOR
THRESHOLDS
V(VReg) - TYPICAL VALUE
TOLERANCE
COMMENTS
Overvoltage
0.8 V × (R1 + R2 + R3) / (R3)
± (V(VRef-tol) + (R1 + R2 / [R1 + R2 + R3])
× (R1-tol + R2-tol + R3-tol)
Overvoltage threshold
setting
Reset
0.8 V × (R1 + R2 + R3) / (R2 + R3)
± (V(VRef-tol) + (R1 / [R1 + R2 + R3]) ×
(R1-tol + R2-tol + R3-tol)
Reset threshold setting
7.3.20 Load Regulation and Line Regulation in Hysteretic Mode
This mode of operation is when a load or line transient step occurs in the application. The converter goes into a
hysteretic mode of operation until the error amplifier stabilizes and controls the output regulation to a tighter
output tolerance. During the load step, V(VReg_OV) sets the regulator upper threshold and the V(VReg_UV) limit sets
the lower threshold.
The converter enters this mode of operation during load- or line-transient events if the main control loop cannot
respond to regulate within the specified tolerances. The regulator exits this mode once the main control loop
responds.
18
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7.3.21 Internal Undervoltage Lockout (UVLO)
On power up, the internal band-gap and bias currents attaining stability, which is typically at
V(VIN) = 3.4 V (minimum), enables the IC. On power down, disabling the internal circuitry occurs at V(VIN) = 2.6
V (maximum).
7.3.22 Loop-Control Frequency Compensation
L
VReg
C8
C7
ESRC4
R9
C5
R6
R4
C4
VSENSE
Error
Amplifier
R5
COMP
Vref = 0.8 V
Figure 20. Type III Compensation
7.3.22.1 Type III Compensation
f(c) = f(SW) × 0.1 (unity-gain frequency is the name of the cutoff frequency when the gain is 1).
f(c) is typically 1/5 to 1/10 of the switching frequency, and is typically greater than five times the double-pole
frequency of the LC filter.
Equation 9 and Equation 10 derive the modulator break frequencies as a function of the output LC filter. The LC
output filter gives a double pole, which has a –180 degree phase shift.
1
f(LC) =
1/2
2p (L ´ C4 )
(9)
The ESR of the output capacitor C gives a zero that has a 90° phase shift.
1
f (ESR) =
2p ´ C4 ´ ESR C4
(10)
(R4 + R5)
V(Vreg) = Vref ´
R5
V(Vreg)
(R4 + R5)
=
0.8 V
R5
(11)
(12)
The V(VIN) / V(ramp) modulator gain is about 10 for 8 V < VIN < 50 V. V(ramp) is fixed at 1 V for V(VIN) < 8 V and at
5 V for V(VIN) > 48 V.
Note that the V(VIN) / V(ramp) gain (A(mod)) is not precise and has a tolerance of about 20%.
V(VIN)
V(ramp) =
10
æ V(VIN) ö
Gain (dB) = 20 ´ log ç
÷
ç V(ramp) ÷
è
ø
Gain = 20 ´ log 10 = 20 dB
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(13)
(14)
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f (p1) =
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(C5 + C8)
2p ´ R6 ´ (C5 ´ C8)
(15)
1
2p ´ R9 ´ C7
1
=
2p ´ R6 ´ C5
1
=
2p ´ (R4 + R9 ) ´ C7
f (p2) =
f (z1)
f (z2)
(16)
(17)
(18)
7.3.23 Bode Plot of Converter Gain
Open-Loop
Error-Amplifier Gain
f(p1) f(p2)
Gain - dB
f(z1) f(z2)
20 log R6 (R4 + R9) / (R4 ´ R9)
20 log (R6 / R4)
20 log (10)
Modulator Gain
Compensation
Gain
Closed-Loop Gain
f(LC)
f(ESR)
Frequency - Hz
Figure 21. Bode Plot of Converter Gain
7.4 Device Functional Modes
7.4.1 Low-Power Mode (LPM)
The TPS54362-Q1 device automatically enters low-power mode once the regulation goes into discontinuous
mode. The internal control circuitry for any transition from low-power mode to high-power mode occurs within 5
μs (typical). In low-power mode, the converter operates as a hysteretic controller with the threshold limits set by
V(VReg_UV) = 0.82 × (R1 + R2 +R3 / (R2 + R3) for the lower limit and approximately V(VReg) for the upper limit. To
ensure tight regulation in the low-power mode, set the R2 and R3 values accordingly.
The device operates with both automatic and digitally controlled low-power mode. The digital low-power mode
can override the automatic low-power mode function by applying the appropriate signal on the LPM pin. The
device goes into active or normal mode for at least 100 μs on the tripping of RST_TH or VREG_UV. Operating in
active mode or normal mode enables all blocks, including the OV function.
Being in LPM disables the OV function.
Active or normal mode: When the device is in DCM with LPM = High or in CCM with LPM = High or Low
LPM: When part is in DCM with LPM = Low
Automatic and Digital
LPM high:
20
LPM high forces the device to normal mode at fixed frequency, even at light load
current (the device does pulse skipping to keep output voltage in regulation at light
loads).
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Device Functional Modes (continued)
LPM low or open:
The device automatically changes between normal and low-power mode depending on
load current.
7.4.2 Buck-Mode Low-Power-Mode Operation
When operating in low-power mode (buck regulator) with the output shorted to ground, the device asserts a
reset. The thermal-shutdown and current-limiting circuitry activates to protect the device.
Low-power-mode operation begins once the converter enters the discontinuous mode of operation.
7.4.3 External LPM Operation
The low-power mode (LPM) is active-low; if there is an open on this pin the IC enters the low-power mode
(internal pulldown).
To allow low-power mode operation, the load current must be low with the LPM pin set to ground.
To inhibit low-power mode, the microcontroller must drive the pin high, and the converter must not be in
discontinuous mode of operation.
The device can only power up in LPM or DCM if V(VReg) < 5.5 V and V(VIN) – V(VReg) > 2.5 V.
In active mode. the device powers up when V(VIN) > 3.6 V (minimum).
NOTE
Being in LPM prevents enabling of the OV_TH circuit.
Active or normal mode: When the device is in CCM or DCM with LPM = High
LPM: When the device is in DCM with LPM = Low
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8 Application and Implementation
8.1 Application Information
This section is a starting point, with theoretical representation of the values used for the application. Improving
the performance of the device may require further optimization of the component values.
8.2 Typical Application
D1
J1
1
C11
3
GND
V(VReg)
GND
GND
V(VReg)
GND
R11
R8
102k
221k
4
5
6
R7
30.1k
7
R12
2k
8
C2
199.2nF
10
NC
VIN
SYNC
VIN
LPM
PH
EN
VReg
RT
COMP
Rslew
VSENSE
RST
RST_TH
CDLY
OV_TH
AGND
SS
20
0.1uF
C3
V(VIN) = 8 V to 28 V
V(VReg)
18
L1 22uH
1
17
16
1
GND
19
V(VReg)
GND
2
D2
R9
C5
15
330pF
C12
2.55k
R4
C8
R6
C7
22pF
274K
220pF
+
0.1uF
14
J2
187k
C4
2
100uF
1
V(VReg) = 3.3 V
IO = 2.5 A
13
GND
12
11
C6
21
GND
9
BOOT
220uF
1
2
NC
2
2
GND
1
PAD
GND
0.1uF
TPS54362x-Q1
C1
2
U1
+
R5
V(VReg)
150 nF
35.7k
R1
GND
82.5k
GND
GND
GND
R2
C10
2.32k
22pF
C9
R3
22pF
15k
GND
GND
Figure 22. Schematic of Typical Application
8.2.1 Design Requirements
For this design example, use the parameters listed in Table 3.
Table 3. Design Parameters
22
DESIGN PARAMETER
EXAMPLE VALUE
Input voltage, VI
8 V to 28 V
Output voltage, VO
5 V ± 2%
Maximum output current, IO-max
3A
Transient response 0.25-A to 2.25-A load step
ΔVO = 5%
Reset threshold
92% of output voltage
Overvoltage threshold
106% of output voltage
Undervoltage threshold
95% of output voltage
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8.2.2 Detailed Design Procedure
8.2.2.1 Selecting the Switching Frequency
The user selects the switching frequency based on the minimum on-time of the internal power switch, the
maximum input voltage, the minimum output voltage, and the frequency-shift limitations. Use Equation 19 to find
the maximum frequency for the regulator. Determine the value of the resistor to connect to the RT pin to set this
frequency from Figure 13.
æ VO(min) ö
ç
÷
ç VI(max) ÷
è
ø (Hz)
f(SW - max) =
t(ON-Min)
where
•
•
f(SW-max) = 770 kHz
t(ON-Min) = 150 ns from the Electrical Characteristics table
(19)
Because the oscillator can vary 10%, decrease the frequency by 10%. Further, to keep the switching frequency
outside the AM band, one can select f(sw) as 400 kHz (500 kHz in the application example).
8.2.2.2 Output Inductor Selection (LO)
Calculate the minimum inductor value using Equation 21.
k(IND) is the coefficient that represents the amount of inductor ripple current relative to the maximum output
current. Calculate the ripple using Equation 20.
The output capacitor filters the inductor ripple current, and so the typical range of this ripple current is in the
range calculated with k(IND) = 0.2 to 0.3, depending on the ESR and the ripple-current rating of the output
capacitor. The minimum inductor value calculated is 14.5 μH; choose an inductor ≈ 22 μH.
I (Ripple) = k (IND) ´ I O
where
•
I(Ripple) = 0.2 × 2.5 = 0.5 A (peak-to-peak)
(20)
Calculate inductor L(O):
L (O-min) =
(VI(max) - VO ) × VO
f (SW) × I(Ripple) × VI(max)
(Henries)
where
•
•
f(SW) is the regulator switching frequency
I(Ripple) = Allowable ripple current in the inductor, typically 20% of maximum IO
(21)
The RMS (root-mean-square) and peak current flowing in the inductor is:
IL(RMS) =
(I O )2 +
(I(Ripple) ) 2
Inductor peak current:
I(Ripple)
IL(pk) = I O +
2
8.2.2.3
12
(Amperes)
(22)
(Amperes)
(23)
Output Capacitor (CO)
The selection of the output capacitor determines several parameters in the operation of the converter, the
modulator pole, voltage droop on the output capacitor, and the output ripple.
During a load step from no load to full load or changes in the input voltage, the output capacitor must hold up the
output voltage above a certain level for a specified time and not issue a reset, until the main regulator control
loop responds to the change. Equation 25 determines the minimum output capacitance required to allow
sufficient droop on the output voltage without issuing a reset.
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The capacitance value determines the modulator pole and the rolloff frequency due to the LC output filter double
pole – Equation 9.
The output ripple voltage is a product of the output capacitor ESR and ripple current – Equation 27.
Using Equation 24, the minimum capacitance needed to maintain the desired output voltage during a high-to-low
load transition and prevent overshoot is 157 μF.
C(O) =
L ´ éI O(max)2 - I O(min)2 ù
ë
û
2
2
VO(max) - VO(min)
(Farads)
where
•
•
IO(max) is the maximum output current
IO(min) is the minimum output current
The difference between the output current maximum-to-minimum is the worst-case load step in the system
•
•
VO(max) is maximum tolerance of regulated output voltage
VO(min) is the minimum tolerance of regulated output voltage
(24)
The calculation of minimum capacitance needed for transient load response, using Equation 25, yields 53 μF.
2 ´ DI O
C(O) >
(Farads)
f (SW ) ´ DVO
(25)
The calculation of minimum capacitance needed for output voltage ripple specification, using Equation 26, yields
1.18 μF.
1
1
´
C(O) >
(Farads)
8 ´ f (SW )
æ VO(Ripple) ö
ç
÷
ç I(Ripple) ÷
è
ø
(26)
The most critical condition based on the foregoing calculations indicates that the output capacitance must be a
minimum of 157 μF to keep the output voltage in regulation during load transients.
Factoring in additional capacitance de-ratings for temperature, aging, and dc bias yields a value of 220 μF.
Equation 27 calculates the ESR required to meet the ripple-voltage tolerance of the system, but for system
stability the ESR should not exceed 100 mΩ.
Maximum ESR of the out capacitor based on output ripple voltage specification is:
VO(Ripple)
R (ESR) <
(Ohms)
I(Ripple)
(27)
Output capacitor root-mean-square (rms) ripple current. This is to prevent excess heating or failure due to high
ripple currents. This parameter is sometimes specified by the manufacturers.
I O(RMS) =
VO ´
(VI(max) ) - VO )
12 ´ VI(max) ´ L (O) ´ f (SW )
(Amperes)
(28)
8.2.2.4 Flyback Schottky Diode
The TPS54362-Q1 device requires an external Schottky diode connected between the PH pin and the power
ground termination. The absolute voltage at the PH pin should not go beyond the values mentioned in Absolute
Maximum Ratings. The Schottky diode conducts the output current during the off state of the internal power
switch. This Schottky diode must have a reverse breakdown higher than the maximum input voltage of the
application. The low forward voltage of a Schottky diode makes it ideal for this situation. Select the Schottky
diode based on the appropriate power rating, which factors in the dc conduction losses and the ac losses due to
the high switching frequencies; Equation 29 determines the power requirement.
24
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P(diode)
æ é VI(max) - VO ù ´ I O ´ V(fd)
û
= çë
ç
VI(max)
è
2
æé
ö
ö
VI - V(fd) ûù ´ f (SW ) ´ C j ÷
ç
ë
÷ +
ç
÷
÷
2
ç
÷
ø
è
ø
(Watts)
where
•
•
VF = forward conducting voltage of Schottky diode
Cj = junction capacitance of the Schottky diode
(29)
The recommended part numbers of the Flyback Schottky diodes are PDS360 and SBR8U60P5.
8.2.2.5 Input Capacitor, C(I)
The requires an input ceramic decoupling capacitor with type X5R or X7R dielectric, and bulk capacitance to
minimize input ripple voltage. The dc voltage rating of this input capacitance must be greater than the maximum
input voltage. The capacitor must have an input ripple current rating higher than the maximum input ripple current
of the converter for the application; Equation 30 determines the ripple current.
The input capacitors for power regulators are chosen to have reasonable capacitance-to-volume ratio and fairly
stable over temperature. The value of the input capacitance also determines the input ripple voltage of the
regulator, shown by Equation 31.
II(RMS) = I O ´
DVI =
VO
VI(min)
´
I O(max) ´ 0.25
VI(min) - VO
VI(min)
(Amperes)
(30)
(Volts)
C(I) ´ f (SW )
(31)
8.2.2.6 Output Voltage and Feedback Resistor Selection
In the design example, the R4 selection is 187 kΩ; using Equation 1, R4 calculates as 35.7 kΩ. To minimize the
effect of leakage current on the VSENSE pin, the current flowing through the feedback network should be greater
than 5 μA in order to maintain output accuracy. Higher resistor values help improve the converter efficiency at
low output currents, but may introduce noise immunity problems.
8.2.2.7 Overvoltage Resistor Selection
Use Equation 6 to determine the value of R3 to set the overvoltage threshold at 1.06 × 5.5 V. The total resistor
network from the VReg output to ground is approximately 100 kΩ (this is R1 + R2 +R3). The calculated value of
R3 is then 15.09 kΩ. Use the nearest standard value, which is 15 kΩ. This pin may require a noise decoupling
capacitor to ensure proper operation; the value chosen for this design is 56 pF.
8.2.2.8 Reset-Threshold Resistor Selection
Using Equation 5, calculate the value of R2 + R3, and then knowing R3 from the OV_TH setting, determine R2.
The value of R2 + R3 yields 17.39 kΩ, which means R2 is approximately 2.32 kΩ. This sets the reset threshold
at 0.92 × 5 V. This pin may require a noise-decoupling capacitor to ensure proper operation; the value chosen for
this design is 15 pF. The value determined for R1 is 82.5 kΩ.
8.2.2.9 Low-Power Mode Threshold
To obtain an approximation of the output load current at which the converter is operating in discontinuous mode,
use Equation 32. The values used in the equation for minimum and maximum input voltage affect the duty cycle
and the overall discontinuous-mode (DCM) load current. With a maximum input voltage of 28 V, the output load
current for DCM is 165.8 mA, and for minimum input voltage of 8 V, the DCM-mode load current is 111.7 mA.
These are nominal values, calculated without taking into consideration other factors like external component
variations with temperature and aging.
IL(DCM) = I L(LPM) =
(1
- D ) ´ VO
2 ´ f (SW ) ´ L (O)
(Amperes) (with ± 30% hysteresis)
where
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•
•
•
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IL(DCM) = Output load current at which the converter is operating in discontinuous mode
IL(LPM) = Output load current at which the converter is operating in low-power mode
D = Duty cycle
(32)
8.2.2.10 Undervoltage Threshold for Low-Power Mode and Load-Transient Operation
Setting the undervoltage threshold above the reset threshold ensures the regulator operates within the specified
tolerances during output load transients of low load to high load and during discontinuous conduction mode.
Using Equation 4, determine the typical voltage threshold.
In this design, the value for this threshold is 0.95 × 5 V.
8.2.2.11 Soft-Start Capacitor
The soft-start capacitor determines the minimum time to reach the desired output voltage during a power-up
cycle. This time is important when a load requires a controlled voltage-slew rate. Soft starting helps to limit the
current draw from the input voltage supply line. This design requires a 4.7-nF capacitor to meet the soft-start
criteria. If the buck converter starts up with output shorted to ground, the circuit requires a TPS54362-Q1 device
and a minimum 150-nF C(SS).
8.2.2.12 Bootstrap Capacitor Selection
Connect a 0.1-μF ceramic capacitor between the PH and BOOT pins for the converter to operate and regulate
the desired output voltage. TI recommends using a capacitor with X5R or better-grade dielectric material, and a
voltage rating on this capacitor of at least 25 V to allow for derating.
8.2.2.13 Guidelines for Compensation Components
1
Make the two zeroes close to the double pole (LC), for example, f(z1) ≈ f(z2) ≈ 2 x p LCO .
1. Make the first zero below the filter double pole (approximately 50% to 75% of f(LC)).
2. Make the second zero at the filter double pole (f(LC)).
Make the two poles above the crossover frequency f(c),
1. Make the first pole at the ESR frequency (f(ESR)).
2. Make the second pole at 0.5 the switching frequency (0.5 × f(SW)).
Select R4 = 187 kW
R5 =
R6 =
(R4 ´ 0.8)
(VO - 0.8)
(33)
f (c) ´ V(ramp) ´ R4
(f (LC) ´ VI )
(34)
Calculate C5 based on placing a zero at 50% to 75% of the output-filter double-pole frequency.
1
C5 =
p ´ R6 ´ f (LC)
(35)
Calculate C8 by placing the first pole at the ESR zero frequency.
C5
C8 =
(2p ´ R6 ´ C5 ´ f (ESR) - 1)
(36)
Set the second pole at 0.5 times the switching frequency, and also set the second zero at the output-filter
double-pole frequency.
R4
R9 =
f(SW)
- 1
2 ´ f(LC)
(37)
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C7 =
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1
p ´ R9 ´ f (SW)
(38)
8.2.2.14 Compensation
8.2.2.14.1 Calculate the Loop Compensation
DC modulator gain (A(mod)) = 8 / V(ramp)
V(ramp) = 0.8 V
A(mod) (dB) = 20 log (10) = 20 dB
Output filter due to LCO poles and CO ESR zeros from Equation 9 and Equation 10.
f(LC) = 3.4 kHz for LCO = 22 µH, CO = 100 µF
f(ESR) = 15.9 kHz for CO = 100 µF, ESR = 100 mΩ
Choose R4 = 187 kΩ.
Calculate the poles and zeros for a type III network using equations Equation 33 to Equation 38.
R5 = 35.7 kΩ (use standard value 35.7 kΩ)
R6 = 276 kΩ (use standard value 274 kΩ)
C5 = 340 pF (use standard value 330 pF)
C8 = 40.6 pF (use standard value 22 pF)
R9 = 2.57 kΩ (use standard value 2.55 kΩ)
C7 = 247 pF (use standard value 220 pF)
Calculate the poles and zeros based on these compensation values, using Equation 15 through Equation 18.
8.2.2.14.2 Power Dissipation
The power dissipation losses applicable for continuous-conduction-mode operation (CCM) are:
V
P(CON) = I O2 ´ rDS(on) ´ O
(Conduction losses)
VI
P(SW) = 1/2 ´ VI ´ I O ´ (tr + t f ) ´ f (SW)
P(Gate) = V(drive) ´ Q g ´ f (SW)
P(IC) = VI ´ I(q-normal)
(Switching losses)
(Gate drive losses) where Q g = 1 ´ 10
(40)
-9
(nC)
(Supply losses)
PT = P(CON) + P(SW) + P(Gate) + P(IC)
(39)
(41)
(42)
(Watts)
(43)
where:
VO = Output voltage
VI = Input voltage
IO = Output current
tr = FET switching rise time (maximum tr = 40 ns)
tf = FET switching fall time
V(drive) = FET gate-drive voltage (typically V(drive) = 6 V and maximum V(drive) = 8 V)
f(sw) = Switching frequency
For a given operating ambient temperature TA
TJ = TA + RqJA ´ PT
(44)
For a given maximum junction temperature TJ-Max = 150°C
TA(Max) = TJ(Max) - RqJA ´ PT
(45)
where:
PT = Total power dissipation (watts)
TA = Ambient temperature in °C
TJ = Junction temperature in °C
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TA(Max) = Maximum ambient temperature in °C
TJ(Max) = Maximum junction temperature in °C
RθJA = Thermal resistance of package in (°C/W)
Other factors not included in the preceding information which affect the overall efficiency and power losses are:
• Inductor ac and dc losses
• Trace resistance and losses associated with the copper trace routing and connections
• Flyback catch diode
The output current rating for the regulator may require derating for ambient temperatures above 85°C. The
derating value depends on the calculated worst-case power dissipation and the thermal management
implementation in the application.
Power Dissipation (W) versus Ambient Temperature (C)
4.00
Power Dissipation (W)
3.50
3.00
2.50
2.00
1.50
1.00
0.50
0.00
-40
-20
0
20
40
60
80
100
120
140
Ambient Temperture (C)
Figure 23. Power Dissipation Derating
8.2.3 Application Curves
CH1: ILOAD = 500 mA/div
CH3: V(VReg) = 1 V/div
CH3: V(VReg) = 1 V/div
CH1: ILOAD = 500 mA/div
CH2: V(VIN) = 5 V/div
CH2: V(VIN) = 5 V/div
CH4: RST = 5 V/div
CH4: RST = 5 V/div
100 μs/div
CH1: IO
CH4: RST
CH2: VIN
V(VIN) = 12 V
100 μs/div
CH3: Vreg
V(VReg) = 5 V
Figure 24. Transition Response (IO from 0 A to 2 A)
28
CH1: IO
CH4: RST
CH2: VIN
V(VIN) = 12 V
CH3: Vreg
V(VReg) = 5 V
Figure 25. Transition Response (IO from 0 A to 3 A)
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CH3: V(VReg) = 1 V/div
CH3: V(VReg) = 1 V/div
CH2: V(VIN) = 5 V/div
CH2: V(VIN) = 5 V/div
CH4: PH = 5 V/div
CH4: PH = 5 V/div
100 ms/div
CH2: VIN
V(VIN) = 12 V
CH3: Vreg
V(VReg) = 5 V
200 μs/div
CH4: PH (100 ms/div)
CH4 = 1.905 kHz
Figure 26. Output Ripple (IO = 0 A)
CH2: VIN
V(VIN) = 12 V
CH3: Vreg
V(VReg) = 5 V
Figure 27. Output Ripple (IO = 100 mA)
CH3: V(VReg) = 1 V/div
CH3: V(VReg) = 1 V/div
CH2: V(VIN) = 5 V/div
CH2: V(VIN) = 5 V/div
CH4: PH = 5 V/div
CH4: PH = 5 V/div
1 μs/div
CH2: VIN
V(VIN) = 12 V
CH4: PH (200 µs/div)
CH4 = 45.7 kHz
CH3: Vreg
V(VReg) = 5 V
1 μs/div
CH4: PH (1 µs/div)
CH4 = 519.1 kHz
CH2: VIN
V(VIN) = 12 V
Figure 28. Output Ripple (IO = 1 A)
CH3: Vreg
V(VReg) = 5 V
CH4: PH (1 µs/div)
CH4 = 523.5 kHz
Figure 29. Output Ripple (IO = 3 A)
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9 Power Supply Recommendations
The design of the TPS54362-Q1 devices is for operation using an input supply range from 3.6 V to 48 V. This
input supply should be well regulated. If there is a possibility for a reverse-voltage condition to occur, place a
series Schottky diode in the power routing.
10 Layout
10.1 Layout Guidelines
TI recommends the following guidelines for PCB layout of the TPS54362-Q1 device.
10.1.1 Inductor
Use a low-EMI inductor with a ferrite-type shielded core. Other types of inductors may be used; however, they
must have low-EMI characteristics and be located away from the low-power traces and components in the circuit.
10.1.2 Input Filter Capacitors
Input ceramic filter capacitors should be located in close proximity to the VIN pin. TI recommends surface-mount
capacitors to minimize lead length and reduce noise coupling.
10.1.3 Feedback
Route the feedback trace such that there is minimum interaction with any noise sources associated with the
switching components. Recommended practice is to place the inductor away from the feedback trace to prevent
EMI noise.
10.1.4 Traces and Ground Plane
All power (high-current) traces should be thick and as short as possible. The inductor and output capacitors
should be as close to each other as possible. This reduces the EMI radiated by the power traces due to high
switching currents.
In a two-sided PCB, TI recommends having ground planes on both sides of the PCB to help reduce noise and
ground-loop errors. Connect the ground connection for the input and output capacitors and IC ground to this
ground plane.
In a multilayer PCB, the ground plane separates the power plane (where high switching currents and
components are placed) from the signal plane (where the feedback trace and components are) for improved
performance.
Also, arrange the components such that the switching-current loops curl in the same direction. Place the highcurrent components such that during conduction the current path is in the same direction. Doing so prevents
magnetic field reversal caused by the traces between the two half cycles, helping to reduce radiated EMI.
30
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10.2 Layout Example
Output
Capacitor
Topside Supply Area
Input Capacitor
Ground
Plane
Catch Diode
NC
BOOT
NC
VIN
Output
Inductor
VIN
SYNC
LPM
PH
EN
VReg
RT
COMP
Rslew
VSENSE
RST
RST_TH
Cdly
OV_TH
GND
Compensation Network
Supervisor Network
SS
Resistor
Divider
Signal via to
Ground Plane
Topside Ground Area
Thermal Via
Signal Via
Figure 30. PCB Layout Example
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11 Device and Documentation Support
11.1 Trademarks
PowerPAD is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.2 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.3 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the mostcurrent data available for the designated devices. This data is subject to change without notice and without
revision of this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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6-Feb-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
MSL Peak Temp
(2)
(6)
(3)
Op Temp (°C)
Device Marking
(4/5)
TPS54362AQPWPRQ1
NRND
HTSSOP
PWP
20
2000
Green (RoHS
& no Sb/Br)
NIPDAU
Level-3-260C-168 HR
-40 to 125
54362AQ1
TPS54362BQPWPRQ1
ACTIVE
HTSSOP
PWP
20
2000
Green (RoHS
& no Sb/Br)
NIPDAU
Level-3-260C-168 HR
-40 to 125
54362BQ
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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