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TPS54678
SLVSBF3B – JUNE 2012 – REVISED MAY 2019
TPS54678 2.95-V to 6-V Input, 6-A Synchronous Step-Down SWIFT™
Converter With Hiccup Current Limit
1 Features
3 Description
•
TPS54678 device is a full-featured 6-V, 6-A,
synchronous step-down current mode converter with
two integrated MOSFETs.
1
•
•
•
•
•
•
•
•
•
•
Two 12-mΩ (typical) MOSFETs for high-efficiency
6-A continuous output current
200-kHz to 2-MHz Switching frequency
0.6-V ±1% Voltage reference over temperature
(–40°C to 150°C)
Synchronizes to external clock
Start-Up with prebiased voltage
Power good output
Adjustable slow start and sequencing
Cycle-by-cycle current limit and hiccup current
protection
Adjustable input voltage UVLO
Thermally enhanced 16-pin 3-mm × 3-mm WQFN
(RTE)
Create a custom design using the TPS54678 with
the WEBENCH® Power Designer
2 Applications
•
•
•
•
•
Low-voltage, high-density power systems
Point-of-load regulation for high-performance
DSPs, FPGAs, ASICs, and microprocessors
Broadband, networking, and optical
Communications infrastructure
Gaming, DTV, and set-top boxes
TPS54678 enables small designs by integrating the
MOSFETs, implementing current mode control to
reduce external component count, reducing inductor
size by enabling up to 2-MHz switching frequency,
and minimizing the IC footprint with a small 3-mm ×
3-mm thermally enhanced WQFN package.
TPS54678 provides accurate regulation for a variety
of loads with an accurate ±1% voltage reference
(VREF) over temperature.
Efficiency is maximized through the integrated 12-mΩ
MOSFETs. Using the enable (EN) pin, shutdown
supply current is reduced by disabling the device.
The output voltage start-up ramp is controlled by the
soft-start (SS) pin that can also be configured for
sequencing or tracking. Monotonic start-up is
achieved with prebiased voltage. Undervoltage
lockout can be increased by programming the
threshold with a resistor divider on the enable pin. An
open-drain power good signal indicates the output is
within 93% to 105% of its nominal voltage.
Cycle-by-cycle current limit, hiccup overcurrent
protection and thermal shutdown protect the device
during an overcurrent condition.
Device Information(1)
PART NUMBER
TPS54678
PACKAGE
WQFN (16)
BODY SIZE (NOM)
3.00 mm × 3.00 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic
VIN
Efficiency vs Output Current
95
TPS54678
VIN
94
BOOT
93
EN
PH
PWRGD
SS
RT/CLK
COMP
VSENSE
GND
AGND
Exposed thermal pad
Efficiency (%)
VOUT
92
91
90
89
88
VOUT = 1.8 V
Fsw = 500 KHz
DCR = 7.5 mΩ
87
VIN = 3.3 V
VIN = 5 V
86
85
1
2
3
4
Current (A)
5
6
G020
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
TPS54678
SLVSBF3B – JUNE 2012 – REVISED MAY 2019
www.ti.com
Table of Contents
1
2
3
4
5
6
7
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
6.1
6.2
6.3
6.4
6.5
6.6
4
4
4
4
5
7
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 11
7.1
7.2
7.3
7.4
Overview .................................................................
Functional Block Diagram .......................................
Feature Description.................................................
Device Functional Modes........................................
11
12
12
17
8
Application and Implementation ........................ 21
8.1 Application Information............................................ 21
8.2 Typical Application .................................................. 21
9 Power Supply Recommendations...................... 32
10 Layout................................................................... 32
10.1 Layout Guidelines ................................................. 32
10.2 Layout Example .................................................... 33
10.3 Power Dissipation Estimate .................................. 33
11 Device and Documentation Support ................. 35
11.1
11.2
11.3
11.4
11.5
11.6
11.7
Device Support ....................................................
Documentation Support ........................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
35
35
35
35
36
36
36
12 Mechanical, Packaging, and Orderable
Information ........................................................... 36
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision A (November 2015) to Revision B
•
Page
Editorial changes only — no technical content changed; added links for WEBENCH .......................................................... 1
Changes from Original (June 2012) to Revision A
Page
•
Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device Functional
Modes, Application and Implementation section, Power Supply Recommendations section, Layout section, Device
and Documentation Support section, and Mechanical, Packaging, and Orderable Information section .............................. 1
•
Changed Thermal Information table values............................................................................................................................ 4
2
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SLVSBF3B – JUNE 2012 – REVISED MAY 2019
5 Pin Configuration and Functions
VIN
EN
PWRGD
BOOT
RTE Package
16-Pin WQFN
Top View
16
15
14
13
VIN 1
12 PH
VIN 2
11 PH
Thermal
Pad
GND 3
10 PH
GND 4
6
7
8
COMP
RT/CLK
AGND
5
VSENSE
9
SS/TR
Pin Functions
PIN
I/O (1)
DESCRIPTION
NAME
NO.
AGND
5
G
Analog ground should be electrically connected to GND close to the device.
BOOT
13
I
A bootstrap capacitor is required between BOOT and PH. If the voltage on this capacitor is below the
minimum required by the BOOT UVLO, the output is forced to switch off until the capacitor is refreshed.
COMP
7
O
Error amplifier output, and input to the output switch current comparator. Connect frequency
compensation components to this pin.
EN
15
I
Enable pin, internal pullup current source. Pull below 1.2 V to disable. Float to enable. Can be used to
set the on and off threshold (adjust UVLO) with two additional resistors.
G
Power ground. This pin should be electrically connected directly to the thermal pad under the device.
O
The source of the internal high-side power MOSFET, and drain of the internal low-side (synchronous)
rectifier MOSFET.
GND
3
4
10
PH
11
12
PWRGD
14
O
An open-drain output asserts low if output voltage is low due to thermal shutdown, overvoltage,
undervoltage, or EN shut down.
RT/CLK
8
I/O
Resistor Timing or External Clock input pin
SS/TR
9
I/O
Slow-start and Tracking. An external capacitor connected to this pin sets the output voltage rise time.
This pin can also be used for tracking.
1
VIN
2
I
Input supply voltage, 2.95 V to 6 V
I
Inverting node of the transconductance (gm) error amplifier
–
GND pin should be connected to the exposed thermal pad for proper operation. This thermal pad should
be connected to any internal PCB ground plane using multiple vias for good thermal performance.
16
VSENSE
Thermal Pad
(1)
6
I = Input, O = Output, G = Ground
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SLVSBF3B – JUNE 2012 – REVISED MAY 2019
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6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
Input voltage
(1)
MIN
MAX
VIN, EN
–0.3
7
RT/CLK, PWRGD
–0.3
6
COMP, SS/TR, VSENSE
–0.3
3
BOOT-PH
Output voltage
Source current
Sink current
UNIT
V
7
PH
–0.7
7
PH (20 ns transent)
–2
10
PH (5 ns transient)
–4
12
V
EN, RT/CLK
100
COMP, SS
100
µA
µA
PWRGD
10
mA
Operating junction temperature, TJ
–40
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6.2 ESD Ratings
Electrostatic
discharge
V(ESD)
(1)
(2)
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins
(1)
VALUE
UNIT
±2000
V
±500
V
Charged device model (CDM), per JEDEC specification JESD22-C101, all pins (2)
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VIN
Input voltage
TJ
Operating junction temperature
MIN
MAX
3
6
UNIT
V
–40
150
°C
6.4 Thermal Information (1)
TPS54678
THERMAL METRIC (2)
RTE (WQFN)
UNIT
16 PINS
RθJA
Junction-to-ambient thermal resistance
43.4
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
44.2
°C/W
RθJB
Junction-to-board thermal resistance
14.6
°C/W
ψJT
Junction-to-top characterization parameter
0.6
°C/W
ψJB
Junction-to-board characterization parameter
14.5
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
4.1
°C/W
(1)
(2)
4
Unless otherwise specified, metrics listed in this table refer to JEDEC high-K board measurements
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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SLVSBF3B – JUNE 2012 – REVISED MAY 2019
6.5 Electrical Characteristics
TJ = –40°C to +150°C, VIN = 2.95 to 6 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE (VIN PIN)
Operating input voltage
Shutdown supply current
2.95
EN = 0 V, 25°C, 2.95 V ≤ VIN ≤ 6 V
Operating non-switching supply current VSENSE = 0.6 V, VIN = 5 V, 25°C, fSW = 500 kHz
6
V
1
3
µA
570
800
µA
ENABLE AND UVLO (EN PIN)
Enable threshold
Rising
1.3
V
Enable threshold
Falling
1.18
V
Input current
Enable threshold + 50 mV
–3.5
µA
Input current
Enable threshold – 50 mV
–0.70
µA
VOLTAGE REFERENCE
Voltage reference
2.95 V ≤ VIN ≤ 6 V, –40°C < TJ < 150°C
0.594
0.600
0.606
V
MOSFET
High-side switch resistance
BOOT-PH = 5 V
12
25
High-side switch resistance
BOOT-PH = 2.95 V
17
33
Low-side switch resistance
BOOT-PH = 5 V
12
25
Low-side switch resistance
BOOT-PH = 2.95 V
17
33
mΩ
mΩ
ERROR AMPLIFIER
Input current
7
Error amplifier transconductance (gm)
–2 µA < I(COMP) < 2 µA V(COMP) = 1 V
Error amplifier transconductance (gm)
during slow-start
–2 µA < I(COMP) < 2 µA V(COMP) = 1 V, V(VSENSE) = 0.4 V
Error amplifier source and sink
V(COMP) = 1 V 100-mV overdrive
COMP to Iswitch gm
nA
245
umhos
80
umhos
±20
µA
20
A/V
CURRENT LIMIT
Current limit threshold
Fs = 500 KHz
9.5
Cycles before entering hiccup during
overcurrent
10.5
11.5
512
Hiccup cycles
cycles
16384
Low-side sourcing current threshold
7
Low-side FET reverse current
protection
8.5
A
cycles
10.5
A
4
A
170
°C
15
°C
THERMAL SHUTDOWN
Thermal shutdown
Hysteresis
TIMING RESISTOR AND EXTERNAL CLOCK (RT/CLK PIN)
Switching frequency range using RT
mode
Switching frequency
200
Rt = 82.5 kΩ
400
Switching frequency range using CLK
mode
300
Minimum CLK pulse width
RT/CLK voltage
500
2000
kHz
600
kHz
2000
kHz
75
R(RT/CLK) = 82.5 kΩ
0.5
RT/CLK high threshold
1.6
RT/CLK low threshold
ns
0.4
V
2.2
V
0.6
V
RT/CLK falling edge to PH rising edge
delay
Measure at 500 kHz with RT resistor in series
55
ns
PLL lock in time
Measure at 500 kHz
40
µs
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Electrical Characteristics (continued)
TJ = –40°C to +150°C, VIN = 2.95 to 6 V (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Measured at 50% points on PH. IOUT = 3 A
85
110
ns
Measured at 50% points on PH. IOUT = 0 A
100
ns
PH (PH PIN)
Minimum ON-time
Minimum OFF-time
Prior to skipping off pulses,
BOOT-PH = 3 V, IOUT = 3 A
70
ns
Rise and fall dV/dT
BOOT-PH = 3 V; IO = 6 A
1.5
V/ns
BOOT (BOOT PIN)
Charging resistor
VIN = 6 V, BOOT-PH = 6 V
BOOT-PH UVLO
VIN = 3.3 V
7
Ω
2.2
V
SLOW START AND TRACKING (SS/TR PIN)
Charge current
V(SS/TR) < 0.15 V
47
V(SS/TR) > 0.15 V
2.2
µA
SS/TR to VSENSE matching
VIN = 3.3 V
60
mV
SS/TR to reference crossover
98% nominal
0.8
V
SS/TR discharge voltage (overload)
VSENSE = 0 V
4.5
mV
SS/TR discharge to current (overload)
VSENSE = 0 V; V(SS/TR) = 4 V
95
µA
SS/TR discharge current (UVLO, EN,
thermal fault)
VIN = 3 V; V(SS/TR) = 4 V
925
µA
91
% VREF
VSENSE rising (Good)
93
% VREF
VSENSE rising (Fault)
105
% VREF
VSENSE falling (Good)
103
% VREF
POWER GOOD (PWRGD PIN)
VSENSE falling (Fault)
VSENSE threshold
Output high leakage
VSENSE = VREF, V(PWRGD) = 5.5 V
ON-Resistance
VIN = 5 V
65
120
Ω
Output low
I(PWRGD) = 2.5 mA
0.2
0.3
V
Minimum VIN for valid output
V(PWRGD) < 0.5 V at 100 µA
1.2
1.5
V
6
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nA
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6.6 Typical Characteristics
1.3
590
VIN = 5 V
VIN = 3.3 V
VIN =5 V
VIN = 3.3 V
Operating Current (µA)
Shutdown Supply Current (µA)
1.4
1.2
1.1
1
0.9
0.8
570
560
0.7
0.6
−40
0
40
80
Temperature (°C)
120
550
−40
160
1.3
−1
EN Pin Current (µA)
−0.5
1.28
1.26
1.24
Rising, VIN = 3.3 V
Rising, VIN = 5 V
Falling, VIN = 3.3 V
Falling, VIN = 5 V
1.2
1.18
−40
0
40
80
Temperature (°C)
120
120
160
G002
−1.5
−2
Threshold−50 mV, VIN = 3.3 V
Threshold−50 mV, VIN = 5 V
Threshold+50 mV, VIN = 3.3 V
Threshold+50 mV, VIN = 5 V
−2.5
−3
−3.5
−4
−40
160
0
G003
Figure 3. EN Pin Voltage vs Junction Temperature
40
80
Temperature (°C)
120
160
G004
Figure 4. EN Pin Current vs Junction Temperature
600.2
22.25
VIN = 5 V
MOSFET Rds(on) (mΩ)
600.1
Voltage Reference (mV)
40
80
Temperature (°C)
Figure 2. VIN Operating Current vs Junction Temperature
1.32
1.22
0
G001
Figure 1. Shutdown Supply Current vs Junction
Temperature
EN Pin Voltage (V)
580
600
599.9
599.8
599.7
20.25
Lowside, VIN = 3.3 V
Highside, VIN = 3.3 V
Lowside, VIN = 5 V
Highside, VIN = 5 V
18.25
16.25
14.25
599.6
599.5
−40
0
40
80
Temperature (°C)
120
160
12.25
−40
G005
Figure 5. Voltage Reference vs Junction Temperature
0
40
80
Temperature (°C)
120
160
G006
Figure 6. MOSFET Rds(on) vs Junction Temperature
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Typical Characteristics (continued)
10.3
258
High Side FET Current Limit (A)
Transconductance (µA/V)
VIN = 3.3 V
VIN = 5 V
248
238
228
10.2
10.2
10.2
10.1
VIN = 3.3 V
218
−40
0
40
80
Temperature (°C)
120
10.1
−40
160
0
40
80
Temperature (°C)
G007
Figure 7. Transconductance vs Junction Temperature
160
G008
Figure 8. High-Side FET Current Limit vs Junction
Temperature
490
1800
VIN = 5 V
RT = 85 kΩ
Switching Frequency (kHz)
1600
Switching Frequency (KHz)
120
1400
1200
1000
800
600
400
488
486
484
482
200
0
0
20
40
60
80
100 120
140 160 180
RT Resistance (KΩ)
480
−40
200
160
G010
VIN = 3.3 V
VIN = 5 V
−2.22
152.6
152.4
152.2
0
−2.23
−2.24
−2.25
VSS
TR
> 0.15 V
−2.26
−2.27
VIN = 3.3 V
VIN = 5 V
152
−40
120
−2.21
152.8
SS Charge Current (µA)
VSS Voltage Threshold VSSTHR (mV)
40
80
Temperature (°C)
Figure 10. Switching Frequency vs Junction Temperature
Figure 9. Switching Frequency vs RT Pin Resistance
40
80
Temperature (°C)
120
160
−2.28
−40
G011
Figure 11. VSS Voltage Threshold vs Junction Temperature
8
0
G009
0
40
80
Temperature (°C)
120
160
G012
Figure 12. SS Charge Current vs Junction Temperature
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Typical Characteristics (continued)
−44.5
80
VIN = 3.3 V
VIN = 5 V
75
PWRGD Rdson (Ω)
SS Charge Current (µA)
−45
−45.5
−46
−46.5
VSS
TR
= < 0.15 V
−47
70
65
60
−47.5
VIN = 5 V
−48
−40
0
40
80
Temperature (°C)
120
55
−40
160
0
40
80
Temperature (°C)
G013
Figure 13. SS Charge Current vs Junction Temperature
120
160
G014
Figure 14. PWRGD Rds(on) vs Junction Temperature
107
4
PWRGD Leakage Current (nA)
PWRGD Threshold (% of vref)
VIN = 5 V
105
103
101
Fault Rising
Good Rising
Fault Falling
Good Falling
99
97
95
93
91
−40
0
40
80
Temperature (°C)
120
2
1
0
−1
−2
−40
160
96
96
94
94
92
Efficiency (%)
92
90
88
86
VOUT = 3.3 V
VOUT = 1.8 V
VOUT = 1.2 V
VOUT = 1 V
82
80
78
1
2
VIN = 5 V
Fsw = 500 KHz
DCR = 7.5 mΩ
TA = 25°C
3
4
Current (A)
40
80
Temperature (°C)
5
160
G016
90
88
86
84
82
78
VIN = 3.3 V
Fsw = 500 KHz
DCR = 7.5 mΩ
TA = 25°C
VOUT = 1.8 V
VOUT = 1.2 V
VOUT = 1 V
80
6
120
Figure 16. PWRGD Leakage Current vs Junction
Temperature
98
84
0
G015
Figure 15. PWRGD Threshold vs Junction Temperature
Efficiency (%)
3
1
G017
Figure 17. Efficiency vs Load Current
2
3
4
Current (A)
5
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G018
Figure 18. Efficiency vs Load Current
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9
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98
96
96
94
94
92
92
Efficiency (%)
Efficiency (%)
Typical Characteristics (continued)
90
88
86
84
VOUT = 3.3 V
VOUT = 1.8 V
VOUT = 1.2 V
VOUT = 1 V
82
80
78
1
2
VIN = 5 V
Fsw = 1 MHz
DCR = 7.5 mΩ
TA = 25°C
3
4
Current (A)
5
90
88
86
84
82
VOUT = 1.8 V
VOUT = 1.2 V
VOUT = 1 V
80
6
78
1
G019
Figure 19. Efficiency vs Load Current
10
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2
VIN = 3.3 V
Fsw = 1 MHz
DCR = 7.5 mΩ
TA = 25°C
3
4
Current (A)
5
6
G020
Figure 20. Efficiency vs Load Current
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7 Detailed Description
7.1 Overview
The TPS54678 is a 6-V, 6-A, synchronous step-down (buck) converter with two integrated N-channel MOSFETs.
To improve the performance during line and load transients the device implements a constant frequency, peak
current mode control which reduces output capacitance and simplifies external frequency compensation design.
The wide switching frequency range of 200 kHz to 2000 kHz allows for efficiency and size optimization when
selecting the output filter components. The switching frequency is adjusted using a resistor to ground on the
RT/CLK pin. The device has an internal phase lock loop (PLL) on the RT/CLK pin that is used to synchronize the
power switch turn on to a falling edge of an external system clock.
The TPS54678 has a typical default start-up voltage of 2.4 V. The EN pin has an internal pullup current source
that can be used to adjust the input voltage undervoltage lockout (UVLO) with two external resistors. In addition,
the pullup current provides a default condition when the EN pin is floating for the device to operate. The total
operating current for the TPS54678 is typically 570 µA when not switching and under no load. When the device
is disabled, the supply current is less than 3 µA.
The integrated 12-mΩ MOSFETs allow for high-efficiency power supply designs with continuous output currents
up to 6 amperes. The TPS54678 reduces the external component count by integrating the boot recharge diode.
The bias voltage for the integrated high-side MOSFET is supplied by a capacitor between the BOOT and PH
pins. The boot capacitor voltage is monitored by an UVLO circuit and turns off the high-side MOSFET when the
voltage falls below a preset threshold. This BOOT circuit allows the TPS54678 to operate approaching 100%.
The output voltage can be stepped down to as low as the 0.60-V reference.
TPS54678 features monotonic start-up under prebias conditions. The low-side FET turns on for a short time
period every cycle before the output voltage reaches the prebiased voltage. This ensures the boot cap has
enough charge to turn on the top FET when the output voltage reaches the prebiased voltage.
The TPS54678 has a power good comparator (PWRGD) with 2% hysteresis.
The TPS54678 minimizes excessive output overvoltage transients by taking advantage of the overvoltage power
good comparator. When the regulated output voltage is greater than 105% of the nominal voltage, the
overvoltage comparator is activated, and the high-side MOSFET is turned off and masked from turning on until
the output voltage is lower than 103%.
The SS/TR (slow-start or tracking) pin is used to minimize inrush currents or provide power supply sequencing
during power up. A small value capacitor should be coupled to the pin for slow-start. The SS/TR pin is
discharged before the output power up to ensure a repeatable restart after an overtemperature fault, UVLO fault
or disabled condition. To optimize the output startup waveform, two levels of SS current are implemented.
To reduce the power dissipation of TPS54678 during overcurrent event, the hiccup protection is implemented
beyond the cycle-by-cycle protection.
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7.2 Functional Block Diagram
PWRGD EN
VIN
i1
93%
iHYS
Logic
Thermal
Shutdown
Enable
Comparator
105%
Shutdown
Logic
Voltage
Reference
VSENSE
UVLO
Enable
Threshold
Boot
Charge
Boot
UVLO
+
+
BOOT
SS
Shutdown
Minimum
COMP Clamp
Logic
PWM
Comparator
COMP
Logic and
PWM Latch
PH
Slope
Compensation
Frequency
Shift
Overload
Recovery
Maximum
Clamp
GND
OSC with
PLL
AGND
PowerPad
RT/CLK
7.3 Feature Description
7.3.1 Fixed Frequency PWM Control
The TPS54678 uses a settable fixed frequency, peak current mode control. The output voltage is compared
through external resistors on the VSENSE pin to an internal voltage reference by an error amplifier which drives
the COMP pin. An internal oscillator initiates the turn on of the high-side power switch. The error amplifier output
is compared to the high-side power switch current. When the power switch current reaches the COMP voltage
level the high-side power switch is turned off and the low-side power switch is turned on. The COMP pin voltage
will increase and decrease as the output current increases and decreases. The device implements a current limit
by clamping the COMP pin voltage to a maximum level and implements a sleep mode with a minimum clamp.
7.3.2 Slope Compensation and Output Current
The TPS54678 adds a compensating ramp to the switch current signal. This slope compensation prevents subharmonic oscillations. The available peak inductor current maintains constant over the full duty cycle range.
7.3.3 Bootstrap Voltage (Boot) and Low Dropout Operation
The TPS54678 has an integrated boot regulator and requires a small ceramic capacitor between the BOOT and
PH pin to provide the gate drive voltage for the high-side MOSFET. The value of the ceramic capacitor should be
0.1 µF. A ceramic capacitor with an X7R or X5R grade dielectric is recommended because of the stable
characteristics over temperature and voltage.
To improve dropout, the TPS54678 is designed to operate at 100% duty cycle as long as the BOOT to PH pin
voltage is greater than 2.2 V. The high-side MOSFET is turned off using an UVLO circuit, allowing for the lowside MOSFET to conduct, when the voltage from BOOT to PH drops below 2.2 V. Because the supply current
sourced from the BOOT pin is low, the high-side MOSFET can remain on for more switching cycles than are
required to refresh the capacitor, thus the effective duty of the switching regulator is high.
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Feature Description (continued)
7.3.4 Error Amplifier
The TPS54678 has a transconductance amplifier for the error amplifier. The error amplifier compares the
VSENSE voltage to the lower of the SS/TR pin voltage or the internal 0.6-V voltage reference. The
transconductance of the error amplifier is 245 µA/V during normal operation. During the slow-start operation, the
transconductance is a fraction of the normal operating gm. The frequency compensation components are added
to the COMP pin to ground.
7.3.5 Voltage Reference
The voltage reference system produces a precise ±1% voltage reference over temperature by scaling the output
of a temperature stable bandgap circuit. During production, the bandgap and scaling circuits are trimmed to
produce 0.6 V at the amplifier output.
7.3.6 Adjusting the Output Voltage
The output voltage is set with a resistor divider from the output node to the VSENSE pin. TI recommends using
1% tolerance or better divider resistors. Start with a 20-kΩ resistor for R1 and use Equation 1 to calculate R2. To
improve efficiency at light loads, consider using larger value resistors. If the values are too high, the regulator will
be more susceptible to noise and voltage errors from the VSENSE input current will be noticeable.
æ
R2 = R1 × ç
0.6 V ö÷
- 0.6 V ÷ø
çV
è O
(1)
TPS54678
VOUT
R1
VSENSE
+
0.6 V
R2
Figure 21. Voltage Divider Circuit
7.3.7 Enable and Adjusting Undervoltage Lockout
The EN pin provides electrical on and off control of the device. Once the EN pin voltage exceeds the threshold
voltage, the device starts operation. If the EN pin voltage is pulled below the threshold voltage, the regulator
stops switching and enters low Iq state. If an application requires controlling the EN pin, use open drain or open
collector output logic to interface with the pin.
For input undervoltage lockout (UVLO), use the EN pin as shown in Figure 22 to set up the UVLO by using the
two external resistors. Once the EN pin voltage exceeds 1.3 V, an additional 2.8 µA of hysteresis is added. This
additional current facilitates input voltage hysteresis. Use Equation 2 to set the external hysteresis for the input
voltage. Use Equation 3 to set the input startup voltage. TI recommends that the minimum input shutdown
voltage be set at 2.45 V or higher to ensure proper operation before shutdown.
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Feature Description (continued)
TPS54678
VIN
Ip
0.7 µA
Ih
2.8 µA
R1
EN
+
R2
Figure 22. Set Up Input Undervoltage Lockout.
VSTART (
R1 =
VEN_FALLING
VEN_RISING
) - VSTOP
V
Ip (1 - EN_FALLING ) + I
h
VEN_RISING
(2)
R1 × VEN_FALLING
R2 =
VSTOP - VEN_FALLING + R1 × (Ip + I )
h
where
•
•
•
•
•
R1 and R2 are in Ω
Ih = 2.8 µA
Ip = 0.7 µA
VEN_RISING = 1.3 V
VEN_FALLING = 1.18 V
(3)
7.3.8 Soft-Start Pin
TPS54678 effectively uses the lower voltage of the internal voltage reference or the SS/TR pin voltage as the
reference voltage of the power supply and regulates the output accordingly. A capacitor on the SS/TR pin to
ground will implement a slow-start time. The TPS54678 has an internal pullup current source of 47 µA when
V(SS/TR) is less than 0.15 V and 2.2 µA when V(SS/TR) is higher than 0.15 V. The ISS charges the external slowstart capacitor. The equation for the slow-start time is shown in Equation 4 considering the fact the first 47 µA
charges the SS to 0.15 V. The 2.2 µA then charges the SS from 0.15 V to about 0.8 V for the handoff of the SS
voltage to reference voltage.
Css( nF ) = 3 ´ Tss( mS )
(4)
If during normal operation, the VIN UVLO is exceeded, EN pin pulled below 1.2 V, or a thermal shutdown event
occurs, the TPS54678 will stop switching and the SS/TR must be discharged to about 60 mV before reinitiating a
powering-up sequence.
The VSENSE voltage will follow the SS/TR pin voltage up to 90% of the internal voltage reference. When the
SS/TR voltage is greater than 90% of the internal voltage, the effective system reference voltage will transit from
the SS/TR voltage to the internal voltage reference.
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Feature Description (continued)
7.3.9 Sequencing
Many of the common power supply sequencing methods can be implemented using the SS/TR, EN and PWRGD
pins. The sequential method can be implemented using an open-drain or collector output of a power on reset pin
of another device. The sequential method is shown in Figure 23. The power good is coupled to the EN pin on the
TPS54678, which will enable the second power supply once the primary supply reaches regulation.
TPS54678
TPS54678
PWRGD
EN
EN
SS
SS/TR
CSS
PWRGD
CSS
Figure 23. Sequential Start-Up Sequence
7.3.10 Constant Switching Frequency and Timing Resistor (RT/CLK Pin)
The switching frequency of the TPS54678 is adjustable over a wide range from approximately 200 kHz to
2000 kHz by placing a maximum of 210 kΩ and minimum of 18 kΩ, respectively, on the RT/CLK pin. The
RT/CLK is typically 0.5 V. To determine the timing resistance for a given switching frequency, use the curve in
Figure 24. To reduce the solution size one would typically set the switching frequency as high as possible, but
tradeoffs of the supply efficiency, maximum input voltage and minimum controllable on time should be
considered.
The minimum controllable on time is typically 85 ns at 3-A current load and 100 ns at no load, and will limit the
maximum operating input voltage or output voltage.
56183
R (kΩ) =
1.052
T
éF
(KHz)ù
ë SW
û
(5)
1800
Switching Frequency (KHz)
1600
1400
1200
1000
800
600
400
200
0
0
20
40
60
80
100 120
140 160 180
RT Resistance (KΩ)
200
G009
Figure 24. Switching Frequency vs RT Set Resistor
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Feature Description (continued)
7.3.11 Overcurrent Protection
The TPS54678 implements current mode control which uses the COMP pin voltage to turn off the high-side
MOSFET and turn on the low-side MOSFET on a cycle-by-cycle basis. Each cycle the switch current and the
COMP pin voltage are compared, when the peak switch current intersects the COMP voltage the high-side
switch is turned off.
7.3.11.1 High-Side Overcurrent Protection
During overcurrent conditions that pull the output voltage low, the error amplifier will respond by driving the
COMP pin high, increasing the switch current. The error amplifier output is clamped internally. This clamp
functions as a high-side switch current limit. When the high-side switch current limit occurs consecutively for 512
CLK cycles, the converter enters hiccup mode in which no switching action happens for about 16000 cycles. This
helps to reduce the power consumption during an overcurrent event.
7.3.11.2 Low-Side Overcurrent Protection
The conduction current of the low-side MOSFET is also monitored by TPS54678. During normal operation, the
low-side sources current into the load. When the sourcing current reaches the internally set low-side sourcing
(forward) current limit, the high-side is not turned on and skipped during the next clock cycle. Under this
condition, the low-side is kept on until the sourcing current becomes less than the internally set current limit and
then the high-side is turned on at the beginning of the following clock cycle. This ensures protection under an
output short condition; thereby, preventing current run-away.
The low-side can also sink current from the load. If the low-side sinking (reverse) current limit is exceeded, the
low-side is turned off immediately for the rest of the clock cycle. Under this condition, both the high-side and lowside are off until the start of the next cycle.
7.3.12 Safe Start-Up into Prebiased Outputs
The TPS54678 allows monotonic start-up into prebiased output. The low-side FET turns on for a short time
period every cycle before the output voltage reaches the prebiased voltage. This ensures the boot cap has
enough charge to turn on the top FET when the output voltage reaches the prebiased voltage.
The TPS54678 also implements low-side current protection by detecting the voltage over the low-side MOSFET.
When the converter sinks current through the low side FET and if the current exceeds 4 A, the control circuit
turns the low-side FET off. Due to the implemented prebias function, the low-side FET reverse current protection
should not be reached, but it provides another layer of protection.
7.3.13 Synchronize Using the RT/CLK Pin
The RT/CLK pin is used to synchronize the converter to an external system clock. See Figure 25. To implement
the synchronization feature in a system connect a square wave to the RT/CLK pin with on time at least 75 ns.
The square wave amplitude at this pin must transition lower than 0.6 V and higher than 1.6 V. The
synchronization frequency range is 300 kHz to 2000 kHz. The rising edge of the PH will be synchronized to the
falling edge of RT/CLK pin.
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Feature Description (continued)
TPS54678
RT/CLK
PLL
RRT
Figure 25. Synchronizing to a System Clock
7.3.14 Power Good (PWRGD Pin)
The PWRGD pin is an open-drain output and pulls the PWRGD pin low when the VSENSE voltage is less than
91% or greater than 105% of the nominal internal reference voltage.
There is a 2% hysteresis, so once the VSENSE pin is within 93% to 103% of the internal voltage reference the
PWRGD pin is de-asserted and the pin floats. TI recommends to use a pullup resistor between the values of
1 kΩ and 100 kΩ to a voltage source that is 5.5 V or less. The PWRGD is in a valid state once the VIN input
voltage is greater than 1.2 V.
7.3.15 Overvoltage Transient Protection
The TPS54678 incorporates an overvoltage transient protection (OVTP) circuit to minimize voltage overshoot
when recovering from output fault conditions or strong unload transients. The OVTP feature minimizes the output
overshoot by implementing a circuit to compare the VSENSE pin voltage to OVTP threshold which is 105% of
the internal voltage reference. If the VSENSE pin voltage is greater than the OVTP threshold, the high-side
MOSFET is disabled preventing current from flowing to the output and minimizing output overshoot. When the
VSENSE voltage drops lower than the OVTP threshold the high-side MOSFET is allowed to turn on the next
clock cycle.
7.3.16 Thermal Shutdown
The device implements an internal thermal shutdown to protect itself if the junction temperature exceeds 170°C.
The thermal shutdown forces the device to stop switching when the junction temperature exceeds thermal trip
threshold. When the die temperature decreases below 155°C, the device reinitiates the power-up sequence by
discharging the SS/TR pin to about 60 mV. The thermal shutdown hysteresis is 15°C.
7.4 Device Functional Modes
7.4.1 Small Signal Model for Loop Response
The Figure 26 shows an equivalent model for the TPS54678 control loop that can be modeled in a circuit
simulation program to check frequency response and dynamic load response. The error amplifier is a
transconductance amplifier with a gm of 245 µA/V. The error amplifier can be modeled using an ideal voltage
controlled current source. The resistor RO and capacitor CO model the open loop gain and frequency response of
the amplifier.
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Device Functional Modes (continued)
TPS54678
PH
VOUT
Power Stage
20 A/V
a
RESR
b
R1
VSENSE
COMP
RLOAD
COUT
c
+
C2
COUT(ea)
R3
ROUT(ea)
0.6 V
gM
245 µA/V
R2
C1
Figure 26. Small Signal Model for Loop Response
7.4.2 Simple Small Signal Model for Peak Current Mode Control
Figure 26 is a simple small signal model that can be used to understand how to design the frequency
compensation. The TPS54678 power stage can be approximated to a voltage controlled current source (duty
cycle modulator) supplying current to the output capacitor and load resistor. The control to output transfer
function is shown in Equation 6 and consists of a DC gain, one dominant pole and one ESR zero. The quotient of
the change in switch current and the change in COMP pin voltage (node c in Figure 26) is the power stage
transconductance. The gm for the TPS54678 is 20 A/V. The low-frequency gain of the power stage frequency
response is the product of the transconductance and the load resistance as shown in Equation 7. As the load
current increases and decreases, the low-frequency gain decreases and increases, respectively. This variation
with load may seem problematic at first glance, but fortunately the dominant pole moves with load current (see
Equation 8). The combined effect is highlighted by the dashed line in Figure 28. As the load current decreases,
the gain increases and the pole frequency lowers, keeping the 0-dB crossover frequency the same for the
varying load conditions, which makes it easier to design the frequency compensation.
VC
RESR
RLOAD
gm(ps)
COUT
Copyright © 2016, Texas Instruments Incorporated
Figure 27. Small Signal Model For Peak Current Mode Control
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Device Functional Modes (continued)
Gain
Adc
fZ
fP
Frequency
Figure 28. Frequency Response Model for Peak Current Mode Control
VO
= Adc x
VC
æ
ç1
è
æ
ç1
è
ö
s
2p ´ fz ÷ø
ö
s
+
2p ´ fp ÷ø
+
(6)
Adc = gmps ´ R L
fp =
fz =
(7)
COUT
1
´ R L ´ 2p
(8)
COUT
1
´ R ESR ´ 2p
(9)
7.4.3 Small Signal Model for Frequency Compensation
The TPS54678 uses a transconductance amplifier for the error amplifier and readily supports two of the
commonly used frequency compensation circuits. The compensation circuits are shown in Figure 29. The Type 2
circuits are normally implemented in high-bandwidth power supply designs using low ESR output capacitors. In
Type 2A, one additional high-frequency pole is added to attenuate high-frequency noise.
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Device Functional Modes (continued)
VOUT
TPS54678
R1
VSENSE
COMP
gM(ea)
R2
+
VREF
R3
ROUT(ea)
COUT(ea)
5 pF
C1
Type IIA
R3
C2
C1
Type IIB
Figure 29. Types of Frequency Compensation
The design guidelines for TPS54678 loop compensation are as follows:
1. Set up crossover frequency fc.
2. R3 can be determined by Equation 10:
2p ´ fc ´ VO ´ COUT
R3 =
gmea ´ VREF ´ gm ps
where
•
•
gmea is the GM amplifier gain,
gmPS is the power stage gain (20 A/V).
(10)
1
´ R L ´ 2p C1 can be determined by
fp =
COUT
3. Place a compensation zero at the dominant pole
Equation 11:
R L ´ COUT
C1 =
R3
4. C2 is optional. It can be used to cancel the zero from ESR of the Co in Equation 12:
R ESR ´ COUT
C2 =
R3
20
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
This example details the design of a high-frequency switching regulator design using ceramic output capacitors.
This design is available as the TPS54678EVM-155 (PWR155) evaluation module (EVM).
8.2 Typical Application
This section details a high-frequency, 1.2-V output power supply design application with adjusted UVLO.
L1
1.2 H
U1
TPS54678RTE
VIN = 3-6V
16
1
C16
220 PF
C1
47 PF
C2
47 PF
C3
47 PF
C4
0.1 PF
2
15
EN
6
VSNS
VIN
7
8
R3
26.7k
R1
14.7k
9
R4
82.5k
C5
Open
R2
12.7k
VIN
VIN
EN
VSNS
COMP
RT/CLK
SS/TR
VOUT = 1.2V, 6A
VOUT
C9
47 PF
C8
0.1 PF
C10
47 PF
C13
47 PF
C12
47 PF
C11
47 PF
R8
51.1
VIN
R7
100k
R9
20.0k
PWRGD
C15
150 pF
PWPD
EN
C14
220 pF
PH 10
11
PH
12
PH
13
BOOT
14
PWRGD
3
GND
4
GND
5
AGND
VIN
C7
0.01 PF
VSNS
17
R10
20.0k
C6
2200 pF
Figure 30. Typical Application Schematic, TPS54678
8.2.1 Design Requirements
Table 1 lists the design parameters of the TPS54678.
Table 1. Design Parameters
PARAMETER
VIN
Input voltage
VOUT
Output voltage
ΔVOUT
Transient response
IOUT(max)
Maximum output current
VOUT(ripple)
Output voltage ripple
fSW
Switching frequency
NOTES AND CONDITIONS
Operating
MIN
TYP
MAX
3
5
6
UNIT
V
1.2
3-A load step
V
5%
6
A
30
mVP-P
500
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8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS54678 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2
Step One: Select the Switching Frequency
The first step is to decide on a switching frequency for the regulator. Typically, it is desirable to choose the
highest switching frequency possible since this produces the smallest component solution size. The high
switching frequency allows for lower value inductors and smaller output capacitors compared to a power supply
that switches at a lower frequency. However, the higher switching frequency causes extra switching losses,
which degrade the performance of the converter. This SWIFT™ converter is capable of running from 200 kHz to
2 MHz. Unless a small solution size is the top priority, a moderate switching frequency of 500 kHz is selected to
achieve both a small solution size and high-efficiency operation. Using Equation 13, RT is calculated to be
81.34 kΩ. A standard 1% 82.5-kΩ value was chosen for the design.
56183
56183
R (kΩ) =
=
= 81.34 kΩ
T
(F )1.052
(500)1.052
SW
where
•
•
RT is in kΩ
FSW is in kHz
(13)
8.2.2.3 Step Two: Select the Output Inductor
The inductor selected works for the entire TPS54678 input voltage range. To calculate the value of the output
inductor, use Equation 14. KIND is a coefficient that represents the amount of inductor ripple current relative to the
maximum output current. The inductor ripple current is filtered by the output capacitor. Therefore, choosing highinductor ripple currents impacts the selection of the output capacitor because the output capacitor must have a
ripple current rating equal to or greater than the inductor ripple current. In general, the inductor ripple value is at
the discretion of the designer, however KIND is usually chosen between 0.1 to 0.3 for the majority of applications.
For this design example, a value of KIND = 0.3 was used at 6 VIN and 6 AOUT, and the inductor value is calculated
to be 1.06 μH. For this design, the nearest standard value of 1.2 μH was chosen. For the output filter inductor, it
is important that the RMS current and saturation current ratings not be exceeded. The RMS and peak inductor
current can be found from Equation 16 and Equation 17.
For this design, the RMS inductor current is 6.02 A and the peak inductor current is 6.8 A. The chosen inductor is
a Coilcraft XAL5030-122ME. It has a saturation current rating 0f 11.8 A (20% inductance loss) and a RMS
current rating of 8.7 A (20°C temperature rise). The series resistance is 6.78 mΩ typical.
The current flowing through the inductor is the inductor ripple current plus the output current. During power up,
faults or transient load conditions, the inductor current can increase above the calculated peak inductor current
level calculated above. In transient conditions, the inductor current can increase up to the switch current limit of
the device. For this reason, the most conservative approach is to specify an inductor with a saturation current
rating equal to or greater than the switch current limit rather than the peak inductor current.
22
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VIN _ MAX - VOUT
VOUT
´
L1 =
Io ´ KIND
VIN _ MAX ´ FSW
IRIPPLE
(14)
VIN _ MAX - VOUT
VOUT
=
´
L1
VIN _ MAX ´ FSW
IIND _ RMS
IIND _ peak
(
æ
1 ç VOUT ´ VIN _ MAX - VOUT
= Io +
´
12 ç
VIN _ MAX ´ L1´ FSW
è
I
= Io + RIPPLE
2
2
(15)
)ö÷
2
÷
ø
(16)
(17)
8.2.2.4 Step Three: Choose the Output Capacitor
There are three primary considerations for selecting the value of the output capacitor. Along with the inductor, the
output capacitor determines the output voltage ripple, and also how the regulator responds to a large change in
load current. The output capacitance needs to be selected based on the more stringent of these two criteria.
The desired response to a large change in the load current is the first criteria. The output capacitor needs to
supply the load with current when the regulator can not due to limited control speed. The regulator is temporarily
not able to supply sufficient change in output current if there is a large, fast increase or decrease in the current
needs of the load such as transitioning from no load to full load. The regulator usually needs two or more clock
cycles for the control loop to see the change in load current and output voltage and adjust the duty cycle to react
to the change. The output capacitor must be sized to supply the extra current to the load until the control loop
responds to the load change, or conversely, absorb the excess current from the inductor. Because the output
voltage is less than half the input voltage, the worst-case deviation in output voltage occurs when the load has an
extremely rapid reduction in current, or a load dump. The desired specification is a 50% or 3-A load step, and a
resulting voltage deviation of no more than 5%, or 60mV. When a load dump occurs, the excess stored current in
the inductor will tend to charge the output capacitors, and the best the converter can achieve to limit the increase
in output voltage is to fold back the duty cycle to zero. Under these circumstances, the amount of rise in output
voltage is defined by the energy from the choke being fully absorbed by the capacitor bank. Equation 18 through
Equation 20 can be used to calculate the required capacitor bank value.
For this example, the transient load response is specified as a 5% change in Vout for a 50% load step from 3 A
to 0 A. So, ΔIOUT = 3 A and ΔVOUT = 0.05 × 1.2 = 0.06 V. Using these numbers gives a minimum capacitance of
73.2 μF. This calculation does not take the ESR of the output capacitor into account in the output voltage
change, and it does not account for latency in control loop speed. For ceramic capacitors, the ESR is usually
small enough to ignore in this calculation.
EnergyIND = 0.5 ´ L ´ I2 = 0.5 ´ 1.2m ´ 32 = 5.4m Joule
(18)
EnergyCAP Initial = 0.5 ´ C ´ V 2 = 0.5 ´ C ´ 1.22
2
EnergyCAP Final = 0.5 ´ C ´ 1.22 + EnergyIND = 0.5 ´ C ´ (1.2 + 0.06 )
(19)
Solving for C:
5.4mJ
= 73.17mF
(0.7938 - 0.72 )
CBank =
(20)
This 73.17 µF defines the minimum capacitance required to meet the transient spec; however, because the
control loop speed is finite, more capacitance than this is required to meet desired performance.
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Equation 21 calculates the minimum output capacitance needed to meet the output voltage ripple specification.
In this case, the maximum output voltage ripple is 60 mV. Under this requirement, Equation 21 yields 13.33 µF.
1
1
´
= 13.33mF
CBank =
(8 ´ FSW ) VRIPPLE
IRIPPLE
where
•
•
•
FSW is the switching frequency,
VRIPPLE is the maximum allowable output voltage ripple,
and Iripple is the inductor ripple current.
(21)
Equation 22 calculates the maximum ESR for the capacitor bank to meet the output voltage ripple specification.
Equation 22 indicates the ESR should be less than 37.5 mΩ. In this case, the ESR of the ceramic capacitor bank
is less than 37.5 mΩ.
V
RESR < RIPPLE
IRIPPLE
(22)
Additional capacitance de-ratings for aging, temperature and DC bias should be factored in which increases the
minimum value calculated in Equation 20. For this example, five 47-μF 10-V X5R ceramic capacitors with 3 mΩ
of ESR are used. The estimated capacitance after derating is 5 × 47 μF × 0.9 = 211.5 μF.
8.2.2.5 Step Four: Select the Input Capacitor
The TPS54678 requires a high-quality ceramic, type X5R or X7R, input-decoupling capacitor of at least 10 μF of
effective capacitance and in some applications a bulk capacitance. The effective capacitance includes any DC
bias effects. The voltage rating of the input capacitor must be greater than the maximum input voltage. The
capacitor must also have a ripple current rating greater than the maximum input current ripple of the TPS54678.
The input ripple current can be calculated using Equation 23.
IRMS = IOUT ´
(
VIN _ MIN - VOUT
VOUT
´
VIN _ MIN
VIN _ MIN
)
(23)
The value of a ceramic capacitor varies significantly over temperature and the amount of DC bias applied to the
capacitor. The capacitance variations due to temperature can be minimized by selecting a dielectric material that
is stable over temperature. X5R and X7R ceramic dielectrics are usually selected for power regulator capacitors
because they have a high capacitance to volume ratio and are fairly stable over temperature. The output
capacitor must also be selected with the DC bias taken into account. The capacitance value of a capacitor
decreases as the DC bias across a capacitor increases.
For this example design, a ceramic capacitor with at least a 10-V voltage rating is required to support the
maximum input voltage. For this example, three 47-μF and one 0.10-μF 10-V capacitors in parallel have been
selected. In addition to these low ESR capacitors, an input bulk cap of 220-µF electrolytic is included so as to
provide low source impedance at low frequencies for instances where the input voltage source is connected with
a lossy feed.
The input capacitance value determines the input ripple voltage of the regulator. The input voltage ripple can be
calculated using Equation 24. Using the design example values, IOUT_MAX = 6 A, CIN = 141 μF (neglecting the
electrolytic due to high ESR), FSW = 500 kHz, yields an input voltage ripple of 21.3 mV and an rms input ripple
current of 2.94 A.
IOUT _ MAX ´ 0.25
DVIN =
CIN ´ FSW
(24)
24
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8.2.2.6 Step Five: Choose the Soft-Start Capacitor
The slow-start capacitor determines the minimum amount of time it takes for the output voltage to reach the
nominal programmed value during power up. This is useful if a load requires a controlled voltage slew rate. This
is also used if the output capacitance is very large and would require large amounts of current to quickly charge
the capacitor to the output voltage level. The large currents necessary to charge the capacitor may cause the
TPS54678 to trip OCP, or excessive current draw from the input power supply may cause the input voltage rail to
sag. Limiting the output voltage slew rate mitigates both of these issues.
The slow-start capacitor value can be calculated using Equation 25. For the example circuit, the slow-start time is
not critical because the output capacitor value is 5 × 47 μF which does not require much current to charge to
1.2 V. The example circuit has the slow-start time set to an arbitrary value of 3.33 ms, which requires a 10-nF
capacitor.
CSS = 3 ´ TSS
(25)
8.2.2.7 Step Six: Select the Bootstrap Capacitor
A 0.1-μF ceramic capacitor must be connected between the BOOT and PH pins for proper operation. TI
recommends to use a ceramic capacitor with X5R or better grade dielectric. The capacitor should have 10 V or
higher voltage rating.
8.2.2.8 Step Eight: Select Output Voltage and Feedback Resistors
For the example design, 20 kΩ was selected for R10. Using Equation 26, R9 is calculated also as 20 kΩ.
æV
ö
R9 = R10 ´ ç OUT - 1÷
V
è REF
ø
(26)
8.2.2.8.1 Output Voltage Limitations
Due to the internal design of the TPS54678, there is a minimum output voltage limit for any given input voltage.
The output voltage can never be lower than the internal voltage reference of 0.6 V. Above 0.6 V, the output
voltage may be limited by the minimum controllable ON-time. The minimum output voltage in this case is given
by Equation 27.
(
)
(
VOUT _ MIN = t ON _ MIN ´ FSW _ MAX VIN _ MAX - IOUT _ MIN ´ RDS(ON)MIN - IOUT _ MIN ´ RL + RDS(ON)MIN
)
where
•
•
•
•
•
•
•
VOUT_MIN = minimum achievable output voltage
tON_MIN = minimum controllable ON-time (100 ns typical, 120 ns no load)
FSW_MAX = maximum switching frequency including tolerance
VIN_MAX = maximum input voltage
IOUT_MIN = minimum load current
RDS(ON)_MIN = minimum high-side MOSFET ON-resistance (see Electrical Characteristics)
RL = series resistance of output inductor
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There is also a maximum achievable output voltage which is limited by the minimum off time. The maximum
output voltage is given by Equation 28.
t OFF _ MAX ö
æ
VOUT _ MAX = VIN ´ çç 1 ÷ - IOUT _ MAX ´ RDS(ON)MAX + RL - 0.7 - IOUT _ MAX ´ RDS(ON)MAX
Period ÷ø
è
æt
ö
´ ç DEAD ÷
Period
è
ø
(
) (
)
where
•
•
•
•
•
•
•
•
VOUT_MAX = maximum achievable output voltage
VIN = minimum input voltage
tOFF_MAX = maximum OFF-time (180 ns typical for adequate margin)
Period = 1/Fs
IOUT_MAX = maximum current
RDS(ON)_MAX = maximum high-side MOSFET ON-resistance (see Electrical Characteristics)
RL = DCR of the inductor
tDEAD = dead time (40 ns)
(28)
8.2.2.9 Step Nine: Select Loop Compensation Components
There are several possible methods to design closed-loop compensation for DC/DC converters. For the ideal
current mode control, the design equations can be easily simplified. The power stage gain is constant at low
frequencies, and rolls off at –20 dB/decade above the modulator pole frequency. The power stage phase is zero
degrees at low frequencies and starts to fall one decade below the modulator pole frequency reaching a
minimum of –90 degrees one decade above the modulator pole frequency. In this case the modulator pole is a
simple pole shown in Equation 29.
1
FPMOD =
2pCOUTRLOAD
(29)
For the TPS54678 most circuits will have relatively high amounts of slope compensation. As more slope
compensation is applied, the power stage characteristics will deviate from the ideal approximations. The phase
loss of the power stage will extend beyond –90 degrees and can approach –180 degrees, making compensation
more difficult. The power stage transfer function can be solved but it is a tedious hand calculation that does not
lend itself to simple approximations. It is easier to either simulate the circuit or to actually measure the plant
transfer function so that a reliable compensation circuit can be designed. The latter technique used in this design
procedure. The power stage plant was measured and is shown in Figure 31.
135
Gain
Phase
Gain (dB)
20
90
10
45
0
0
−10
−45
−20
−90
−30
−135
−40
−180
−50
100
1000
10000
Frequency (Hz)
Phase (°)
30
−225
1000000
100000
G001
Figure 31. Measured Plant Bode
For this design, the desired crossover frequency Fc is 50 kHz. From the power stage gain and phase plot above,
the gain at 50 kHz is –10.6 dB and the phase is –123.3 degrees. Because the plant phase loss is greater than
–90 degrees, to achieve at least 60 degrees of phase margin, additional phase boost from a feedforward
capacitor in parallel with the upper resistor of the voltage set point divider is required.
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See the schematic in Figure 30. R3 sets the gain of the compensated error amplifier to be equal and opposite (in
dB) to the power stage gain at Fc, so 10.6 dB is needed. The required value of R3 can be calculated from
Equation 30.
æ -GPlant ö
ç 20 ÷
ø
VOUT
10è
´
R3 =
gmEA
VREF
(30)
The compensator zero formed by R3 and C6 is placed at the plant pole, as shown approximately 2.5 kHz. The
required value for C6 is given by Equation 31.
1
C6 =
2pR3Fplant pole
(31)
The high-frequency noise pole formed by C5 and R3 is not used in this design. If the resulting design shows
noise susceptibility, the value of C5 can be calculated per Equation 32.
1
C5 =
2pR3Fpole
(32)
To avoid a penalty in loop phase, the Fpole in Equation 32 should be placed a decade above Fc or higher, and is
intended to reject noise at FSW.
The feedforward capacitor C15 is used to increase the phase boost at crossover above what is normally
available from Type II compensation. It places an additional zero/pole pair with the zero located at Equation 33
and the pole at Equation 34.
1
Fz =
2pC15R9
(33)
Fp =
1
2pC15 (R9 || R10 )
(34)
This zero and pole pair is not independent since R9 and R10 are set by the desired VOUT. Once the zero location
is chosen, the pole is fixed as well. For optimum performance, the zero and pole should be located symmetrically
about the intended crossover frequency. The required value for C15 can be calculated from Equation 35.
1
C15 =
VREF
2pR9Fc
VOUT
(35)
Table 2 lists the values the compensation equations yield.
Table 2. Frequency Compensation Component Values
REF DES
CALCULATED VALUE
CHOSEN VALUE
R3
19.6 kΩ
26.7 kΩ
C6
2.38 nF
2.2 nF
C15
225 pF
150 pF
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8.2.3 Application Curves
Figure 32 through Figure 47 were measured on the TPS54678 Evaluation Module. More explanation of
waveforms, as well as a schematic document can be found in the TPS54678EVM-155 6-A, SWIFT™ Regulator
Evaluation Module user guide, SLVU747.
8.2.3.1 Additional Information About Application Curves
8.2.3.1.1 Efficiency
System efficiency may be lower than shown in Figure 32 at higher ambient temperatures, due to temperature
variation in the drain-to-source resistance RDS(ON) of the internal MOSFETs.
8.2.3.1.2 Voltage Ripple Measurements
Probe placement and noise pickup can give unreliable voltage ripple results. Figure 37 and Figure 38 show the
output voltage ripple of the converter, measured directly across the output capacitors. Likewise, Figure 39 and
Figure 40 show the input voltage ripple of the converter, measured directly across the input capacitors.
8.2.3.1.3 Start-Up and Shutdown Waveforms
Figure 41 and Figure 42 show the start-up waveforms for the TPS54678EVM-155. In Figure 41, the output
voltage ramps up as soon as the input voltage reaches the UVLO threshold as set by the R1 and R2 resistor
divider network. In Figure 42, the input voltage is initially applied and the output is inhibited by using a jumper at
JP1 to tie EN to GND. When the jumper is removed, EN is released. When the EN voltage reaches the enablethreshold voltage, the start-up sequence begins and the output voltage ramps up to the externally set value of
1.2 V.
The TPS54678 is designed to start up into prebiased outputs. Figure 43 shows the output voltage start-up
waveform when the output is prebiased with 550 mV at no load.
Figure 44 and Figure 45 show the shutdown waveforms for the TPS54678EVM-155. In Figure 44, the output
voltage ramps down as soon as the input voltage falls below the UVLO stop threshold as set by the R1 and R2
resistor divider network. At the point of shutdown, the input voltage rises slightly due to the resistive drop in the
input feed impedance. In Figure 45, the output is inhibited by using a jumper at JP1 to tie EN to GND.
8.2.3.1.4 Hiccup Mode Current Limit
The TPS54678 has hiccup mode current limit. When the peak switch current exceeds the current limit threshold,
the device shuts down and restarts. Hiccup mode current limit operation is shown in Figure 46 and Figure 47.
Figure 46 shows the hiccup mode current limit with a slight resistive overload. When the peak current limit is
exceeded, the output voltage is disabled. Figure 47 shows the operation of the TPS54678 with the output
shorted to ground. The device continuously resets until the fault condition is removed.
96
1.2100
3V
4V
5V
6V
Efficiency (%)
92
1.2090
90
88
86
84
1.2085
1.2080
1.2075
1.2070
1.2065
1.2060
82
80
3V
4V
5V
6V
1.2095
Output Voltage (V)
94
1.2055
0
0.5
1
1.5
2
2.5 3 3.5 4
Output Current (A)
4.5
5
5.5
6
1.2050
0
G002
Figure 32. Efficiency vs Load Current
28
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0.5
1
1.5
2
2.5 3 3.5 4
Output Current (A)
4.5
5
5.5
6
G003
Figure 33. Load Regulation
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1.2100
3A
4A
5A
6A
1.2095
Output Voltage (V)
1.2090
1.2085
VOUT = 50 mV / div (ac coupled)
1.2080
1.2075
1.2070
1.2065
IOUT = 1 A / div
Load step = 0 - 3 A
1.2060
1.2055
1.2050
3
3.5
4
4.5
5
Input Voltage (V)
5.5
6
Time = 100 ms/div
G003
Figure 35. Transient Response
Figure 34. Line Regulation
150
Gain
Phase
Gain (dB)
40
VIN = 3 V
IOUT = 6 A
120
30
90
20
60
10
30
0
0
Phase (°)
50
VOUT = 20 mV / div (ac coupled)
−30
−10
−20
100
1000
10000
Frequency (Hz)
100000
−60
1000000
SW Node = 5 V / div
G005
Time = 500 ns/div
VIN = 3 V, IOUT = 6 A
Figure 36. Loop Response
Figure 37. Output Ripple
VIN = 3 V
IOUT = 6 A
VIN = 6 V
IOUT = 6 A
VIN = 50 mV / div
(ac coupled)
20 MHZ BW Limited
VOUT = 20 mV / div (ac coupled)
SW Node = 5 V / div
SW Node = 5 V / div
Time = 500 ns/div
Time = 500 ns/div
VIN = 6 V, IOUT = 6 A
VIN = 3 V, IOUT = 6 A
Figure 38. Output Ripple
Figure 39. Input Ripple
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VIN = 50 mV / div (ac coupled)
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VIN = 6 V
IOUT = 6 A
IOUT = 1 A / div (inverted for clarity)
VIN = 1 V / div
SW Node = 5 V / div
VOUT = 200 mV / div
IOUT = 4.6 A
Time = 500 ms/div
Time = 500 ns/div
VIN = 6 V, IOUT = 6 A
Figure 41. Start-Up Relative to VIN
Figure 40. Input Ripple
VIN = 1 V / div
VIN = 1 V / div
IOUT = 1 A / div
(inverted for clarity)
VOUT = 200 mV / div
IOUT = 1 A / div
(inverted for clarity)
IOUT = 4.6 A
VOUT = 200 mV / div
Time = 500 ms/div
IOUT = 4.6 A
Time = 500 ms/div
Figure 43. Start-Up into Prebias
Figure 42. Start-Up Relative to Enable
VOUT = falling, 200 mV / div
VOUT = falling, 200 mV / div
VIN = 1 V / div
VIN = 1 V / div (near 2.7 V)
VIN = 5 V
IOUT = 6.6 A
IOUT = 6.6 A
IOUT = 1 A / div (inverted for clarity)
IOUT = 1 A / div (invertey for clarity)
Time = 100 ms/div
Time = 100 ms/div
Figure 44. Shutdown Relative to VIN
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VIN = 1 V / div
VIN = 6 V
IOUT = short
VIN = 5 V
IOUT = 9.2 A
VOUT = 100 mV / div
IOUT = 1 A / div
IOUT = 5 A / div
VOUT = 200 mV / div
Time = 5 ms/div
Time = 5 ms/div
Figure 46. Hiccup Mode Current Limit Shutdown
Figure 47. Hiccup Mode Current Limit Restart into Short
Circuit
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9 Power Supply Recommendations
These devices are designed to operate from an input voltage supply between 2.95 V and 6 V. This supply must
be well regulated. Proper bypassing of input supplies and internal regulators is also critical for noise
performance, as is PCB layout and grounding scheme. See the recommendations in the Layout Guidelines
section.
10 Layout
10.1 Layout Guidelines
Layout is a critical portion of good power supply design. There are several signal paths that conduct fast
changing currents or voltages that can interact with stray inductance or parasitic capacitance to generate noise
or degrade the power supplies performance.
• Minimize the loop area formed by the bypass capacitor connections and the VIN pins. See Figure 48 for a
PCB layout example.
• Tie the GND pins and AGND pin directly to the thermal pad under the TPS54678 device. Connect the thermal
pad to any internal PCB ground planes using multiple vias directly under the device. Additional vias can be
used to connect the top-side ground area to the internal planes near the input and output capacitors. For
operation at full rated load, the top-side ground area along with any additional internal ground planes must
provide adequate heat dissipating area.
• Place the input bypass capacitor as close as possible to the device.
• Route the PH pin to the output inductor. Because the PH connection is the switching node, place the output
inductor close to the PH pins. Minimize the area of the PCB conductor to prevent excessive capacitive
coupling.
• The boot capacitor must also be located close to the device.
• The sensitive analog ground connections for the feedback voltage divider, compensation components, softstart capacitor and frequency set resistor must be connected to a separate analog ground trace as shown in
Figure 48.
• The RT/CLK pin is particularly sensitive to noise so the RT resistor should be located as close as possible to
the device and routed with minimal trace lengths.
• The additional external components can be placed approximately as shown. It is possible to obtain
acceptable performance with alternate PCB layouts, however, this layout has been shown to produce good
results and can be used as a guide.
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10.2 Layout Example
Figure 48. TPS54678 Layout Example
10.3 Power Dissipation Estimate
The following formulas show how to estimate the IC power dissipation under continuous conduction mode (CCM)
operation. The power dissipation of the IC (Ptot) includes conduction loss (Pcon), dead time loss (Pd), switching
loss (Psw), gate drive loss (Pgd) and supply current loss (Pq).
Pcon = IO2 × RDS(on) (temperature dependent)
where
• IO is the output current (A)
• RDS(on) is the on-resistance of the high-side MOSFET with given temperature (Ω)
Pd = ƒsw × IO × 0.7 × (20 nS + 20 nS)
(36)
where
• IO is the output current (A)
• ƒsw is the switching frequency (Hz)
Psw = 0.5 × VIN × IO × ƒsw× 7 × 10–9
(37)
where
• IO is the output current (A)
• VIN is the input voltage (V)
• ƒsw is the switching frequency (Hz)
Pgd = 2 × VIN × ƒsw× 6 × 10–9
(38)
where
• VIN is the input voltage (V)
• ƒsw is the switching frequency (Hz)
Pq = VIN × 500 × 10–6
(39)
where
•
VIN is the input voltage (V)
(40)
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Power Dissipation Estimate (continued)
So
Ptot = Pcon + Pd + Psw + Pgd + Pq
where
•
Ptot is the total device power dissipation (W)
(41)
For given TA,
TJ = TA + Rth × Ptot
where
•
•
•
•
Ptot is the total device power dissipation (W)
TA is the ambient temperature (°C)
TJ is the junction temperature (°C)
Rth is the thermal resistance of the package (°C/W)
(42)
For given TJ max = 150°C
TA max = TJ max – Rth × Ptot
where
•
•
•
•
Ptot is the total device power dissipation (W)
Rth is the thermal resistance of the package (°C/W)
TJ max is maximum junction temperature (°C)
TA max is maximum ambient temperature (°C)
(43)
There are additional power losses in the regulator circuit due to the inductor AC and DC losses and trace
resistance that impact the overall efficiency of the regulator.
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
For more SWIFTTM documentation, see the TI website at www.ti.com/swift.
11.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS54678 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Documentation Support
11.2.1 Related Documentation
For related documentation, see the following:
TPS54678EVM-155 6-A, SWIFT™ Regulator Evaluation Module
11.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
11.4 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
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11.5 Trademarks
SWIFT, E2E are trademarks of Texas Instruments.
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.6 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.7 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
36
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Copyright © 2012–2019, Texas Instruments Incorporated
Product Folder Links: TPS54678
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS54678RTER
ACTIVE
WQFN
RTE
16
3000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 150
54678
TPS54678RTET
ACTIVE
WQFN
RTE
16
250
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 150
54678
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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