TPS61089, TPS610891
SLVSD38C – NOVEMBER 2015 – REVISED AUGUST 2021
TPS61089x 12.6-V, 7-A Fully-Integrated Synchronous Boost Converters
in 2.0-mm x 2.5-mm VQFN Package
1 Features
•
•
•
•
•
•
•
•
•
•
•
•
•
Input voltage range: 2.7 V to 12 V
Output voltage range: 4.5 V to 12.6 V
Up to 90% efficiency at VIN = 3.3 V, VOUT = 9 V,
and IOUT = 2 A
Resistor-programmable peak current limit up to 10
A for high pulse current
Adjustable switching frequency: 200 kHz to 2.2
MHz
4-ms built-in soft start time
PFM operation mode at light load (TPS61089)
Forced PWM operation mode at light load
(TPS610891)
Internal output overvoltage protection at 13.2 V
Cycle-by-cycle overcurrent protection
Thermal shutdown
2.00-mm × 2.50-mm VQFN HotRod™ package
Create a custom design using the TPS61089x with
the WEBENCH® Power Designer
2 Applications
•
•
•
Bluetooth™ speaker
Quick charge power bank
Portable POS terminal
solution for portable equipment. The TPS61089x
features a wide input voltage range from 2.7 V
to 12 V to support applications powered with
single-cell or two-cell Lithium ion/polymer batteries.
The TPS61089x has 7-A continuous switch current
capability and provides output voltage up to 12.6 V.
The TPS61089x uses adaptive constant off-time peak
current control topology to regulate the output voltage.
In moderate to heavy load condition, the TPS61089x
works in the pulse width modulation (PWM) mode.
In light load condition, the TPS61089 works in the
pulse frequency modulation (PFM) mode to improve
the efficiency, while the TPS610891 still works
in the PWM mode to avoid application problems
caused by low switching frequency. The switching
frequency in PWM mode is adjustable from 200
kHz to 2.2 MHz. The TPS61089x also implements
a built-in 4-ms soft start function and an adjustable
peak switch current limit function. In addition, the
device provides 13.2-V output overvoltage protection,
cycle-by-cycle overcurrent protection, and thermal
shutdown protection.
The TPS61089x is available in an extremely compact
size of a 2.0-mm × 2.5-mm 11-pin VQFN package.
3 Description
Device Information Table
The TPS61089x represents the TPS61089 and the
TPS610891. The TPS61089x is a fully-integrated
synchronous boost converter with a 19-mΩ main
power switch and a 27-mΩ rectifier switch. The device
provides a high-efficiency and small-size power
(1)
PART NUMBER
PACKAGE(1)
BODY SIZE (NOM)
TPS61089x
VQFN (11)
2.00 mm × 2.50 mm
For all available packages, see the orderable addendum at
the end of the data sheet.
L1
VIN
VOUT
SW
C4
R3
C1
VOUT
C2
BOOT
GND
R1
FSW
VIN
FB
EN
COMP
ON
R2
OFF
C6
VCC
R5
ILIM
C3
R4
C5
Typical Application Circuit
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. UNLESS OTHERWISE NOTED, this document contains PRODUCTION
DATA.
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Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Device Comparison Table...............................................3
6 Pin Configuration and Functions...................................4
7 Specifications.................................................................. 5
7.1 Absolute Maximum Ratings........................................ 5
7.2 ESD Ratings............................................................... 5
7.3 Recommended Operating Conditions.........................5
7.4 Thermal Information....................................................5
7.5 Electrical Characteristics.............................................6
7.6 Typical Characteristics................................................ 7
8 Detailed Description........................................................9
8.1 Overview..................................................................... 9
8.2 Functional Block Diagram........................................... 9
8.3 Feature Description...................................................10
8.4 Device Functional Modes..........................................11
9 Application and Implementation.................................. 13
9.1 Application Information............................................. 13
9.2 Typical Application.................................................... 13
10 Power Supply Recommendations..............................21
11 Layout........................................................................... 22
11.1 Layout Guidelines................................................... 22
11.2 Layout Example...................................................... 22
12 Device and Documentation Support..........................24
12.1 Device Support....................................................... 24
12.2 Receiving Notification of Documentation Updates..24
12.3 Support Resources................................................. 24
12.4 Trademarks............................................................. 24
12.5 Electrostatic Discharge Caution..............................24
12.6 Glossary..................................................................24
13 Mechanical, Packaging, and Orderable
Information.................................................................... 25
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision B (July 2016) to Revision C (August 2021)
Page
• Updated the numbering format for tables, figures and cross-references throughout the document. .................1
• Corrected grammar and numeric format throughout document..........................................................................1
• Added WEBENCH links......................................................................................................................................1
Changes from Revision A (April 2016) to Revision B (July 2016)
Page
• Changed x axis in .............................................................................................................................................. 7
• Changed x axis in .............................................................................................................................................. 7
2
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5 Device Comparison Table
(1)
PART NUMBER
OPERATION MODE AT LIGHT LOAD
TPS61089RNR
PFM
TPS610891RNR(1)
Forced PWM
Product Preview. Contact TI factory for more information.
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SW
6 Pin Configuration and Functions
FSW
BOOT
VCC
VIN
FB
ILIM
VOUT
EN
GND
COMP
Figure 6-1. 11-Pin VQFN With Thermal Pad RNR Package (Top View)
Table 6-1. Pin Functions
PIN
NAME
4
NUMBER
I/O
DESCRIPTION
FSW
1
I
The switching frequency is programmed by a resister between this pin and the SW pin.
VCC
2
O
Output of the internal regulator. A ceramic capacitor of more than 1.0 µF is required between
this pin and ground.
FB
3
I
Output voltage feedback
COMP
4
O
Output of the internal error amplifier. The loop compensation network should be connected
between this pin and the GND pin.
GND
5
PWR
Ground
VOUT
6
PWR
Boost converter output
EN
7
I
Enable logic input. Logic high level enables the device. Logic low level disables the device
and turns it into shutdown mode.
ILIM
8
O
Adjustable switching peak current limit. An external resister should be connected between
this pin and the GND pin.
VIN
9
I
IC power supply input
BOOT
10
O
Power supply for high-side MOSFET gate driver. A capacitor must be connected between
this pin and the SW pin
SW
11
PWR
The switching node pin of the converter. It is connected to the drain of the internal low-side
power MOSFET and the source of the internal high-side power MOSFET.
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature (unless otherwise noted)(1)
MIN
MAX
BOOT
–0.3
SW + 7
VIN, SW, FSW, VOUT
–0.3
14.5
EN, VCC, COMP
–0.3
7
ILIM, FB
–0.3
3.6
Operating junction temperature, TJ
–40
150
°C
Storage temperature, Tstg
–65
150
°C
Voltage at terminals(2)
(1)
(2)
UNIT
V
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress
ratings only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under
Recommended Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device
reliability.
All voltage values are with respect to network ground terminal.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic
discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins
(1)
UNIT
±2000
Charged device model (CDM), per JEDEC specification JESD22-C101, all pins (2)
V
±500
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VIN
Input voltage range
VOUT
Output voltage range
L
Inductance, effective value
CIN
Input capacitance, effective value
10
CO
Output capacitance, effective value
10
TJ
Operating junction temperature
NOM
MAX
2.7
12
4.5
0.47
12.6
UNIT
V
V
2.2
10
µH
47
1000
µF
125
°C
µF
–40
7.4 Thermal Information
TPS61089x
THERMAL METRIC(1)
RNR (VQFN)
UNIT
11 PINS
RθJA
Junction-to-ambient thermal resistance
53.4
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
59.2
°C/W
RθJB
Junction-to-board thermal resistance
9.6
°C/W
ψJT
Junction-to-top characterization parameter
0.5
°C/W
ψJB
Junction-to-board characterization parameter
9.5
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
0.7
°C/W
RθJA(EVM) (2)
Junction-to-ambient thermal resistance on EVM
39.2
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
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(2)
The EVM board is a 4-layer PCB of 76-mm x 52-mm size. The copper thickness of top layer and bottom layer is 2 oz. The copper
thickness of inner layers is 1 oz.
7.5 Electrical Characteristics
VIN = 2.7 V to 5.5 V, VOUT = 9 V, TJ = –40°C to 125°C. Typical values are at TJ = 25°C, unless otherwise noted.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
POWER SUPPLY
VIN
Input voltage range
2.7
VIN rising
12
V
2.7
V
2.5
V
VIN_UVLO
Input voltage undervoltage lockout
(UVLO) threshold
VIN_HYS
VIN UVLO hysteresis
200
mV
VCC
VCC regulation voltage
ICC = 2 mA, VIN = 8 V
5.8
V
VCC_UVLO
VCC UVLO threshold
VCC falling
2.1
V
Quiescent current into VIN pin
IC enabled, No load, VIN = 2.7 V to 5.5 V, VFB = 1.3
V, VOUT = 12 V, TJ ≤ 85°C
1
3
µA
Quiescent current into VOUT pin
IC enabled, No load, VIN = 2.7 V to 5.5 V, VFB = 1.3
V, VOUT = 12 V, TJ ≤ 85°C
100
180
µA
Shutdown current into VIN pin
IC disabled, VIN = 2.7 V to 5.5 V, TJ ≤ 85°C
1
3
µA
12.6
V
1.212
1.236
V
IQ
ISD
VIN falling
2.4
OUTPUT
VOUT
Output voltage range
4.5
PWM mode
VREF
Reference voltage at FB pin
IFB_LKG
Leakage current into FB pin
VFB = 1.2 V
VOVP
Output overvoltage protection
threshold
VOUT rising
VOVP_HYS
Output overvoltage protection
hysteresis
VOUT falling below VOVP
tSS
Soft startup time
COUT(effective) = 47 µF, IOUT = 0 A
1.188
PFM mode
1.224
12.7
13.2
V
100
nA
13.6
V
0.25
2
4
V
6
ms
ERROR AMPLIFIER
ISINK
COMP pin sink current
VFB = VREF + 200 mV, VCOMP = 1.9 V
20
µA
ISOURCE
COMP pin source current
VFB = VREF – 200 mV, VCOMP = 1.9 V
20
µA
VCCLP_H
High clamp voltage at the COMP pin
VFB = 1 V, RILIM = 127 kΩ
2.3
V
VCCLP_L
Low clamp voltage at the COMP pin
VFB = 1.4 V, RILIM = 127 kΩ
1.4
V
GEA
Error amplifier transconductance
VCOMP = 1.9 V
190
µS
POWER SWITCH
RDS(on)
High-side MOSFET on-resistance
VCC = 6 V
27
44
mΩ
Low-side MOSFET on-resistance
VCC = 6 V
19
31
mΩ
SWITCHING FREQUENCY
fSW
Switching frequency
tON_min
Minimum on time
RFSW = 301 kΩ
500
RFSW = 46.4 kΩ
2000
VCC = 6 V
kHz
kHz
90
180
ns
CURRENT LIMIT
ILIM
Peak switch current limit, TPS61089
VILIM
Internal reference voltage at ILIM pin
RILIM = 127 kΩ
7.3
8.1
8.9
A
RILIM = 100 kΩ
9.0
10
11
A
1.212
V
EN LOGIC INPUT
VEN_H
EN Logic high threshold
VEN_L
EN Logic Low threshold
REN
EN pulldown resistor
1.2
0.4
V
V
800
kΩ
150
°C
20
°C
PROTECTION
6
TSD
Thermal shutdown threshold
TJ rising
TSD_HYS
Thermal shutdown hysteresis
TJ falling below TSD
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7.6 Typical Characteristics
100
100
90
90
80
80
70
70
Efficiency (%)
Efficiency (%)
VIN = 3.6 V, VOUT = 9 V, TJ = 25°C, unless otherwise noted
60
50
40
30
40
20
VIN = 3 V
VIN = 3.6 V
VIN = 4.2 V
10
0.001
0.01
0.1
Output Current (A)
TPS61089
1
VOUT = 5 V
VOUT = 9 V
VOUT = 12 V
10
0
0.0001
10
0.001
0.01
0.1
Output Current (A)
D001
VOUT = 9 V
TPS61089
Figure 7-1. Load Efficiency with Different Input
Voltage
1
10
D001
VIN = 3.6 V
Figure 7-2. Load Efficiency with Different Output
Voltage
12
2500
10
2000
Frequency (kHz)
Current Limit (A)
50
30
20
0
0.0001
60
8
6
4
1500
1000
500
2
0
100
0
150
200
250
300
Resistance (k:)
350
0
400
100
200
300
400 500 600
Resistance (k:)
700
800
900
D004
D003
TPS61089
Figure 7-4. Switching Frequency Setting
Figure 7-3. Switching Peak Current Limit Setting
160
1.22
140
Quiescent Current (PA)
Reference Voltage (V)
1.216
1.212
1.208
1.204
120
100
80
60
40
1.2
-40
-20
0
20
40
60
80
Temperature (°C)
100
120
140
20
-40
-20
D005
Figure 7-5. Reference Voltage vs Temperature
0
20
40
Temperature (°C)
60
80
100
D006
Figure 7-6. Quiescent Current vs Temperature
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2
Shutdown Current (PA)
1.6
1.2
0.8
0.4
0
-40
-20
0
20
40
Temperature (°C)
60
80
100
D007
Figure 7-7. Shutdown Current vs Temperature
8
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8 Detailed Description
8.1 Overview
The TPS61089x is a synchronous boost converter, integrating a 19-mΩ main power switch and a 27-mΩ rectifier
switch with adjustable switch current up to 10 A. It is capable to output continuous power more than 18 W from
input of a single cell Lithium-ion battery or two-cell Lithium-ion batteries in series. The TPS61089x operates
at a quasi-constant frequency pulse-width modulation (PWM) at moderate to heavy load currents. At light load
current, the TPS61089 operates in PFM mode and the TPS610891 operates in forced PWM (FPWM) mode. The
PFM mode brings high efficiency over the entire load range, and the FPWM mode can avoid the acoustic noise
and switching frequency interference at light load. The converter uses the constant off-time peak current mode
control scheme, which provides excellent line and load transient response with minimal output capacitance. The
external loop compensation brings flexibility to use different inductors and output capacitors. The TPS61089x
supports adjustable switching frequency ranging from 200 kHz to 2.2 MHz. The device implements cycle-bycycle current limit to protect the device from overload conditions during boost switching. The current limit is set
by an external resistor.
8.2 Functional Block Diagram
L1
VIN
C4
C1
SW
BOOT
VIN
VOUT
VOUT
deadme
control logic
C2
Q
VCC
Shutdown
LDO
C3
R1
GND
S
R
Comp
Comp
CLIMIT
FSW
FB
Comp
Gm
R2
R3
Vref
1/K
SW
VIN
COMP
R5
EN
C5
ON/
OFF
Shutdown
Control
VOUT
OVP
VIN
UVLO
Shutdown
Vref
CLIMIT
Thermal
shutdown
ILIM
R4
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8.3 Feature Description
8.3.1 Undervoltage Lockout (UVLO)
An undervoltage lockout (UVLO) circuit stops the operation of the converter when the input voltage drops below
the typical UVLO threshold of 2.5 V. A hysteresis of 200 mV is added so that the device cannot be enabled again
until the input voltage goes up to 2.7 V. This function is implemented to prevent the device from malfunctioning
when the input voltage is between 2.5 V and 2.7 V.
8.3.2 Enable and Disable
When the input voltage is above maximal UVLO rising threshold of 2.7 V and the EN pin is pulled above the
high threshold, the TPS61089x is enabled. When the EN pin is pulled below the low threshold, the TPS61089x
goes into shutdown mode. The device stops switching in shutdown mode and consumes less than 3-µA current.
Because of the body diode of the high-side rectifier FET, the input voltage goes through the body diode and
appears at the VOUT pin at shutdown mode.
8.3.3 Soft Start
The TPS61089x implements the soft start function to reduce the inrush current during start-up. The TPS61089x
begins soft start when the EN pin is pulled to logic high voltage. The soft start time is typically 4 ms.
8.3.4 Adjustable Switching Frequency
The TPS61089x features a wide adjustable switching frequency ranging from 200 kHz to 2.2 MHz. The switching
frequency is set by a resistor connected between the FSW pin and the SW pin of the TPS61089x. Do not leave
the FSW pin open. Use Equation 1 to calculate the resistor value required for a desired frequency.
4u(
RFREQ
1
¦SW
tDELAY u
VOUT
)
9IN
CFREQ
(1)
where
•
•
•
•
•
•
RFREQ is the resistance connected between the FSW pin and the SW pin
CFREQ = 24 pF
ƒSW is the desired switching frequency
tDELAY = 86 ns
VIN is the input voltage
VOUT is the output voltage
8.3.5 Adjustable Peak Current Limit
To avoid an accidental large peak current, an internal cycle-by-cycle current limit is adopted. The low-side switch
turns off immediately as long as the peak switch current touches the limit. The peak inductor current can be
set by selecting the correct external resistor value correlating with the required current limit. Use Equation 2 to
calculate the correct resistor value for the TPS61089.
ILIM =
1030000
RILIM
(2)
where
•
•
RILIM is the resistance connected between the ILIM pin and ground
ILIM is the switch peak current limit
For a typical current limit of 8 A, the resistor value is 127 kΩ for the TPS61089.
8.3.6 Overvoltage Protection
If the output voltage at the VOUT pin is detected above the overvoltage protection threshold of 13.2 V (typical
value), the TPS61089x stops switching immediately until the voltage at the VOUT pin drops the hysteresis
10
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voltage lower than the output overvoltage protection threshold. This function prevents overvoltage on the output
and secures the circuits connected to the output from excessive overvoltage.
8.3.7 Thermal Shutdown
A thermal shutdown is implemented to prevent damage due to excessive heat and power dissipation. Typically,
the thermal shutdown happens at the junction temperature of 150°C. When the thermal shutdown is triggered,
the device stops switching until the junction temperature falls below typically 130°C, then the device starts
switching again.
8.4 Device Functional Modes
8.4.1 Operation
The TPS61089x synchronous boost converter operates at a quasi-constant frequency pulse width modulation
(PWM) in moderate to heavy load condition. Based on the VIN to VOUT ratio, a circuit predicts the required
off-time of the switching cycle. At the beginning of each switching cycle, the low-side N-MOSFET switch, shown
in Section 8.2, is turned on, and the inductor current ramps up to a peak current that is determined by the output
of the internal error amplifier. After the peak current is reached, the current comparator trips, and turns off the
low-side N-MOSFET switch and the inductor current goes through the body diode of the high-side N-MOSFET
in a dead-time duration. After the dead-time duration, the high-side N-MOSFET switch is turned on. Since the
output voltage is higher than the input voltage, the inductor current decreases. The high-side switch is not turned
off until the fixed off-time is reached. After a short dead-time duration, the low-side switch is turned on again and
the switching cycle is repeated.
In light load condition, the TPS61089 implements PFM mode for applications requiring high efficiency at light
load. And the TPS610891 implements forced PWM mode for applications requiring fixed switching frequency to
avoid unexpected switching noise interference.
8.4.1.1 Forced PWM Mode
In forced PWM mode, the TPS610891 keeps the switching frequency unchanged in light load condition. When
the load current decreases, the output of the internal error amplifier decreases as well to keep the inductor
peak current down, delivering less power from input to output. When the output current further reduces, the
current through the inductor will decrease to zero during the off-time. The high-side N-MOSFET is not turned
off even if the current through the MOSFET is zero. Thus, the inductor current changes its direction after it runs
to zero. The power flow is from output side to input side. The efficiency will be low in this mode. But with the
fixed switching frequency, there is no audible noise and other problems which might be caused by low switching
frequency in light load condition.
8.4.1.2 PFM Mode
The TPS61089 improves the efficiency at light load with PFM mode. When the converter operates in light load
condition, the output of the internal error amplifier decreases to make the inductor peak current down, delivering
less power to the load. When the output current further reduces, the current through the inductor will decrease
to zero during the off-time. Once the current through the high-side N-MOSFET is zero, the high-side MOSFET
is turned off until the beginning of the next switching cycle. When the output of the error amplifier continuously
goes down and reaches a threshold with respect to the peak current of ILIM / 10, the output of the error amplifier
is clamped at this value and does not decrease any more. If the load current is smaller than what the TPS61089
delivers, the output voltage increases above the nominal setting output voltage. The TPS61089 extends its off
time of the switching period to deliver less energy to the output and regulate the output voltage to 1.0% higher
than the nominal setting voltage. With the PFM operation mode, the TPS61089 keeps the efficiency above 70%
even when the load current decreases to 1 mA. At light load, the output voltage ripple is much smaller due to low
peak inductor current. Refer to Figure 8-1.
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Output
Voltage
PFM mode at light load
1.01 x VOUT_NOM
VOUT_NOM
PWM mode at heavy load
Figure 8-1. Output Voltage in PWM Mode and PFM Mode
12
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9 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
9.1 Application Information
The TPS61089x is designed for outputting voltage up to 12.6 V with 7-A continuous switch current capability to
deliver more than 18-W power. The TPS61089x operates at a quasi-constant frequency pulse-width modulation
(PWM) in moderate to heavy load condition. In light load condition, the TPS61089 operates in PFM mode
and the TPS610891 operates in forced PWM mode. The PFM mode brings high efficiency over entire load
range, while PWM mode can avoid the acoustic noise as the switching frequency is fixed. In PWM mode, the
TPS61089x converter uses the adaptive constant off-time peak current control scheme, which provides excellent
transient line and load response with minimal output capacitance. The TPS61089x can work with a different
inductor and output capacitor combination by external loop compensation. It also supports adjustable switching
frequency ranging from 200 kHz to 2.2 MHz.
9.2 Typical Application
L1
1.8µH
VIN = 3.0V to 4.35V
C1
22µF
VOUT = 9V
SW
C4
0.1µF BOOT
R3
301k
VOUT
C2
3 x 22µF
GND
R1
681k
FSW
VIN
FB
EN
COMP
ON
OFF
VCC
R2
107k
C6
ILIM
C3
2.2µF
R4
127k
R5
17.4k
C5
4.7nF
Figure 9-1. TPS61089x Single Cell Li-ion Battery to 9-V/2-A Output Converter
9.2.1 Design Requirements
Table 9-1. Design Parameters
DESIGN PARAMETERS
EXAMPLE VALUES
Input voltage range
3.0 to 4.35 V
Output voltage
9V
Output voltage ripple
100 mV peak to peak
Output current rating
2A
Operating frequency
500 kHz
Operation mode at light load
PFM
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9.2.2 Detailed Design Procedure
9.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS61089x device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
9.2.2.2 Setting Switching Frequency
The switching frequency is set by a resistor connected between the FSW pin and the SW pin of the TPS61089x.
The resistor value required for a desired frequency can be calculated using Equation 3.
4u(
RFREQ
1
¦SW
tDELAY u
VOUT
)
9IN
CFREQ
(3)
where
•
•
•
•
•
•
RFREQ is the resistance connected between the FSW pin and the SW pin
CFREQ = 24 pF
ƒSW is the desired switching frequency
tDELAY = 86 ns
VIN is the input voltage
VOUT is the output voltage
9.2.2.3 Setting Peak Current Limit
The peak input current is set by selecting the correct external resistor value correlating to the required current
limit. Use Equation 4 to calculate the correct resistor value:
ILIM =
1030000
RILIM
(4)
where
•
•
RILIM is the resistance connected between the ILIM pin and ground
ILIM is the switching peak current limit
For a typical current limit of 8 A, the resistor value is 127 kΩ. Considering the device variation and the tolerance
over temperature, the minimum current limit at the worst case can be 0.8 A lower than the value calculated by
Equation 4. The minimum current limit must be higher than the required peak switch current at the lowest input
voltage and the highest output power to make sure the TPS61089x does not hit the current limit and still can
regulate the output voltage in these conditions.
14
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9.2.2.4 Setting Output Voltage
The output voltage is set by an external resistor divider (R1, R2 in TPS61089x Single Cell Li-ion Battery to
9-V/2-A Output Converter). Typically, a minimum current of 10 μA flowing through the feedback divider gives
good accuracy and noise covering. A resistor of less than 120 kΩ is typically selected for low-side resistor R2.
When the output voltage is regulated, the typical voltage at the FB pin is VREF. Thus, the value of R1 is
calculated as:
R1
(VOUT
VREF ) u R2
VREF
(5)
9.2.2.5 Inductor Selection
Because the selection of the inductor affects the steady state operation of the power supply, transient behavior,
loop stability, and boost converter efficiency, the inductor is the most important component in switching power
regulator design. Three most important specifications to the performance of the inductor are the inductor value,
DC resistance, and saturation current.
The TPS61089x is designed to work with inductor values between 0.47 µH and 10 µH. A 0.47-µH inductor is
typically available in a smaller or lower-profile package, while a 10-µH inductor produces lower inductor current
ripple. If the boost output current is limited by the peak current protection of the IC, using a 10-µH inductor can
maximize the controller’s output current capability.
Inductor values can have ±20% or even ±30% tolerance with no current bias. When the inductor current
approaches saturation level, its inductance can decrease 20% to 35% from the value at 0-A current depending
on how the inductor vendor defines saturation. When selecting an inductor, make sure its rated current,
especially the saturation current, is larger than its peak current during the operation.
Follow Equation 6 to Equation 7 to calculate the peak current of the inductor. To calculate the current in the worst
case, use the minimum input voltage, maximum output voltage, and maximum load current of the application.
To leave enough design margin, TI recommends using the minimum switching frequency, the inductor value with
–30% tolerance, and a low-power conversion efficiency for the calculation.
In a boost regulator, calculate the inductor DC current as in Equation 6.
IDC
VOUT u IOUT
VIN u K
(6)
where
•
•
•
•
VOUT is the output voltage of the boost regulator
IOUT is the output current of the boost regulator
VIN is the input voltage of the boost regulator
η is the power conversion efficiency
Calculate the inductor current peak-to-peak ripple as in Equation 7.
1
IPP
/u
1
VOUT
VIN
1
u ¦SW
VIN
(7)
where
•
•
•
•
•
IPP is the inductor peak-to-peak ripple
L is the inductor value
ƒSW is the switching frequency
VOUT is the output voltage
VIN is the input voltage
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Therefore, the peak current, ILpeak, seen by the inductor is calculated with Equation 8.
ILpeak
IDC
IPP
2
(8)
Set the current limit of the TPS61089x higher than the peak current ILpeak. Then select the inductor with
saturation current higher than the setting current limit.
Boost converter efficiency is dependent on the resistance of its current path, the switching loss associated with
the switching MOSFETs, and the core loss of the inductor. The TPS61089x has optimized the internal switch
resistance. However, the overall efficiency is affected significantly by the DC resistance (DCR) of the inductor,
equivalent series resistance (ESR) at the switching frequency, and the core loss. Core loss is related to the core
material and different inductors have different core loss. For a certain inductor, larger current ripple generates
higher DCR and ESR conduction losses and higher core loss. Usually, a data sheet of an inductor does not
provide the ESR and core loss information. If needed, consult the inductor vendor for detailed information.
Generally, TI would recommend an inductor with lower DCR and ESR. However, there is a tradeoff among the
inductance of the inductor, DCR and ESR resistance, and its footprint. Furthermore, shielded inductors typically
have higher DCR than unshielded inductors. Table 9-2 lists recommended inductors for the TPS61089x. Verify
whether the recommended inductor can support the user's target application with the previous calculations and
bench evaluation. In this application, the Sumida inductor CDMC8D28NP-1R8MC is selected for its small size
and low DCR.
Table 9-2. Recommended Inductors
SATURATION CURRENT /
HEAT RATING CURRENT (A)
SIZE MAX
(L × W × H mm)
VENDOR
12.6
9.4 / 9.3
9.5 x 8.7 x 3.0
Sumida
7.2
14.0 / 11.0
7.3 x 7.2 x 4.0
WurthElektronik
12.5
13.0 / 9.0
7.3 × 7.2 × 4.0
WurthElektronik
9.0
16 / 13
11.2 × 10.3 × 3.0
Cyntec
12.5
12 / 10.5
7.4 × 6.8 × 5.0
Cyntec
PART NUMBER
L (µH)
DCR MAX (mΩ)
CDMC8D28NP-1R8MC
1.8
744311150
1.5
744311220
2.2
PIMB103T-2R2MS
2.2
PIMB065T-2R2MS
2.2
9.2.2.6 Input Capacitor Selection
For good input voltage filtering, TI recommends low-ESR ceramic capacitors. The VIN pin is the power supply for
the TPS61089x. A 0.1-μF ceramic bypass capacitor is recommended as close as possible to the VIN pin of the
TPS61089x. The VCC pin is the output of the internal LDO. A ceramic capacitor of more than 1.0 μF is required
at the VCC pin to get a stable operation of the LDO.
For the power stage, because of the inductor current ripple, the input voltage changes if there is parasitic
inductance and resistance between the power supply and the inductor. It is recommended to have enough input
capacitance to make the input voltage ripple less than 100 mV. Generally, 10-μF input capacitance is sufficient
for most applications.
Note
DC bias effect: High-capacitance ceramic capacitors have a DC bias effect, which has a strong
influence on the final effective capacitance. Therefore, the right capacitor value must be chosen
carefully. The differences between the rated capacitor value and the effective capacitance result from
package size and voltage rating in combination with material. A 10-V rated 0805 capacitor with 10 μF
can have an effective capacitance of less 5 μF at an output voltage of 5 V.
9.2.2.7 Output Capacitor Selection
For small output voltage ripple, TI recommends a low-ESR output capacitor like a ceramic capacitor. Typically,
three 22-μF ceramic output capacitors work for most applications. Higher capacitor values can be used to
improve the load transient response. Take care when evaluating a capacitor’s derating under DC bias. The bias
16
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can significantly reduce capacitance. Ceramic capacitors can lose most of their capacitance at rated voltage.
Therefore, leave margin on the voltage rating to ensure adequate effective capacitance. From the required
output voltage ripple, use the following equations to calculate the minimum required effective capacitance CO:
(VOUT
Vripple _ dis
VIN _ MIN ) u IOUT
9OUT u ¦SW u &O
Vripple _ ESR
(9)
ILpeak u RESR
(10)
where
•
•
•
•
•
•
•
•
Vripple_dis is output voltage ripple caused by charging and discharging of the output capacitor.
Vripple_ESR is output voltage ripple caused by ESR of the output capacitor.
VIN_MIN is the minimum input voltage of boost converter.
VOUT is the output voltage.
IOUT is the output current.
ILpeak is the peak current of the inductor.
ƒSW is the converter's switching frequency.
RESR is the ESR of the output capacitors.
9.2.2.8 Loop Stability
The TPS61089x requires external compensation, which allows the loop response to be optimized for each
application. The COMP pin is the output of the internal error amplifier. An external compensation network
comprised of resistor R5, ceramic capacitors C5 and C6 is connected to the COMP pin.
The power stage small signal loop response of constant off time (COT) with peak current control can be modeled
by Equation 11.
GPS (S)
5O u
§
¨1
'
©
u
2 u Rsense
·§
·
S
S
¸¨ 1
¸
u S u ¦ESRZ ¹ ©
u S u ¦RHPZ ¹
S
1
u S u ¦P
(11)
where
•
•
•
•
•
•
D is the switching duty cycle
RO is the output load resistance
Rsense is the equivalent internal current sense resistor, which is 0.08 Ω
ƒP is the pole's frequency
ƒESRZ is the zero's frequency
ƒRHPZ is the right-half-plane-zero's frequency
The D, ƒP, ƒESRZ, and ƒRHPZ can be calculated by following equations:
D
1
VIN u K
VOUT
(12)
where
•
η is the power conversion efficiency
¦P
2
2S u RO u CO
(13)
where
•
CO is effective capacitance of the output capacitor
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¦ESRZ
1
2S u RESR u CO
www.ti.com
(14)
where
•
RESR is the equivalent series resistance of the output capacitor
¦RHPZ
RO u 1 D
2
(15)
2S u L
The COMP pin is the output of the internal transconductance amplifier. Equation 16 shows the small signal
transfer function of compensation network.
Gc(S)
GEA u REA u VREF
u
VOUT
§
¨1
©
§
·
S
¨1
¸
u S u ¦COMZ ¹
©
·§
·
S
S
¸¨ 1
¸
u S u ¦COMP1 ¹©
u S u ¦COMP2 ¹
(16)
where
•
•
•
•
•
•
GEA is the amplifier’s transconductance
REA is the amplifier’s output resistance
VREF is the refernce voltage at the FB pin
VOUT is the output voltage
ƒCOMP1, ƒCOMP2 are the poles' frequency of the compensation network
ƒCOMZ is the zero's frequency of the compensation network
The next step is to choose the loop crossover frequency, ƒC. The higher in frequency that the loop gain stays
above zero before crossing over, the faster the loop response is. It is generally accepted that the loop gain cross
over no higher than the lower of either 1/10 of the switching frequency, ƒSW, or 1/5 of the RHPZ frequency,
ƒRHPZ.
At the crossover frequency, the loop gain is 1. Thus the value of R5 can be calculated by Equation 17, then set
the values of C5 and C6 (in TPS61089x Single Cell Li-ion Battery to 9-V/2-A Output Converter) by Equation 18
and Equation 19.
R5
S u 9OUT u 5sense u ¦C u &O
± ' u 9REF u *EA
(17)
where
•
ƒC is the selected crossover frequency
The value of C5 can be set by Equation 18.
C5
RO u CO
2R5
(18)
The value of C6 can be set by Equation 19.
C6
RESR u CO
R5
(19)
If the calculated value of C6 is less than 10 pF, it can be left open.
18
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Designing the loop for greater than 45° of phase margin and greater than 10-dB gain margin eliminates output
votlage ringing during the line and load transient.
9.2.3 Application Curves
Vout (AC)
100 mV/div
Vout (AC)
20 mV/div
Inductor
Current
2 A/div
Inductor
Current
1 A/div
SW
3 V/div
SW
3 V/div
VIN = 3.6 V
VOUT = 9 V
IOUT = 2 A
Figure 9-2. Switching Waveforms in CCM
VIN = 3.6 V
EN
1 V/div
Inductor
Current
600 mA/div
Vout
2 V/div
VIN = 3.6 V
IOUT = 200 mA
Figure 9-3. Switching Waveforms in DCM
Vout (AC)
10 mV/div
SW
3 V/div
VOUT = 9 V
Inductor
Current
2 A/div
VOUT = 9 V
IOUT = 20 mA
Figure 9-4. Switching Waveforms in PFM Mode
EN
1 V/div
Figure 9-5. Start-up Waveforms
Output
Current
500 mA/div
Vout
2 V/div
Vout (AC)
500 mV/div
Inductor
Current
2 A/div
VIN = 3.6 V
Figure 9-6. Shutdown Waveforms
VOUT = 9V
IOUT = 1 A to 2 A
Figure 9-7. Load Transient
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Input
Voltage
500 mV/div
Vout (AC)
200 mV/div
VIN = 3.3 V to 4.0
V
VOUT = 9 V
IOUT = 2 A
Figure 9-8. Line Transient
20
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10 Power Supply Recommendations
The device is designed to operate from an input voltage supply range between 2.7 V to 12 V. This input supply
must be well regulated. If the input supply is located more than a few inches from the converter, additional bulk
capacitance can be required in addition to the ceramic bypass capacitors. A typical choice is an electrolytic or
tantalum capacitor with a value of 47 μF.
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11 Layout
11.1 Layout Guidelines
As for all switching power supplies, especially those running at high switching frequency and high currents,
layout is an important design step. If layout is not carefully done, the regulator could suffer from instability and
noise problems. To maximize efficiency, switching rise time and fall time are very fast. To prevent radiation
of high-frequency noise (for example, EMI), proper layout of the high-frequency switching path is essential.
Minimize the length and area of all traces connected to the SW pin, and always use a ground plane under the
switching regulator to minimize interplane coupling. The input capacitor needs to be close to the VIN pin and
GND pin to reduce the input supply current ripple.
The most critical current path for all boost converters is from the switching FET, through the rectifier FET, then
the output capacitors, and back to ground of the switching FET. This high current path contains nanosecond rise
time and fall time, and should be kept as short as possible. Therefore, the output capacitor needs not only to be
close to the VOUT pin, but also to the GND pin to reduce the overshoot at the SW pin and VOUT pin.
11.2 Layout Example
trace on bottom layer
ILIM
VIN
BOOT
GND
VOUT
EN
SW
VOUT
SW
GND
FB
VCC
FSW
COMP
L
CIN
VIN
COUT
GND
Figure 11-1. Layout Example
11.2.1 Thermal Considerations
The maximum IC junction temperature should be restricted to 125°C under normal operating conditions.
Calculate the maximum allowable dissipation, PD(max), and keep the actual power dissipation less than or equal
to PD(max). The maximum-power-dissipation limit is determined using Equation 20.
PD(max)
125 TA
RTJA
(20)
where
•
•
22
TA is the maximum ambient temperature for the application
RθJA is the junction-to-ambient thermal resistance given in the Thermal Information table
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The TPS61089x comes in a thermally-enhanced VQFN package. The pads underneath the package improve
the thermal capabilities of the package. The real junction-to-ambient thermal resistance of the package greatly
depends on the PCB type, layout, and pad connection. Using thick PCB copper and soldering the SW pin,
VOUT pin, and GND pin to large copper plate enhances the thermal performance. Using more vias connects
the ground plate on the top layer and bottom layer around the IC without solder mask also improves the thermal
capability.
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12 Device and Documentation Support
12.1 Device Support
12.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
12.1.2 Development Support
12.1.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the TPS61089x device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
12.3 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
12.4 Trademarks
HotRod™ and TI E2E™ are trademarks of Texas Instruments.
WEBENCH® is a registered trademark of Texas Instruments.
All trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
12.6 Glossary
TI Glossary
24
This glossary lists and explains terms, acronyms, and definitions.
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13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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11-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
TPS61089RNRR
ACTIVE
VQFN-HR
RNR
11
3000
RoHS & Green
Call TI | NIPDAU
Level-1-260C-UNLIM
-40 to 85
ZGOI
Samples
TPS61089RNRT
ACTIVE
VQFN-HR
RNR
11
250
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
ZGOI
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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