TPS92020
www.ti.com
SLUSA54 – JULY 2010
Resonant-Switching Driver Controller for LED Lighting
Check for Samples: TPS92020
FEATURES
1
•
•
•
•
•
•
•
•
•
•
•
•
DESCRIPTION
LLC Resonant Switching Driver Controller for
Multi-String LED Lighting Applications
Half-Bridge Topology
Fixed or Variable Switching Frequency Control
Programmable Soft-Start Time
Programmable Dead Time for Best Efficiency
Easy ON/OFF Control
Overcurrent Protection
Over-Temperature Protection
Bias Voltage UVLO and OVP
Integrated Gate Driver With 0.4-A Source and
0.8-A Sink Capability
Operating Temperature Range: –40°C to 125°C
SOIC 8-Pin Package
The TPS92020 is a high-performance resonantswitching LED driver controller. It is designed for use
in higher power LED lighting systems. The TPS92020
uses resonant switching in an LLC topology to
achieve a very high efficiency compared to traditional
half-bridge converters.
The programmable dead time enables zero-voltage
switching with minimum magnetizing current,
maximizing system efficiency across a variety of
applications.
The TPS92020 can operate in two switching
frequency modes. Fixed frequency allows for simple
design when the load current is constant while
variable switching allows for optimal closed-loop
control for loads with varying currents. The internal
oscillator supports the switching frequencies from 30
kHz to 350 kHz. This high-accuracy oscillator realizes
the minimum switching frequency limiting with 4%
tolerance, allowing the designer to avoid over-design
of the power stage and, thus, further reducing overall
system cost.
APPLICATIONS
•
•
•
•
•
•
•
•
•
Commercial / Industrial LED Lighting Drivers
High Bay LED Lighting
Low Bay LED Lighting
Street LED Lighting
Area LED Lighting
Stadium LED Lighting
LED Wall Washing
LED DTV and Monitor Back-lighting
Electronic Lighting Ballasts
UCC28810
Bias
TPS92020
1
2
3
4
VSENSE
VDD
EAOUT
GDRV
VINS
GND
8
3
OC VCC
7
2
RT
6
1
DT
ISENSE
TZE
4
5
7
GD1
8
GND
6
GD2
5
SS
UDG-10043
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2010, Texas Instruments Incorporated
TPS92020
SLUSA54 – JULY 2010
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
DESCRIPTION (CONTINUED)
The programmable soft-start timer maximizes design flexibility demanded by the varied requirements of end
equipments utilizing a half-bridge topology. The TPS92020 incorporates a 0.4-A source and 0.8-A sink for driving
a low-cost gate driver transformer, delivering complete system protection functions including overcurrent, UVLO,
bias supply OVP and OTP.
Table 1. PACKAGE INFORMATION (1)
(1)
DEVICE
PACKAGE
OPERATING FREQUENCY
OPERATING
TEMPERATURE
TPS92020D
8-Pin SOIC
Variable
-40°C to 125°C
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or visit the
device product folder on ti.com.
ABSOLUTE MAXIMUM RATINGS (1)
(2) (3) (4)
over operating free-air temperature range (unless otherwise noted)
VALUE
MIN
VCC
Voltage range
Gate drive current – continuous
Current range
UNITS
MAX
22
GD1, GD2
-0.5 VVCC + 0.5
GD1, GD2
± 25
RT
–5
DT
-0.7
Operating junction temperature
TJ
−40
125
Storage temperature
Tstg
−65
150
Human Body Model (HBM)
Electrostatic Discharge
2,000
Charged Device Model (CDM)
500
Lead temperature (10 seconds)
(1)
(2)
(3)
(4)
V
mA
°C
V
260
These are stress limits. Stress beyond these limits may cause permanent damage to the device. Functional operation of the device at
these or any conditions beyond those indicated under RECOMMENDED OPERATING CONDITIONS is not implied. Exposure to
absolute maximum rated conditions for extended periods of time may affect device reliability.
All voltages are with respect to GND.
All currents are positive into the terminal, negative out of the terminal.
In normal use, terminals GD1 and GD2 are connected to an external gate driver and are internally limited in output current.
DISSIPATION RATINGS
PACKAGE
THERMAL IMPEDANCE,
JUNCTION-TO-AMBIENT
8-Pin SOIC
(1)
(2)
2
150°C/watt
(1) (2)
T A = 25°C POWER RATING
667 mW
(1)
TA = 85°C POWER RATING
267 mW
(1)
Thermal resistance is a strong function of board construction and layout. Air flow will reduce thermal resistance. This number is only a
general guide.
Thermal resistance calculated with a low-K methodology.
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RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
MIN
TYP
MAX
UNIT
VVCC
VCC input voltage from a low-impedance source
11.5
18.0
RRT
RT resistor
1
8.666
RDT
DT resistor
3.3
39
CSS
SS capacitor
0.01
1
V
kΩ
mF
ELECTRICAL CHARACTERISTICS
over operating free-air temperature range, −40°C < TA < 125°C, TJ = TA, VVCC = 12 V, GND = 0 V, RRT = 4.7 kΩ, RDT = 16.9
kΩ, CVCC = 1 mF, (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
BIAS SUPPLY (VCC)
1
1.5
2.5
5
7.5
100
400
Measured at VCC rising
9.9
10.5
11.1
UVLO turn-off threshold
Measured at VCC falling
8.9
9.5
10.1
UVLO hysteresis
Measured at VCC
0.7
1
1.3
OVP turn-off threshold
Measured at VCC rising
18
20
22
OVP turn-on threshold
Measured at VCC falling
16
18
20
OVP hysteresis
Measured at VCC
1.5
2
2.5
Dead time
RDT = 16.9 kΩ
390
420
450
fSW(min)
Minimum switching frequency at GD1,
GD2
-40°C ≤ TA ≤ 125°C
40.04
41.70
43.36
-20°C ≤ TA ≤ 105°C
40.45
41.70
42.95
KICO
Switching frequency gain/I (RT)
RRT = 4.7 kΩ, IRT = 0 to 1 mA
60
80
100
Hz/mA
t
GD1, GD2 on- time mismatching
50
ns
fSW(bm)
Switching frequency starting burst
mode
VSS = 5 V
300
350
400
Switching frequency to come out of
burst mode
VSS = 5 V
280
330
380
-40°C ≤ TA ≤ 125°C
122
142.5
162
-20°C ≤ TA ≤ 105°C
125
142.5
160
VUVLO
VOVP
VCC current, disabled
SS = 0 V
VCC current, enabled
SS = 5 V, CGD1 = CGD2 = 1 nF
VCC current, UVLO
VCC = 9 V
UVLO turn-on threshold
mA
mA
V
DEAD TIME (DT)
tDT
ns
OSCILLATOR
fSW(start)
Switching frequency at soft start
-50
kHz
kHz
EXTERNAL DISABLE/SOFT START
ISS
Enable threshold
Measure at SS rising
1.1
1.2
1.3
Disable threshold
Measured at SS falling
0.85
1
1.1
Disable hysteresis
Measured at SS
0.15
Disable prop. delay
Measured between SS (falling)
and GD2 (falling)
250
500
750
Source current on SS pin
VSS = 0.5 V
-225
-175
-125
Source current on SS pin
VSS = 1.35 V
-5.5
-5
-4.5
0.35
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V
ns
mA
3
TPS92020
SLUSA54 – JULY 2010
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ELECTRICAL CHARACTERISTICS (continued)
over operating free-air temperature range, −40°C < TA < 125°C, TJ = TA, VVCC = 12 V, GND = 0 V, RRT = 4.7 kΩ, RDT = 16.9
kΩ, CVCC = 1 mF, (unless otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
PEAK CURRENT LIMIT
VOC1(off)
Level 1 over current threshold – VOC
rising
0.9
1
1.1
VOC2(off)
Level 2 overcurrent latch threshold –
VOC rising
1.8
2.0
2.2
VOC1(on)
Level 1 over current threshold – VOC
falling
0.5
0.6
0.7
tdOC
Propagation delay
60
200
500
ns
IOC
OC bias current
VOC = 0.8 V
200
nA
GD1, GD2 output voltage high
IGD1 = −20 mA, IGD2 = −20 mA
GD1, GD2 on-resistance high
IGD1 = −20 mA, IGD2 = −20 mA
GD1, GD2 output voltage low
GD1, GD2 on-resistance low
tRISE
Rise time GDx
tFALL
-200
V
GATE DRIVE
9
11
V
12
30
Ω
IGD1 = −20 mA, IGD2 = 20 mA
0.08
0.2
V
IGD1 = −20 mA, IGD2 = 20 mA
4
10
Ω
VVCC rising from 1 V to 9 V,
CLOAD = 1 nF
18
35
Fall time GDx
VVCC falling from 9 V to 1 V,
CLOAD = 1 nF
12
25
GD1, GD2 output voltage during UVLO
VVCC = 6 V, IGD1 = 1.2 mA,
IGD2 = 1.2 mA
ns
0.5
1.75
V
THERMAL SHUTDOWN
TSD
4
Thermal shutdown threshold
160
Thermal shutdown recovery threshold
140
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°C
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DEVICE INFORMATION
TPS92020 (Top View)
DT
1
8
GD1
RT
2
7
VCC
OC
3
6
GND
SS
4
5
GD2
TERMINAL FUNCTIONS
TERMINAL
NAME
NO.
DESCRIPTION
I/O
Sets the dead time of high-side and low-side switch driving signals. Connect a resistor to ground. With internal
2.25-V voltage reference, the current flowing through the resistor sets the dead time. To prevent shoot through
when this pin is accidentally short to ground, the minimum dead time is set to 120 ns. Any dead time setting
less than 120 ns defaults to 120-ns dead time.
DT
1
I
GD1
8
O
GD2
5
O
High-side and low-side switch gate driver. Connect gate driver transformer primary side to these two pins to
drive the half bridge.
GND
6
-
Ground.
OC
3
I
Overcurrent protection. When the voltage on this pin is above 1 V, gate driver signals are actively pulled low.
After the voltage falls below 0.6 V, the gate driver signal recovers with soft start. When OC pin voltage is
above 2 V, the device is latched off. Bringing VCC below UVLO level resets the overcurrent latch off.
RT
2
I
The current flowing out of this pin sets the frequency of the gate driver signals. Connect the opto-coupler
collector to this pin to control the switching frequency for regulation purpose. Parallel a resistor to ground to
set the minimum current flowing out of the pin and set the minimum switching frequency. To set the maximum
switching frequency limiting, place a resistor in series with the opto-coupler transistor. This resistor sets the
maximum current flowing out of the pin and limits the maximum switching frequency.
SS
4
I
Soft-start. This pin sets the soft-start time of the system. Connect a capacitor to ground. Pulling this pin below
1 V disables the device to allow easy ON/OFF control. The soft-start function is enabled after all fault
conditions, including bias supply OV, UVLO, overcurrent protection and over-temperature protection.
VCC
7
-
Bias supply. Connect this pin to a power supply less than 20 V. Place a 1-mF capacitor in parallel to ground to
filter out noise.
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BLOCK DIAGRAM
2.25V
DT
T
J
160oC/140oC
+
TSD
Thermal
ShutDown
1
OV
+
10.5V
9.5V
Dead time
generator
RDT
20V
18V
UVLO
+
7 VCC
2.5V
Feed
back
RT
2
Ic
8 GD1
OSC
Vss
UVLO
OV
Q
OC
TSD
SET
VCC
D
5 GD2
FAULT
Q
CLR
5uA
6 GND
GD_Stop
6V
170uA
OC
SS
3
+
OC
1V
Vss
4
Css
OC_latch
+
+
2V
1.2V/1V
FAULT
S
R
6
SET
CLR
Q
Q
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TYPICAL CHARACTERISTICS
At VVCC = 12 V, RRT = 4.7 kΩ, RDT = 16.9 kΩ, VSS = 5 V, VOC = 0 V; all voltages are with respect to GND, TJ = TA
= 25°C, unless otherwise noted.
1.0
350
0.9
300
Switching Frequency (kHz)
Bias Supply Current (mA)
0.8
0.7
0.6
0.5
0.4
0.3
250
200
150
100
0.2
–40 °C
50
0.1
25 °C
VOC = OPEN
125 °C
0
0
6
7
8
9
10
11
12
13
14
0
0.5
Bias Supply Voltage (V)
1.0 1.5
2.0
2.5
3.0 3.5
4.0 4.5
5.0
Timing Resistance Current (mA)
Figure 1. Bias Supply Current vs. Bias Supply Voltage
Figure 2. Switching Frequency vs. Timing Resistance
1000
1000
–40 °C
900
900
25 °C
125 °C
800
800
700
Dead Time (ns)
Dead Time (ns)
700
600
500
400
600
500
400
300
300
200
200
100
100
0
0
–40 °C
25 °C
0
100
200
300
400
500
600
700
125 °C
0
5
Dead Time Current (mA)
10
15
20
25
30
35
40
45
Dead Time Resistance (kW)
Figure 3. Dead Time vs. Dead Time Current
Figure 4. Dead Time vs. Dead Time Resistance
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TYPICAL CHARACTERISTICS (continued)
0.8
VCC = 15 V
12
1.2
12
0.6
10
1.0
10
0.5
8
0.8
8
0.4
6
0.6
6
0.3
4
0.4
4
0.2
2
Gate Drive Voltage (V)
14
Gate Drive Current (A)
1.4
14
Gate Drive Voltage (V)
16
1.6
VCC = 15 V
Gate Drive Voltage
0.2
2
0
0
–0.2
600
–2
Gate Drive Voltage
Gate Drive Current
0.7
Gate Drive Current (A)
16
0.1
Gate Drive Current
0
–2
0
100
200
300
400
500
0
0
200
400
Time (ns)
600
800
–0.1
1000
Time (ns)
Figure 5. Gate Drive Voltage vs. Gate Drive Current vs.
Time
Figure 6. Gate Drive Voltage vs. Gate Drive Current vs.
Time
12.0
300
UVLO-On Threshold (VCC Rising)
UVLO-Off Threshold (VCC Falling)
11.5
UVLO Threshold Voltage (V)
Propagation Delay Time (ns)
250
200
150
100
11.0
10.5
10.0
9.5
9.0
50
8.5
0
–60 –40 –20
8
0
20
40
60
80
100 120 140
8.0
–60 –40 –20
0
20
40
60
80
100 120 140
Junction Temperature (°C)
Junction Temperature (°C)
Figure 7. Propagation Delay Time vs. Temperature
Figure 8. UVLO Threshold Voltage vs. Temperature
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TYPICAL CHARACTERISTICS (continued)
22.0
2.4
OVP-Off Threshold (VCC Rising)
OVP-On Threshold (VCC Falling)
21.5
2.2
Overcurrent Threshold Voltage (V)
Overvoltage Threshold (V)
21.0
20.5
20.0
19.5
19.0
18.5
18.0
17.5
17.0
2.0
1.8
OC Off Threshold (VOC Rising)
OC On Threshold (VOC Falling)
OC Latch Threshold (VOC Rising)
1.6
1.4
1.2
1.0
0.8
0.6
16.5
16.0
–60 –40 –20
0
20
40
60
80
0.4
–60 –40 –20
100 120 140
0
20
40
60
80
100 120 140
Junction Temperature (°C)
Junction Temperature (°C)
Figure 9. Overvoltage Threshold vs. Temperature
Figure 10. Overcurrent Threshold Voltage vs. Temperature
100
90
On-Time Mismatch Time (ns)
80
70
60
50
40
30
20
10
0
0
50
100
150
200
250
300
350
Switching Frequency (kHz)
Figure 11. On-Time Mismatch vs. Switching Frequency
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APPLICATION INFORMATION
Principle of Operation
The soft-switching capability, high efficiency and long holdup time make the LLC resonant converter attractive for
many applications, such as digital TV, ac/dc adapters and computer power supplies. Figure 12 shows the
schematic of the LLC resonant converter.
The LLC resonant converter is based on the series resonant converter (SRC). By using the transformer
magnetizing inductor, zero-voltage switching can be achieved over a wide range of input voltage and load. As a
result of multiple resonances, zero-voltage switching can be maintained even when the switching frequency is
higher or lower than resonant frequency. This simplifies the converter design to avoid the zero-current switching
region, which can lead to system damage. The converter achieves the best efficiency when operated close to its
resonant frequency at a nominal input voltage. As the switching frequency is lowered the voltage gain is
significantly increased. This allows the converter to maintain regulation when the input voltage falls low. These
features make the converter ideally suited to operate from the output of a high-voltage boost PFC pre-regulator,
allowing it to hold up through brief periods of ac line-voltage dropout.
Due to the nature of resonant converter, all the voltages and currents on the resonant components are
approximately sinusoidal. The gain characteristic of LLC resonant converter is analyzed based on the First
Harmonic Approximation (FHA), which means all the voltages and currents are treated as sinusoidal shape with
the frequency same as switching frequency.
According to the operation principle of the converter, the LLC resonant converter can be draw as the equivalent
circuit as shown in Figure 13.
CR
CR
LR
LR
n:1:1
VGE
LM
LM
RE
VOE
UDG-10044
UDG-10045
Figure 12. LLC Resonant Converter
Figure 13. LLC Resonant Converter Equivalent
Circuit
In this equivalent circuit, the Vge and Voe are the fundamental harmonics of the voltage generated by the half
bridge and the voltage on the transformer primary side, respectively. These voltages can be calculated through
Fourier analysis. The load resistor Re is the equivalent resistor of the load, and it can be calculated as:
æ
ö
8 ÷
2
RE = ç
´ (n ) ´ R
2
ç p ÷
è( ) ø
(1)
Based on this equivalent circuit, the converter gain at different switching frequencies can be calculated as:
æ
ö
ç
÷
ç VOUT ÷ =
ç æ VDC ö ÷
çç
÷÷
èè 2 øø
jw ´ LM ´ RE
(jw ´ LM ) + RE
jw ´ LM ´ RE
1
+
+ jw ´ LR
(jw ´ LM ) + RE jw ´ CR
where
•
10
VDC/2 is the equivalent input voltage due to the half-bridge structure
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Table 2. Circuit Definition Calculations
RESONANT
FREQUENCY
NORMALIZED GAIN
æ
ö
ç
÷
V
M = ç OUT ÷
ç æ VDC ö ÷
çç
÷÷
èè 2 øø
f0 =
NORMALIZED
FREQUENCY
QUALITY FACTOR
1
LR
2p ´ LR ´ CR
QE =
(4)
CR
RE
(5)
æ f ö
fn = ç ÷
è f0 ø
INDUCTOR RATIO
æL ö
Ln = ç M ÷
è LR ø
(6)
(7)
(3)
Following the definitions in Table 2, the converter gain at different switching frequencies can be calculated in
Equation 8.
2
M=
Ln ´ (fn )
2
Ln ´ (fn ) + (fn - 1)´ (fn + 1 + j ´ fn ´ Ln ´ Qe )
where
•
•
•
•
M is the converter voltage gain
Ln is the ratio of the magnetizing inductance to the resonant inductance
fn is the normalized switching frequency
Qe is the quality factor
(8)
Because of the FHA, Equation 8 is an approximation. When the switching frequency moves away from the
resonant frequency, the error becomes larger. However, this equation can be used as the design tool. The final
results need to be verified by the time based simulation or hardware test.
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From Equation 8, when switching frequency is equal to resonant frequency, fn = 1 and converter voltage gain is
equal to 1. Converter gain at different loads and inductor ratio conditions are shown in Figure 14 through
Figure 17.
2
2
Qe = 0.1
Qe = 0.2
Qe = 0.5
Qe = 1
Qe = 2
Qe = 5
1.5
M
1.5
M
1
0.5
0
0.1
Qe = 0.1
Qe = 0.2
Qe = 0.5
Qe = 1
Qe = 2
Qe = 5
1
0.5
0.5
1
1.5
0
0.1
2
0.5
1
fn
Figure 14. Normalized Switching Frequency
vs.Converter Voltage Gain, Ln=1
2
Qe = 0.1
Qe = 0.2
Qe = 0.5
Qe = 1
Qe = 2
Qe = 5
1.5
Qe = 0.1
Qe = 0.2
Qe = 0.5
Qe = 1
Qe = 2
Qe = 5
1.5
M
1
0.5
0
0.1
2
Figure 15. Normalized Switching Frequency
vs.Converter Voltage Gain, Ln=5
2
M
1.5
fn
1
0.5
0.5
1
1.5
2
0
0.1
fn
0.5
1
1.5
2
fn
Figure 16. Normalized Switching Frequency
vs.Converter Voltage Gain, Ln=10
Figure 17. Normalized Switching Frequency
vs.Converter Voltage Gain, Ln=20
Based on its theory of operation the LLC resonant converter is controlled through Pulse Frequency Modulation
(PFM). The output voltage is regulated by adjusting the switching frequency according to the input and output
conditions. Optimal efficiency is achieved at the nominal input voltage by setting the switching frequency close to
the resonant frequency. When the input voltage droops low the switching frequency is decreased to boost the
gain and maintain regulation.
The TPS92020 resonant half-bridge controller uses variable switching frequency control to adjust the resonant
tank impedance and regulate output voltage. This 8-pin package device integrates the critical functions for
optimizing the system performance while greatly simplifying the design and layout.
12
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Adjustable Dead Time
Resonant half-bridge converter relies on the resonant tank current at MOSFETs turn-off to achieve soft switching
and reduce switching loss. Higher turn-off current provides more energy to discharge the junction capacitor, while
it generates more turn-off loss. Smaller turn-off current reduces turn-off loss, but it requires longer time to
discharge MOSFETs junction capacitors and achieve soft switching. By choosing an appropriate dead time,
turn-off current is minimized while still maintaining zero-voltage switching, and best system performance is
realized.
In TPS92020, dead time can be adjusted through a single resistor from DT pin to ground. With internal 2.25-V
voltage reference, the current flow through the resistor sets the dead time.
tD = 20ns + RDT ´ 24ns
(9)
To prevent shoot through when the DT pin accidentally connects to ground, the two gate driver outputs limit the
dead-time to a minimum of 120-ns. Any dead-time setting less than 120-ns, defaults to the minimum 120-ns limit.
Oscillator
With variable switching frequency control, TPS92020 relies on the internal oscillator to vary the switching
frequency. The oscillator is controlled by the current flowing out of RT pin. Except during soft start, the
relationship between the gate signal frequency and the current flowing out of RT pin can be represented in
Equation 10.
1
1
» IRT ´ 83Hz mA
fSW = ´
2 æ 6ns ´ 1A ö
+
150ns
ç
÷
è IRT
ø
(10)
Since the switching frequency is proportional to the current, by limiting the maximum and minimum current
flowing out of RT pin, the minimum and maximum switching frequency of the converter could be easily limited.
As shown in Figure 18, putting a resistor from RT pin to ground limits the minimum current and putting a resistor
in series with the opto-coupler limits the maximum current.
Maximum
Frequency Limiting
TPS92020
R1
2
Minimum
Frequency Limiting
RT
R2
UDG-10046
Figure 18. Maximum and Minimum Frequency Setting for TPS92020
The frequency limiting resistor can be calculated in Equation 11 through Equation 14.
IF(max ) =
IF(min ) =
6ns
æ
ö
1
çç
÷÷ - 150ns
è (2 ´ fMAX ) ø
6ns
æ
ö
1
çç
÷÷ - 150ns
è (2 ´ fMIN ) ø
(11)
æ 1
1 ö
IF(max ) = 2.5 V ç
+
÷
è R1 R2 ø
IF(min ) =
2.5 V
R2
(14)
(13)
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Soft Start
During start up and fault recovery conditions, soft start is always implemented to prevent excessive resonant tank
current and ensure Zero-Voltage Switching (ZVS). During soft start, the switching frequency is increased. The
soft-start time can be programmed by placing a capacitor from SS pin to ground.
The soft-start pin also serves as an ON/OFF control pin of the device. By actively pulling the SS pin below 1 V,
the device is disabled. When the pull down is removed, SS pin voltage is increased because of internal charging
current. Once SS pin becomes above 1.2 V, the device starts to generated gate-driver signal and enters
soft-start mode. The time sequence of soft start is shown in Figure 19.
4V
1.2 V
VSS
Gate Driver
tSS
tSS(delay)
UDG-10047
Figure 19. Soft-Start Sequence
To prevent a long delay between the ON command and appearance of a gate driver signal, the SS pin current is
set as two different levels. When SS pin voltage is below 1.2 V, its output current is 175 mA. This high current
could charge the soft-start pin capacitor to 1.2 V in a short period of time, and reduces the time delay. This time
delay is calculated in Equation 15.
1.2 V
´ CSS
tSS(delay ) =
175 mA
(15)
The switching frequency during soft start is determined by both the current flowing out of the RT pin and the
voltage on SS pin. The switching frequency can be calculated based on the Equation 16.
fSW =
1
´
2
1
6ns ´ 1A
V
æ
ö
IRT + ç 1.81mA - VSS ÷
2.2k
W
è
ø
+ 150ns
(16)
After SS pin voltage reaches 4 V, soft-start period is finished and switching frequency becomes the same as
demanded by the RT pin current. The time used to charge SS pin from 1.2 V to 4 V is defined as soft-start time
and can be calculated in Equation 17.
2.8 V
tSS =
´ CSS
5 mA
(17)
To ensure reliable operation, the gate drivers restart with GD2 turning high. This prevents uncertainty during
system start up.
14
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Burst-Mode Operation
During light load condition, the resonant converter tends to increase its switching frequency and maintain the
output voltage regulation. However, due to ringing caused by transformer parasitic capacitor and the
rectification-diode-junction capacitors, the energy could be directly transferred to the load through these
capacitors. When this power becomes more than the load requires, output voltage become higher than the
regulation level. In this case, further increasing the switching frequency will not help the situation because energy
transfer to the load is not through the power stage itself.
To prevent output over voltage during this condition, the TPS92020 includes the burst-mode operation function.
When the control loop demands switching frequency higher than 350 kHz, the gate driver is disabled and the
power stage stops switching. When the output voltage drops, the control loop begins to demand switching
frequency less than 330 kHz, the gate driver recovers and the power stage begins to deliver power again. This
allows output voltage to be regulated.
This burst mode can be easily disabled by limiting the maximum switching frequency to less than 350 kHz. In this
way, the control loop never demands a switching frequency higher than 350 kHz and as a result, burst mode
operation does not occur.
Overcurrent Protection
To prevent power stage failure under excessive load current condition, the TPS92020 includes an overcurrent
protection function. With a dedicated OC pin, the power stage is shut down when OC pin voltage is above 1 V.
Once the OC pin voltage falls below 0.6 V, the gate driver recovers with a soft start. To enhance system safety,
the TPS92020 latches up the entire system when the OC pin voltage rises above 2 V. Bringing the VCC voltage
below the UVLO voltage level resets the device.
The current can be indirectly sensed through the voltage across resonant capacitor by using the sensing network
shown in Figure 20.
LR
From half-bridge
TR
LM
D2
To OC
CP
RP
RS
CS
D1
CR
UDG-10048
Figure 20. Current Sensing for LLC Resonant Converter
The general concept of this sensing method is that the ac voltage across the resonant capacitor is proportional to
load current.
According to the FHA model, peak voltage of the ac component on the resonant capacitor can be calculated in
Equation 18.
VCR(pk ) =
jwn ´ Ln ´ (Qe + 1)
4
´ n ´ VOUT
p
(w )2 ´ L
n
n
(18)
Therefore, the resonant capacitor voltage reaches its maximum value at the minimum switching frequency and
maximum load. According to Equation 18, the current sensing network components can be calculated. Due to the
nature of FHA, the final circuit parameters must be verified through actual hardware test.
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Table 3. Calculated Current Sensing Network Components
SYMBOL
RS
CS
RP
CP
FUNCTION
Transfer ac voltage across resonant capacitor into current source
Blocking dc voltage on resonant capacitor
Load resistor of the current source
DESIGN EQUATION
Rs
(V
=
CS =
RP =
CP =
Filter capacitor
CR(pk )MAX
2
)
2 ´ PRS(max )
(19)
10
RS ´ fMIN
(20)
RS
p
´
VCR(pk )MAX 2
(21)
10
(RP´ fMIN )
(22)
Gate Driver
Half-bridge resonant converter is controlled by the nearly 50% duty cycle variable frequency square wave
voltage. This allows the half bridge to be easily driven by the gate-driver transformer. Compared with a
half-bridge driver device, a gate-driver transformer provides a simple and reliable solution, which:
• Eliminate the need for gate driver power supply
• Enable simplified layout
• Preventing shoot through due to the transformer coupling
• No latch up
The TPS92020 integrates two-gate drivers with 0.4-A source and 0.8-A sink capability to directly drive the gate
driver transformer.
For LLC resonant converter, it is critical for the gate-driver signal to be precisely symmetrical. Otherwise, the
resonant tank operation will be symmetrical. The load current distribution will be unbalanced for the output
rectifiers, which in turn requires over design of the power stages and thermal management.
In TPS92020, the gate-driver output is precisely trimmed to have less than 50 ns mismatch. Although the
gate-driver signal is quite symmetrical, it is still recommended to insert the dc blocking capacitor in the
gate-driver transformer primary side to prevent transformer saturation during fast transients.
VCC Pin
Connect a regulated bias supply to VCC pin. When VCC becomes above 10.5 V the device is enabled and after
all fault conditions are cleared the gate driver starts with soft start. When the VCC voltage drops below 9.5 V, the
device enters UVLO protection mode and both gate drivers are actively pulled low. When VCC rises above 20 V
the device enters VCC overvoltage protection mode and the device is disabled with both gate drivers actively
pulled low. VCC over-voltage protection will recover with soft start when the VCC voltage returns below 18 V.
Over-Temperature Protection
TPS92020 continuously senses its junction temperature. When the junction temperature rises above 160°C the
device enters over-temperature protection mode with both gate drivers actively pulled low. When junction
temperature drops below 140°C, gate driver restarts with soft start.
16
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS92020D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
92020D
TPS92020DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
92020D
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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