TPS92075
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SLUSB88B – DECEMBER 2012 – REVISED JANUARY 2014
Non-Isolated, Phase Dimmable, Buck PFC LED Driver
with Digital Reference Control
Check for Samples: TPS92075
FEATURES
DESCRIPTION
•
•
•
•
•
•
•
The TPS92075 is a hybrid power factor controller
(PFC) with a built-in phase dimming decoder. The
device analyzes line cycles continuously using an
internal, low-power, digital controller for shape and
symmetry. The power converter stage generates an
analog current reference and uses it to regulate the
output current. The device uses control algorithms to
manipulate the analog reference. These algorithms
optimize dimmer compatibility, power factor and total
harmonic distortion (THD).
1
•
•
•
•
•
•
•
Controlled Reference Derived PFC
Integrated Digital Phase-Angle Decoder
Digital 50/60 Hz Synchronization
Phase-Symmetry Balancing
Constant LED current operation
Fast Start-up
Dimming Implemented Via Analog Reference
Control
Smooth Dimming Transitions
Overvoltage Protection
Feedback Short-Circuit Protection
Leading and Trailing Edge Dimmer
Compatibility
Low BOM Cost and Small PCB Footprint
Patent Pending Digital Architecture
Available in 8-Pin SOIC and 6-Pin TSOT
Using a constant off-time control, the solution
achieves low component count, high efficiency and
inherently provides variation in the switching
frequency. This variation creates an emulated spread
spectrum effect easing the converters EMI signature
and allowing a smaller input filter.
The TPS92075 also includes standard features:
current limit, overvoltage protection, thermal shutdown, and VCC undervoltage lockout, all in packages
utilizing only 6 pins.
APPLICATIONS
•
•
•
Bulb Replacement
Area Lighting
Dimmable and Non-Dimmable LED Lamps
SIMPLIFIED APPLICATION DIAGRAM
A
VIN
AC
A
TPS92075
EMI
Filter
Triac
Dimmer
ASNS COFF
GND
VCC
ISNS
GATE
OR
B
B
Buck-Boost
Buck
UDG-12144
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2012–2014, Texas Instruments Incorporated
TPS92075
SLUSB88B – DECEMBER 2012 – REVISED JANUARY 2014
www.ti.com
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
ORDERING INFORMATION (1)
TEMPERATURE RANGE (TJ)
PACKAGE (2)
PINS
–40 to 125°C
SOIC
8
–40 to 125°C
(1)
(2)
2
TSOT
6
ORDERABLE
DEVICE NUMBER
TRANSPORT
MEDIUM
QUANTITY
TPS92075D
Rail
95
TPS92075DR
Tape and Reel
2500
TPS92075DDC
Tape and Mini-Reel
1000
TPS92075DDCR
Tape and Reel
3000
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
web site at www.ti.com.
Package drawings, thermal data, and symbolization are available at www.ti.com/packaging.
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ABSOLUTE MAXIMUM RATINGS (1)
All voltages are with respect to GND, –40°C < TJ = TA < 125°C, all currents are positive into and negative out of the specified
terminal (unless otherwise noted)
VALUE
MIN
Input voltage range
Bias and ISNS
MAX
VCC
–0.3
22
ASNS, COFF
–0.3
6.0
IQ bias current (non-switching)
Gate
mA
2.5
V
–0.3
18
V
–2.5
20.5
V
–0.3
GATE - continuous
GATE - 100 ns
Internally Limited
Electrostatic discharge
Human Body Model (HBM)
2
Field Induced Charged Device Model (FICDM)
Operating junction temperature, TJ (3)
Storage temperature range, Tstg
–65
Lead temperature, soldering, 10s
(1)
(2)
(3)
V
2.5
ISNS (2) to Ground
Continuous power dissipation
UNIT
kV
750
V
160
°C
150
°C
260
°C
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
ISNS can sustain –2 V for 100 ns without damage.
Maximum junction temperature is internally limited.
THERMAL INFORMATION
TPS92075
THERMAL METRIC (1)
Junction-to-ambient thermal resistance (2)
θJA
(3)
SOIC
(D)
TSOT
(DDC)
8 PINS
6 PINS
112.3
165.5
θJCtop
Junction-to-case (top) thermal resistance
58.4
28.8
θJB
Junction-to-board thermal resistance (4)
52.5
24.6
ψJT
Junction-to-top characterization parameter (5)
12.5
0.3
ψJB
Junction-to-board characterization parameter (6)
51.9
23.8
NA
NA
θJCbot
(1)
(2)
(3)
(4)
(5)
(6)
(7)
Junction-to-case (bottom) thermal resistance
(7)
UNITS
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
The junction-to-ambient thermal resistance under natural convection is obtained in a simulation on a JEDEC-standard, high-K board, as
specified in JESD51-7, in an environment described in JESD51-2a.
The junction-to-case (top) thermal resistance is obtained by simulating a cold plate test on the package top. No specific JEDECstandard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
The junction-to-board thermal resistance is obtained by simulating in an environment with a ring cold plate fixture to control the PCB
temperature, as described in JESD51-8.
The junction-to-top characterization parameter, ψJT, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA, using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-board characterization parameter, ψJB, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA , using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-case (bottom) thermal resistance is obtained by simulating a cold plate test on the exposed (power) pad. No specific
JEDEC standard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
Spacer
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RECOMMENDED OPERATING CONDITIONS (1)
Unless otherwise noted, all voltages are with respect to GND, –40°C < TJ = TA < 125°C.
MIN
Supply input voltage range
VCC
11
Operating junction temperature
(1)
TYP
–40
MAX
UNIT
18
V
125
°C
Operating Ratings are conditions under which operation of the device is specified and do not imply assured performance limits. For
specified performance limits and associated test conditions, see the Electrical Characteristics table.
ELECTRICAL CHARACTERISTICS
Unless otherwise specified –40°C ≤ TJ = TA ≤ 125°C, VCC = 14 V, CVCC = 10 µF CGATE = 2.2 nF
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
SUPPLY VOLTAGE INPUT (VCC)
IQ
VCC quiescent current
Not switching
1.3
2.5
mA
IQ_SD
VCC low power mode current
VCC < VCC(UVLO)
120
250
µA
VVCC
Input range
VCC ≤ VCC(OVP)
18
V
VCC(OVP)
Overvoltage protection threshold
VCC > VCC(OVP)
20.0
V
VCC(UVLO)
VCC UVLO threshold
10.5
V
VCC(HYS)
VCC UVLO hysteresis
18.0
VCC rising
VCC falling
9.8
5.75
6.40
V
3.3
V
ANGLE DEMODULATION
ASNSTH-Hi
Angle detect rising threshold
0.9
1.0
1.1
V
ASNSTH-Low
Angle detect falling threshold
0.465
0.500
0.540
V
1.14
OFF-TIME CONTROL
VCOFF
OFF capacitor threshold
1.20
1.285
V
RCOFF
OFF capacitor pull-down resistance
33
60
Ω
tOFF-max
Maximum off-time
280
μs
GATE DRIVER OUTPUT (GATE)
RGATE(H)
Gate sourcing resistance
3
8
Ω
RGATE(L)
Gate sinking resistance
3
8
Ω
500
555
mV
CURRENT SENSE
VISNS
Average ISNS limit threshold
VCL
Current Limit
1.2
V
Leading edge blanking
240
ns
Current limit reset delay
280
µs
ISNS limit to GATE delay
33
ns
OFF capacitor limit to GATE delay
33
ns
tISNS
tCOFF_DLY
DAC: 63/127
445
THERMAL SHUTDOWN
TSD
Thermal limit threshold
160
°C
THYS
Thermal limit hysteresis
20
°C
4
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SLUSB88B – DECEMBER 2012 – REVISED JANUARY 2014
DEVICE INFORMATION
FUNCTIONAL BLOCK DIAGRAM
VCC
VCC
Regulator
ASNS
Filter
Internal
Regulator
VCC OVP
VCC UVLO
DAC
Logic
Thermal
Shutdown
0V to 1V (Analog)
Standby
PWM
+
ISNS
ILIM
+
240 ns
Delay
1.2 V
COFF
GATE
Control
Logic
+
1.2 V
280 Ps
Max Offtimer
GND
TPS92075
UDG-12177
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TPS92075
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SOIC (D) PACKAGE
8 PINS
(TOP VIEW)
GND
1
8
ASNS
COFF
2
7
NC
VCC
3
6
NC
GATE
4
5
ISNS
TSOT (DDC) PACKAGE
6 PINS
(TOP VIEW)
ASNS
1
6
COFF
GND
2
5
VCC
ISNS
3
4
GATE
PIN DESCRIPTIONS
PIN NUMBERS
6
NAME
SIOC
(D)
TSOT
(DDC)
I/O
ASNS
8
1
I
The phase of the TRIAC is detected through this pin and is then fed to the digital decoder.
Sensing thresholds are 1V rising and 0.5V falling – nominal.
COFF
2
6
I
Used to set the converter constant off-time. A current and capacitor connected from the output
to this pin sets the constant off-time of the switching controller.
GATE
4
4
O
Power MOSFET driver pin. This output provides the gate drive for the power switching
MOSFET.
GND
1
2
—
Circuit ground connection
ISNS
5
3
I
VCC
3
5
—
DESCRIPTION
LED current sense pin. Connect a resistor from main switching MOSFET source to GND to set
the maximum switching cycle LED current. Connect ISNS to the switching FET source.
Input voltage pin. This pin provides the power for the internal control circuitry and gate driver.
VCC undervoltage lockout has been implemented with a wide range: 10V rising, 6V falling to
ensure operation with start-up methods that allow elimination of the linear pass device. This
includes using a coupled inductor with resistive start-up.
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TYPICAL CHARACTERISTICS
1.1630
1400
1.1625
1200
VCC Input Current (µA)
COFF Voltage Threshold (V)
Unless otherwise stated, –40°C ≤ TA = TJ ≤ 125°C, VCC = 14 V, CVCC = 10 µF CGATE = 2.2 nF
1.1620
1.1615
1.1610
1.1605
1.1600
−40 −25 −10
800
600
400
200
5
20 35 50 65
Temperature (°C)
80
95
0
110 125
0
2
4
6
8
10
12
14
VCC Input Voltage (V)
16
18
20
G000
Figure 2. VCC Input Current vs Vcc Input Voltage
6.48
9.82
6.46
UVLO Threshold (V)
9.84
9.80
9.78
9.76
9.74
6.44
6.42
6.40
6.38
9.72
9.70
−40 −25 −10
VCC Rising
VCC Falling
G000
Figure 1. COFF Threshold Voltage vs Temperature
UVLO Threshold (V)
1000
5
20 35 50 65
Temperature (°C)
80
95
110 125
6.36
−40 −25 −10
5
20 35 50 65
Temperature (°C)
G000
Figure 3. Input Voltage (UVLO Rising) vs Junction
Temperature
80
95
110 125
G000
Figure 4. Input Voltage (UVLO Falling) vs Junction
Temperature
50
45
Number of Devices
40
35
30
25
20
15
10
5
0.
47 0 0
.5
20
0
ISNS Mid-Scale Voltage Range (V)
Figure 5. ISNS 0.5V Threshold Distribution
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APPLICATION INFORMATION
The TPS92075 is an AC-DC power factor correction (PFC) controller for phase-cut dimmer-compatible, LED
lighting applications. A hysteretic, peak current, constant off-time approach implements the conversion.
Rectified AC
A
A
Vcc
C3
C3
TPS92075
ASNS COFF
GND
VCC
ISNS
GATE
D2
C8
D2
B
B
Buck-Boost
Buck
Q2
R7
UDG-12180
Figure 6. Simplified TPS92075 Schematic
The TPS92075 controls the inductor current by controlling two features: (A) The peak inductor current, and (B)
The cycle off-time. The following items summarize the basics of the switch operation in this hysteretic controller.
• The main switch Q2 turns on and current ramps in the inductor.
• The Q2 current flows through the sense resistor R7. The R7 voltage is compared to a reference voltage at
ISNS. The Q2 on-time ends when the voltage on R7 is equal to a controlled reference voltage and the
inductor current has reached its set peak current level for that switching cycle.
• Q2 is turned off and a constant off-time timer begins. Voltage begins ramping on C8.
• The next cycle begins when the voltage on C8 reaches 1.2 V. This ends the constant off-time and discharges
C8.
• Capacitor C3 eliminates most of the ripple current seen in the LEDs.
iL
(A) Peak Inductor Current
'iL-PP
(B)
tOFF
tON
(constant)
Ts
0
Time
UDG-12176
Figure 7. Current Regulation Method
The TPS92075 incorporates a patent-pending control methodology to generate the reference for the conversion
stage. The controlled reference used for the comparison of the ISNS signal may be DC or another shape
depending on the mode of operation. Each mode controls the peak current level using a different methodology.
8
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SLUSB88B – DECEMBER 2012 – REVISED JANUARY 2014
Initial Start-Up
The TPS92075 is designed to achieve instant turn-on using an external linear regulator circuit. The start-up
sequence is internally controlled by a VCC under-voltage lockout (UVLO) circuit. Sufficient headroom has been
incorporated to support the use of an auxiliary winding with start-up linear, resistive or coupled capacitor start-up
methods.
VCC Bias Supply
The TPS92075 can be configured to use a linear regulator with or without the use of an auxiliary winding. Using
a linear regulator to provide VCC incurs more losses than an auxiliary winding, but has several advantages:
• allows the use of inexpensive off-the-shelf inductors as the main magnetic
• speeds start-up time under deep dimming conditions
• can reduce the size of the required VCC capacitor
• the extra current draw when dimming can improve dimming compatibility
Another consideration when selecting a bias method involves the OVP configuration. Because the feature is
enabled via the VCC pin, an auxiliary winding provides the simplest implementation of output over-voltage
protection.
A typical start-up sequence begins with VCC input voltage below the UVLO threshold and the device operating in
low-power, shut-down mode. The VCC input voltage increases to the UVLO threshold of 9.8V typical. At this point
all of the device features are enabled. The device loads the initial start-up value as the output reference and
switching begins. The device operates until the VCC level falls below the VCC(UVLO) falling threshold. (6.4V typical)
When VCC is below this threshold, the device enters low-power shut-down mode.
Angle Sense Operation
The ASNS (angle sense) pin is the only input to the digital controller. The time between the rising edge and the
falling edge of the signal determines converter functions. The pin incorporates internal analog and digital filtering
so that any transition that remains beyond the threshold for more than approximately 150 µs will cause the
device to record a change-of-state.
V
Signal at ASNS pin
ASNS Internal Signal
1.0 9
0.5 ;
tASNS1
tASNS2
TL
UDG-12179
Figure 8. Angle Sense Operation
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Controller
Basic Operation and Modes
The controller continuously monitors the line cycle period and the present conduction angle length to determine
the state of operation and configure other control features. Control algorithms use a normalized line period of 256
samples from ASNS fall to ASNS fall and a normalized converter reference control of 127 levels over a range of
0V to 1V .
The four main controller states are:
• Start-up
• Non-Dimming
• Dimming
• ASNS signal lost
With the exception of start-up, the controller can enter any of the states at any time as conditions demand.
The two primary modes of controlling the converter reference are:
• DC mode
• Ramp mode
During active dimming, a DC control reference increases or decreases depending on the input AC duty cycle
derived from the ASNS signal. The relationship follows the algorithm: (ASNS Length + Fixed Offset) = Output Set
point. When the conduction angle is long enough, the converter reference is changed to a triangular ramp to
achieve a high power factor. The ramp is generated gradually over several cycles ensuring the implementation is
undetectable. The controller maintains the ramp between the rising and falling ASNS signals.
The controller also sets DC reference levels during start-up and when the ASNS signal is lost. Active states in
the controller and controlled ranges are shown in Table 1.
Table 1. Control States and Controlled Reference Values
MODE
LINE DUTY CYCLE
Start-up
Any
CONTROLLED REFERENCE VALUE
(value / 127 ) X 1V = reference
50
> 70%, typical average
55
> 70%, typical ramp range
22 to 127
Dimming
≤ 70%
35 to 63
No ASNS
Any
42
Non-Dimming
Initial Start-up
Line Synchronization
When the device reaches the turn-on UVLO threshold, the output current reference resets to 0.393V (50/127)
and switching begins. The controller samples the line for approximately 80 ms (t1 to t2 , Figure 9) to determine
the line frequency and establish the present state of operation. After determining the line frequency, the controller
uses the information to calibrate the internal oscillator. The controller supports line frequencies from 45Hz to
65Hz. After determining frequency and duty cycle, the controller enters the appropriate control state.
10
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VREC
VASNS
Controlled Reference
VISNS(peak)
VGATE
Time
t0
t1
t2
UDG-12168
Figure 9. Line Synchronization
Non-Dimming Ramp Mode
When the conduction angle is greater than 70%, the controller begins to create a triangular ramp that is
synchronized to the line and is centered between rising and falling edges of the ASNS signal as shown in
Figure 10. The triangular shape is much easier to generate than a sine wave while maintaining a high power
factor and low THD. The edges of the ramp do not decrease completely to zero to ensure compatibility with
TRIAC dimmers that can provide conduction angles approaching 100%.
V
Rectified AC
ASNS
Controlled Reference
Internal
Time
UDG-12169
Figure 10. Controlled Reference Output, Non-dimming
When changing between dimming mode and non-dimming mode, the ramp is created over 127 line cycles (see
Figure 11) or approximately 1 second (t2 to t3 ≈ 1 second). Because the output level before and after the change
is very similar and the change very gradual, it is impossible for the user to perceive a change in output level. The
ramp morphs from a DC level to a ramp using a method that further ensures transparency to the user. Ramp
transition occurs during construction and deconstruction of the ramp and is reversed if the conduction angle
changes sufficiently during the change process. A hysteresis in angle length is also built in to the change-toramp-mode and change-from-ramp-mode transition.
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VREC
...
VASNS
...
VISNS(peak)
...
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Time
t2
t3
UDG-12178
Figure 11. Transition Stages of the Controlled Reference
Dimming Mode
When the conduction angle is reduced below the 70% threshold the output is controlled with a DC reference
level based on: (angle sense rise to fall length count) / 2 + 35, ≤ 63. The control level clamps at both the high
and low end of the range to increase TRIAC dimmer compatibility. Rather than adding passive (heat generating)
hold current or implementing other means to draw sufficient current from the TRIAC dimmer to maintain optimal
operation, the TPS92075 implements a translation that shifts output demand higher, lower in the dimming range.
The effect is that more current is drawn at low angles, eliminating the need for hold circuitry. A net reduction in
light output occurs because of the energy transfer relation. As the phase-dimmer conduction decreases, the time
during which the converter can provide output power during each cycle decreases, and a reduction in light output
follows.
Triac Asymmetry Balancing
Triacs are two silicon-controlled rectifiers (SCRs) configured so that one device conducts current in the positive
AC cycle and the other device conducts current in the negative AC cycle. It is common for the devices to have
different trigger levels and this leads to differences in conduction angle for each of the positive and negative AC
cycles. The amount of variation between each cycle varies greatly between dimmer brands, makes and models.
In all single stage TRIAC compatible dimming solutions, the ability of the converter to provide output power
depends on the length of the conduction time. If the output current demand remains constant during each cycle
and if there is a difference in TRIAC conduction angles, the result is a difference in light output for each cycle.
The TPS92075 incorporates a balancing algorithm to reduce the difference in LED current (and light output)
between cycles that have a conduction angle difference greater than 20%.
ILED
Active Hold
VRECT AC
VISNS Pk
Offset
Figure 12. LED current variation, Constant
Reference
Figure 13. LED current variation, TPS92075 with
Balancing
When the difference in conduction becomes greater than 20%, the controller begins to adjust the controller
reference line-cycle by line-cycle to balance the energy provided to the LEDs. In this example the difference in
conduction angles is 800 μs and flicker was visible with the constant reference (Figure 12). With the TPS92075
balancing feature the peaks in the LED current have been equalized and flicker cannot be seen (Figure 13).
12
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Lossless or 'Active' Hold
When used in the buck configuration, the converter enters a drop-out condition each cycle as the input AC line
drops below the LED stack voltage. When this occurs, a resonance in the input filter can be excited causing a
ring in the input current at the end of the conduction cycle. This can lead to output flicker if not controlled. One
method of eliminating this is to modify the control method to send the energy that would otherwise affect the
ringing to the output. To do this, the controller increases the output set-point at the end of each cycle after the
ASNS fall (< 0.5 V) signal is received. The increse in set point can be seen in Figure 14.
ILED
Active Hold
VRECT AC
VISNS Pk
Offset
Figure 14. TPS92075 Reference Control - Active Hold
Another benefit of the active hold is that a low impedance path is created to the LED stack. This ensures the
current demand is as high as possible for as long as possible before the converter fully enters drop-out.
Active Hold, ASNS, and Buck-Boost Topology
When using the converter in a buck-boost configuration attention must be given to the configuration of the ASNS
signal to ensure there is some added delay in the signal crossing the 0.5V threshold. Because the converter can
continue to provide energy to the output below the LED stack voltage, it is best to configure the ASNS signal to
fall when the rectified AC signal is as close to zero as possible.
Rectified AC
R9
D4
ASNS
D3
R2
C7
UDG-12171
Figure 15. Buck-Boost Angle Sense Circuit
This can be implemented by adding an additional zener and capacitor on the ASNS pin. Capacitance between
2200 pF and 4700 pF provides a good balance between allowing the ASNS signal to fall below 0.5V and
extending the ASNS time. The D4 zener allows the ASNS signal to be widened further. This component can be
the same type of zener selected for the input voltage linear supply, in many prototyping examples a 15V zener
diode is used. The buck-boost configuration tends to provide greater dimmer compatibility because of its ability to
continue to draw power below the LED stack. This increases the time the converter can provide output current
and increases the light output at a given dimmer setting. A higher light output for a given dimmer setting is an
important control technique which increases the probability that the design will remain flicker-free over its lifetime
and range of installations. This trade-off between dimming ratio, dimmer compatibility and component count
make the components a desirable addition.
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Loss of Angle Sense
When using a dimmer that can control the phase angle to very short conduction times (< 250 µs), the ASNS
signal may become so narrow that the controller cannot determine its length. When this occurs the controller
simply sets the reference to a default value 0.33V (42/127) and waits for the ASNS signal to return.
A simplified version of the TPS92075 circuit can be implemented by grounding the ASNS signal if minimum
component count and size are essential design criteria. In this configuration balancing, ramp mode and active
hold are not implemented. The output is controlled with a default, static reference of 0.394V (50/127). If used in
conjunction with an on-time clamp, good dimming and power factors (>0.9) can still be achieved.
Thermal Shutdown
The TPS92075 includes thermal shutdown protection. If the die temperature reaches approximately 160°C the
device stops switching (GATE pin low). When the die temperature cools to approximately 140°C, the device
resumes normal operation.
If thermal fold back is desired at levels below the IC thermal shut down, application circuits have been created to
implement this feature. The simplest of these is the addition of a thermistor in the off-time circuitry.
Thermal Foldback
To implement thermal foldback, adjust the resistance of an existing circuit resistor with the use of an NTC
(negative temperature coefficient) thermistor.
For example, a resistor combination creating a dominant effect when the thermistor reaches the desired
temperature and resistance can be incorporated by paralleling a thermistor and another resistor with R10 (Figure
17). This circuit option creates a shorter on-time as the temperature increases, reducing the output current. The
use of a thermistor in these types of circuit implementations is simple and saves costly added circuitry and
additional device pins.
Overvoltage Protection (OVP)
The implementation of overvoltage protection is simple and built-in if using a two-coil magnetic (coupled inductor)
to derive VCC. If the LED string is opened the auxiliary VCC rises and reaches the VCC(OVP) trip point. This action
disables and grounds the gate pin, preventing the converter from switching. The converter remains disabled until
VCC drops 0.5V after a 1 second time-out. If an inductor is used, implement other discrete circuits to disable the
converter.
Output Bulk Capacitor
The required output bulk capacitor, CBULK, stores energy during the input voltage zero crossing interval and limits
twice the line frequency ripple component flowing through the LEDs. Equation 1 describes the calculation of the
of output capacitor value.
PIN
CBULK ³
4p ´ fL ´ RLED ´ VLED ´ ILED(ripple )
where
•
•
•
RLED is the dynamic resistance of LED string
ILED(ripple) is the peak to peak LED ripple current
and fL is line frequency
(1)
RLED is found by computing the difference in LED forward voltage divided by the difference in LED current for a
given LED using the manufacturer’s VF vs. IF curve. For a rough initial estimate a typical value of 0.25Ω per LED
can be used. More detail can be found in Application Note 1656.
In typical applications, the solution size becomes a limiting factor and dictates the maximum dimensions of the
bulk capacitor. When selecting an electrolytic capacitor, manufacturer recommended de-rating factors should be
applied based on the worst case capacitor ripple current, output voltage and operating temperature to achieve
the desired operating lifetime. It should also be a consideration to provide a minimum load at the output of the
driver to discharge the capacitor after the power is switched off or during LED open circuit failures.
14
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Design Guidelines
This TPS92075 application design requires the selection of components for the power conversion stage and
angle sensing. Output inductor, sense resistor and switching frequency are the key aspects of the power stage
design. Another important consideration is the inclusion of an on-time clamp. The combination of the line voltage
going to zero each cycle and the hysteretic control method can lead to large increases in current draw at the
start and end of each cycle. The components required for the on-time clamp are very inexpensive and return
results that make their inclusion a common choice for LED driver designers. This simplified design procedure
assumes the use of an on-time clamp in the design.
UDG-12183
iL(ave)
Peak Inductor Current follows
this Controlled Reference.
Ipk(t) = VISNS(t)/RSENSE
RSENSE adjusts
the average,
peak inductor
current
The Inductance (L)
defines 'iL(P-P)
ûiL-PP = (VLED * toff )
L
Rectified
AC
Inductor
Current
Ripple
The average output current =
the average peak ± ½ the
peak to peak inductor ripple
iL(ave)= iLave(pk) ± ûiL(P-P)
2
iLave(pk) = VISNS(ave)
RSENSE
Time
tON
Time
tOFF
Figure 16. TPS92075 Output Current Control
The mode of operation that determines average continuous output current is non-dimming, during which the
reference is a triangular waveform.
The device uses the controller reference every switching cycle to set the peak current through the main switch
and sense resistor. The average value of this reference and the inductor ripple current can be used to calculate
the average output current. Another consideration is the length of time the converter is providing power to the
LEDs. A conversion factor (CF) that accounts for a lower level of power conversion at the ends of each cycle is
used to provide a more accurate sense resistor value. The lower level of power conversion in these areas also
helps to increase the power factor. For the RSENSE calculation use VISNS (ave) = 0.433V (55/127). The CF
calculation involves computing the normalized time length of the angle sense pulse using a formula shown in
Equation 3. Simplified design expressions are provided below. For a more comprehensive approach refer to the
TPS92075 Design Spreadsheet.
To calculate RSENSE, use Equation 2.
æ
ö
ç V
÷
ISNS
ave
( ) ÷
RSENSE = ç
´ CF
ç
DiL(P-P ) ÷
ç ILED +
÷
2
è
ø
(2)
To calculate the conversion factor, use Equation 3.
æ
æ VLED
ç sin-1 ç
ç 2´V
ç
RMS
è
CF = 1 - ç
90
ç
ç
è
ö
ö
÷
÷
÷ 3÷
ø´
÷
2÷
÷
ø
(3)
To calculate inductance ripple, use Equation 4.
´t
æV
ö
DiL(P-P ) = ç LED OFF ÷
L
è
ø
(4)
To calculate the constant off-time, use Equation 5
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æ æ æ 1.2
ööö
tOFF = ç ln ç - ç
- 1÷ ÷ ÷ ´ (-CTOFF ´ RCOFF )
÷
ç ç è VVCC
ø ø ÷ø
è è
(5)
To calculate the average switching frequency, use Equation 6.
æ
ö
1
fSW = ç
÷
ç tOFF + (tOFF ´ CF ) ÷
è
ø
(6)
On-Time Clamp
The use of an on-time clamp (Figure 17) provides a soft-start and soft-stop action to the conversion each line
cycle. It also adds a means to control the energy in these conversion areas to optimize dimming performance.
For example, cutting the energy conversion in these areas in half maintains strong current pull through these
critical TRIAC regions, but is not high enough to excite circuit resonances.
TPS92075
ISNS
GATE
R10
R11
D5b
D5a
ISNS
ISNS
R8
C10
UDG-12172
Figure 17. On-time Clamp Circuitry
The circuit uses the gate drive output to generate a ramp. The ramp increases at a rate to reach the current
sense trip point at the desired maximum conduction time. The gate signal, resistor R10 and capacitor C10 create
the ramp. Diode D5b resets the ramp for each switching cycle. Resistor R11 provides an impedance so this
signal can override ISNS.
In the regions at the start and end of a line cycle the current sense reference is controlled to 0.173V (22/127). To
select an R-C to reach this point in the desired time use Equation 7. A good starting estimate for the maximum
on-time clamp is ~tOFF/2 . For example, choosing 33 nF as the value of capacitor C10, and assuming VGATE ≈
VCC, R10 (Rton(max)) is calculated in Equation 7.
tOFF
R ton(max ) =
é æ æ 0.173
ö öù
- 1÷ ÷ ú ´ -Cton(max )
2 ´ êln ç - ç
ç
÷
êë è è VGATE
ø ø úû
(7)
16
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Angle Sense Circuitry and Minimum ASNS Signal Length
If implementing a buck converter, select the divider so the falling 0.5V ASNS threshold is reached when the
rectified AC voltage is at the LED stack voltage. For example, if the LED stack is 20V and the top resistor is 400
kΩ, the bottom resistor should be 10.25 kΩ to provide a falling ASNS signal at 0.5V when the rectified AC
reaches 20V. A 20V ASNS falling signal will mean a 40V ASNS rising threshold because of the 2:1 hysteresis.
This will provide an ASNS signal length of ~7.4 ms, adequate to activate the ramp mode when not connected to
a dimmer. This buck configuration and ASNS divider will activate the hold feature each time the rectified AC
reaches the LED stack voltage. This method is shown in Figure 18. Regardless of the ASNS connection method
used, the divider must ensure an adequate angle sense length (tASNS > 5.9 ms) when non-dimming to activate
the creation of the ramp if this is desired. For example, if a straight resistor divider (Figure 18) is implemented
and the design LED stack is more than 42V, the ASNS conduction time may not be adequate to activate the use
of the ramp reference.
LED(+)
LED(-)
LED(–)
ASNS
ASNS
ASNS
UDG-12173
Figure 18. Angle Sense for Low
Voltage Buck Applications
UDG-12175
UDG-12174
Figure 19. Angle Sense for Buck
Applications up to 65V
Figure 20. Angle Sense for BuckBoost Applications
For LED stack voltages between 3V and 65V, use an alternate method that senses from LED(–). Because
LED(–) reaches ground each line cycle, the absolute ASNS comparison limits of 0.5V and 1V can be used,
providing extra conduction time for the ASNS signal as shown in Figure 19. Beyond a ~65V LED stack, alternate
ASNS methods utilizing a bridge tap can be used. For buck-boost applications, implement the circuit shown in
Figure 20.
A capacitor on the ASNS pin may be required, depending on operating conditions.
EMI Filtering: AC versus DC side of the rectifier bridge
The TPS92075 requires a minimal amount of EMI filtering to pass conducted and radiated emissions levels to
comply with agency requirements. Applications have been tested with the filter on the AC or DC side of the diode
bridge and have obtained passing results. The use of an R-C snubber to damp filter resonances and optimize
TRIAC compatibility is strongly recommended. The EMI filter design involves optimizing several factors and
design considerations, including:
• the use of ‘X’ versus non-X rated filter capacitors
• the use of ceramic versus film capacitors
• component rating requirements when on the AC or DC side of the diode bridge
• filtering on the AC or DC side of the bridge and the effect on the TRIAC firing angle and dimming range
• snubber time constant and position in the design schematic
• filter design choices and audible noise
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Application Circuits
+
TPS92075
ASNS COFF
Triac
Dimmer
GND
VCC
ISNS
GATE
UDG-12181
Figure 21. TPS92075 Application Circuit for Buck Topology with AC Side Filter
+
TPS92075
ASNS COFF
GND
VCC
ISNS
GATE
Triac
Dimmer
ISNS
ISNS
UDG-12182
Figure 22. TPS92075 Application Circuit for Buck-Boost Topology with DC Side Filter
18
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+
TPS92075
ASNS COFF
GND
VCC
ISNS
GATE
Triac
Dimmer
ISNS
ISNS
UDG-12184
Figure 23. TPS92075 Application Circuit for Buck-Boost with Resistive Start-up and AUX Supply
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REVISION HISTORY
Changes from Revision A (JANUARY 2013) to Revision B
•
Page
Changed TSOT package availability status in FEATURES section ..................................................................................... 1
Changes from Original (DECEMBER 2012) to Revision A
Page
•
Changed title of Figure 7 ...................................................................................................................................................... 8
•
Changed Figure 17 to correct resistor position ................................................................................................................... 16
•
Changed Figure 22 to correct resistor position ................................................................................................................... 18
•
Changed Figure 23 to correct resistor position ................................................................................................................... 19
20
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
TPS92075D/NOPB
ACTIVE
SOIC
D
8
95
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
T92075
TPS92075DDC/NOPB
ACTIVE
SOT-23-THIN
DDC
6
1000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
SN8B
TPS92075DDCR/NOPB
ACTIVE
SOT-23-THIN
DDC
6
3000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
SN8B
TPS92075DR/NOPB
ACTIVE
SOIC
D
8
2500
RoHS & Green
SN
Level-1-260C-UNLIM
-40 to 125
T92075
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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