UC1625
UC2625
UC3625
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
Brushless DC Motor Controller
Check for Samples: UC1625, UC2625, UC3625
FEATURES
DESCRIPTION
•
The UC3625 family of motorcontroller devices
integrate most of the functions required for highperformance brushless dc-motor control into one
package. When coupled with external power
MOSFETs or Darlingtons, these devices perform
fixed-frequency PWM motor control in either voltage
or current mode while implementing closed-loop
speed control and braking with smart noise rejection,
safe direction reversal, and cross–conduction
protection.
1
•
•
•
•
•
•
•
•
•
•
Drives Power MOSFETs or Power Darlingtons
Directly
50-V Open Collector High-Side Drivers
Latched Soft Start
High-Speed Current-Sense Amplifier with Ideal
Diode
Pulse-by-Pulse and Average Current Sensing
Over-Voltage and Under-Voltage Protection
Direction Latch for Safe Direction Reversal
Tachometer
Trimmed Reference Sources 30 mA
Programmable Cross-Conduction Protection
Two-Quadrant and Four-Quadrant Operation
Although specified for operation from power supplies
between 10 V and 18 V, the UC1625 controls higher
voltage power devices with external level-shifting
components. The UC1625 contains fast, high-current
push-pull drivers for low-side power devices and 50-V
open-collector outputs for high-side power devices or
level-shifting circuitry.
The UC1625 is characterized for operation over the
military temperature range of –55°C to +125°C, while
the UC2625 is characterized from –40°C to +105°C
and the UC3625 is characterized from 0°C to +70°C.
(Note: ESD Protection to 2 kV)
TYPICAL APPLICATION
VREF
+15V
+5V TO HALL
SENSORS
100nF
100nF
20mF
QUAD
ROSC
33kW
10kW
2
2N3904
10W
19
11
22
DIR
3kW
1k
17
1
18
UC3625
28
100nF
14
27
13
25
2200pF
COSC
2N3906
IRF9350
16
6
4kW
TO
MOTOR
TO OTHER
CHANNELS
10W
15
REQUIRED
FOR BRAKE
AND FAST
REVERSE
TO OTHER
CHANNELS
12
BRAKE
+
100mF
3kW
+
20mF
10kW
3kW
VMOTOR
IRF532
20
21
3nF
CT
26
68kW
RT
3
24
23
8
9
10
4
5
7
100nF
5nF
FROM
HALL
SENSORS
100nF
2nF
51kW
2nF
2nF
VREF
10kW
240W
5nF
240W
0.02
W
RS
REQUIRED
FOR
AVERAGE
CURRENT
SENSING
0.02
W
RD
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2003–2013, Texas Instruments Incorporated
UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
www.ti.com
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
VALUE
VCC
Pwr
VCC
UNIT
20
Supply voltage
20
PWM IN
–0.3 to +6
E/A IN(+), E/A IN(–)
–0.3 to +12
ISENSE1, ISENSE2
–1.3 to +6
OV–COAST, DIR, SPEED-IN, SSTART, QUAD SEL
–0.3 to +8
H1, H2, H3
–0.3 to +12
PU Output Voltage
–0.3 to +50
PU
+200 continuous
PD
±200 continuous
E/A
±10
ISENSE
Output Current
–10
Tach
Out
Tstg
(1)
2
mA
±10
VREF
TJ
V
–50 continuous
Operating Temperature Range UC1625
–55 to +125
Operating Temperature Range UC2625
–40 to +105
Operating Temperature Range UC3625
0 to +70
Storage Temperature
–65 to +150
°C
°C
Currents are positive into and negative out of the specified terminal.
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UC3625
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
Figure 1. CONNECTION DIAGRAM
A.
E/A IN(+)
1
28
E/A IN(-)
VREF
2
27
E/A OUT
ISENSE
3
26
PWM IN
ISENSE1
4
25
RC-OSC
ISENSE2
5
24
SSTART
DIR
6
23
OV-COAST
SPEED-IN
7
22
QUAD SEL
H1
8
21
RC-BRAKE
H2
9
20
TACH-OUT
H3
10
19
VCC
PWR VCC
11
18
PUA
PDC
12
17
PUB
PDB
13
16
PUC
PDA
14
15
GND
This pinout applies to the SOIC (DW), PLCC (Q), and LCC (L) packages (for example: pin 22 has the same function
on all packages.)
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Product Folder Links: UC1625 UC2625 UC3625
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UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
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ELECTRICAL CHARACTERISTICS
Unless otherwise stated, these specifications apply for: TA = 25°C; Pwr VCC = VCC = 12 V; ROSC = 20 kΩ to VREF; COSC = 2 nF;
RTACH = 33 kΩ; CTACH = 10 nF; and all outputs unloaded. TA = TJ.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Overall
Supply current
14.5
30.0
8.65
8.95
9.45
7.75
8.05
8.55
OV-COAST inhibit threshold
1.65
1.75
1.85
OV-COAST restart threshold
1.55
1.65
1.75
0.05
0.10
0.15
-10
-1
0
0.8
1.0
1.2
1.6
1.9
2.0
-400
-250
-120
0.8
1.4
2.0
VCC turn-on threshold
Over operating range
VCC turn-off threshold
mA
V
Overvoltage/Coast
Over operating range
OV-COAST hysteresis
OV-COAST input current
V
μA
Logic Inputs
H1, H2, H3 low threshold
Over operating range
H1, H2, H3 high threshold
H1, H2, H3 input current
Over operating range, to 0 V
QUAD SEL, dir thresholds
Over operating range
QUAD SEL hysteresis
70
DIR hysteresis
V
μA
V
mV
0.6
V
QUAD SEL input current
-30
50
150
DIR input current
-30
-1
30
μA
PWM Amp/Comparator
E/A IN(+), E/A IN(–) input current
To 2.5 V
-5.0
-0.1
5.0
PWM IN input current
To 2.5 V
0
3
30
Error amp input offset
0 V < VCOMMON-MODE < 3 V
-10
Error amp voltage gain
70
E/A OUT range
SSTART
10
90
0.25
μA
mV
dB
3.50
V
Pull-up current
To 0 V
-16
-10
-5
μA
Discharge current
To 2.5 V
0.1
0.4
3.0
mA
0.1
0.2
0.3
V
1.75
1.95
2.15
V/V
2.4
2.5
2.65
0.14
0.20
0.26
0.26
0.30
0.36
-850
-320
0
±2
±12
Restart threshold
Current Amp
Gain
ISENSE1 = 0.3 V, ISENSE2 = 0.5 V to
0.7 V
Level shift
ISENSE1 = 0.3 V, ISENSE2 = 0.3 V
Peak current threshold
Over current threshold
ISENSE1, ISENSE2 input current
ISENSE1, ISENSE2 offset current
ISENSE1 = 0 V, force ISENSE2
To 0 V
Range ISENSE1, ISENSE2
-1
2
V
μA
V
Tachometer/Brake
TACH-OUT high level
TACH-OUT low level
4.7
Over operating range, 10 kΩ to 2.5
V
170
Over operating range
RC-BRAKE input current
To 0 V
Threshold to brake, RC-brake
Over operating range
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280
-4.0
-1.9
0.8
1.0
μs
mA
1.2
0.09
Over operating range
SPEED-IN input current
4
220
V
0.1%
Brake hysteresis, RC-brake
SPEED-IN threshold
5.3
0.2
On time
On time change with temp
5
V
220
257
290
mV
-30
-5
30
μA
Copyright © 2003–2013, Texas Instruments Incorporated
Product Folder Links: UC1625 UC2625 UC3625
UC1625
UC2625
UC3625
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
ELECTRICAL CHARACTERISTICS (continued)
Unless otherwise stated, these specifications apply for: TA = 25°C; Pwr VCC = VCC = 12 V; ROSC = 20 kΩ to VREF; COSC = 2 nF;
RTACH = 33 kΩ; CTACH = 10 nF; and all outputs unloaded. TA = TJ.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
Low-Side Drivers
Voh, –1 mA, down from VCC
1.60
2.1
V Voh, –50 mA, down from VCC
1.75
2.2
0.05
0.4
0.36
0.8
Over operating range
Vol, 1 mA
Vol, 50 mA
Rise/fall time
10% to 90% slew time, into 1 nF
50
V
ns
High-Side Drivers
Vol, 1 mA
Over operating range
Vol, 50 mA
Leakage current
Output voltage = 50 V
Fall time
10% to 90% slew time, 50 mA load
0.1
0.4
1.0
1.8
25
50
V
μA
ns
Oscillator
40
Frequency
Over operating range
50
35
60
65
kHz
Reference
4.9
5.0
5.1
Over operating range
4.7
5.0
5.3
Load regulation
0 mA to –20 mA load
-40
-5
Line regulation
10 V to 18 V VCC
-10
-1
10
Short circuit current
Over operating range
50
100
150
Output voltage
V
mV
mA
Miscellaneous
Output turn-on delay
1
Output turn-off delay
1
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μs
5
UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
www.ti.com
BLOCK DIAGRAM
Quad Sel
22
RC-Osc
25
PWM In
26
E/A Out
27
E/A In(+)
1
E/A In (–)
28
SSTART
24
ISENSE
3
S
OSC
VCC
19
OV-Coast
23
Speed-In
R
0.2V
2.5V
250W
9V
16
PUC
11
Pwr Vcc
14
PDA
13
PDB
12
PDC
15
GND
20
Tach-Out
PWM
CLOCK
6
DIRECTION
LATCH
7
PWM CLOCK
DIR
8
D
+5V
Q
H1
Q
H2
COAST
CHOP
QUAD
CROSS
CONDUCTION
PROTECTION
LATCHES
L
9
D
+5V
H3
PUB
S
3.1V
+5V
H2
17
Q1
Q
0.25V
H1
PUA
10mA
1.75V
Dir
18
2.9V
2X
5
VREF
R
4
ISENSE2
2
PWM CLOCK
ABS VALUE
ISENSE1
5V
REFERENCE
Q
DECODER
L
10
D
L
Q
H3
BRAKE
EDGE
DETECT
+5V
2k
RC-Brake
21
ONE
SHOT
1V
6
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UC2625
UC3625
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
DEVICE INFORMATION
Terminal Functions
TERMINAL
NAME
DIR, SPEED-IN
E/A IN(+), E/A IN(–), E/A
OUT, PWM IN
GND
H1, H2, H3
ISENSE1, ISENSE2,
ISENSE
OV-COAST
NO.
I/O
DESCRIPTION
6, 7
The position decoder logic translates the Hall signals and the DIR signal to the
correct driver signals (PUs and PDs). To prevent output stage damage,the signal
on DIR is first loaded into a direction latch, then shifted through a two-bit register.
As long as SPEED-IN is less than 250 mV, the direction latch is transparent. When
SPEED-IN is higher than 250 mV, the direction latch inhibits all changes
indirection. SPEED-IN can be connected to TACH-OUT through a filter, so that the
direction latch is only transparent when the motor is spinning slowly, and has too
little stored energy to damage power devices.
Additional circuitry detects when the input and output of the direction latch are
different, or when the input and output of the shift register are different, and inhibits
all output drives during that time. This can be used to allow the motor to coast to a
safe speed before reversing.
The shift register ensures that direction can not be changed instantaneously. The
register is clocked by the PWM oscillator, so the delay between direction changes
is always going to be between one and two oscillator periods. At 40 kHz, this
corresponds to a delay of between 25 µs and 50 µs. Regardless of output stage,
25-µs deadtime should be adequate to ensure no overlap cross-conduction.
Toggling DIR causes an output pulse on TACH-OUT regardless of motor speed.
1, 28, 27,
26
E/A IN(+) and E/A IN(–) are not internally committed to allow for a wide variety of
uses. They can be connected to the ISENSE, to TACH-OUT through a filter, to an
external command voltage, to a D/A converter for computer control, or to another
op amp for more elegant feedback loops. The error amplifier is compensated for
unity gain stability, so E/A OUT can be tied to E/A IN(–) for feedback and major
loop compensation.
E/A OUT and PWM In drive the PWM comparator. For voltage-mode PWM
systems, PWM In can be connected to RC-OSC. The PWM comparator clears the
PWM latch, commanding the outputs to chop.
The error amplifier can be biased off by connecting E/A IN(–) to a higher voltage
than /EA IN(+). When biased off, E/A OUT appears to the application as a resistor
to ground. E/A OUT can then be driven by an external amplifier.
All thresholds and outputs are referred to the GND pin except for the PD and PU
outputs.
15
8, 9, 10
The three shaft position sensor inputs consist of hysteresis comparators with input
pull-up resistors. Logic thresholds meet TTL specifications and can be driven by 5V CMOS, 12-V CMOS, NMOS, or open-collectors.
Connect these inputs to motor shaft position sensors that are positioned 120
electrical degrees apart. If noisy signals are expected, zener clamp and filter these
inputs with 6-V zeners and an RC filter. Suggested filtering components are 1 kΩ
and 2 nF. Edge skew in the filter is not a problem, because sensors normally
generate modified gray code with only one output changing at a time, but rise and
fall times must be shorter than 20 µs for correct tachometer operation. Motors with
60 electrical degree position sensor coding can be used if one or two of the
position sensor signals is inverted.
3, 4, 5
The current sense amplifier has a fixed gain of approximately two. It also has a
built-in level shift of approximately 2.5 V. The signal appearing on ISENSE is:
ISENSE = 2.5 V + (2 × ABS ( ISENSE1 - ISENSE2) )
ISENSE1 and ISENSE2 are interchangeable and can be used as differential inputs.
The differential signal applied can be as high as ±0.5 V before saturation.
If spikes are expected on ISENSE1 or ISENSE2, they are best filtered by a
capacitor from ISENSE to ground. Filtering this way allows fast signal inversions to
be correctly processed by the absolute value circuit. The peak-current comparator
allows the PWM to enter a current-limit mode with current in the windings never
exceeding approximately 0.2 V / RSENSE. The overcurrent comparator provides a
fail-safe shutdown in the unlikely case of current exceeding 0.3 V / RSENSE. Then,
softstart is commanded, and all outputs are turned off until the high current
condition is removed. It is often essential to use some filter driving ISENSE1 and
ISENSE2 to reject extreme spikes and to control slew rate. Reasonable starting
values for filter components might be 250-Ω series resistors and a 5-nF capacitor
between ISENSE1 and ISENSE2. Input resistors should be kept small and
matched to maintain gain accuracy.
This input can be used as an over-voltage shut-down input, as a coast input, or
both. This input can be driven by TTL, 5-V CMOS, or 12-V CMOS.
23
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UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
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Terminal Functions (continued)
TERMINAL
NAME
PDA, PDB, PDC
PUA, PUB, PUC
PWR VCC
QUAD SEL
RC-BRAKE
RC-OSC
8
NO.
I/O
DESCRIPTION
12, 13, 14
These outputs can drive the gates of N-channel power MOSFETs directly or they
can drive the bases of power Darlingtons if some form of current limiting is used.
They are meant to drive low-side power devices in high-current output stages.
Current available from these pins can peak as high as 0.5 A. These outputs feature
a true totem-pole output stage. Beware of exceeding device power dissipation
limits when using these outputs for high continuous currents. These outputs pull
high to turn a “low-side” device on (active high).
16, 17, 18
These outputs are open-collector, high-voltage drivers that are meant to drive highside power devices in high-current output stages. These are active low outputs,
meaning that these outputs pull low to command a high-side device on. These
outputs can drive low-voltage PNP Darlingtons and P-channel MOSFETs directly,
and can drive any high-voltage device using external charge pump techniques,
transformer signal coupling, cascode level-shift transistors, or opto-isolated drive
(high-speed opto-devices are recommended). (See applications).
11
This supply pin carries the current sourced by the PD outputs. When connecting
PD outputs directly to the bases of power Darlingtons, the PWR VCC pin can be
current limited with a resistor. Darlington outputs can also be "Baker Clamped" with
diodes from collectors back to PWR VCC. (See Applications)
22
The device can chop power devices in either of two modes, referred to as “twoquadrant” (Quad Sellow) and four quadrant (Quad Sel high). When two-quadrant
chopping, the pull-down power devices are chopped by the output of the PWM
latch while the pull-up drivers remain on. The load chops into one commutation
diode, and except for back-EMF, will exhibit slow discharge current and faster
charge current. Two-quadrant chopping can be more efficient than four-quadrant.
When four-quadrant chopping, all power drivers are chopped by the PWM latch,
causing the load current to flow into two diodes during chopping. This mode
exhibits better control of load current when current is low, and is preferred in servo
systems for equal control over acceleration and deceleration. The QUAD SEL input
has no effect on operation during braking.
21
Each time the TACH-OUT pulses, the capacitor tied to RC-BRAKE discharges
from approximately 3.33 V down to 1.67 V through a resistor. The tachometer
pulse width is approximately T = 0.67 RT CT, where RT and CT are a resistor and
capacitor from RC-BRAKE to ground. Recommended values for RT are 10 kΩ to
500 kΩ, and recommended values for CT are 1 nF to 100 nF, allowing times
between 5 μs and 10 ms. Best accuracy and stability are achieved with values in
the centers of those ranges.
RC-BRAKE also has another function. If RC-BRAKE pin is pulled below the brake
threshold, the device enters brake mode. This mode consists of turning off all three
high-side devices, enabling all three low-side devices, and disabling the
tachometer. The only things that inhibit low-side device operation in braking are
low-supply, exceeding peak current, OV-COAST command, and the PWM
comparator signal. The last of these means that if current sense is implemented
such that the signal in the current sense amplifier is proportional to braking current,
the low-side devices will brake the motor with current control. (See applications)
Simpler current sense connections results in uncontrolled braking and potential
damage to the power devices.
25
The UC3625 can regulate motor current using fixed-frequency pulse width
modulation (PWM). The RC-OSC pin sets oscillator frequency by means of timing
resistor ROSC from the RC-OSC pin to VREF and capacitor COSC from RC-OSC
to Gnd. Resistors 10 kΩ to 100 kΩ and capacitors 1 nF to 100 nF works the best,
but frequency should always be below 500 kHz. Oscillator frequency is
approximately:
F = 2 / (ROSC × COSC )
Additional components can be added to this device to cause it to operate as a
fixed off-time PWM rather than a fixed frequency PWM, using the RC-OSC pin to
select the monostable time constant.
The voltage on the RC-OSC pin is normally a ramp of about 1.2-V peak-to-peak,
centered at approximately 1.6 V. This ramp can be used for voltage-mode PWM
control, or can be used for slope compensation in current-mode control.
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
Terminal Functions (continued)
TERMINAL
NAME
SSTART
TACH-OUT
VCC
VREF
I/O
NO.
DESCRIPTION
24
Any time that VCC drops below threshold or the sensed current exceeds the overcurrent threshold, the soft-start latch is set. When set, it turns on a transistor that
pulls down on SSTART. Normally, a capacitor is connected to this pin, and the
transistor will completely discharge the capacitor. A comparator senses when the
NPN transistor has completely discharged the capacitor, and allows the soft-start
latch to clear when the fault is removed. When the fault is removed, the soft-start
capacitor charges from the on-chip current source.
SSTART clamps the output of the error amplifier, not allowing the error amplifier
output voltage to exceed SSTART regardless of input. The ramp on RC-OSC can
be applied to PWM In and compared to E/A OUT. With SSTART discharged below
0.2 V and the ramp minimum being approximately 1.0 V, the PWM comparator
keeps the PWM latch cleared and the outputs off. As SSTART rises, the PWM
comparator begins to duty-cycle modulate the PWM latch until the error amplifier
inputs overcome the clamp. This provides for a safe and orderly motor start-up
from an off or fault condition. A 51-kΩ resister is added between VREF and
SSTART to ensure switching.
20
Any change in the H1, H2, or H3 inputs loads data from these inputs into the
position sensor latches. At the same time data is loaded, a fixed-width 5-V pulse is
triggered on TACH-OUT. The average value of the voltage on TACH-OUT is
directly proportional to speed, so this output can be used as a true tachometer for
speed feedback with an external filter or averaging circuit which usually consists of
a resistor and capacitor.
Whenever TACH-OUT is high, the position latches are inhibited, such that during
the noisiest part of the commutation cycle, additional commutations are not
possible. Although this effectively sets a maximum rotational speed, the maximum
speed can be set above the highest expected speed, preventing false commutation
and chatter.
19
This device operates with supplies between 10 V and 18 V. Under-voltage lockout
keeps all outputs off below 7.5 V, insuring that the output transistors never turn on
until full drive capability is available. Bypass VCC to ground with an 0.1-μF ceramic
capacitor. Using a 10-μF electrolytic bypass capacitor as well can be beneficial in
applications with high supply impedance.
2
This pin provides regulated 5 V for driving Hall-effect devices and speed control
circuitry. VREF reaches 5 V before VCC enables, ensuring that Hall-effect devices
powered from VREF becomes active before the UC3625 drives any output.
Although VREF is current limited, operation over 30 mA is not advised. For proper
performance VREF should be bypassed with at least a 0.1-μF capacitor to ground.
TYPICAL CHARACTERISTICS
Oscillator Frequency
vs
COSC and ROSC
Tachometer on Time
vs
RT and CT
Oscillator Frequency
1MHz
100kHz
10kHz
µ
1kHz
100Hz
0.001
µ
µ
0.01
µ
0.1
Figure 2.
µ
Figure 3.
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TYPICAL CHARACTERISTICS (continued)
10
Supply Current
vs
Temperature
Soft-Start Pull-Up Current
vs
Temperature
Figure 4.
Figure 5.
Soft-Start Discharge Current
vs
Temperature
Current Sense Amplifier Transfer Function
vs
ISENSE2 - ISENSE1
Figure 6.
Figure 7.
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
APPLICATION INFORMATION
Cross Conduction Prevention
The UC3625 inserts delays to prevent cross conduction due to overlapping drive signals. However, some thought
must always be given to cross conduction in output stage design because no amount of dead time can prevent
fast slewing signals from coupling drive to a power device through a parasitic capacitance.
The UC3625 contains input latches that serve as noise blanking filters. These latches remain transparent through
any phase of a motor rotation and latch immediately after an input transition is detected. They remain latched for
two cycles of the PWM oscillator. At a PWM oscillator speed of 20 kHz, this corresponds to 50 μs to 100 μs of
blank time which limits maximum rotational speed to 100 kRPM for a motor with six transitions per rotation or 50
kRPM for a motor with 12 transitions per rotation.
This prevents noise generated in the first 50 μs of a transition from propagating to the output transistors and
causing cross-conduction or chatter.
The UC3625 also contains six flip flops corresponding to the six output drive signals. One of these flip flops is set
every time that an output drive signal is turned on, and cleared two PWM oscillator cycles after that drive signal
is turned off. The output of each flip flop is used to inhibit drive to the opposing output (Figure 8). In this way, it is
impossible to turn on driver PUA and PDA at the same time. It is also impossible for one of these drivers to turn
on without the other driver having been off for at least two PWM oscillator clocks.
EDGE
FINDER
SHIFT
REG
PWM
CLK
S
Q
R
Q
S
Q
R
Q
PUA
PULL UP
FROM
DECODER
PULL
DOWN
PDA
Figure 8. Cross Conduction Prevention
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UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
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Power Stage Design
The UC3625 is useful in a wide variety of applications, including high-power in robotics and machinery. The
power output stages used in such equipment can take a number of forms, according to the intended performance
and purpose of the system. Figure 9 show four different power stages with the advantages and disadvantages of
each.
For high-frequency chopping, fast recovery circulating diodes are essential. Six are required to clamp the
windings. These diodes should have a continuous current rating at least equal to the operating motor current,
since diode conduction duty-cycle can be high. For low-voltage systems, Schottky diodes are preferred. In higher
voltage systems, diodes such as Microsemi UHVP high voltage platinum rectifiers are recommended.
In a pulse-by-pulse current control arrangement, current sensing is done by resistor RS, through which the
transistor's currents are passed (Figure A, Figure B, and Figure C of Figure 9). In these cases, RD is not needed.
The low-side circulating diodes go to ground and the current sense terminals of the UC3625 (ISENSE1 and ISENSE2)
are connected to RS through a differential RC filter. The input bias current of the current sense amplifier causes a
common mode offset voltage to appear at both inputs, so for best accuracy, keep the filter resistors below 2 kΩ
and matched.
The current that flows through RS is discontinuous because of chopping. It flows during the on time of the power
stage and is zero during the off time. Consequently, the voltage across RS consists of a series of pulses,
occurring at the PWM frequency, with a peak value indicative of the peak motor current.
To sense average motor current instead of peak current, add another current sense resistor (RD in Figure D of
Figure 9) to measure current in the low-side circulating diodes, and operate in four quadrant mode (pin 22 high).
The negative voltage across RD is corrected by the absolute value current sense amplifier. Within the limitations
imposed by Table 1, the circuit of Figure B can also sense average current.
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
FIGURE A
FIGURE B
TO
MOTOR
TO
MOTOR
RS
RS
FIGURE C
FIGURE D
TO
MOTOR
TO
MOTOR
RS
RD
RS
Figure 9. Four Power Stage Designs
Table 1. Imposed limitations for Figure 9
2 QUADRANT
4 QUADRANT
SAFE
BRAKING
POWER REVERSE
Figure A
Yes
No
No
CURRENT SENSE
Pulse-by-Pulse
Average
N0
Yes
No
Yes
Figure B
Yes
Yes
No
In 4-quad mode only
Yes
Figure C
Yes
Yes
Yes
In 4-quad mode only
Yes
No
Figure D
Yes
Yes
Yes
In 4-quad mode only
Yes
Yes
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UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
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For drives where speed is critical, P-Channel MOSFETs can be driven by emitter followers as shown in
Figure 10. Here, both the level shift NPN and the PNP must withstand high voltages. A zener diode is used to
limit gate-source voltage on the MOSFET. A series gate resistor is not necessary, but always advisable to control
overshoot and ringing.
High-voltage optocouplers can quickly drive high-voltage MOSFETs if a boost supply of at least 10 V greater
than the motor supply is provided (See Figure 11) To protect the MOSFET, the boost supply should not be
higher than 18 V above the motor supply.
For under 200-V 2-quadrent applications, a power NPN driven by a small P-Channel MOSFET performs well as
a high-side driver as in Figure 12. A high voltage small-signal NPN is used as a level shift and a high voltage
low-current MOSFET provides drive. Although the NPN will not saturate if used within its limitations, the baseemitter resistor on the NPN is still the speed limiting component.
Figure 13 shows a power NPN Darlington drive technique using a clamp to prevent deep saturation. By limiting
saturation of the power device, excessive base drive is minimized and turn-off time is kept fairly short. Lack of
base series resistance also adds to the speed of this approach.
Figure 10. Fast High-Side P-Channel Driver
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Figure 11. Optocoupled N-Channel High-Side
Driver
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UC2625
UC3625
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
Figure 12. Power NPN High-Side Driver
Figure 13. Power NPN Low-Side Driver
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UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
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Fast High-Side N-Channel Driver with Transformer Isolation
A small pulse transformer can provide excellent isolation between the UC3625 and a high-voltage N-Channel
MOSFET while also coupling gate drive power. In this circuit (shown in Figure 14), a UC3724 is used as a
transformer driver/encoder that duty-cycle modulates the transformer with a 150-kHz pulse train. The UC3725
rectifies this pulse train for gate drive power, demodulates the signal, and drives the gate with over 2-A peak
current.
+12V
VMOTOR
3
33kW
6
7
PUA
2
7
UC3724N
UC3725N
4
4
8
1:2
8
1
5kW
2
5
1
1nF
6
3
100nF
TO MOTOR
Figure 14. Fast High-Side N-Channel Driver with Transformer Isolation
Both the UC3724 and the UC3725 can operate up to 500 kHz if the pulse transformer is selected appropriately.
To raise the operating frequency, either lower the timing resistor of the UC3724 (1 kΩ min), lower the timing
capacitor of the UC3724 (500 pF min) or both.
If there is significant capacitance between transformer primary and secondary, together with very high output
slew rate, then it may be necessary to add clamp diodes from the transformer primary to 12 V and ground.
General purpose small signal switching diodes such as 1N4148 are normally adequate.
The UC3725 also has provisions for MOSFET current limiting. Consult the UC3725 data sheet for more
information on implementing this.
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
Computational Truth Table
Table 2 shows the outputs of the gate drive and open collector outputs for given hall input codes and direction
signals. Numbers at the top of the columns are pin numbers.
These devices operate with position sensor encoding that has either one or two signals high at a time, never all
low or all high. This coding is sometimes referred to as 120° Coding because the coding is the same as coding
with position sensors spaced 120 magnetic degrees about the rotor. In response to these position sense signals,
only one low-side driver turns on (go high) and one high-side driver turns on (pull low) at any time.
Table 2. Computational Truth Table
INPUTS
OUTPUTS
DIR
H1
H2
H3
Low-Side
High-Side
6
8
9
10
12
13
14
16
17
18
1
0
0
1
L
H
L
L
H
H
1
0
1
1
L
L
H
L
H
H
1
0
1
0
L
L
H
H
L
H
1
1
1
0
H
L
L
H
L
H
1
1
0
0
H
L
L
H
H
L
1
1
0
1
L
H
L
H
H
L
0
1
0
1
L
L
H
H
L
H
0
1
0
0
L
L
H
L
H
H
0
1
1
0
L
H
L
L
H
H
0
0
1
0
L
H
L
H
H
L
0
0
1
1
H
L
L
H
H
L
0
0
0
1
H
L
L
H
L
H
X
1
1
1
L
L
L
H
H
H
X
0
0
0
L
L
L
H
H
H
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UC1625
UC2625
UC3625
SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
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VREF
+15V
+5V TO HALL
SENSORS
100nF
100nF
20mF
ROSC
33kW
3kW
10kW
2
2N3904
10W
19
11
22
DIR
3kW
1k
17
1
18
UC3625
28
100nF
14
27
13
25
2200pF
COSC
2N3906
IRF9350
16
6
4kW
TO
MOTOR
TO OTHER
CHANNELS
10W
15
IRF532
20
21
3nF
CT
REQUIRED
FOR BRAKE
AND FAST
REVERSE
TO OTHER
CHANNELS
12
BRAKE
+
100mF
3kW
+
20mF
10kW
QUAD
VMOTOR
26
68kW
RT
3
24
23
8
9
10
4
5
7
10kW
100nF
5nF
FROM
HALL
SENSORS
100nF
2nF
51kW
240W
5nF
2nF
2nF
VREF
240W
0.02
W
RS
REQUIRED
FOR
AVERAGE
CURRENT
SENSING
0.02
W
RD
Figure 15. 45-V/8-A Brushless DC Motor Drive Circuit
N-Channel power MOSFETs are used for low-side drivers, while P-Channel power MOSFETs are shown for
high-side drivers. Resistors are used to level shift the UC3625 open-collector outputs, driving emitter followers
into the MOSFET gate. A 12-V zener clamp insures that the MOSFET gate-source voltage never exceeds 12 V.
Series 10-Ω gate resistors tame gate reactance, preventing oscillations and minimizing ringing.
The oscillator timing capacitor should be placed close to pins 15 and 25, to keep ground current out of the
capacitor. Ground current in the timing capacitor causes oscillator distortion and slaving to the commutation
signal.
The potentiometer connected to pin 1 controls PWM duty cycle directly, implementing a crude form of speed
control. This control is often referred to as "voltage mode" because the potentiometer position sets the average
motor voltage. This controls speed because steady-state motor speed is closely related to applied voltage.
Pin 20 (Tach-Out) is connected to pin 7 (SPEED IN) through an RC filter, preventing direction reversal while the
motor is spinning quickly. In two-quadrant operation, this reversal can cause kinetic energy from the motor to be
forced into the power MOSFETs.
A diode in series with the low-side MOSFETs facilitates PWM current control during braking by insuring that
braking current will not flow backwards through low-side MOSFETs. Dual current-sense resistors give continuous
current sense, whether braking or running in four-quadrant operation, an unnecessary luxury for two-quadrant
operation.
The 68-kΩ and 3-nF tachometer components set maximum commutation time at 140 μs. This permits smooth
operation up to 35,000 RPM for four-pole motors, yet gives 140 μs of noise blanking after commutation.
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SLUS353C – NOVEMBER 2003 – REVISED JUNE 2013
REVISION HISTORY
Changes from Revision B (April 2007) to Revision C
•
Page
Added Storage Temperature and temperature range to Absolute Maximum Ratings Table ............................................... 2
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PACKAGE OPTION ADDENDUM
www.ti.com
13-Aug-2021
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UC2625DW
ACTIVE
SOIC
DW
28
20
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
UC2625DW
UC2625DWG4
ACTIVE
SOIC
DW
28
20
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
UC2625DW
UC2625DWTR
ACTIVE
SOIC
DW
28
1000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 85
UC2625DW
UC3625DW
ACTIVE
SOIC
DW
28
20
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
0 to 70
UC3625DW
UC3625DWTR
ACTIVE
SOIC
DW
28
1000
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
0 to 70
UC3625DW
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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