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UCx84x Current-Mode PWM Controllers
The UCx84x family offers a variety of package
options, temperature range options, choice of
maximum duty cycle, and choice of turnon and turnoff
thresholds and hysteresis ranges. Devices with higher
turnon or turnoff hysteresis are ideal choices for offline power supplies, while the devices with a narrower
hysteresis range are suited for DC-DC applications.
The UC184x devices are specified for operation from
–55°C to 125°C, the UC284x series is specified for
operation from –40°C to 85°C, and the UC384x series
is specified for operation from 0°C to 70°C.
1 Features
•
•
•
•
•
•
•
•
•
•
•
Optimized for off-line and DC-to-DC converters
Low start-up current (< 1 mA)
Automatic feedforward compensation
Pulse-by-pulse current limiting
Enhanced load-response characteristics
Undervoltage lockout with hysteresis
Double-pulse suppression
High-current totem-pole output
Internally trimmed bandgap reference
Up to 500-kHz operation
Error amplifier with low output resistance
Device Information(1)
2 Applications
•
•
Switching regulators of any polarity
Transformer-coupled DC-DC converters
3 Description
PART NUMBER
PACKAGE (PIN)
BODY SIZE (NOM)
CDIP (8)
9.60 mm × 6.67 mm
UC184x
LCCC (20)
8.89 mm × 8.89 mm
UC284x
The UCx84x series of control integrated circuits
provide the features that are necessary to implement
off-line or DC-to-DC fixed-frequency current-mode
control schemes, with a minimum number of external
components. The internally implemented circuits
include an undervoltage lockout (UVLO), featuring a
start-up current of less than 1 mA, and a precision
reference trimmed for accuracy at the error amplifier
input. Other internal circuits include logic to ensure
latched operation, a pulse-width modulation (PWM)
comparator that also provides current-limit control,
and a totem-pole output stage that is designed to
source or sink high-peak current. The output stage,
suitable for driving N-channel MOSFETs, is low when
it is in the off state.
UC384x
(1)
CFP (8)
9.21 mm × 5.97 mm
SOIC (8)
4.90 mm × 3.91 mm
SOIC (14)
8.65 mm × 3.91 mm
PDIP (8)
9.81 mm × 6.35 mm
SOIC (8)
4.90 mm × 3.91 mm
SOIC (14)
8.65 mm × 3.91 mm
PDIP (8)
9.81 mm × 6.35 mm
CFP (8)
9.21 mm × 5.97 mm
For all available packages, see the orderable addendum at
the end of the datasheet.
VIN
VCC
OUTPUT
VREF
ISENSE
UC2843
VFB
RT/CT
GROUND
COMP
Copyright © 2016, Texas Instruments Incorporated
Simplified Application
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
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Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Device Comparison Table...............................................3
6 Pin Configuration and Functions...................................3
7 Specifications.................................................................. 6
7.1 Absolute Maximum Ratings........................................ 6
7.2 ESD Ratings............................................................... 6
7.3 Recommended Operating Conditions.........................6
7.4 Thermal Information....................................................6
7.5 Electrical Characteristics.............................................7
7.6 Typical Characteristics................................................ 9
8 Detailed Description...................................................... 11
8.1 Overview................................................................... 11
8.2 Functional Block Diagrams....................................... 11
8.3 Feature Description...................................................12
8.4 Device Functional Modes..........................................20
9 Application and Implementation.................................. 21
9.1 Application Information............................................. 21
9.2 Typical Application.................................................... 21
10 Power Supply Recommendations..............................34
11 Layout........................................................................... 35
11.1 Layout Guidelines................................................... 35
11.2 Layout Example...................................................... 36
12 Device and Documentation Support..........................37
12.1 Receiving Notification of Documentation Updates..37
12.2 Support Resources................................................. 37
12.3 Trademarks............................................................. 37
12.4 Electrostatic Discharge Caution..............................37
12.5 Glossary..................................................................37
13 Mechanical, Packaging, and Orderable
Information.................................................................... 37
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision F (April 2020) to Revision G (July 2022)
Page
• Updated the numbering format for tables, figures and cross-references throughout the document...................1
Changes from Revision E (January 2017) to Revision F (April 2020)
Page
• Changed UVLO Table updated ..........................................................................................................................7
2
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5 Device Comparison Table
UVLO
TURNON AT 16 V
TURNOFF AT 10 V
SUITABLE FOR OFF-LINE
APPLICATIONS
TURNON AT 8.4 V
TURNOFF AT 7.6 V
SUITABLE FOR DC-DC
APPLICATIONS
UC1842
UC1843
–55°C to 125°C
UC2842
UC2843
–40°C to 85°C
UC3842
UC3843
0°C to 70°C
UC1844
UC1845
–55°C to 125°C
UC2844
UC2845
–40°C to 85°C
UC3844
UC3845
0°C to 70°C
TEMPERATURE RANGE
MAX DUTY CYCLE
Up to 100%
Up to 50%
6 Pin Configuration and Functions
COMP
1
8
VREF
VFB
2
7
VCC
ISENSE
3
6
OUTPUT
RT/CT
4
5
GROUND
COMP
1
14
VREF
NC
2
13
NC
VFB
3
12
VCC
NC
4
11
VC
ISENSE
5
10
OUTPUT
NC
6
9
GROUND
RT/CT
7
8
PWRGND
Figure 6-1. D, JG, and P Packages 8-Pin SOIC,
CDIP, and PDIP Top View
NC
2
1 20 19
NC
COMP
3
VREF
NC
Figure 6-2. D and W Packages 14-Pin SOIC and
CFP Top View
NC
4
18
VCC
VFB
5
17
VC
NC
6
16
NC
ISENSE
7
15
OUTPUT
NC
8
14
NC
GROUND
NC
PWRGND
NC
RT/CT
9 10 11 12 13
Figure 6-3. FK Package 20-Pin LCCC Top View
Table 6-1. Pin Functions
PIN
NAME
COMP
SOIC,
CDIP,
PDIP
(8)
1
SOIC, CFP
(14)
1
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LCCC
(20)
2
TYPE
DESCRIPTION
O
Error amplifier compensation pin. Connect external compensation
components to this pin to modify the error amplifier output. The error
amplifier is internally current-limited so the user can command zero
duty cycle by externally forcing COMP to GROUND.
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Table 6-1. Pin Functions (continued)
PIN
SOIC,
CDIP,
PDIP
(8)
SOIC, CFP
(14)
LCCC
(20)
TYPE
GROUND
5
9
13
G
Analog ground. For device packages without PWRGND, GROUND
functions as both power ground and analog ground.
PWRGND
—
8
12
G
Power ground. For device packages without PWRGND, GROUND
functions as both power ground and analog ground
ISENSE
3
5
7
I
Primary-side current sense pin. Connect to current sensing resistor.
The PWM uses this signal to terminate the OUTPUT switch
conduction. A voltage ramp can be applied to this pin to run the
device with a voltage-mode control configuration.
NC
—
2, 4, 6, 13
1, 3, 4, 6,
8, 9, 11,
14, 16, 19
—
Do not connect
O
OUTPUT is the gate drive for the external MOSFET. OUTPUT is
the output of the on-chip driver stage intended to directly drive a
MOSFET. Peak currents of up to 1 A are sourced and sunk by this
pin. OUTPUT is actively held low when VCC is below the turnon
threshold.
NAME
OUTPUT
6
10
15
DESCRIPTION
Fixed frequency oscillator set point. Connect timing resistor, RRT, to
VREF and timing capacitor, CCT, to GROUND from this pin to set the
switching frequency. For best performance, keep the timing capacitor
lead to the device GROUND as short and direct as possible. If
possible, use separate ground traces for the timing capacitor and
all other functions.
The frequency of the oscillator can be estimated with the following
equations:
RT/CT
4
7
10
I/O
1.72
fOSC =
RRT × CCT
(1)
where fOSC is in Hertz, RRT is in Ohms and CCT is in Farads. Never
use a timing resistor less than 5 kΩ. The frequency of the OUTPUT
gate drive of the UCx842 and UCx843, fSW, is equal to fOSC at up to
100% duty cycle; the frequency of the UCx844 and UCx845 is equal
to half of the fOSC frequency at up to 50% duty cycle.
VC
—
11
17
I
Bias supply input for the output gate drive. For PWM controllers that
do not have this pin, the gate driver is biased from the VCC pin. VC
must have a bypass capacitor at least 10 times greater than the gate
capacitance of the main switching FET used in the design.
Analog controller bias input that provides power to the device. Total
VCC current is the sum of the quiescent VCC current and the
average OUTPUT current. Knowing the switching frequency and the
MOSFET gate charge, Qg, the average OUTPUT current can be
calculated from:
VCC
7
12
18
I
IOUTPUT = Q g × fSW
(2)
A bypass capacitor, typically 0.1 µF, connected directly to GROUND
with minimal trace length, is required on this pin. An additional
bypass capacitor at least 10 times greater than the gate capacitance
of the main switching FET used in the design is also required on
VCC.
VFB
4
2
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3
5
I
Inverting input to the internal error amplifier. VFB is used to control
the power converter voltage-feedback loop for stability.
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Table 6-1. Pin Functions (continued)
PIN
NAME
VREF
SOIC,
CDIP,
PDIP
(8)
8
SOIC, CFP
(14)
14
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LCCC
(20)
20
TYPE
O
DESCRIPTION
5-V reference voltage. VREF is used to provide charging current
to the oscillator timing capacitor through the timing resistor. It is
important for reference stability that VREF is bypassed to GROUND
with a ceramic capacitor connected as close to the pin as possible.
A minimum value of 0.1-µF ceramic is required. Additional VREF
bypassing is required for external loads on VREF.
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)(1)
MIN
MAX
UNIT
30
V
6.3
V
Low impedance source
VVCC
IVCC < 30 mA
Self Limiting
VVFB and VISENSE
Analog input voltage
VVC
Input Voltage, Q and D Package only
30
V
IOUTPUT
Output drive current
±1
A
ICOMP
Error amplifier output sink current
10
mA
EOUTPUT
Output energy (capacitive load)
5
µJ
TJ
Junction temperature
150
°C
Tstg
Storage temperature
150
°C
(1)
–0.3
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Section 7.3.
Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins(1)
±3000
Charged device model (CDM), per JEDEC specification JESD22-C101, all pins(2)
±3000
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VVCC and VVC (1)
Supply voltage
VVFB
VISENSE
IVCC
Supply current, externally limited
IOUTPUT
Average output current
IVREF
Reference output current
fOSC
Oscillator frequency
TA
Operating free-air temperature
(1)
TYP
MAX
12
UNIT
28
V
Input voltage
2.5
V
Input voltage
1
V
25
mA
200
mA
100
–20
mA
500
kHz
UC184x
–55
125
UC284x
–40
85
UC384x
0
70
°C
These recommended voltages for VC and POWER GROUND apply only to the D package.
7.4 Thermal Information
UCx84x
THERMAL
6
METRIC(1)
D (SOIC)
D (SOIC)
P (PDIP)
FK (LCCC)
8 PINS
14 PINS
8 PINS
20 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
104.8
78.2
53.7
—
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
47.3
37.1
46.7
36.2
°C/W
RθJB
Junction-to-board thermal resistance
45.9
32.6
31
35.4
°C/W
ψJT
Junction-to-top characterization parameter
8.2
7.3
17.1
—
°C/W
ψJB
Junction-to-bottom characterization parameter
45.2
32.4
30.9
—
°C/W
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7.4 Thermal Information (continued)
UCx84x
THERMAL
RθJC(bottom)
(1)
METRIC(1)
D (SOIC)
D (SOIC)
P (PDIP)
FK (LCCC)
8 PINS
14 PINS
8 PINS
20 PINS
—
—
—
4.1
Junction-to-case (bottom) thermal resistance
UNIT
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
7.5 Electrical Characteristics
over operating free-air temperature range (unless otherwise noted) –55°C ≤ TA ≤ 125°C for the UC184x; –40°C ≤ TA ≤ 85°C
for the UC284x, 0°C ≤ TA ≤ 70°C for the UC384x, VVCC = 15 V(2); 0.1 µF capacitor from VCC to GROUND, 0.1 µF capacitor
from VREF to GROUND, RRT = 10 kΩ; CCT = 3.3 nF, TJ = TA.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UC184x
and
UC284x
4.95
5
5.05
UC384x
4.9
5
5.1
6
20
UNIT
REFERENCE SECTION
VVREF
Reference voltage
IVREF = 1 mA, TJ = 25°C
Line regulation
12 ≤ VCC ≤ 25 V
Load regulation
1 ≤ IVREF ≤ 20 mA
Temperature stability
See (1) (3)
Line, load, temperature
(1)
Output noise voltage
10 Hz ≤ fOSC ≤ 10 kHz,
(1)
Long term stability
TA = 125°C, 1000 Hrs (1)
Total output variation
6
25
mV
0.4
mV/°C
4.9
5.1
UC384x
4.82
5.18
Output short circuit
50
–30
mV
0.2
UC184x
and
UC284x
TJ = 25°C
V
V
μV
5
25
mV
–100
–180
mA
52
57
kHz
0.2%
1%
OSCILLATOR SECTION
fOSC
VRT/CT
Initial accuracy
TJ = 25°C(5)
Voltage stability
12 ≤ VCC ≤ 25 V
Temperature stability
TMIN ≤ TA ≤ TMAX (1)
5%
Amplitude
Peak-to-peak (1)
1.7
47
V
ERROR AMPLIFIER SECTION
VVFB
IVFB
Input voltage
VCOMP = 2.5 V
Input bias current
AVOL
UC184x
and
UC284x
2.45
2.5
2.55
UC384x
2.42
2.5
2.58
UC184x
and
UC284x
–1
UC384x
–2
V
µA
2 ≤ VCOMP ≤ 4 V
65
90
dB
Unity gain bandwidth
TJ = 25°C (1)
0.7
1
MHz
PSRR
Power supply rejection ratio
12 ≤ VCC ≤ 25 V
60
70
dB
I(snk)
COMP sink current
VVFB = 2.7 V, VCOMP = 1.1 V
2
6
I(src)
COMP source current
VVFB = 2.3 V, VCOMP = 5 V
–0.5
–0.8
VCOMP
High
High-level output voltage
VVFB = 2.3 V, RL = 15-kΩ COMP to GROUND
5
6
VCOMP Low Low-level output voltage
VVFB = 2.7 V, RL = 15-kΩ COMP to VREF
0.7
mA
V
1.1
CURRENT SENSE SECTION
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7.5 Electrical Characteristics (continued)
over operating free-air temperature range (unless otherwise noted) –55°C ≤ TA ≤ 125°C for the UC184x; –40°C ≤ TA ≤ 85°C
for the UC284x, 0°C ≤ TA ≤ 70°C for the UC384x, VVCC = 15 V(2); 0.1 µF capacitor from VCC to GROUND, 0.1 µF capacitor
from VREF to GROUND, RRT = 10 kΩ; CCT = 3.3 nF, TJ = TA.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
2.85
3
3.15
V/V
0.9
1
1.1
V
–2
–10
µA
VISENSE stepped from 0 V to 2 V (1)
150
300
ns
ISINK = 20 mA
0.1
0.4
ISINK = 200 mA
1.5
2.2
(4) (6)
ACS
Gain
See
VISENSE
Maximum input signal
VCOMP = 5 V (4)
PSRR
Power supply rejection ratio
IISENSE
Input bias current
tDLY
Delay to output
12 V ≤ VVCC ≤ 25 V
(1) (4)
70
dB
OUTPUT SECTION
VOUT Low
Low-level OUTPUT voltage
VOUT High
High-level OUTPUT voltage
tRISE
Rise time (1)
COUTPUT = 1 nF, TJ = 25°C
50
150
ns
tFALL
Fall time (1)
COUTPUT = 1 nF, TJ = 25°C,
50
150
ns
16
17
ISOURCE = 20 mA
13
13.5
ISOURCE = 200 mA
12
13.5
V
V
UNDERVOLTAGE LOCKOUT (UVLO)
UC1842/4 and UC2842/4
VCCON
Enable threshold
UC3842/4
14.5
16
17.5
UCx843/5
7.8
8.4
9
UC1842/4 and UC2842/4
VCCOFF
UVLO off threshold
15
9
10
11
UC3842/4
8.5
10
11.5
UCx843/5
7
7.6
8.2
UCx842/3
95%
97%
100%
UC1844/5 and UC2844/5
46%
48%
50%
UC3844/5
47%
48%
50%
V
V
PWM
DMAX
DMIN
Maximum duty cycle
Minimum duty cycle
0%
TOTAL STANDBY CURRENT
IVCC
Start-up current
IVCC
Operating supply current
VVFB = VISENSE= 0 V
VCC Zener voltage
IVCC = 25 mA
(1)
(2)
(3)
(6)
8
1
11
17
34
mA
V
Specified by design. Not production tested.
Adjust VCC above the start threshold before setting at 15 V
Temperature stability, sometimes referred to as average temperature coefficient, is described by the equation:
Temp Stability =
(4)
(5)
30
0.5
VREF:max ; F VREF:min ;
TJ:max ; F TJ:min ;
VREFmin and VREFmax are the maximum and minimum reference voltages measured over the
appropriate temperature range. Note that the extremes in voltage do not necessarily occur at the extremes in temperature.
Parameter measured at trip point of latch with VFB = 0 V.
OUTPUT switching frequency, fSW, equals the oscillator frequency, fOSC, for the UCx842 and UCx843. OUTPUT switching frequency,
fSW, is one half oscillator frequency, fOSC, for the UCx844 and UCx845.
Gain defined as: A = ΔVCOMP/ΔVISENSE, 0 V ≤ VISENSE ≤ 0.8 V.
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7.6 Typical Characteristics
1.1
VTH, Current Sense Input Threshold (V)
9.2
9
8.6
8.4
8.2
8
7.8
7.6
-25
0
25
50
75
Temperature (C)
100
125
0.5
0.4
0.3
0.2
TA = 125qC
TA = 25qC
TA = 55qC
0.1
1
60
150
100
40
50
20
0
0
-50
1000
10000 100000 1000000
Freq (Hz)
D003
9
-2
8
-3
-6
7
Source Saturation at 25 C
Source Saturation at -55 C 6
Sink Saturation at -55 C
5
Sink Saturation at 25 C
4
-7
3
-8
2
-9
1
-4
-5
0
Reference Voltage Delta (mV)
60
40
-75
100
200
300
400
500
600
IO, Output Load Current (mA)
700
0
800
D005
Figure 7-4. OUTPUT Saturation Voltage vs Load Current for
VCC = 15 V with 5-ms Input Pulses
-10
80
D005
10
160
100
6
0
0
120
5.5
-1
180
140
2.5
3
3.5
4
4.5
5
VO, Error Amp Output Voltage (V)
-10
-100
1E+7
Figure 7-3. Error Amplifier Open-Loop Gain and Phase vs
Frequency, VCC = 15 V, RL = 100 kΩ, and TA = 25 °C
2
Figure 7-2. Current Sense Input Threshold vs Error Amplifier
Output Voltage for VCC = 15 V
200
Gain
Phase
100
1.5
D001
80
Gain (dB)
0.6
150
100
ISC (mA)
0.7
0
-50
Figure 7-1. Oscillator Discharge Current vs Temperature for
VCC = 15 V and VOSC = 2 V
-20
10
0.8
Sink Saturation Voltage (V)
7.4
-75
Source Saturation Voltage (V)
IDISCHARGE (mA)
8.8
1
0.9
Ta = 125 C
Ta = 25 C
Ta = -40 C
-20
-30
-40
-50
-60
-50
-25
0
25
50
75
Temperature (C)
100
125
150
D006
Figure 7-5. VREF Short-Circuit Current vs Temperature for VCC
= 15 V
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0
20
40
60
80
100
Source Current (mA)
120
140
160
D007
Figure 7-6. VREF Voltage vs Source Current
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7.6 Typical Characteristics (continued)
5.2
4
Source, TA = 25°C
5.15
Sink, TA = 25°C
Saturation Voltage (V)
VREF (V)
5.1
5.05
5
4.95
4.9
Source, TA = ±55°C
3
Sink, TA = ±55°C
2
1
4.85
4.8
-75
-50
-25
0
25
50
75
Temperature (C)
100
125
0
0.01
150
0.1
D008
1
Output Current (A)
Figure 7-7. VREF Voltage vs Temperature
Figure 7-8. Output Saturation
30
tDEADTIME (Ps)
10
5
1
0.5
0.3
1
5
10
CCT (nF)
50
100
D006
Figure 7-9. Dead Time vs Timing Capacitance, CCT
Timing Resistance (NŸ)
100
50
20
CCT (nF)
10
100
47
22
10
4.7
2.2
1
5
2
1
100
1000
10 k
100 k
1M
Frequency (Hz)
Figure 7-10. Timing Resistance, RRT, vs Frequency
10
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8 Detailed Description
8.1 Overview
The UCx84x series of control integrated circuits provide the features necessary to implement AC-DC or DC-toDC fixed-frequency current-mode control schemes with a minimum number of external components. Protection
circuitry includes undervoltage lockout (UVLO) and current limiting. Internally implemented circuits include a
start-up current of less than 1 mA, a precision reference trimmed for accuracy at the error amplifier input, logic
to ensure latched operation, a pulse-width modulation (PWM) comparator that also provides current-limit control,
and a totem-pole output stage designed to source or sink high-peak current. The output stage, suitable for
driving N-channel MOSFETs, is low when it is in the off-state.
Major differences between members of these series are the UVLO thresholds, acceptable ambient temperature
range, and maximum duty-cycle. Typical UVLO thresholds of 16 V (ON) and 10 V (OFF) on the UCx842 and
UCx844 devices make them ideally suited to off-line AC-DC applications. The corresponding typical thresholds
for the UCx843 and UCx845 devices are 8.4 V (ON) and 7.6 V (OFF), making them ideal for use with
regulated input voltages used in DC-DC applications. The UCx842 and UCx843 devices operate to duty cycles
approaching 100%. The UCx844 and UCx845 obtain a duty-cycle range of 0% to 50% by the addition of an
internal toggle flip-flop, which blanks the output off every other clock cycle.
The UC184x-series devices are characterized for operation from –55°C to 125°C. UC284x-series devices are
characterized for operation from −40°C to 85°C. The UC384x devices are characterized for operation from 0°C
to 70°C.
8.2 Functional Block Diagrams
VCC
UVLO
34 V
5-V
Reference
EN
GROUND
VREF
Internal
Bias
VC
2.5 V
VREF Good
Logic
RT/CT
VFB
Osc
OUTPUT
S
2R
+
E/A
PWRGND
R
R
1V
PWM
Latch
PWM
Comparator
COMP
ISENSE
UCx842
UCx843
Copyright © 2016, Texas Instruments Incorporated
Figure 8-1. UCx842 and UCx843 Block Diagram, No Toggle
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VCC
UVLO
34 V
5-V
Reference
EN
GROUND
VREF
Internal
Bias
VC
2.5 V
VREF Good
Logic
RT/CT
Osc
VFB
S
2R
+
E/A
OUTPUT
T
PWRGND
R
R
1V
PWM
Latch
PWM
Comparator
COMP
ISENSE
UCx844
UCx845
Copyright © 2016, Texas Instruments Incorporated
Figure 8-2. UCx844 and UCx845 Block Diagram, Toggle
8.3 Feature Description
8.3.1 Detailed Pin Description
8.3.1.1 COMP
The error amplifier in the UCx84x family is an open collector in parallel with a current source, with a unity-gain
bandwidth of 1 MHz. The COMP terminal can both source and sink current. The error amplifier is internally
current-limited, so that one can command zero duty cycle by externally forcing COMP to GROUND.
8.3.1.2 VFB
VFB is the inverting input of the error amplifier. VFB is used to control the power converter voltage-feedback loop
for stability. For best stability, keep VFB lead length as short as possible and VFB stray capacitance as small as
possible.
8.3.1.3 ISENSE
The UCx84x current sense input connects to the PWM comparator. Connect ISENSE to the MOSFET source
current sense resistor. The PWM uses this signal to terminate the OUTPUT switch conduction. A voltage
ramp can be applied to this pin to run the device with a voltage mode control configuration or to add slope
compensation. To prevent false triggering due to leading edge noises, an RC current sense filter may be
required. The gain of the current sense amplifier is typically 3 V/V.
8.3.1.4 RT/CT
RT/CT is the oscillator timing pin. For fixed frequency operation, set the timing capacitor charging current
by connecting a resistor from VREF to RT/CT. Set the frequency by connecting timing capacitor from RT/CT
to GROUND. For the best performance, keep the timing capacitor lead to GROUND as short and direct as
possible. If possible, use separate ground traces for the timing capacitor and all other functions.
The UCx84x’s oscillator allows for operation to 500 kHz. The device uses an external resistor to set the charging
current for the external capacitor, which determines the oscillator frequency. The recommended range of timing
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resistor values is between 5 kΩ and 100 kΩ; the recommended range of timing capacitor values is between 1 nF
and 100 nF.
1.72
fOSC =
RRT × CCT
(3)
In this equation, the switching frequency, fSW is in Hz, RRT is in Ω, and CCT is in Farads.
8.3.1.5 GROUND
GROUND is the signal and power returning ground. TI recommends separating the signal return path and the
high current gate driver path so that the signal is not affected by the switching current.
8.3.1.6 OUTPUT
The high-current bipolar totem-pole output of the UCx84x devices sinks or sources up to 1-A peak of current.
The OUTPUT pin can directly drive a MOSFET. The OUTPUT of the UCx842 and UCx843 devices switches
at the same frequency as the oscillator and can operate near 100% duty cycle. In the UCx844 and UCx845
devices, the switching frequency of OUTPUT is one-half that of the oscillator due to an internal T flipflop. This
limits the maximum duty cycle in the UCx844 and UCx845 to < 50%. Schottky diodes may be necessary on
the OUTPUT pin to prevent overshoot and undershoot due to high impedance to the supply rail and to ground,
respectively. A bleeder resistor, placed between the gate and the source of the MOSFET, should be used to
prevent activating the power switch with extraneous leakage currents during undervoltage lockout. An external
clamp circuit may be necessary to prevent overvoltage stress on the MOSFET gate when VCC exceeds the gate
voltage rating.
8.3.1.7 VCC
VCC is the power input connection for this device. In normal operation, power VCC through a current-limiting
resistor. Although quiescent VCC current is only 0.5 mA, the total supply current is higher, depending on
the OUTPUT current. Total VCC current is the sum of quiescent VCC current and the average OUTPUT
current. Knowing the operating frequency and the MOSFET gate charge (Qg), average OUTPUT current can be
calculated from Equation 4.
IOUTPUT = Q g × fSW
(4)
The UCx84x has a VCC supply voltage clamp of 34 V typical, but the absolute maximum value for VCC from
a low-impedance source is 30 V. For applications that have a higher input voltage than the recommended VCC
voltage, place a resistor in series with VCC to increase the source impedance. The maximum value of this
resistor is calculated with Equation 5.
R VCC :max ; =
VIN :min ; F VVCC :max ;
IVCC + kQ g × fSW o
(5)
In Equation 5, VIN(min) is the minimum voltage that is used to supply VCC, VVCC(max) is the maximum VCC clamp
voltage and IVCC is the IC supply current without considering the gate driver current and Qg is the external power
MOSFET gate charge and fSW is the switching frequency.
The turnon and turnoff thresholds for the UCx84x family are significantly different: 16 V and 10 V for the UCx842
and UCx844; 8.4 V and 7.6 V for the UCx843 and UCx855. To ensure against noise related problems, filter VCC
with an electrolytic and bypass with a ceramic capacitor to ground. Keep the capacitors close to the IC pins.
8.3.1.8 VREF
VREF is the voltage reference for the error amplifier and also for many other internal circuits in the IC. The
high-speed switching logic uses VREF as the logic power supply. The 5-V reference tolerance is ±2% for the
UCx84x family. The output short-circuit current is 30 mA. For reference stability and to prevent noise problems
with high-speed switching transients, bypass VREF to ground with a ceramic capacitor close to the IC package.
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A minimum of 0.1-µF ceramic capacitor is required. Additional VREF bypassing is required for external loads on
the reference. An electrolytic capacitor may also be used in addition to the ceramic capacitor.
When VCC is greater than 1 V and less than the UVLO threshold, a 5-kΩ resistor pulls VREF to ground. VREF
can be used as a logic output indicating power-system status because when VCC is lower than the UVLO
threshold, VREF is held low.
8.3.2 Pulse-by-Pulse Current Limiting
Pulse-by-pulse limiting is inherent in the current mode control scheme. An upper limit on the peak current can be
established by simply clamping the error voltage. Accurate current limiting allows optimization of magnetic and
power semiconductor elements while ensuring reliable supply operation.
8.3.3 Current-Sense
An external series resistor, RCS, senses the current and converts this current into a voltage that becomes the
input to the ISENSE pin. The ISENSE pin is the noninverting input to the PWM comparator. The ISENSE input is
compared to a signal proportional to the error amplifier output voltage; the gain of the current sense amplifier is
typically 3 V/V. The peak ISENSE current is determined by Equation 6:
ISENSE =
VISENSE
R CS
(6)
The typical value for VISENSE is 1 V. A small RC filter, RCSF and CCSF, may be required to suppress switch
transients caused by the reverse recovery of a secondary side diode or equivalent capacitive loading in addition
to parasitic circuit impedances. The time constant of this filter should be considerably less than the switching
period of the converter.
Error
Amplifier
COMP
2R
R
ISENSE
RCSF
RCS
1V
PWM
Comparator
ISENSE
CCSF
GROUND
Copyright © 2016, Texas Instruments Incorporated
Figure 8-3. Current-Sense Circuit Schematic
8.3.4 Error Amplifier With Low Output Resistance
The error amplifier output is an open collector in parallel with a current source. With a low output resistance,
various impedance networks may be used on the compensation pin input for error amplifier feedback. The
error amplifier output, COMP, is frequently used as a control port for secondary-side regulation by using an
external secondary-side adjustable voltage regulator, such as a TL431, to send an error signal across the
secondary-to-primary isolation boundary through an opto-isolator, in this configuration connect the COMP pin
directly to the opto-isolator feedback. On the primary side, the inverting input to the UCx48x error amplifier, VFB,
should be connected to GROUND. With VFB tied to GROUND, the error amplifier output, COMP, is forced to its
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high state and sources current, typically 0.8 mA. The opto-isolator must overcome the source current capability
to control the COMP pin below the error amplifier output high level, VOH.
For primary-side regulation, configure the inverting input to the error amplifier, VFB, with a resistor divider
to provide a signal that is proportional to the converter output voltage being regulated. Add the voltage loop
compensation components between VFB and COMP. The internal noninverting input to the error amplifier
is trimmed to 2.5 V. For best stability, keep VFB lead length as short as possible and minimize the stray
capacitance on VFB.
The internal resistor divider on COMP is maintained at an R:2R ratio, the specific values of these internal
resistors should not be critical in any application.
0.5 mA
2.5 V
+
2R
Error
Amplifier
R
ZI
s
VFB
ZF
1V
PWM
Comparator
COMP
ISENSE
Error amplifier can source or sink up to 0.5 mA.
Figure 8-4. Error-Amplifier Configuration Schematic
8.3.5 Undervoltage Lockout
The UCx84x devices feature undervoltage lockout protection circuits for controlled operation during power-up
and power-down sequences. The UVLO circuit insures that VCC is adequate to make the UCx84x fully
operational before enabling the output stage. Undervoltage lockout thresholds for the UCx842, UCx843,
UCx844, and UCx845 devices are optimized for two groups of applications: off-line power supplies and DC-DC
converters. The 6-V hysteresis in the UCx842 and UCx844 devices prevents VCC oscillations during power
sequencing. This wider VCCON to VCCOFF range, make these devices ideally suited to off-line AC input
applications. The UCx843 and UCx845 controllers have a much narrower VCCON to VCCOFF hysteresis and
may be used in DC to DC applications where the input is considered regulated.
Start-up current is less than 1 mA for efficient bootstrapping from the rectified input of an off-line converter, as
illustrated by Figure 8-7. During normal circuit operation, VCC is developed from auxiliary winding NA with D BIAS
and C VCC. At start-up, however, CVCC must be charged to 16 V through RSTART. With a start-up current of 1 mA,
RSTART can be as large as 100 kΩ and still charge CVCC when VAC = 90 V RMS (low line). Power dissipation in
RSTART would then be less than 350 mW even under high line (VAC= 130 V RMS) conditions.
During UVLO the IC draws less than 1 mA of supply current. Once crossing the turnon threshold the IC supply
current increases to a maximum of 17 mA, typically 11 mA, During undervoltage lockout, the output driver is
biased to a high impedance state and sinks minor amounts of current. A bleeder resistor, placed between the
gate and the source of the MOSFET should be used to prevent activating the power switch with extraneous
leakage currents during undervoltage lockout.
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VCC
< 17 mA
ON/OFF Command
to rest of device
7
IVCC
UCx842 UCx843
UCx844 UCx845
VON (V)
16
8.4
VOFF (V)
10
7.6
< 1 mA
VOFF
VON
VVCC
Figure 8-6. UVLO ON and OFF Profile
Copyright © 2016, Texas Instruments Incorporated
Figure 8-5. UVLO Threshold
NP
NS
RSTART
NA
DBIAS
IVCC • 1mA
VAC
CIN
VCC
OUTPUT
CVCC
0.1 PF
GROUND
RCS
Figure 8-7. Providing Power to UCx84x
8.3.6 Oscillator
The oscillator allows for up to 500-kHz switching frequency. The OUTPUT gate drive is the same frequency as
the oscillator in the UCx842 and UCx843 devices and can operate near 100% duty cycle. In the UCx844 and
UCx845 devices, the frequency of OUTPUT is one-half that of the oscillator due to an internal T flipflop that
blanks the output off every other clock cycle, resulting in a maximum duty cycle for these devices of < 50% of
the switching frequency. An external resistor, RRT, connected from VREF to RT/CT sets the charging current
for the timing capacitor, CCT, which is connected from RT/CT to GROUND. An RRT value greater than 5 kΩ is
recommended on RT/CT to set the positive ramp time of the internal oscillator. Using a value of 5 kΩ or greater
for RRT maintains a favorable ratio between the internal impedance and the external oscillator set resistor and
results in minimal change in frequency over temperature. Using a value of less the recommended minimum
value may result in frequency drift over temperature, part tolerances, or process variations.
The peak-to-peak amplitude of the oscillator waveform is 1.7 V in UCx84x devices. The UCx842 and UCx843
have a maximum duty cycle of approximately 100%, whereas the UCx844 and UCx845 are clamped to 50%
maximum by an internal toggle flip flop. This duty cycle clamp is advantageous in most flyback and forward
converters. For optimum IC performance the dead-time should not exceed 15% of the oscillator clock period.
The discharge current, typically 6 mA at room temperature, sets the dead time, see Figure 7-9. During the
discharge, or dead time, the internal clock signal blanks the output to the low state. This limits the maximum duty
cycle DMAX to:
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DMAX = 1 F :t DEADTIME × fOSC ;
(7)
Equation 8 applies to UCx842 and UCx843 units because the OUTPUT switches at the same frequency as the
oscillator and the maximum duty cycle can be as high as 100%.
DMAX = 1 F lt DEADTIME ×
fOSC
p
2
(8)
Equation 8 applies to UCx844 and UCx845 units because the OUTPUT switches at half the frequency as the
oscillator and the maximum duty cycle can be as high as 50%.
When the power transistor turns off, a noise spike is coupled to the oscillator RT/CT terminal. At high duty
cycles, the voltage at RT/CT is approaching its threshold level (approximately 2.7 V, established by the internal
oscillator circuit) when this spike occurs. A spike of sufficient amplitude prematurely trips the oscillator. To
minimize the noise spike, choose CCT as large as possible, remembering that dead time increases with CCT. It is
recommended that CCT never be less than approximately 1000 pF. Often the noise which causes this problem is
caused by the OUTPUT being pulled below ground at turnoff by external parasitics. This is particularly true when
driving MOSFETs. A Schottky diode clamp from GROUND to OUTPUT prevents such output noise from feeding
to the oscillator.
VREF
RRT
RT/CT
CCT
GROUND
Copyright © 2016, Texas Instruments Incorporated
1.72
For RRT > 5 kΩ:
fOSC =
RRT × CCT
Figure 8-8. Oscillator Section Schematic
8.3.7 Synchronization
The simplest method to force synchronization uses the timing capacitor, CCT, in near standard configuration.
Rather than bring CCT to ground directly, a small resistor is placed in series with CCT to ground. This resistor
serves as the input for the sync pulse which raises the CCT voltage above the oscillator’s internal upper
threshold. The PWM is allowed to run at the frequency set by RRT and CCT until the sync pulse appears.
This scheme offers several advantages including having the local ramp available for slope compensation. The
UC3842/3/4/5 oscillator must be set to a lower frequency than the sync pulse stream, typically 20% with a 0.5-V
pulse applied across the resistor.
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VREF
RRT
RT/CT
CCT
SYNC
24 O
GROUND
Figure 8-9. Synchronizing the Oscillator
8.3.8 Shutdown Technique
The PWM controller (see Figure 8-10) can be shut down by two methods: either raise the voltage at ISENSE
above 1 V or pull the COMP terminal below a voltage two diode drops above ground. Either method causes the
output of the PWM comparator to be high (see Figure 8-10). The PWM latch is reset dominant so that the output
remains low until the next clock cycle after the shutdown condition at the COMP or ISENSE terminal is removed.
In one example, an externally latched shutdown can be accomplished by adding an SCR that resets by cycling
VCC below the lower UVLO threshold. At this point, the reference turns off, allowing the SCR to reset.
1 kO
VREF
COMP
SHUTDOWN
30 O
SHUTDOWN
ISENSE
500 O
To Current
Sense Resistor
Figure 8-10. Shutdown Techniques
8.3.9 Slope Compensation
A fraction of the oscillator ramp can be summed resistively with the current-sense signal to provide slope
compensation for converters requiring duty cycles over 50% (see Figure 8-11). Note that capacitor CCSF forms a
filter with RCSF to suppress the leading-edge switch spikes.
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UCx842
UCx843
VREF
0.1 µF
RRT
RT/CT
CCT
RRAMP
RCSF
ISENSE
ISENSE
RCS
CCSF
Copyright © 2016, Texas Instruments Incorporated
Figure 8-11. Slope Compensation
8.3.10 Soft Start
Upon power up, it is desirable to gradually widen the PWM pulse width starting at zero duty cycle. The
UCx84x devices do not have internal soft-start control, but this can be easily implemented externally with
three components. An R/C network is used to provide the time constant to control the error amplifier output. A
transistor is also used to isolate the components from the normal operation of either node. It also minimizes the
loading effects on the RT/CT time constant by amplification through the transistors gain.
VREF
RSS
COMP
CSS
Figure 8-12. Soft-Start Circuitry
8.3.11 Voltage Mode
In duty cycle control (voltage mode), pulse width modulation is attained by comparing the error amplifier output
to an artificial ramp. The oscillator timing capacitor CCT is used to generate a sawtooth waveform on both current
or voltage mode ICs. To use the UCx84x in a voltage mode configuration, this sawtooth waveform will be input
to the current sense input, ISENSE, for comparison to the error voltage at the PWM comparator. This sawtooth
is used to determine pulse width instead of the actual primary current in this method. Loop compensation
is similar to that of voltage mode controllers with subtle differences due to the low output resistance voltage
amplifier in the UCx84x as opposed to a transconductance (current) type amplifier used in traditional voltage
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mode controllers. For further reference on topologies and compensation, consult Closing the Feedback Loop
(SLUP068).
VREF
1N4148
2N2907
RT/CT
2N2222
2.7 k
ISENSE
1k
CCT
Figure 8-13. Current Mode PWM Used as a Voltage Mode PWM
8.4 Device Functional Modes
8.4.1 Normal Operation
During normal operating mode, the IC can be used in peak current mode or voltage mode control. When the
converter is operating in peak current mode, the controller regulates the converter's peak current and duty cycle.
When the IC is used in voltage mode control, the controller regulates the power converter's duty cycle. The
regulation of the system's peak current and duty cycle can be achieved with the use of the integrated error
amplifier and external feedback circuitry.
8.4.2 UVLO Mode
During the system start-up, VCC voltage starts to rise from 0 V. Before the VCC voltage reaches its
corresponding turn on threshold, the IC is operating in UVLO mode. In this mode, the VREF pin voltage is
not generated. When VCC is above 1 V and below the turnon threshold, the VREF pin is actively pulled low
through a 5-kΩ resistor. This way, VREF can be used as a logic signal to indicate UVLO mode. If the bias
voltage to VCC drops below the UVLO-off threshold, PWM switching stops and VREF returns to 0 V. The device
can be restarted by applying a voltage greater than the UVLO-on threshold to the VCC pin.
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9 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
9.1 Application Information
The UCx84x controllers are peak current mode pulse width modulators. These controllers have an onboard
amplifier and can be used in isolated and non-isolated power supply design. There is an onboard totem pole
gate driver capable of delivering 1 A of peak current. This is a high-speed PWM capable of operating at
switching frequencies up to 500 kHz.
9.1.1 Open-Loop Test Fixture
The following application is an open-loop laboratory test fixture. This circuit demonstrates the setup and use of
the UCx84x devices and their internal circuitry.
In the open-loop laboratory test fixture (see Figure 9-1), high peak currents associated with loads necessitate
careful grounding techniques. Timing and bypass capacitors should be connected close to the GROUND
terminal in a single-point ground. The transistor and 5-kΩ potentiometer sample the oscillator waveform and
apply an adjustable ramp to the ISENSE terminal.
VREF
R1
4.7 NŸ
UCx842
100 NŸ
1
VREF
8
VCC
1 NŸ
E/A
Adjust
COMP
2
VFB
VCC
7
1 NŸ
5 NŸ
3
ISENSE
OUTPUT
6
0.1 PF
ISENSE
Adjust
4.7 NŸ
OUTPUT
0.1 PF
4
RT/CT
GROUND
5
GROUND
CRTCT
Copyright © 2016, Texas Instruments Incorporated
Figure 9-1. Open-Loop Laboratory Test Fixture
9.2 Typical Application
A typical application for the UC2842 in an off-line flyback converter is shown in Figure 9-2. The UC2842 uses
an inner current control loop that contains a small current sense resistor which senses the primary inductor
current ramp. This current sense resistor transforms the inductor current waveform to a voltage signal that is
input directly into the primary side PWM comparator. This inner loop determines the response to input voltage
changes. An outer voltage control loop involves comparing a portion of the output voltage to a reference voltage
at the input of an error amplifier. When used in an off-line isolated application, the voltage feedback of the
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isolated output is accomplished using a secondary-side error amplifier and adjustable voltage reference, such as
the TL431. The error signal crosses the primary to secondary isolation boundary using an opto-isolator whose
collector is connected to the VREF pin and the emitter is connected to VFB. The outer voltage control loop
determines the response to load changes.
DCLAMP
~
VIN = 85 VAC
to 265 VAC
±
DBRIDGE
CSS
CSNUB
10 nF
RSNUB
50 k
DOUT
+
CIN
180 µF
~
RSTART
100 k
NS
NP
DBIAS
VOUT
12 V,
4A
COUT
2200 µF
RVCC
22
NA
CVCC
120 µF
RSS
LP =1.5 mH
NP:NS = 10
NP:NA = 10
UC2842
RCOMPp
10 k
1 COMP
CCOMPp
10 nF
VREF 8
2 VFB
3
RRT
15.4 k
VCC 7
ISENSE
QSW
4 RT/CT
CRAMP
10 nF
RG
10
OUTPUT 6
GROUND 5
CVCCbp
0.1 µF
CCT
1000 pF
CVREF
1 µF
RBLEEDER
10 k
RCSF
4.2 k
RRAMP
24.9 k
CCSF
100 pF
RP
Not Populated
RCS
0.75
RLED
1.3 k
RTLbias
1k
OPTOCOUPLER
10 V
RFBG
4.99 k
RCOMPz
88.7 k
RFBU
9.53 k
CCOMPz
0.01 µF
ROPTO
1k
TL431
RFBB
2.49 k
Copyright © 2016, Texas Instruments Incorporated
Figure 9-2. Typical Application Design Example Schematic
9.2.1 Design Requirements
Table 9-1 illustrates a typical set of performance requirements for an off-line flyback converter capable of
providing 48 W at 12-V output voltage from a universal AC input. The design uses peak primary current control
in a continuous current mode PWM converter.
Table 9-1. Performance Requirements
PARAMETER
TEST CONDITIONS
MIN
NOM
MAX
UNIT
85
115/230
265
VRMS
47
50/60
63
Hz
11.75
12
12.25
V
VIN
Input Voltage
fLINE
Line Frequency
VOUT
Output Voltage
IOUT(min) ≤ IOUT ≤ IOUT(max)
VRIPPLE
Output Ripple
Voltage
IOUT(min) ≤ IOUT ≤ IOUT(max)
IOUT
Output Current
fSW
Switching
Frequency
100
η
Efficiency
85%
22
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100
0
4
mVpp
A
kHz
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9.2.2 Detailed Design Procedure
This procedure outlines the steps to design an off-line universal input continuous current mode (CCM) flyback
converter using the UC2842. See Figure 9-2 for component names referred to in the design procedure.
9.2.2.1 Input Bulk Capacitor and Minimum Bulk Voltage
Bulk capacitance may consist of one or more capacitors connected in parallel, often with some inductance
between them to suppress differential-mode conducted noise. The value of the input capacitor sets the minimum
bulk voltage; setting the bulk voltage lower by using minimal input capacitance results in higher peak primary
currents leading to more stress on the MOSFET switch, the transformer, and the output capacitors. Setting the
bulk voltage higher by using a larger input capacitor results in higher peak current from the input source and
the capacitor itself will be physically larger. Compromising between size and component stresses determines
the acceptable minimum input voltage. The total required value for the primary-side bulk capacitance, CIN, is
selected based upon the power level of the converter, POUT, the efficiency target, η, the minimum input voltage,
VIN(min), and is chosen to maintain an acceptable minimum bulk voltage level, VBULK(min), using Equation 9.
CIN =
2 × PIN × F0.25 +
VBULK (min )
1
× arcsin F
GG
N
¾2 × VIN (min )
2
2
k2 × VIN
(min ) F VBULK (min ) o × fLINE (min )
(9)
In this equation, VIN(min) is the RMS value of the minimum AC input voltage, 85 VRMS, whose minimum line
frequency is denoted as fLINE(min), equal to 47 Hz. Based on the CIN equation, to achieve a minimum bulk voltage
of 75 V, assuming 85% converter efficiency, the bulk capacitor should be larger than 126 µF; 180 µF was chosen
for the design, taking into consideration component tolerances and efficiency estimation.
9.2.2.2 Transformer Turns Ratio and Maximum Duty Cycle
The transformer design starts with selecting a suitable switching frequency for the given application. The
UC2842 is capable of switching up to 500 kHz but considerations such as overall converter size, switching
losses, core loss, system compatibility, and interference with communication frequency bands generally
determine an optimum frequency that should be used. For this off-line converter, the switching frequency, fSW, is
selected to be 110 kHz as a compromise to minimize the transformer size and the EMI filter size, and still have
acceptable losses.
The transformer primary to secondary turns ratio, NPS, can be selected based on the desired MOSFET voltage
rating and the secondary diode voltage rating. Because the maximum input voltage is 265 VRMS, the peak bulk
input voltage can be calculated as shown in Equation 10.
VBULK (max ) = ¾2 × VIN (max ) N 375 V
(10)
To minimize the cost of the system, a readily available 650-V MOSFET is selected. Derating the maximum
voltage stress on the drain to 80% of its rated value and allowing for a leakage inductance voltage spike of up
to 30% of the maximum bulk input voltage, the reflected output voltage should be less than 130 V as shown in
Equation 11.
VREFLECTED
0.8 u VDS(rated) 1.3 u VBULK(max)
130.2 V
(11)
The maximum primary to secondary transformer turns ratio, NPS, for a 12 V output can be selected as
NPS =
VREFLECTED
= 10.85
VOUT
(12)
A turns ratio of NPS = 10 is used in the design example.
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The auxiliary winding is used to supply bias voltage to the UC2842. Maintaining the bias voltage above the VCC
minimum operating voltage after turn on is required for stabile operation. The minimum VCC operating voltage
for the UC2842 version of the controller is 10 V. The auxiliary winding is selected to support a 12-V bias voltage
so that it is above the minimum operating level but still keeps the losses low in the IC. The primary to auxiliary
turns ratio, NPA, can be calculated from Equation 13:
NPA = NPS ×
VOUT
= 10
VBIAS
(13)
The output diode experiences a voltage stress that is equal to the output voltage plus the reflected input voltage:
VDIODE =
VBULK :max ;
+ VOUT = 49.5 V
NPS
(14)
To allow for voltage spikes due to ringing, a Schottky diode with a rated blocking voltage of greater than 60 V is
recommended for this design. The forward voltage drop, VF, of this diode is estimated to be equal to 0.6 V
To avoid high peak currents, the flyback converter in this design operates in continuous conduction mode. Once
NPS has been determined, the maximum duty cycle, DMAX, can be calculated using the transfer function for a
CCM flyback converter:
DMAX
1
VOUT + VF
p×l
p
=l
NPS
1 F DMAX
VBULK :min ;
DMAX
NPS u VOUT
VF
VBULK(min) NPS u VOUT
VF
(15)
0.627
(16)
Because the maximum duty cycle exceeds 50%, and the design is an off-line (AC-input) application, the UC2842
is best suited for this application.
9.2.2.3 Transformer Inductance and Peak Currents
For this design example, the transformer magnetizing inductance is selected based upon the CCM condition. An
inductance value that allows the converter to stay in CCM over a wider operating range before transitioning into
discontinuous current mode is used to minimize losses due to otherwise high currents and also to decrease the
output ripple. The design of the transformer in this example sizes the inductance so the converter enters CCM
operation at approximately 10% load and minimum bulk voltage to minimize output ripple.
The inductor, LP for a CCM flyback can be calculated using Equation 17.
2
LP =
1
×
2
2
NPS × VOUT
p
VBULK :min ; + NPS × VOUT
0.1 × PIN × fSW
kVBULK :min ; o × l
(17)
In Equation 17, the input power, PIN, is estimated by dividing the maximum output power, POUT, by the target
efficiency, η, and fSW is the switching frequency; for the UC2842 the switching frequency is equal to the oscillator
frequency and is set to 110 kHz. Therefore, the transformer inductance should be approximately 1.8 mH; a
1.5-mH inductance is chosen as the magnetizing inductance value for this design.
Based on calculated inductor value and the switching frequency, the current stress of the MOSFET and output
diode can be calculated.
The peak current in the primary-side MOSFET of a CCM flyback can be calculated as shown in Equation 18.
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IPK MOSFET
NPS × VOUT
VBULK (min ) VBULK :min ; + :NPS × VOUT ;
PIN
=
+n
×
r
NPS × VOUT
2 × Lm
fSW
VBULK :min ; ×
VBULK :min ; + :NPS × VOUT ;
(18)
The MOSFET peak current is 1.36 A. The RMS current of the MOSFET is calculated to be 0.97 A as shown in
Equation 19. Therefore, IRFB9N65A is selected to be used as the primary-side switch.
IRM S MOSFET
DMAX 2 × IPK MOSFET × VBULK (min )
VBULK (min ) 2
DMAX 3
p FF
×l
G + kDMAX × IPK MOSFET 2 o
=¨
LP × fSW
LP × fSW
3
(19)
The output diode peak current is equal to the MOSFET peak current reflected to the secondary side.
IPK DIODE = NPS × IPK MOSFET = 13.634 A
(20)
The diode average current is equal to the total output current, 4 A; combined with a required 60-V rating and
13.6-A peak current requirement, a 48CTQ060-1 is selected for the output diode.
9.2.2.4 Output Capacitor
The total output capacitance is selected based upon the output voltage ripple requirement. In this design, 0.1%
voltage ripple is assumed. Based on the 0.1% ripple requirement, the capacitor value can be selected using
Equation 21.
NPS × VOUT
VBULK :min ; + NPS × VOUT
= 1865 JF
0.001 × VOUT × fSW
IOUT ×
COUT R
(21)
To design for device tolerances, a 2200-µF capacitor was selected.
9.2.2.5 Current Sensing Network
The current sensing network consists of the primary-side current sensing resistor, RCS, filtering components
RCSF and CCSF, and optional RP. Typically, the direct current sense signal contains a large amplitude leading
edge spike associated with the turnon of the main power MOSFET, reverse recovery of the output rectifier, and
other factors including charging and discharging of parasitic capacitances. Therefore, CCSF and RCSF form a
low-pass filter that provides immunity to suppress the leading edge spike. For this converter, CCSF is chosen to
be 100 pF.
Without RP, RCS sets the maximum peak current in the transformer primary based on the maximum amplitude of
the ISENSE pin, which is specified to be 1 V. To achieve 1.36-A primary side peak current, a 0.75-Ω resistor is
chosen for RCS.
The high current sense threshold of ISENSE helps to provide better noise immunity to the system but also
results in higher losses in the current sense resistor. These current sense losses can be minimized by injecting
an offset voltage into the current sense signal using RP. RP and RCSF form a resistor divider network from the
current sense signal to the device’s reference voltage, VREF, which adds an offset to the current sense voltage.
This technique still achieves current mode control with cycle-by-cycle over-current protection. To calculate
required offset value (VOFFSET), use Equation 22.
VOFFSET =
R CSF
× VREF
R CSF + R P
(22)
Once RP is added, adjust the current sense resistor, RCS, accordingly.
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9.2.2.6 Gate Drive Resistor
RG is the gate driver resistor for the power switch, QSW. The selection of this resistor value must be done in
conjunction with EMI compliance testing and efficiency testing. Using a larger resistor value for RG slows down
the turnon and turnoff of the MOSFET. A slower switching speed reduces EMI but also increases the switching
loss. A trade-off between switching loss and EMI performance must be carefully performed. For this design, a
10-Ω resistor was chosen for the gate drive resistor.
9.2.2.7 VREF Capacitor
A precision 5-V reference voltage performs several important functions. The reference voltage is divided down
internally to 2.5 V and connected to the error amplifier’s noninverting input for accurate output voltage regulation.
Other duties of the reference voltage are to set internal bias currents and thresholds for functions such as
the oscillator upper and lower thresholds. Therefore, the reference voltage must be bypassed with a ceramic
capacitor (CVREF), a 1-µF, 16-V ceramic capacitor was selected for this converter. Placement of this capacitor
on the physical printed-circuit board layout must be as close as possible to the respective VREF and GROUND
pins.
9.2.2.8 RT/CT
The internal oscillator uses a timing capacitor (CCT) and a timing resistor (RRT) to program the oscillator
frequency and maximum duty cycle. The operating frequency can be programmed based the curves in Section
9.2.3, where the timing resistor can be found once the timing capacitor is selected. It is best for the timing
capacitor to have a flat temperature coefficient, typical of most COG or NPO type capacitors. For this converter,
15.4 kΩ and 1000 pF were selected for RRT and CCT to operate at 110-kHz switching.
9.2.2.9 Start-Up Circuit
At start-up, the IC gets its power directly from the high-voltage bulk, through a high-voltage resistor RSTART.
The selection of the start-up resistor is the trade-off between power loss and start-up time. The current flowing
through RSTART at the minimum input voltage must be higher than the VCC current under UVLO conditions (1
mA at its maximum value). A resistance of 100-kΩ was chosen for RSTART, providing 1 mA of start-up current at
low-line conditions. The start-up resistor is physically comprised of two 50-kΩ resistors in series to meet the high
voltage requirements and power rating at high-line.
After VCC is charged up above the UVLO-on threshold, the UC2842 starts to consume full operating current.
The VCC capacitor is required to provide enough energy to prevent its voltage from dropping below the UVLOoff threshold during start-up, before the output is able to reach its regulated level. A large bulk capacitance would
hold more energy but would result in slower start-up time. In this design, a 120-µF capacitor is chosen to provide
enough energy and maintain a start-up time of approximately 2 seconds.
9.2.2.10 Voltage Feedback Compensation
Feedback compensation, also called closed-loop control, can reduce or eliminate steady state error, reduce
the sensitivity of the system to parametric changes, change the gain or phase of a system over some desired
frequency range, reduce the effects of small signal load disturbances and noise on system performance, and
create a stable system from an unstable system. A system is stable if its response to a perturbation is that the
perturbation eventually dies out. A peak current mode flyback uses an outer voltage feedback loop to stabilize
the converter. To adequately compensate the voltage loop, the open-loop parameters of the power stage must
be determined.
9.2.2.10.1 Power Stage Poles and Zeroes
The first step in compensating a fixed frequency flyback is to verify if the converter is continuous conduction
mode (CCM) or discontinuous conduction mode (DCM). If the primary inductance, LP, is greater than the
inductance for DCM/CCM boundary mode operation, called the critical inductance, or LPcrit, then the converter
operates in CCM:
LP > LPcrit , then CCM
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2
R OUT × :NPS ;2
VIN
p
=
×l
2 × fSW
VIN + VOUT × NPS
LPcrit
(24)
For the entire input voltage range, the selected inductor has value larger than the critical inductor. Therefore, the
converter operates in CCM and the compensation loop requires design based on CCM flyback equations.
The current-to-voltage conversion is done externally with the ground-referenced current sense resistor, RCS, and
the internal resistor divider of 2R/R which sets up the internal current sense gain, ACS = 3. Note that the exact
value of these internal resistors is not critical but the IC provides tight control of the resistor divider ratio, so
regardless of the actual resistor value variations their relative value to each other is maintained.
The DC open-loop gain, GO, of the fixed-frequency voltage control loop of a peak current mode control CCM
flyback converter shown in Equation 25 is approximated by first using the output load, ROUT, the primary to
secondary turns ratio, NPS, the maximum duty cycle, D, calculated in Equation 25.
GO =
R OUT × NPS
1
×
2
:1 F D;
R CS × ACS
+ :2 × M; + 1
RL
(25)
In Equation 25, D is calculated with Equation 26, τL is calculated with Equation 27, and M is calculated with
Equation 28.
D=
NPS × VOUT
VBULKmin + :NPS × VOUT ;
RL =
M=
(26)
2 × LP × fSW
R OUT × :NPS ;2
(27)
VOUT × NPS
VBULKmin
(28)
For this design, a converter with an output voltage VOUT of 12 V, and 48 W relates to an output load, ROUT, equal
to 3 Ω at full load. With a maximum duty cycle calculated to be 0.627, a current sense resistance, RCS, of 0.75 Ω,
and a primary to secondary turns-ratio, NPS, of 10, the open-loop gain calculates to 3.082, or 9.776 dB.
A CCM flyback has two zeroes that are of interest. The ESR and the output capacitance contribute a left-half
plane zero, ωESRz, to the power stage, and the frequency of this zero, fESRz, are calculated with Equation 30.
XESRz =
R ESR
1
× COUT
(29)
1
fESRz =
2 × N × R ESR × COUT
(30)
The fESRz zero for an output capacitance of 2200 µF and a total ESR of 43 mΩ is located at 1.682 kHz.
CCM flyback converters have a zero in the right-half plane, RHP, in their transfer function. A RHP zero has the
same 20 dB/decade rising gain magnitude with increasing frequency just like a left-half plane zero, but it adds a
90° phase lag instead of lead. This phase lag tends to limit the overall loop bandwidth. The frequency location,
fRHPz, of the RHP zero, ωRHPz, is a function of the output load, the duty cycle, the primary inductance, LP, and
the primary to secondary side turns ratio, NPS.
XRHPz
R OUT × :1 F D;2 × :NPS ;2
=
LP × D
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fRHPz =
R OUT × :1 F D;2 × :NPS ;2
2 × N × LP × D
(32)
The right-half plane zero frequency increases with higher input voltage and lighter load. Generally, the design
requires consideration of the worst case of the lowest right-half plane zero frequency and the converter must be
compensated at the minimum input and maximum load condition. With a primary inductance of 1.5 mH, at 75-V
DC input, the RHP zero frequency, fRHPz, is equal to 7.07 kHz at maximum duty cycle, full load.
The power stage has one dominate pole, ωP1, which is in the region of interest, located at a lower frequency, fP1,
which is related to the duty cycle, D, the output load, and the output capacitance, calculated with Equation 34.
There is also a double pole placed at half the switching frequency of the converter, fP2 calculated with Equation
36. For this example, pole fP1 is located at 40.37 Hz and fP2 is at 55 kHz.
XP1
fP1
:1 F D;3
+1+D
RL
=
R OUT × COUT
(33)
:1 F D;3
+1+D
RL
=
2 × N × R OUT × COUT
(34)
XP2 = N × fSW
(35)
fSW
2
(36)
fP2 =
9.2.2.10.2 Slope Compensation
Slope compensation is the large signal sub-harmonic instability that can occur with duty cycles that may extend
beyond 50% where the rising primary side inductor current slope may not match the falling secondary side
current slope. The sub-harmonic oscillation would result in an increase in the output voltage ripple and may even
limit the power handling capability of the converter.
The target of slope compensation is to achieve an ideal quality coefficient, QP , to be equal to 1 at half of the
switching frequency. The QP is calculated with Equation 37.
QP =
1
N × >MC × :1 F D; F 0.5?
(37)
In Equation 37, D is the primary side switch duty cycle and MC is the slope compensation factor, which is defined
with Equation 38.
MC =
Se
+1
Sn
(38)
In Equation 38, Se is the compensation ramp slope and the Sn is the inductor rising slope. The optimal goal
of the slope compensation is to achieve QP equal to 1; upon rearranging Equation 38 the ideal value of slope
compensation factor is determined:
Mideal
1
+ 0.5
N
=
1FD
(39)
For this design to have adequate slope compensation, MC must be 2.193 when D reaches it maximum value of
0.627.
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The inductor rising slope, Sn, at the ISENSE pin is calculated with Equation 40.
Sn =
VINmin × R CS
V
= 0.038
Js
LP
(40)
The compensation slope, Se, is calculated with Equation 41.
Se = :MC F 1; × Sn = 44.74
mV
Js
(41)
The compensation slope is added into the system through RRAMP and RCSF. The CRAMP is an AC-coupling
capacitor that allows the voltage ramp of the oscillator to be used without adding an offset to the current
sense; select a value to approximate high frequency short circuit, such as 10 nF as a starting point and make
adjustments if required. The RRAMP and RCSF resistors form a voltage divider from the oscillator charge slope
and this proportional ramp is injected into the ISENSE pin to add slope compensation. Choose the value of
RRAMP to be much larger than the RRT resistor so that it does not load down the internal oscillator and result in a
frequency shift. The oscillator charge slope is calculated using the peak-to-peak voltage of the RT/CT sawtooth
waveform, VOSCpp, equal to 1.7 V, and the minimum on-time, as shown in Equation 43.
D
t ONmin =
SOSC =
fSW
(42)
VOSCpp
1.7 V
mV
=
= 298
5.7 Js
Js
t ONmin
(43)
To achieve a 44.74-mV/µs compensation slope, RCSF resistor is calculated with Equation 44. In this design,
RRAMP is selected as 24.9 kΩ, a 4.2-kΩ resistor was selected for RCSF.
R CSF =
R RAMP
SOSC
F1
Se
(44)
9.2.2.10.3 Open-Loop Gain
Once the power stage poles and zeros are calculated and the slope compensation is determined, the power
stage open-loop gain and phase of the CCM flyback converter can be plotted as a function of frequency. The
power stage transfer function can be characterized with Equation 45.
HOPEN :s; = G0 ×
l1 +
s:f;
s:f;
p × l1 F
p
1
XESRz
XRHPz
×
s:f;
s:f;
s:f;2
1+
1
+
+
XP1
XP2 × Q P :XP2 ;2
(45)
The bode for the open-loop gain and phase can be plotted by using Equation 46.
GainOPEN :s; = 20 × log: HOPEN :s; ;
(46)
(see Figure 9-3 and Figure 9-4).
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10
0
5
-45
Phase (q)
Gain (dB)
0
-5
-10
-15
-90
-135
-20
-180
-25
1
10
100
1000
frequency (Hz)
10000
100000
D001
Figure 9-3. Converter Open-Loop Bode Plot - Gain
1
10
100
1000
frequency (Hz)
10000
100000
D002
Figure 9-4. Converter Open-Loop Bode Plot Phase
9.2.2.10.4 Compensation Loop
The design of the compensation loop involves selecting the appropriate components so that the required gain,
poles, and zeros can be designed to result in a stabile system over the entire operating range. There are
three distinct portions of the loop: the TL431, the opto-coupler, and the error amplifier. Each of these stages is
combined with the power stage to result in a stable robust system.
For good transient response, the bandwidth of the finalized design should be as large as possible. The
bandwidth of a CCM flyback, fBW, is limited to ¼ of the RHP zero frequency, or approximately 1.77 kHz using
Equation 47.
fBW =
fRHPz
4
(47)
The gain of the open-loop power stage at fBW can be calculated using Equation 46 or can be observed on the
Bode plot (Figure 9-3 ) and is equal to –19.55 dB and the phase at fBW is equal to –58°.
The secondary side portion of the compensation loop begins with establishing the regulated steady state output
voltage. To set the regulated output voltage, a TL431 adjustable precision shunt regulator is ideally suited for
use on the secondary side of isolated converters due to its accurate voltage reference and internal op amp.
The resistors used in the divider from the output terminals of the converter to the TL431 REF pin are selected
based upon the desired power consumption. Because the REF input current for the TL431 is only 2 µA, selecting
the resistors for a divider current, IFB_REF, of 1 mA results in minimal error. The top divider resistor, RFBU, is
calculated using Equation 48:
R FBU =
VOUT F REFTL431
IFB _REF
(48)
The TL431 reference voltage, REFTL431, has a typical value of 2.495 V. A 9.53-kΩ resistor is chosen for RFBU. To
set the output voltage to 12 V, 2.49 kΩ is used for RFBB.
R FBB =
REFTL431
× R FBU
VOUT F REFTL431
(49)
For good phase margin, a compensator zero, fCOMPz, is needed and should be placed at 1/10th the desired
bandwidth:
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fCOMPz =
fBW
10
(50)
XCOMPz = 2 × N × fCOMPz
(51)
With this converter, fCOMPz should be set at approximately 177 Hz. A series resistor, RCOMPz, and capacitor,
CCOMPz, placed across the TL431 cathode to REF sets the compensator zero location. Setting CCOMPz to 0.01
µF, RCOMPz is calculated using Equation 52:
R COMPz =
XCOMPz
1
× CCOMPz
(52)
Using a standard value of 88.7 kΩ for RZ and a 0.01 µF for CZ results in a zero placed at 179 Hz.
Referring to Figure 9-2, RTLbias provides cathode current to the TL431 from the regulated voltage provided from
the Zener diode, DREG. For robust performance, 10 mA is provided to bias the TL431 by way of the 10-V Zener
and 1-kΩ resistor is used for RTLbias.
The gain of the TL431 portion of the compensation loop can be written as:
GTL431 :s; = lR COMPz +
1
1
p×
R FBU
s(f) × CZCOMPz
(53)
A compensation pole is needed at the frequency of right half plane zero or the ESR zero, whichever is lowest.
Based previous the analysis, the right half plane zero, fRHPz, is located at 7.07 kHz and the ESR zero, fESRz, is at
1.68 kHz; therefore, for this design, the compensation pole must be put at 1.68 kHz. The opto-coupler contains a
parasitic pole that is difficult to characterize over frequency so the opto-coupler is set up with a pulldown resistor,
ROPTO equal to 1 kΩ, which moves the parasitic opto-coupler pole further out and beyond the range of interest
for this design.
The required compensation pole can be added to the primary side error amplifier using RCOMPp and CCOMPp.
Choosing RCOMPp as 10 kΩ, the required value of CCOMPp is determined using Equation 54.
CCOMPp =
1
2 × N × fESRz × R COMPp
= 9.46 nF
(54)
A 10-nF capacitor is used for CCOMPp setting the compensation pole at 1.59 kHz.
Adding a DC gain to the primary side error amplifier may be required to obtain the required bandwidth and
helps to adjust the loop gain as needed. Using a 4.99 kΩ for RFBG sets the DC gain on the error amplifier to
2. At this point the gain transfer function of the error amplifier stage, GEA(s), of the compensation loop can be
characterized:
GEA :s; = l
R COMPp
1
p×F
G
1 + s:f; × CCOMPp × R COMPp
R FBG
(55)
Using an opto-coupler whose current transfer ratio (CTR) is typically at 100% in the frequency range of interest
so that CTR = 1, the transfer function of the opto-coupler stage, GOPTO(s), is equal to:
GOPTO (s) =
CTR × R OPTO
R LED
(56)
The bias resistor, RLED, to the internal diode of the opto-coupler, and the pulldown resistor on the opto emitter,
ROPTO, sets the gain across the isolation boundary. ROPTO has already been set to 1 kΩ but the value of RLED
has not yet been determined.
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SLUS223G – APRIL 1997 – REVISED JULY 2022
The total closed-loop gain, GTOTAL(s), is the combination of the open-loop power stage, Ho(s), the opto gain,
GOPTO(s), the error amplifier gain, GEA(s), and the gain of the TL431 stage, GTL431(s):
GTOTAL :s; = HOPEN :s; × GOPTO :s; × GEA :s; × GTL431 :s;
(57)
The required value for RLED can be selected to achieve the desired crossover frequency, fBW. By setting the total
loop gain equal to 1 at the desired crossover frequency and rearranging Equation 57, the optimal value for RLED
can be determined, as shown in Equation 58.
R LED Q HOPEN :s; × CTR × COPTO × GEA :s; × GTL431 :s;
(58)
A 1.3-kΩ resistor suits the requirement for RLED.
Based on the compensation loop structure, the entire compensation loop transfer function is written as Equation
59.
GCLOSED :s; = HOPEN :s; × l
×n
R COMPz
R COMPp
CTR × R OPTO
1
p×l
p×F
G
R LED
R FBG
1 + ks × CCOMPp × R COMPp o
1
A
+@
s × CCOMPz
r
R FBU
(59)
The final closed-loop bode plots are show in Figure 9-5 and Figure 9-6. The converter achieves a crossover
frequency of approximately 1.8 kHz and has a phase margin of approximately 67o.
TI recommends checking the loop stability across all the corner cases including component tolerances to ensure
system stability.
80
0
60
-45
Degrees (q)
Gain (dB)
40
20
-90
0
-135
-20
-180
-40
1
10
100
1000
frequency (Hz)
10000
100000
D003
Figure 9-5. Converter Closed-Loop Bode Plot –
Gain
32
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1
10
100
1000
frequency (Hz)
10000
100000
D001
D004
Figure 9-6. Converter Closed-Loop Bode Plot –
Phase
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9.2.3 Application Curves
Figure 9-7. Primary Side MOSFET Drain to Source
Voltage at 240-V AC Input (100 V/div)
Figure 9-8. Primary Side MOSFET Drain to Source
Voltage at 120-V AC Input (100 V/div)
Figure 9-9. Output Voltage During 0.9-A to 2.7-A
Load Transient (CH1: Output Voltage AC Coupled,
200 mV/div; CH4: Output Current, 1 A/div)
Figure 9-10. Output Voltage Ripple at Full Load
(100 mV/div)
Figure 9-11. Output Voltage Behavior at Full Load Start-Up (5 V/div)
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10 Power Supply Recommendations
It is important to bypass the ICs supply (VCC) and reference voltage (VREF) pins with a 0.1-µF to 1-µF ceramic
capacitor to ground. The capacitors must be placed as close to the actual pin connections as possible for optimal
noise filtering. A second, larger filter capacitor may also be required in offline applications to hold the supply
voltage (VCC) above the UVLO turnoff threshold during start-up.
To prevent false triggering due to leading edge noises, an RC current sense filter may be required on ISENSE.
Keep the time constant of the RC filter well below the minimum on-time pulse width.
Schottky diodes may be necessary on the OUTPUT pin to prevent overshoot and undershoot due to the high
impedance to the supply rail and to ground, respectively. A bleeder resistor, placed between the gate and the
source of the MOSFET should be used to prevent activating the power switch with extraneous leakage currents
during undervoltage lockout.
To prevent noise problems with high-speed switching transients, bypass VREF to ground with a ceramic
capacitor close to the IC package. A minimum of 0.1-µF ceramic capacitor is required. Additional VREF
bypassing is required for external loads on the reference. An electrolytic capacitor may also be used in addition
to the ceramic capacitor.
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SLUS223G – APRIL 1997 – REVISED JULY 2022
11 Layout
11.1 Layout Guidelines
11.1.1 Feedback Traces
Try to run the feedback trace as far from the inductor and noisy power traces as possible. Be as direct as
possible with the feedback trace and somewhat thick. These two sometimes involve a trade-off, but keeping it
away from EMI and other noise sources is the more critical of the two. If possible, run the feedback trace on the
side of the PCB opposite of the inductor with a ground plane separating the two.
11.1.2 Bypass Capacitors
When using a low value ceramic bypass capacitor, it should be located as close to the VCC pin of the device
as possible. This eliminates as much trace inductance effects as possible and gives the internal device rail a
cleaner voltage supply. Using surface mount capacitors also reduces lead length and lessens the chance of
noise coupling into the effective antenna created by through-hole components.
11.1.3 Compensation Components
For best stability, external compensation components should be placed close to the IC. Keep VFB lead
length as short as possible and VFB stray capacitance as small as possible. Surface mount components are
recommended here as well for the same reasons discussed for the filter capacitors. These should not be located
very close to traces with high switching noise.
11.1.4 Traces and Ground Planes
Make all of the power (high current) traces as short, direct, and thick as possible. It is good practice on a
standard PCB board to make the traces an absolute minimum of 15 mils (0.381 mm) per Ampere. The inductor,
output capacitors, and output diode should be as close to each other possible. This helps reduce the EMI
radiated by the power traces due to the high switching currents through them. This also reduces lead inductance
and resistance as well, which in turn reduces noise spikes, ringing, and resistive losses that produce voltage
errors.
The grounds of the IC, input capacitors, output capacitors, and output diode (if applicable) should be connected
close together directly to a ground plane. It would also be a good idea to have a ground plane on both sides
of the PCB. This reduces noise as well by reducing ground loop errors as well as by absorbing more of the
EMI radiated by the inductor. For multi-layer boards with more than two layers, a ground plane can be used
to separate the power plane (where the power traces and components are) and the signal plane (where the
feedback and compensation and components are) for improved performance. On multi-layer boards the use of
vias is required to connect traces and different planes. It is good practice to use one standard via per 200 mA of
current if the trace needs to conduct a significant amount of current from one plane to the other.
Arrange the components so that the switching current loops curl in the same direction. Due to the way switching
regulators operate, there are two power states. One state when the switch is on and one when the switch is off.
During each state there is a current loop made by the power components that are currently conducting. Place
the power components so that during each of the two states the current loop is conducting in the same direction.
This prevents magnetic field reversal caused by the traces between the two half-cycles and reduces radiated
EMI.
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11.2 Layout Example
MOSFET Heatsink
Track To
FBead
D
½ PRI Winding
RSNUB
CSNUB
Track To
CCT
Aux Cap
1
CVCC1
ROPTO
22AWG Jumper
Wires
E
K
OPTO-ISOLATOR
C
A
PCB Bottom-side View
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Figure 11-1. UCx84x Layout Example
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12 Device and Documentation Support
12.1 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
12.2 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
12.3 Trademarks
TI E2E™ is a trademark of Texas Instruments.
All trademarks are the property of their respective owners.
12.4 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
12.5 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, see the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
5962-8670401PA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670401PA
UC1842
Samples
5962-8670401VPA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670401VPA
UC1842
Samples
5962-8670401XA
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670401XA
UC1842L/
883B
5962-8670402PA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670402PA
UC1843
5962-8670402XA
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670402XA
UC1843L/
883B
5962-8670403PA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670403PA
UC1844
5962-8670403VXA
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670403VXA
UC1844L
QMLV
5962-8670403XA
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670403XA
UC1844L/
883B
5962-8670404DA
ACTIVE
CFP
W
14
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
5962-8670404PA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
5962-8670404VPA
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
5962-8670404VXA
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
Addendum-Page 1
-55 to 125
-55 to 125
Samples
Samples
Samples
Samples
Samples
Samples
5962-8670404DA
UC1845W/883B
Samples
8670404PA
UC1845
Samples
8670404VPA
UC1845
Samples
59628670404VXA
UC1845L
QMLV
Samples
PACKAGE OPTION ADDENDUM
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Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
5962-8670404XA
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670404XA
UC1845L/
883B
UC1842J
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1842J
Samples
UC1842J883B
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670401PA
UC1842
Samples
UC1842L883B
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670401XA
UC1842L/
883B
UC1842W
ACTIVE
CFP
W
14
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1842W
Samples
UC1843J
ACTIVE
CDIP
JG
8
50
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1843J
Samples
UC1843J883B
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670402PA
UC1843
Samples
UC1843L
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1843L
Samples
UC1843L883B
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670402XA
UC1843L/
883B
UC1844J
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1844J
Samples
UC1844J883B
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670403PA
UC1844
Samples
UC1844L883B
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670403XA
UC1844L/
883B
UC1845J
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1845J
Samples
UC1845J883B
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
8670404PA
UC1845
Samples
Addendum-Page 2
Samples
Samples
Samples
Samples
PACKAGE OPTION ADDENDUM
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Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UC1845L
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1845L
UC1845L883B
ACTIVE
LCCC
FK
20
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
59628670404XA
UC1845L/
883B
UC1845W
ACTIVE
CFP
W
14
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
-55 to 125
UC1845W
Samples
UC1845W883B
ACTIVE
CFP
W
14
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
5962-8670404DA
UC1845W/883B
Samples
UC2842D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2842D
Samples
UC2842D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2842
Samples
UC2842D8G4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2842
Samples
UC2842D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2842
Samples
UC2842DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2842D
Samples
UC2842N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2842N
Samples
UC2842NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2842N
Samples
UC2843D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843D
Samples
UC2843D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843
Samples
UC2843D8G4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843
Samples
UC2843D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843
Samples
UC2843D8TRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843
Samples
UC2843DG4
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843D
Samples
UC2843DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2843D
Samples
UC2843N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2843N
Samples
Addendum-Page 3
Samples
Samples
PACKAGE OPTION ADDENDUM
www.ti.com
17-Nov-2022
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UC2843NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2843N
Samples
UC2844D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2844D
Samples
UC2844D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2844
Samples
UC2844D8G4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2844
Samples
UC2844D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2844
Samples
UC2844DG4
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2844D
Samples
UC2844DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2844D
Samples
UC2844N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2844N
Samples
UC2844NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2844N
Samples
UC2845D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845D
Samples
UC2845D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845
Samples
UC2845D8G4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845
Samples
UC2845D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845
Samples
UC2845D8TRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845
Samples
UC2845DG4
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845D
Samples
UC2845DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 85
UC2845D
Samples
UC2845N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2845N
Samples
UC2845NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
-40 to 85
UC2845N
Samples
UC3842D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3842D
Samples
UC3842D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3842
Samples
UC3842D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3842
Samples
Addendum-Page 4
PACKAGE OPTION ADDENDUM
www.ti.com
17-Nov-2022
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UC3842DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3842D
Samples
UC3842N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3842N
Samples
UC3842NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3842N
Samples
UC3843D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843D
Samples
UC3843D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843
Samples
UC3843D8G4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843
Samples
UC3843D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843
Samples
UC3843D8TRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843
Samples
UC3843DG4
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843D
Samples
UC3843DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3843D
Samples
UC3843N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3843N
Samples
UC3843NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3843N
Samples
UC3844D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3844D
Samples
UC3844D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3844
Samples
UC3844D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3844
Samples
UC3844DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3844D
Samples
UC3844DTRG4
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3844D
Samples
UC3844N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3844N
Samples
UC3844NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3844N
Samples
UC3845AJ
ACTIVE
CDIP
JG
8
1
Non-RoHS
& Green
SNPB
N / A for Pkg Type
0 to 70
UC3845AJ
Samples
UC3845D
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845D
Samples
Addendum-Page 5
PACKAGE OPTION ADDENDUM
www.ti.com
17-Nov-2022
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UC3845D8
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845
Samples
UC3845D8G4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845
Samples
UC3845D8TR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845
Samples
UC3845D8TRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845
Samples
UC3845DG4
ACTIVE
SOIC
D
14
50
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845D
Samples
UC3845DTR
ACTIVE
SOIC
D
14
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
UC3845D
Samples
UC3845N
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3845N
Samples
UC3845NG4
ACTIVE
PDIP
P
8
50
RoHS & Green
NIPDAU
N / A for Pkg Type
0 to 70
UC3845N
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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