UCC2583D

UCC1583 UCC2583 UCC3583 Switch Mode Secondary Side Post Regulator FEATURES DESCRIPTION • Precision Secondary Side Post Regulation for Multiple Output Power Supplies The UCC3583 is a synchronizable secondary side post regulator for precision regulation of the auxiliary outputs of multiple output power supplies. It contains a leading edge pulse width modulator, which generates the gate drive signal for a FET power switch connected in series with the rectifying diode. The turn-on of the power switch is delayed from the leading edge of the secondary power pulse to regulate the output voltage. The UCC3583 contains a ramp generator slaved to the secondary power pulse, a voltage error amplifier, a current error amplifier, a PWM comparator and associated logic, a gate driver, a precision reference, and protection circuitry. • Useful for Both Single Ended and Center Tapped Secondary Circuits • Ideal Replacement for Complex Magnetic Amplifier Regulated Circuits • Leading Edge Modulation • Does Not Require Gate Drive Transformer • High Frequency (>500kHz) Operation • Applicable for Wide Range of Output Voltages • High Current Gate Driver (0.5A Sink/1.5A Source) • Average Current Limiting Loop The ramp discharge and termination of the gate drive signal are triggered by the synchronization pulse, typically derived from the falling edge of the transformer secondary voltage. The ramp starts charging again once its low threshold is reached. The gate drive signal is turned on when the ramp voltage exceeds the control voltage. This leading edge modulation technique prevents instability when the UCC3583 is used in peak current mode primary controlled systems. The controller operates from a floating power supply referenced to the output voltage being controlled. It features an undervoltage lockout (UVLO) circuit, a soft start circuit, and an averaging current limit amplifier. The current limit can be programmed to be proportional to the output voltage, thus achieving foldback operation to minimize the dissipation under short circuit conditions. (continued) TYPICAL APPLICATION AND BLOCK DIAGRAM Note: Pin connections shown for 14-pin packages. SLUS299B - OCTOBER 1998 - REVISED JANUARY 2005 UDG-96201-2 UCC1583 UCC2583 UCC3583 CONNECTION DIAGRAMS ABSOLUTE MAXIMUM RATINGS VDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15V IVDD . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15mA RAMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3V to VDD + 1V IRAMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5mA IREF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –30mA PCOM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.2V to 0.2V IGATE (twp < 1µS and Duty Cycle < 10%) . . . . . . –0.8A to 1.8A ICOMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –5mA to 5mA ICAO . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –5mA to 5mA VSYNC . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.6V to VREF +0.3V ISYNC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –05mA to 5mA INV, SS, ILIM, ISENSE. . . . . . . . . . . . . . –0.3V to VREF + 0.3V Storage Temperature . . . . . . . . . . . . . . . . . . . –65°C to +150°C Junction Temperature . . . . . . . . . . . . . . . . . . . –55°C to +150°C Lead Temperature (Soldering, 10 sec.) . . . . . . . . . . . . . +300°C DIL-14, SOIC-14 (Top View) J, N, or D Packages All voltages are with respect to the COM terminal unless otherwise stated. Currents are positive into, negative out of the specified terminal. Consult Packaging Section of Databook for thermal limitations and considerations of packages. PLCC-20 (Top View) Q Package THERMAL IMPEDANCE qja 90 90-120 50-120 43-75 PACKAGE N-14 J-14 D-14 PLCC-20 qjc 45 28 35 34 Note 1. qja (junction to ambient) is for devices mounted to 5 in2 FR4 PC board with one ounce copper where noted. When resistance range is given, lower values are for 5 in2 aluminum PC board. Test PWB was .062 in thick and typically used 0.635 mm trace widths for power pkgs and 1.3 mm trace widths for non-power pkgs with a 100x100 mil probe land area at the end of each trace Note 2. qjc data values stated were derived from MIL-STD-1835B. MIL-STD-1835B states that “The baseline values shown are worst case (mean + 2s) for a 60x60 mil microcircuit device silicon die and applicable for devices with die sizes up to 14400 square mils. For device die sizes greater than14400 square mils use the following values; dual-in-line, 11°C/W; flat pack, 10°C/W; pin grid array, 10°C/W”. ELECTRICAL CHARACTERISTICS:Unless otherwise stated, these specifications apply for TA = –55°C to 125°C for UCC1583, –40°C to 85°C for UCC2583, and 0°C to 70°C for UCC3583; VDD = 12V, RT = 60k, CT = 100pF, TA = TJ. PARAMETER TEST CONDITIONS MIN TYP MAX UNITS Ramp Generation and Synchronization Maximum Input Operating Frequency Ramp Frequency, Free Running For input with 5% to 90% duty cycle (Note 1) 500 kHz TA = 25°C 95 100 105 kHz TA = -55°C to 125°C 90 100 110 kHz Ramp Discharge Current VRAMP = 0.5V 2.0 3.6 Low Threshold Voltage No min, no max, 0=TYP High Threshold Voltage Synchronizing Threshold Voltage (On) 3.75 (Note 1) Synchronizing Comparator Hysteresis 2 mA 0 4 V 4.25 V 1 V 1 V UCC1583 UCC2583 UCC3583 ELECTRICAL CHARACTERISTICS:Unless otherwise stated, these specifications apply for TA = –55°C to 125°C for UCC1583, –40°C to 85°C for UCC2583, and 0°C to 70°C for UCC3583; VDD = 12V, RT = 60k, CT = 100pF, TA = TJ. PARAMETER TEST CONDITIONS MIN TYP MAX UNITS 0 % Output Duty Cycle Minimum Duty Cycle Output D/C = Output PW / Input PW Maximum Duty Cycle Output D/C = Output PW / Input PW 100 % Voltage Error Amplifier VINV VCOMP = VINV, 0°C to 70°C (UCC3583) 2.462 VCOMP = VINV, All Other Temperature Ranges 2.45 IINV VCOMP = VINV VCOMP Low VINV = 2.6V, ICOMP = 100µA 2.5 2.538 V 2.5 2.55 V 300 500 nA 450 700 mV 6.0 VCOMP High VINV = 2.4V, ICOMP = –100µA 5.0 5.5 AVOL No Load 70 90 dB GBW Product At f = 100kHz, TA = 25°C (Note 1) 3 5 MHz V Current Error Amplifier Input Offset Voltage Input CM Low Common Mode for CS and ILIM (Note 1) Input CM High Common Mode for CS and ILIM (Note 1) VCAO Low V+IN = 0V, V–IN = 0.1V, ICAO = 100µA VCAO High V+IN = 0V, V–IN = 0.1V, ICAO = –100µA Input Current (ILIM and CS Pins) 10 mV 0 V 2 5.0 V 250 500 mV 5.5 6.0 V 50 nA –50 0 AVOL No Load 70 90 dB GBW Product At f = 100kHz, TA = 25°C 2 4 MHz Soft Start Current 10 25 µA V UVLO VDD On Threshold Voltage 8.5 9.0 9.5 VDD Off Threshold Voltage 7.9 8.4 8.9 V UVLO Hysteresis 0.3 0.6 0.9 V 13 14 15 V 3 5 mA V Bias Supply Supply Clamp Voltage Supply Current (VDD) f = 100kHz With No Gate Output Load Output Driver VSAT High IGATE = –150mA 0.6 1.0 VSAT Low IGATE = 50mA 0.4 0.75 V Rise Time CGATE = 1nF 50 75 ns Fall Time CGATE = 330pF 20 40 ns V Reference VREF IREF = 0, 0°C to 70°C (UCC3583) 4.925 5 5.075 IREF = 0, All Other Temperature Ranges 4.900 5 5.100 V Line Regulation VDD = 10V to 14V 2 30 mV Load Regulation IREF = 0mA to 2mA 1 20 mV Note 1: Ensured by design. Not 100% tested in production. 3 UCC1583 UCC2583 UCC3583 PIN DESCRIPTIONS CAO: Output of the current error amplifier. Averaging of the sensed current signal is provided by connecting an integrating capacitor between ILIM and CAO. CAO feeds into the PWM comparator input and controls the loop when its voltage is higher than the voltage at COMP (output of the voltage error amplifier). 1mA discharges RAMP when synchronization signal appears or when RAMP crosses a 4V threshold. In the intended mode of operation, the switching frequency is determined by the secondary power pulse. The RC components at RAMP should be selected to give an appropriately sized ramp signal. In the absence of a synchronizing pulse, these RC components determine the free running frequency of the controller. COM: Signal ground for the chip. It is connected to the positive terminal of the output voltage being regulated by the IC. REF: Precision 5V reference pin. REF stays off until VDD exceeds 9V and turns off again when VDD drops below 8.4V. Bypass REF to COM. COMP: Output of the voltage error amplifier fed into the PWM comparator. Loop compensation components are connected between COMP and INV. SS: This pin provides a soft start function. A capacitor to REF programs the soft start time. During soft start, the PWM comparator is controlled by the soft start voltage resulting in a slow increase in output duty cycle. Once the soft start capacitor is discharged, output control is dictated by the larger of the output at CAO or COMP. CS: Non-inverting input of the current error amplifier. The sensed current signal from the current sense resistor is connected to this pin. By making the signal at CS proportional to the output voltage, effective current foldback limiting can be provided. SYNC: Synchronization input pin. It is connected to a signal representative of the secondary power pulse. One possible implementation is to use a resistive divider between terminal S2 of the secondary winding shown in Figure 1 and REF for generating the input to the SYNC pin. The synchronizing comparator is referenced to 0.5V and has ±500mV of hysteresis. The trip levels are approximate 1.0V and 0.0V. The designer should prevent the SYNC pin from exceeding 0.3V below ground as this will turn on the ESD diode. GATE: Gate drive output for the power switch FET. The drive pin has a 0.5A sink/1.5A source capability and very low output off-state impedance. ILIM: Inverting input of the current error amplifier. It sets the DC limit for the output current. INV: Inverting input of the voltage error amplifier. The feedback signal is connected to this pin using a resistive divider between REF and –VO. PCOM: Power ground for the chip. It is connected to the source terminal of the MOSFET being regulated by the IC. VD: Power supply for the output driver. VD should be tied to VDD in the application. VDD: Power supply for the chip. VDD should be bypassed to COM. VDD has to be 9V for the IC to start and 8.4V for it to remain operational. A shunt clamp from VDD to COM limits the supply voltage to 14V. RAMP: This pin is the input to the PWM comparator and provides a ramp signal for generation of the PWM signal. A capacitor to COM and a resistor to REF set the charging rate for the ramp. An internal current source of APPLICATION INFORMATION Power Stage Circuit Configuration and terminations easier to implement. Typical set-up and circuit waveforms of the UCC3583 system application are shown in Figure 1. Figure 2 shows waveforms for a single ended output rectifier application of the UCC3583 shown on page 1. The UCC3583 can also be used in half bridge rectifier applications as shown by the circuit and waveforms depicted in Figures 3 and 4. Referencing the IC to the positive output terminal creates a requirement for a floating bias voltage for the IC which can be referenced to the same positive voltage terminal. Possible implementations of deriving the floating bias voltage are shown in Figure 5. The UCC3583 is designed for use in a post regulator application for tightly regulating auxiliary outputs in a multiple output converter. The post regulation is applied to the secondary side power pulse of a power transformer where the power pulse is controlled by the feedback signal from the main output. In order to simplify the application of the UCC3583, it is required that the IC be referenced to the positive output terminal and the output filter inductor be placed in the return path. The placement of the inductor in the return path facilitates better EMI performance, in addition to making magnetic designs 4 UCC1583 UCC2583 UCC3583 APPLICATION INFORMATION (cont.) For the circuit shown in Figure 5a, CC1 is charged when the transformer voltage is positive and the synchronous switch is on. During the off period of Q-SYNC, the charge is transferred to CC2 through diode DC2. Diode DC3 charges CC2 during the blocking interval of Q-SYNC. This method is preferable when the transformer positive voltage is high enough to generate the required bias voltage. For the circuit shown in Figure 5b, CC1 is charged during the period when reverse (reset) voltage appears across the secondary. The charge on CC1 is transferred to CC2 through DC2 when Q-SYNC turns on. This method is preferable when the reverse voltage is high enough to generate the required bias voltage. The series resistor should be chosen to handle the required voltage drop at full IC operating current when the zener clamp across VDD and COM is activated. t DIS = I R AMP ( dis ) ≈ 3000 • CT The values of RT and CT are also dictated by the fact that the ramp is discharged through an internal impedance of 2k. The value of RT needs to be at least 50k to ensure that the internal discharge current is the current through RT during the entire discharge period. This results in making the value of CT relatively small for a desired frequency of operation. When the synchronizing signal is available, the oscillator frequency should be programmed to be lower than the synchronizing frequency to ensure proper operation. A large difference in self-running and synchronizing frequencies leads to smaller ramp amplitude and higher noise sensitivity. The ramp capacitor is discharged when the synchronization signal arrives and begins charging when the low threshold is crossed. The following is a description of the major functional blocks of the UCC3583. Refer to Figure 6 (Typical Application Circuit) for component designations. There are two methods to synchronize to the secondary pulse. One method is to use the rising edge of the secondary pulse, which reduces the maximum duty cycle available. Subsequently, the post regulator switch cannot be turned on during the CT discharge time. The other method is to use the falling edge of the secondary pulse for synchronization. This method is preferable because it allows a slower discharge of the ramp capacitor without affecting the maximum available duty cycle of the post regulator. The UCC3583 SYNC input needs to reach a fixed threshold (1.0V typical) for synchronization to take effect. Hence the IC is usable with either method of synchronization. However, the UCC3583 oscillator configuration is better suited for synchronization to the falling edge. A recommended method to implement the synchronization is shown in Figure 6. By connecting SYNC to a resistive divider between REF and the secondary terminal S2, the synchronization is achieved whenever the voltage on S2 goes from a negative value to zero. RA and RB should be selected so that the voltage on the SYNC pin varies from 0V to 1V. Placement of a Schottky diode from SYNC to COM prevents the voltage at SYNC from going negative. The internal hysteretic SYNC comparator has an inverting input set to 0.5V with about ±0.5V hysteresis. UVLO and Start Up The UCC3583 has an internal undervoltage lockout circuit which keeps the internal circuitry inactive until VDD exceeds the upper threshold (9V). Once the chip is activated, VDD has to be above the lower UVLO threshold (8.4V) for it to remain functional. The IC requires a low startup current of only 100µA when VDD is under the UVLO threshold. VDD has an internal clamp of 14V which can sink up to 10mA. Measures must be taken not to exceed this current. The internal reference (REF) is brought up when the UVLO on threshold is exceeded. The soft start pin provides an effective means to start the IC in a controlled manner. An internal current of 10µA starts discharging a capacitor connected to SS when the UVLO conditions have been removed. The voltage on SS controls the duty cycle of the output during the discharge period. Synchronizing Circuit and Oscillator UCC3583 is primarily intended for synchronizable operation where its switching frequency is determined by the secondary pulse of the power transformer. However, it has an internal oscillator which allows it to operate in free-running mode when an external synchronization pulse is not available. The switching frequency is determined by resistor RT connected between REF and RAMP and capacitor CT connected from RAMP to GND. The frequency is given by: fre q = CT • VR AMP ( p – p) PWM Comparator The UCC3583 uses a leading edge PWM scheme. In a leading edge PWM, the output pulse (gate signal) is turned on when the error amplifier crosses the PWM ramp and turned off by the clock/oscillator. Leading edge modulation is naturally provided by magamp type post regulators and is an essential feature for post regulators. Without the leading edge modulation in a multiple output 1 whe re t CH = 1. 56 • RT • CT t CH + t DIS and 5 UCC1583 UCC2583 UCC3583 APPLICATION INFORMATION (cont.) converter with post regulation on one or more outputs, the primary current shape does not remain monotonic and can lead to instability when the primary current is used for current mode control or current limiting. When compared to conventional trailing edge PWMs, the leading edge modulation leads to a phase inversion that needs to be accounted for in the feedback loop. For the UCC3583, this inversion is automatically provided since the sensed voltage at the power supply output negative terminal has a negative polarity with respect to the chip common. Thus, UCC3583 does not require inverting buffers which would otherwise be needed. pacitor controls the pulse width. The third control loop is provided by the average current amplifier. By sensing the instantaneous inductor current and filtering/averaging it with the current error amplifier, accurate current limiting is achieved. This loop is in effect only during the overcurrent mode and provides a more accurate and noise free control of the maximum output current compared to conventional peak current limiting circuits. The current limit is set by programming the voltage at ILIM based on the current sense resistor chosen. In addition, the current limit can be made proportional to the output voltage in order to limit the power dissipation under short circuit conditions. This is implemented by inserting a bias voltage on CS which is proportional to the output voltage. Error Signal Generation and Current Limiting The PWM comparator in the UCC3583 is controlled by three parallel loops with only one of them in effect at a time. During normal operation, the voltage error amplifier output is fed to the PWM comparator. The voltage error amplifier can be compensated using commonly used feedback techniques to achieve the desired dynamic performance. The ouput drive capability of the voltage amplifier is limited to 100µA, so appropriately high impedances should be used to utilize the full output swing of the amplifier. During startup, the soft start ca- Gate Drive Circuit The gate drive circuit of the UCC3583 provides high current drive capability and is very easy to implement as a result of tying the chip common to the source of the switching device. Turn on current is higher (1.5A) as fast turn on is essential for low losses and effective operation. During the turn off, the drain voltage disappears, so turn off time can be slower without increasing switching losses. PRIMARY CLOCK ISOLATION PRIMARY PULSE MAIN OUTPUT PRIMARY CURRENT S1 PRIMARY SIDE CONTROLLER CHIP COMMON VGS S2 + VO AUXILIARY OUTPUT – SYNC UCC3583 SSPR CLOCK RAMP ISENSE VOLTAGE FEEDBACK Figure 1. UCC3583 SSPR system application and typical waveforms. 6 SSPR POWER PULSE UDG-98195 UCC1583 UCC2583 UCC3583 APPLICATION INFORMATION (cont.) Note: All waveforms are referenced to chip common. UDG-96141-1 Figure 2. Single ended post regulator waveforms. UDG-96142-1 Figure 3. Half-bridge synchronous post regulator application. 7 UCC1583 UCC2583 UCC3583 APPLICATION INFORMATION (cont.) UDG-96143-1 Figure 4. Half-bridge synchronous post regulator to waveforms. 8 UCC1583 UCC2583 UCC3583 APPLICATION INFORMATION (cont.) UDG-96175-1 Figure 5. Possible implementation for floating bias voltage generation. U N ITROD RB UDG-96072-2 Figure 6. Typical application circuit. 9 PACKAGE OPTION ADDENDUM www.ti.com 10-Dec-2020 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Pins Package Drawing Qty Eco Plan (2) Lead finish/ Ball material MSL Peak Temp Op Temp (°C) Device Marking (3) (4/5) (6) UCC2583D ACTIVE SOIC D 14 50 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 UCC2583D UCC2583DTR ACTIVE SOIC D 14 2500 RoHS & Green NIPDAU Level-2-260C-1 YEAR -40 to 85 UCC2583D UCC2583QTR ACTIVE PLCC FN 20 1000 RoHS & Green SN Level-2-260C-1 YEAR -40 to 85 UCC2583Q UCC3583D ACTIVE SOIC D 14 50 RoHS & Green NIPDAU Level-2-260C-1 YEAR 0 to 70 UCC3583D UCC3583DTR ACTIVE SOIC D 14 2500 RoHS & Green NIPDAU Level-2-260C-1 YEAR 0 to 70 UCC3583D (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may reference these types of products as "Pb-Free". RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption. Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of