UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
INTERLEAVING CONTINUOUS CONDUCTION MODE PFC CONTROLLER
Check for Samples: UCC28070-Q1
FEATURES
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DESCRIPTION
Qualified for Automotive Applications
Interleaved Average Current-Mode PWM
Control with Inherent Current Matching
Advanced Current Synthesizer Current
Sensing for Superior Efficiency
Highly-Linear Multiplier Output with Internal
Quantized Voltage Feed-Forward Correction
for Near-Unity PF
Programmable Frequency (30 kHz to 300 kHz)
Programmable Maximum Duty-Cycle Clamp
Programmable Frequency Dithering Rate and
Magnitude for Enhanced EMI Reduction
– Magnitude: 3 kHz to 30 kHz
– Rate: Up to 30 kHz
External Clock Synchronization Capability
Enhanced Load and Line Transient Response
through Voltage Amplifier Output Slew-Rate
Correction
Programmable Peak Current Limiting
Bias-Supply UVLO, Over-Voltage Protection,
Open-Loop Detection, and PFC-Enable
Monitoring
External PFC-Disable Interface
Open-Circuit Protection on VSENSE and
VINAC pins
Programmable Soft Start
20-Lead TSSOP Package
The UCC28070 is an advanced power factor
correction device that integrates two pulse-width
modulators (PWMs) operating 180° out of phase.
This
interleaved
PWM
operation
generates
substantial reduction in the input and output ripple
currents, and the conducted-EMI filtering becomes
easier and less expensive. A significantly improved
multiplier design provides a shared current reference
to two independent current amplifiers that ensures
matched average current mode control in both PWM
outputs while maintaining a stable, low-distortion
sinusoidal input line current.
The UCC28070 contains multiple innovations
including current synthesis and quantized voltage
feed-forward to promote performance enhancements
in PF, efficiency, THD, and transient response.
Features including frequency dithering, clock
synchronization, and slew rate enhancement further
expand the potential performance enhancements.
The UCC28070 also contains a variety of protection
features including output over-voltage detection,
programmable peak-current limit, under-voltage
lockout, and open-loop protection.
Simplified Application Diagram
VIN
L1
D1
+
VOUT
COUT
–
12V to 21V
To CSB
CCDR
RRDM
RA
RB
1 CDR
DMAX 20
2 RDM
RT 19
3 VAO
SS 18
4 VSENSE
GDB 17
5 VINAC
GND 16
RIMO
6 IMO
RSYN
7 RSYNTH
T1
RS
RDMX
RRT
CSS
M1
VCC 15
GDA 14
L2
8 CSB
VREF 13
9 CSA
CAOA 12
D2
To CSA
10 PKLMT
CAOB 11
RS
From Ixfrms
CZV
RPK1
CZC
CPC
CPV
T2
RA
CZC
CREF
CPC
M2
RPK2
RZV
RZC
RZC
RB
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
ORDERING INFORMATION
TA
–40°C to 125°C
PACKAGE
TSSOP – PW
ORDERABLE PART NUMBER
Reel of 2000
ABSOLUTE MAXIMUM RATINGS (1)
UCC28070QPWRQ1
TOP-SIDE MARKING
28070Q
(2) (3) (4)
over operating free-air temperature range (unless otherwise noted)
PARAMETER
LIMIT
UNIT
Supply voltage: VCC
22
V
Supply current: IVCC
20
mA
Voltage: GDA, GDB
−0.5 to VCC+0.3
V
Gate drive current – continuous: GDA, GDB
+/− 0.25
A
Gate drive current – pulsed: GDA, GDB
+/− 0.75
A
−0.5 to +7
V
−0.5
mA
10
mA
Operating junction temperature, TJ
−40 to +140
°C
Storage temperature, TSTG
−65 to +150
°C
Voltage: DMAX, RDM, RT, CDR, VINAC, VSENSE, SS, VAO, IMO, CSA, CSB,
CAOA, CAOB, PKLMT, VREF
Current: RT, DMAX, RDM, RSYNTH
Current: VREF, VAO, CAOA, CAOB, IMO
Lead temperature (10 seconds)
(1)
(2)
(3)
(4)
260
These are stress limits. Stress beyond these limits may cause permanent damage to the device. Functional operation of the device at
these or any conditions beyond those indicated under RECOMMENDED OPERATING CONDITIONS is not implied. Exposure to
absolute maximum rated conditions for extended periods of time may affect device reliability.
All voltages are with respect to GND.
All currents are positive into the terminal, negative out of the terminal.
In normal use, terminals GDA and GDB are connected to an external gate driver and are internally limited in output current.
DISSIPATION RATINGS
THERMAL
IMPEDANCE
JUNCTION-TOAMBIENT
PACKAGE
20-Pin TSSOP (PW)
(1)
(2)
125°C/W
(1)
and
(2)
TA = 25°C POWER
RATING
800 mW
(1)
TA = 85°C POWER RATING
320 mW
TA = 125°C POWER
RATING
(1)
120 mW
(1)
Thermal resistance is a strong function of board construction and layout. Air flow reduces thermal resistance. This number is only a
general guide.
Thermal resistance calculated with a low-K methodology.
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
PARAMETER
VCC Input Voltage (from a low-impedance source)
MIN
MAX
VUVLO + 1 V
VREF Load Current
UNIT
21
V
2
mA
VINAC Input Voltage Range
0
3
IMO Voltage Range
0
3.3
PKLMT, CSA, CSB Voltage Range
0
3.6
RSYNTH Resistance (RSYN)
15
750
RDM Resistance (RRDM)
30
330
2
V
kΩ
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
ELECTRICAL CHARACTERISTICS
over operating free-air temperature range −40°C < TA < 125°C, VCC = 12 V, GND = 0 V, RRT = 75 kΩ,
RDMX = 68.1 kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 µF, CVCC = 1 µF, IVREF = 0 mA (unless otherwise
noted)
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
23
25
27
UNIT
Bias Supply
VCCSHUNT VCC shunt voltage
(1)
VCC current, disabled
VSENSE = 0 V
7
VCC current, enabled
VSENSE = 3 V (switching)
9
VCC current, UVLO
VUVLO
IVCC = 10 mA
VCC = 9 V
Measured at VCC (rising)
UVLO hysteresis
Measured at VCC (falling)
VREF enable threshold
Measured at VCC (rising)
9.8
mA
200
µA
4
6
mA
10.2
10.6
VCC = 7 V
UVLO turn-on threshold
12
V
1
V
7.5
8
8.5
5.82
6
6.18
Linear Regulator
VREF voltage, no load
IVREF = 0 mA
VREF load rejection
Measured as the change in VREF,
(IVREF = 0 mA and −2 mA)
VREF line rejection
Measured as the change in VREF, TA = 25°C
(VCC = 11V and 20 V, IVREF = 0
TA = -40°C to 125°C
µA)
Enable threshold
Measured at VSENSE (rising)
-12
12
-12
12
-16
16
V
mV
PFC Enable
VEN
0.65
Enable hysteresis
0.75
0.85
0.15
V
External PFC Disable
Disable threshold
Measured at SS (falling)
Hysteresis
VSENSE > 0.85 V
0.5
0.6
Output phase shift
Measured between GDA and GDB
179
180
181
°
Measured at DMAX, RT, & RDM
2.91
3
3.09
V
94
100
105
270
290
330
92%
95%
98%
50
150
250
1
3
4.3
23
30
36
V
0.15
Oscillator
VDMAX,VRT
, and
Timing regulation voltages
VRDM
fPWM
PWM switching frequency
RRT = 75 kΩ, RDMX = 68.1 kΩ,
VRDM = 0 V, VCDR = 6 V
RRT = 24.9 kΩ, RDMX = 22.6 kΩ,
VRDM = 0 V, VCDR = 6 V
Duty-cycle clamp
RRT = 75 kΩ, RDMX = 68.1 kΩ,
VRDM = 0 V, VCDR = 6 V
Minimum programmable
off-time
RRT = 24.9 kΩ, RDMX = 22.6 kΩ,
VRDM = 0 V, VCDR = 6 V
fDM
Frequency dithering magnitude
change in fPWM
RRDM = 316 kΩ, RRT = 75 kΩ
fDR
Frequency dithering rate rate of CCDR = 2.2 nF, RRDM = 100 kΩ
change in fPWM
CCDR = 0.3 nF, RRDM = 100 kΩ
DMAX
ICDR
(1)
RRDM = 31.6 kΩ, RRT = 24.9 kΩ
Dither rate current
Measure at CDR (sink and source)
Dither disable threshold
Measured at CCDR (rising)
kHz
3
ns
kHz
20
±10
5
µA
5.25
V
Excessive VCC input voltage and/or current damages the device. This clamp will not protect the device from an unregulated supply. If
an unregulated supply is used, a series-connected fixed positive voltage regulator such as a UA78L15A is recommended. See the
Absolute Maximum Ratings section for the limits on VCC voltage and current.
Copyright © 2010–2011, Texas Instruments Incorporated
3
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
over operating free-air temperature range −40°C < TA < 125°C, VCC = 12 V, GND = 0 V, RRT = 75 kΩ,
RDMX = 68.1 kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 µF, CVCC = 1 µF, IVREF = 0 mA (unless otherwise
noted)
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
5
5.25
V
ns
Clock Synchronization
VCDR
SYNC enable threshold
Measured at CDR (rising)
SYNC propagation delay
VCDR = 6 V, Measured from RDM (rising) to GDx (rising)
50
100
SYNC threshold (Rising)
VCDR = 6 V, Measured at RDM
1.2
1.5
SYNC threshold (Falling)
VCDR = 6 V, Measured at RDM
0.4
Positive pulse width
0.2
SYNC pulses
Maximum duty cycle
(2)
0.7
V
μs
50
%
Voltage Amplifier
gMV
VSENSE voltage
In regulation, TA = 25°C
2.94
3
3.06
VSENSE voltage
In regulation
2.84
3
3.10
VSENSE input bias current
In regulation
250
500
VAO high voltage
VSENSE = 2.9 V
5
5.2
VAO low voltage
VSENSE = 3.1 V
0.05
0.50
VAO transconductance
2.8 V < VSENSE < 3.2 V, VAO = 3 V
70
VAO sink current, overdriven
limit
VSENSE = 3.5 V, VAO = 3 V
30
4.8
ISRC
nA
V
μs
−30
VAO source current, overdriven VSENSE = 2.5 V, VAO = 3 V, SS = 3 V
V
μA
VAO source current,
overdriven limit + ISRC
VSENSE = 2.5 V, VAO = 3 V
Slew-rate correction threshold
Measured as VSENSE (falling) / VSENSE (regulation)
Slew-rate correction hysteresis
Measured at VSENSE (rising)
Slew-rate correction current
Measured at VAO, in addition to VAO source current.
Slew-rate correction enable
threshold
Measured at SS (rising)
VAO discharge current
VSENSE = 0.5 V, VAO = 1 V
SS source current
VSENSE = 0.9 V, SS = 1 V
−10
Adaptive source current
VSENSE = 2.0 V, SS = 1 V
−1.5
-2.5
mA
Adaptive SS disable
Measured as VSENSE – SS
-30
0
30
mV
SS sink current
VSENSE = 0.5 V, SS = 0.2 V
0.5
0.9
−130
92
93
95
%
3
9
mV
−100
μA
4
V
10
μA
Soft Start
ISS
(2)
4
μA
mA
Due to the influence of the synchronization pulse width on the programmability of the maximum PWM switching duty cycle (DMAX) it is
recommended to minimize the synchronization signal's duty cycle.
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
over operating free-air temperature range −40°C < TA < 125°C, VCC = 12 V, GND = 0 V, RRT = 75 kΩ,
RDMX = 68.1 kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 µF, CVCC = 1 µF, IVREF = 0 mA (unless otherwise
noted)
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
104
106
108
%
Over Voltage
VOVP
OVP threshold
Measured as VSENSE (rising) / VSENSE (regulation)
OVP hysteresis
Measured at VSENSE (falling)
100
OVP propagation delay
Measured between VSENSE (rising) and GDx (falling)
0.2
Zero-power detect threshold
Measured at VAO (falling)
mV
0.3
μs
Zero-Power
VZPWR
0.65
Zero-power hysteresis
0.75
V
0.15
Multiplier
kMULT
IIMO
Gain constant
Output current: zero
VAO ≥ 1.5 V, TA = 25°C
15.4
17
20
VAO = 1.2 V, TA = 25°C
20.5
13.5
17.0
VAO ≥ 1.5 V
14
17
22
VAO = 1.2 V
12
17
22.5
VINAC = 0.9 VPK, VAO = 0.8 V
-0.2
0
0.2
VINAC = 0 V, VAO = 5 V
-0.2
0
0.2
0.6
0.7
0.8
μA
Quantized Voltage Feed Forward
(3)
VLVL1
Level 1 threshold
VLVL2
Level 2 threshold
1
VLVL3
Level 3 threshold
1.2
VLVL4
Level 4 threshold
VLVL5
Level 5 threshold
VLVL6
Level 6 threshold
1.95
VLVL7
Level 7 threshold
2.25
VLVL8
Level 8 threshold
2.6
1.4
Measured at VINAC (rising)
V
1.65
Current Amplifiers
CAOx high voltage
5.75
6
CAOx low voltage
gMC
0.1
CAOx transconductance
50
CAOx source current,
overdriven
0
3.6
RSYNTH = 6 V, TA = 25°C
-16
-8
0
RSYNTH = 6 V
-50
-8
40
-50
−8
40
0
12
Input offset voltage
(3)
μA
−50
Input common mode range
Input offset Voltage
μs
100
CAOx sink current, overdriven
V
TA = 25°C
Phase mismatch
Measured as Phase A's input
offset minus Phase B's input offset TA = -40°C to 125°C
-12
CAOx pulldown current
VSENSE = 0.5 V, CAOx = 0.2 V
0.5
-20
V
mV
mV
14
0.9
mA
The Level 1 threshold represents the "zero-crossing detection" threshold above which VINAC must rise to initiate a new input half-cycle,
and below which VINAC must fall to terminate that half-cycle.
Copyright © 2010–2011, Texas Instruments Incorporated
5
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
ELECTRICAL CHARACTERISTICS (continued)
over operating free-air temperature range −40°C < TA < 125°C, VCC = 12 V, GND = 0 V, RRT = 75 kΩ,
RDMX = 68.1 kΩ, RRDM = RSYN = 100 kΩ, CCDR = 2.2 nF, CSS = CVREF = 0.1 µF, CVCC = 1 µF, IVREF = 0 mA (unless otherwise
noted)
SYMBOL
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
VSENSE = 3 V, VINAC = 0 V
2.91
3
3.09
VSENSE = 3 V, VINAC = 2.85 V
0.10
0.15
0.20
5
5.25
0.250
0.500
μA
3.3
3.33
V
60
100
ns
3.8
4.0
4.2
0.55
0.7
Current Synthesizer
VRSYNTH
Regulation voltage
Synthesizer disable threshold
Measured at RSYNTH (rising)
VINAC input bias current
V
Peak Current Limit
Peak current limit threshold
PKLMT = 3.30 V, measured at CSx (rising)
Peak current limit propagation
delay
3.27
Measured between CSx (rising) and GDx (falling) edges
PWM Ramp
VRMP
PWM ramp amplitude
PWM ramp offset voltage
TA = 25°C, RRT = 75 kΩ
PWM ramp offset temperature
coefficient
−2
V
mV/°C
Gate Drive
GDA, GDB output voltage,
high, clamped
VCC = 20 V, CLOAD = 1 nF
GDA, GDB output voltage, High CLOAD = 1 nF
11.5
13
10
10.5
15
V
GDA, GDB output voltage, Low
CLOAD = 1 nF
0.2
0.3
Rise time GDx
1 V to 9 V, CLOAD = 1 nF
18
30
Fall time GDx
9 V to 1 V, CLOAD = 1 nF
12
25
GDA, GDB output voltage,
UVLO
VCC = 0 V, IGDA, IGDB = 2.5 mA
0.7
2
ns
V
Thermal Shutdown
6
Thermal shutdown threshold
160
Thermal shutdown recovery
140
°C
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
DEVICE INFORMATION
PW PACKAGE
(TOP VIEW)
CDR
RDM
VAO
VSENSE
VINAC
IMO
RSYNTH
CSB
CSA
PKLMT
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
DMAX
RT
SS
GDB
GND
VCC
GDA
VREF
CAOA
CAOB
TERMINAL FUNCTIONS
NAME
CDR
PIN #
1
I/O
DESCRIPTION
I
Dither Rate Capacitor. Frequency-dithering timing pin. An external capacitor to GND programs
the rate of oscillator dither. Connect the CDR pin to the VREF pin to disable dithering.
RDM
(SYNC)
2
I
Dither Magnitude Resistor. Frequency-dithering magnitude and external synchronization pin. An
external resistor to GND programs the magnitude of oscillator frequency dither. When frequency
dithering is disabled (CDR > 5 V), the internal master clock will synchronize to positive edges
presented on the RDM pin. Connect RDM to GND when dithering is disabled and synchronization
is not desired.
VAO
3
O
Voltage Amplifier Output. Output of transconductance voltage error amplifier. Internally
connected to Multiplier input and Zero-Power comparator. Connect the voltage regulation loop
compensation components between this pin and GND.
VSENSE
4
I
Output Voltage Sense. Internally connected to the inverting input of the transconductance
voltage error amplifier in addition to the positive terminal of the Current Synthesis difference
amplifier. Also connected to the OVP, PFC Enable, and slew-rate comparators. Connect to PFC
output with a resistor-divider network.
VINAC
5
I
Scaled AC Line Input Voltage. Internally connected to the Multiplier and negative terminal of the
Current Synthesis difference amplifier. Connect a resistor-divider network between VIN, VINAC,
and GND identical to the PFC output divider network connected at VSENSE.
IMO
6
O
Multiplier Current Output. Connect a resistor between this pin and GND to set the multiplier
gain.
RSYNTH
7
I
Current Synthesis Down-Slope Programming. Connect a resistor between this pin and GND to
set the magnitude of the current synthesizer down-slope. Connecting RSYNTH to VREF will
disable current synthesis and connect CSA and CSB directly to their respective current amplifiers.
CSB
8
I
Phase B Current Sense Input. During the on-time of GDB, CSB is internally connected to the
inverting input of Phase B's current amplifier through the current synthesis stage.
CSA
9
I
Phase A Current Sense Input. During the on-time of GDA, CSA is internally connected to the
inverting input of Phase A's current amplifier through the current synthesis stage.
PKLMT
10
I
Peak Current Limit Programming. Connect a resistor-divider network between VREF and this
pin to set the voltage threshold of the cycle-by-cycle peak current limiting comparators. Allows
adjustment for desired ΔILB.
CAOB
11
O
Phase B Current Amplifier Output. Output of phase B's transconductance current amplifier.
Internally connected to the inverting input of phase B's PWM comparator for trailing-edge
modulation. Connect the current regulation loop compensation components between this pin and
GND.
CAOA
12
O
Phase A Current Amplifier Output. Output of phase A's transconductance current amplifier.
Internally connected to the inverting input of phase A's PWM comparator for trailing-edge
modulation. Connect the current regulation loop compensation components between this pin and
GND.
VREF
13
O
6-V Reference Voltage and Internal Bias Voltage. Connect a 0.1-μF ceramic bypass capacitor
as close as possible to this pin and GND.
GDA
14
O
Phase A's Gate Drive. This limited-current output is intended to connect to a separate gate-drive
device suitable for driving the Phase A switching component(s). The output voltage is typically
clamped to 13.5 V.
VCC
15
I
Bias Voltage Input. Connect a 0.1-μF ceramic bypass capacitor as close as possible to this pin
and GND.
Copyright © 2010–2011, Texas Instruments Incorporated
7
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
TERMINAL FUNCTIONS (continued)
NAME
8
PIN #
I/O
DESCRIPTION
GND
16
I/O
Device Ground Reference. Connect all compensation and programming resistor and capacitor
networks to this pin. Connect this pin to the system through a separate trace for high-current
noise isolation.
GDB
17
O
Phase B's Gate Drive. This limited-current output is intended to connect to a separate
gate-drivedevice suitable for driving the Phase B switching component(s). The output voltage is
typically clamped to 13.5 V.
SS
18
I
Soft-Start and External Fault Interface. Connect a capacitor to GND on this pin to set the
soft-start slew rate based on an internally-fixed 10-μA current source. The regulation reference
voltage for VSENSE is clamped to VSS until VSS exceeds 3 V. Upon recovery from certain fault
conditions a 1-mA current source is present at the SS pin until the SS voltage equals the
VSENSE voltage. Pulling the SS pin below 0.6 V immediately disables both GDA and GDB
outputs.
RT
19
I
Timing Resistor. Oscillator frequency programming pin. A resistor to GND sets the running
frequency of the internal oscillator.
DMAX
20
I
Maximum Duty-Cycle Resistor. Maximum PWM duty-cycle programming pin. A resistor to GND
sets the PWM maximum duty-cycle based on the ratio of RDMX/RRT.
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
Functional Block Diagram
+
OVP
VCC 15
25V
VREF 13 6V
Linear EN
Regulator
160 On o
140 Off
C
+
ReStart
ThermSD
0.75V
0.60V
+
R Q
UVLO
10.2V
9.2V
+
S Q
VSENSE
GND 16
Ext.Disable
SS
8V
0.75V
0.60V
VSENSE
3.18V
3.08V
Fault
ZeroPwr
+
0.90V
0.75V
+
VAO
6
IMO
5 VINAC
DMAX 20
Voltage
FeedForward
CLKA
Oscillator w/
Freq. Dither
RT 19
CLKB
IIMO =
VVINAC * (VVAO – 1)
KVFF
OffA
* 17uA
250nA
KVFF
x
OffB
Mult.
/
3 VAO
x
RDM/
2
SYNC
SYNC
Logic
CDR 1
SYNC Dither
Enable Disable
+
ReStart
100uA
5V
+
SS
4V
Slew Rate
Correction
+
10uA
2.8V
5V
GmAmp
VA
+
+
4 VSENSE
3V
250nA
Adaptive SS
PKLMT 10
IpeakA
1mA
ReStart
ISS
10uA
+
+
Control
Logic
ReStart
Ext.Disable
IpeakB
CSA 9
+
+
18 SS
PWM1
CA1
GmAmp
+
S Q
OutA
CSB 8
Current
Synthesizer
RSYNTH 7
5V
CLKA
R Q
+
VINAC
VSENSE
CAOA 12
VCC
S Q
+
OffB
IpeakB
Fault
CAOB 11
Copyright © 2010–2011, Texas Instruments Incorporated
14 GDA
PWM2
CA2
GmAmp
Driver
GND
Fault
Disable
+
OffA
IpeakA
OutB
VCC
(Clamped at 13.5V)
CLKB
R Q
(Clamped at 13.5V)
Driver
17 GDB
GND
9
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
TYPICAL CHARACTERISTICS
SUPPLY CURRENT
vs
TEMPERATURE
REFERENCE VOLTAGE
vs
TEMPERATURE
12
6.18
6.12
IVCC, VCC = 12 V, enabled
VREF - Reference Voltage - V
IVCC - Supply Current - mA
10
8
IVCC, VCC = 12 V, disabled
6
4
2
6.06
VREF (IVREF = 0 mA)
6.00
5.94
5.88
0
5.82
-60 -40
-20
0
20
40
60
80
100 120 140
-60
-10
TJ - Temperature - 0C
40
90
140
TJ - Temperature - 0C
Figure 1.
Figure 2.
IVSENSE BIAS CURRENT
vs
TEMPERATURE
VSENSE REGULATION
vs
TEMPERATURE
0.50
3.06
0.45
0.40
IVSENSE - Bias Current - mA
VSENSE Regulation - V
3.04
3.02
3.00
2.98
0.35
0.30
0.25
0.20
0.15
0.10
2.96
0.05
2.94
0
-60 -40
-20
0
20
40
60
80 100
TJ - Temperature - 0C
Figure 3.
10
120 140
-60
-10
40
TJ - Temperature -
90
140
0C
Figure 4.
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
TYPICAL CHARACTERISTICS (continued)
MULTIPLIER OUTPUT CURRENT
vs
VOLTAGE AMPLIFIER OUTPUT
MULTIPIER CONSTANT
vs
TEMPERATURE
180
20
QVFF Level
Level 1
19
Level 2
140
Level 3
120
Multiplier Constant - mA
IMO - Multiplier Output Current - mA
160
Level 4
Level 5
100
Level 6
80
Level 7
Level 8
60
18
VAO = 3.0 V
VAO = 5.0 V
17
16
VAO = 1.5 V
40
15
VAO = 1.2 V
20
14
0
0
1
3
2
4
5
-60 -40
6
-20
0
VAO - Voltage Amplifier Output - V
Figure 5.
40
60
80
100 120 140
Figure 6.
IVINAC BIAS CURRENT
vs
TEMPERATURE
SWITCHING FREQUENCY (normalized change)
vs
TEMPERATURE
1.0
Normalized Change in Switching Frequency - %
0.50
0.45
0.40
IVINAC - Bias Current - mA
20
TJ - Temperature - 0C
VINAC = 2.85 V
VINAC = 2.5 V
0.35
0.30
0.25
0.20
VINAC = 1.0 V
0.15
VINAC = 0.2 V
Typical Frequency= 30 kHz
RT = 249 k?
0.8
0.5
Typical Frequency= 100 kHz
RT = 75 k?
0.3
0
-0.3
Typical Frequency= 290 kHz
RT = 24.9 k?
-0.5
-0.8
VINAC = 2.0 V
0.10
-1.0
-60
0.05
-40
-20
0
20
40
60
80
100 120 140
TJ - Temperature - 0C
0.00
-60
-10
40
90
140
TJ - Temperature - 0C
Figure 7.
Copyright © 2010–2011, Texas Instruments Incorporated
Figure 8.
11
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
TYPICAL CHARACTERISTICS (continued)
VOLTAGE AMPLIFIER TRANSFER FUNCTION
vs
VSENSE
VOLTAGE AMPLIFIER TRANSCONDUCTANCE
vs
TEMPERATURE
40
IVAO - Voltage Amplifier Output Current - mA
VAO - Voltage Amplifier Transconductance - mS
80
75
70
65
60
55
50
20
0
-20
-40
-60
-80
-100
-120
-140
-60 -40
-20
0
20
60
40
80 100
120 140
2.6
2.5
2.7
2.8
2.9
TJ - Temperature - 0C
3.0
3.1
3.2 3.3
3.4
3.5
VSENSE - V
Figure 9.
Figure 10.
CURRENT AMPLIFIER TRANSCONDUCTANCE
vs
TEMPERATURE
110
CAOx Transconductance - mS
105
100
95
90
85
80
-60 -40
-20
0
20
40
60
80 100
120 140
TJ - Temperature - 0C
Figure 11.
12
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
TYPICAL CHARACTERISTICS (continued)
CAx INPUT OFFSET VOLTAGE
vs
TEMPERATURE (at 0.8 V common mode)
CA1 TO CA2 RELATIVE OFFSET
vs
TEMPERATURE (at 0.8 V common mode)
5
CAx +3s
0
CAx Input Offset - mV
CA1 to CA2 Relative Offset Voltage - mV
15
-5
CAx AVG
-10
CAx -3s
-15
-20
10
A-B +3s
5
0
A-B AVG
-5
A-B -3s
-10
-15
-60 -40
-20
0
20
40
60
80 100
120 140
-60 -40
-20
TJ - Temperature - 0C
20
40
60
80 100
120 140
TJ - Temperature - 0C
Figure 12.
Figure 13.
CAx INPUT OFFSET VOLTAGE
vs
TEMPERATURE (at 2.0 V common mode)
CA1 TO CA2 RELATIVE OFFSET
vs
TEMPERATURE (at 2.0 V common mode)
5
CA1 to CA2 Relative Offset Voltage - mV
15
0
CAx Input Offset - mV
0
CAx +3s
-5
CAx AVG
-10
-15
CAx -3s
-20
10
A-B +3s
5
0
A-B AVG
-5
-10
A-B -3s
-15
-60 -40
-20
0
20
40
60
TJ - Temperature -
80 100
0C
Figure 14.
Copyright © 2010–2011, Texas Instruments Incorporated
120 140
-60 -40
-20
0
20
40
60
TJ - Temperature -
80 100
120 140
0C
Figure 15.
13
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
TYPICAL CHARACTERISTICS (continued)
CAx INPUT OFFSET VOLTAGE
vs
TEMPERATURE (at 3.6 V common mode)
CA1 TO CA2 RELATIVE OFFSET
vs
TEMPERATURE (at 3.6 V common mode)
5
CA1 to CA2 Relative Offset Voltage - mV
15
CAx Input Offset - mV
0
CAx +3s
-5
CAx AVG
-10
CAx -3s
-15
-20
A-B +3s
5
0
-5
A-B AVG
A-B -3s
-10
-15
-60 -40
-20
0
20
40
60
80 100
TJ - Temperature - 0C
Figure 16.
14
10
120 140
-60 -40
-20
0
20
40
60
80 100
120 140
TJ - Temperature - 0C
Figure 17.
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
APPLICATION INFORMATION
THEORY OF OPERATION
Interleaving
One of the main benefits from the 180° interleaving of phases is significant reductions in the high-frequency
ripple components of both the input current and the current into the output capacitor of the PFC pre-regulator.
Compared to that of a single-phase PFC stage of equal power, the reduced ripple on the input current eases the
burden of filtering conducted-EMI noise and helps reduce the EMI filter and CIN sizes. Additionally, reduced
high-frequency ripple current into the PFC output capacitor, COUT, helps to reduce its size and cost. Furthermore,
with reduced ripple and average current in each phase, the boost inductor size can be smaller than in a
single-phase design [1].
Ripple current reduction due to interleaving is often referred to as "ripple cancellation", but strictly speaking, the
peak-to-peak ripple is completely cancelled only at 50% duty-cycle in a 2-phase system. At duty-cycles other
than 50%, ripple reduction occurs in the form of partial cancellation due to the superposition of the individual
phase currents. Nevertheless, compared to the ripple currents of an equivalent single-phase PFC pre-regulator,
those of a 2-phase interleaved design are extraordinarily smaller [1]. Independent of ripple cancellation, the
frequency of the interleaved ripple, at both the input and output, is 2 x fPWM.
On the input, 180° interleaving reduces the peak-to-peak ripple amplitude to 1/2 or less of the ripple amplitude of
the equivalent single-phase current.
On the output, 180° interleaving reduces the rms value of the PFC-generated ripple current in the output
capacitor by a factor of slightly more than √2, for PWM duty-cycles > 50%.
This can be seen in the following derivations, adapting the method by Erickson [2].
In a single-phase PFC pre-regulator, the total rms capacitor current contributed by the PFC stage at all
duty-cycles can be shown to be approximated by:
æ I ö æ æ 16VO
iCRMS 1j = ç O ÷ ç ç
è h ø èç è 3p VM
ö
ö
2
÷ - 1h ÷÷
ø
ø
(1)
In a dual-phase interleaved PFC pre-regulator, the total rms capacitor current contributed by the PFC stage for D
> 50% can be shown to be approximated by:
æ I ö æ æ 16VO
iCRMS 2j = ç O ÷ ç ç
è h ø çè è 6p VM
ö
ö
2
÷ - 1h ÷÷
ø
ø
(2)
In these equations, IO = average PFC output load current, VO = average PFC output voltage, VM = peak of the
input ac-line voltage, and η = efficiency of the PFC stage at these conditions. It can be seen that the quantity
under the radical for iCrms2φ is slightly smaller than 1/2 of that under the radical for iCrms1φ. The rms currents
shown contain both the low-frequency and the high-frequency components of the PFC output current.
Interleaving reduces the high-frequency component, but not the low-frequency component.
Copyright © 2010–2011, Texas Instruments Incorporated
15
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
Programming the PWM Frequency and Maximum Duty-Cycle Clamp
The PWM frequency and maximum duty-cycle clamps for both GDx outputs of the UCC28070 are set through
the selection of the resistors connected to the RT and DMAX pins, respectively. The selection of the RT resistor
(RRT) directly sets the PWM frequency (fPWM).
RRT (k W ) =
7500
f PWM (kHz )
(3)
Once RRT has been determined, the DMAX resistor (RDMX) may be derived.
RDMX = RRT ´ (2 ´ DMAX - 1)
(4)
where DMAX is the desired maximum PWM duty-cycle.
Frequency Dithering (Magnitude and Rate)
Frequency dithering refers to modulating the switching frequency to achieve a reduction in conducted-EMI noise
beyond the capability of the line filter alone. The UCC28070 implements a triangular modulation method which
results in equal time spent at every point along the switching frequency range. This total range from minimum to
maximum frequency is defined as the dither magnitude, and is centered around the nominal switching frequency
fPWM set with RRT. For example, a dither magnitude of 20 kHz on a nominal fPWM of 100 kHz results in a
frequency range of 100 kHz ±10 kHz. Furthermore, the programmed duty-cycle clamp set by RDMX remains
constant at the programmed value across the entire range of the frequency dithering.
The rate at which fPWM traverses from one extreme to the other and back again is defined as the dither rate. For
example, a dither rate of 1 kHz would linearly modulate the nominal frequency from 110 kHz to 90 kHz to 110
kHz once every millisecond. A good initial design target for dither magnitude is ±10% of fPWM. Most boost
components can tolerate such a spread in fPWM. The designer can then iterate around there to find the best
compromise between EMI reduction, component tolerances, and loop stability.
The desired dither magnitude is set by a resistor from the RDM pin to GND, of value calculated by the following
equation:
RRDM (k W ) =
937.5
f DM (kHz )
(5)
Once the value of RRDM is determined, the desired dither rate may be set by a capacitor from the CDR pin to
GND, of value calculated by the following equation:
æ R ( kW ) ö
CCDR ( pF ) = 66.7 ´ ç RDM
÷
è f DR ( kHz ) ø
(6)
Frequency dithering may be fully disabled by forcing the CDR pin > 5 V or by connecting it to VREF (6 V) and
connecting the RDM pin directly to GND. (If populated, the relatively high impedance of the RDM resistor may
allow system switching noise to couple in and interfere with the controller timing functions if not bypassed with a
low impedance path when dithering is disabled.)
If an external frequency source is used to synchronize fPWM and frequency dithering is desired, the external
frequency source must provide the dither magnitude and rate functions as the internal dither circuitry is disabled
to prevent undesired performance during synchronization. (See following section for more details.)
16
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
External Clock Synchronization
The UCC28070 has also been designed to be easily synchronized to almost any external frequency source. By
disabling frequency dithering (pulling CDR > 5 V), the UCC28070's SYNC circuitry is enabled permitting the
internal oscillator to be synchronized with pulses presented on the RDM pin. In order to ensure a precise 180
degree phase shift is maintained between the GDA and GDB outputs, the frequency (fSYNC) of the pulses
presented at the RDM pin needs to be at twice the desired fPWM. For example, if a 100-kHz switching frequency
is desired, the fSYNC should be 200 kHz.
f PWM =
f SYNC
2
(7)
In order to ensure the internal oscillator does not interfere with the SYNC function, RRT should be sized to set the
internal oscillator frequency at least 10% below the fSYNC.
RRT (k W ) =
15000
´ 1 .1
f SYNC (kHz )
(8)
It must be noted that the PWM modulator gain will be reduced by a factor equivalent to the scaled RRT due to a
direct correlation between the PWM ramp current and RRT. Adjustments to the current loop gains should be
made accordingly.
It must also be noted that the maximum duty cycle clamp programmability is affected during external
synchronization. The internal timing circuitry responsible for setting the maximum duty cycle is initiated on the
falling edge of the synchronization pulse. Therefore, the selection of RDMX becomes dependent on the
synchronization pulse width (tSYNC).
DSYNC = tSYNC ´ f SYNC
(9)
For use in RDMX equation immediately below.
æ 15000 ö
RDMX (k W ) = ç
÷ ´ (2 ´ DMAX - 1 - DSYNC )
è f SYNC ( kHz ) ø
(10)
Consequently to minimize the impact of the tSYNC it is clearly advantageous to utilize the smallest synchronization
pulse width feasible.
NOTE
When external synchronization is used, a propagation delay of approximately 50 ns to 100
ns exists between internal timing circuits and the SYNC signal's falling edge, which may
result in reduced off-time at the highest of switching frequencies. Therefore, RDMX should
be adjusted downward slightly by (TSYNC-0.1 μs)/TSYNC to compensate. At lower SYNC
frequencies, this delay becomes an insignificant fraction of the PWM period, and can be
neglected.
Copyright © 2010–2011, Texas Instruments Incorporated
17
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
Multi-phase Operation
External synchronization also facilitates using more than 2 phases for interleaving. Multiple UCC28070s can
easily be paralleled to add an even number of additional phases for higher-power applications. With appropriate
phase-shifting of the synchronization signals, even more input and output ripple current cancellation can be
obtained. (An odd number of phases can be accommodated if desired, but the ripple cancellation would not be
optimal.) For 4-, 6-, or any 2 x n-phases (where n = the number of UCC28070 controllers), each controller should
receive a SYNC signal which is 360/n degrees out of phase with each other. For a 4-phase application
interleaving with two controllers, SYNC1 should be 180° out of phase with SYNC2 for optimal ripple cancellation.
Similarly for a 6-phase system, SYNC1, SYNC2, and SYNC3 should be 120° out of phase with each other for
optimal ripple cancellation.
In a multi-phase interleaved system, each current loop is independent and treated separately, however there is
only one common voltage loop. To maintain a single control loop, all VSENSE, VINAC, SS, IMO and VAO
signals are paralleled, respectively between the n controllers. Where current-source outputs are combined (SS,
IMO, VAO), the calculated load impedances must be adjusted by 1/n to maintain the same performance as with
a single controller.
Figure 18 illustrates the paralleling of two controllers for a 4-phase 90°-interleaved PFC system.
VSENSE and VINAC Resistor Configuration
The primary purpose of the VSENSE input is to provide the voltage feedback from the output to the voltage
control loop. Thus, a traditional resistor-divider network needs to be sized and connected between the output
capacitor and the VSENSE pin to set the desired output voltage based on the 3-V regulation voltage on
VSENSE.
A unique aspect of the UCC28070 is the need to place the same resistor-divider network on the VIN side of the
inductor to the VINAC pin. This provides the scaled input voltage monitoring needed for the linear multiplier and
current synthesizer circuitry. It is not required that the actual resistance of the VINAC network be identical to the
VSENSE network, but it is necessary that the attenuation (kR) of the two divider networks be equivalent for
proper PFC operation.
kR =
RB
(RA + RB )
(11)
In noisy environments, it may be beneficial for small filter capacitors to be applied to the VSENSE and VINAC
inputs to avoid the destabilizing effects of excessive noise on these inputs. If applied, the RC time-constant
should not exceed 100μs on the VSENSE input to avoid significant delay in the output transient response. The
RC time-constant should also not exceed 100 μs on the VINAC input to avoid degrading of the wave-shape
zero-crossings. Usually, a time constant of 3/fPWM is adequate to filter out typical noise on VSENSE and VINAC.
Some design and test iteration may be required to find the optimal amount of filtering required in a particular
application.
VSENSE and VINAC Open Circuit Protection
Both the VSENSE and VINAC pins have been designed with an internal 250-nA current sink to ensure that in the
event of an open circuit at either pin, the voltage is not left undefined, and the UCC28070 remains in a "safe"
operating mode.
18
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
V IN
–
L1
D1
+
To CSB1
VREF1
RDMX1
1 CDR
DMAX 20
2 RDM
RT 19
3 VAO
SS 18
T1
RS1
RRT1
RA
4 VSENSE
GDB 17
5 VINAC
GND 16
6 IMO
VCC 15
7 RSYNTH
GDA 14
M1
12V to 21V
RB
CSB1
8 CSB
VREF 13
9 CSA
CAOA 12
L2
VREF1
D2
From Ixfrms
CSA1
To CSA1
10 PKLMT
CAOB 11
T2
RSYN1
RS 2
M2
CZV
CZC
RPK1
CREF
RIMO
CPV
CPC
RPK2
CZC
CPC
CSS
RZV
RZC
RZC
VOUT
RZC
RZC
RA
COUT
CPC
CZC
CPC
CZC
RB
CREF
Vin
L3
D3
RSYN2
To CSA2
10 PKLMT
T3
RS 3
CAOB 11
CSB2
9 CSA
CAOA 12
8 CSB
VREF 13
From Ixfrms
VREF2
CSA2
M3
7 RSYNTH
GDA 14
6 IMO
VCC 15
5 VINAC
GND 16
4 VSENSE
GDB 17
3 VAO
SS 18
2 RDM
RT 19
1 CDR
DMAX 20
12V to 21V
L4
D4
RRT2
To CSB2
RDMX2
Synchronized
Clocks
w/ 180 o
Phase Shift
RS 4
T4
M4
Figure 18. Simplified Four-Phase Application Diagram Using Two UCC28070
Copyright © 2010–2011, Texas Instruments Incorporated
19
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
Current Synthesizer
One of the most prominent innovations in the UCC28070 design is the current synthesizer circuitry that
synchronously monitors the instantaneous inductor current through a combination of on-time sampling and
off-time down-slope emulation.
During the on-time of the GDA and GDB outputs, the inductor current is recorded at the CSA and CSB pins
respectively via the current transformer network in each output phase. Meanwhile, the continuous monitoring of
the input and output voltage via the VINAC and VSENSE pins permits the UCC28070 to internally recreate the
inductor current's down-slope during each output's respective off-time. Through the selection of the RSYNTH
resistor (RSYN), based on the equation below, the internal response may be adjusted to accommodate the wide
range of inductances expected across the wide array of applications.
During inrush surge events at power-up and ac drop-out recovery, VSENSE < VINAC, so the synthesized down
slope becomes zero. In this case, the synthesized inductor current will remain above the IMO reference and the
current loop drives the duty cycle to zero. This avoids excessive stress on the MOSFETS during the surge event.
Once VINAC falls below VSENSE the duty cycle increases until steady-state operation resumes.
Waveform at
CSx input
Synthesized
down-slope
Current Synthesizer
output to CA
Figure 19. Inductor Current's Down Slope
RSYN (k W ) =
(10 ´ N
CT
´ LB (m H )´ k R )
RS (W )
(12)
Variables
• LB = Nominal Zero-Bias Boost Inductance (μH),
• RS = Sense Resistor (Ω),
• NCT = Current-sense Transformer turns ratio,
• kR = RB/(RA+RB) = the resistor-divider attenuation at the VSENSE and VINAC pins.
20
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
Programmable Peak Current Limit
The UCC28070 has been designed with a programmable cycle-by-cycle peak current limit dedicated to disabling
either GDA or GDB output whenever the corresponding current-sense input (CSA or CSB respectively) rises
above the voltage established on the PKLMT pin. Once an output has been disabled via the detection of peak
current limit, the output remains disabled until the next clock cycle initiates a new PWM period. The programming
range of the PKLMT voltage extends to upwards of 4 V to permit the full utilization of the 3-V average current
sense signal range, however it should be noted that the linearity of the current amplifiers begin to compress
above 3.6 V.
A resistor-divider network from VREF to GND can easily program the peak current limit voltage on PKLMT,
provided the total current out of VREF is less than 2 mA to avoid drooping of the 6-V VREF voltage. A load of
less than 0.5 mA is suggested, but if the resistance on PKLMT is very high, a small filter capacitor on PKLMT is
recommended to avoid operational problems in high-noise environments.
Externally Programmable Peak
Current Limit level (PKLMT)
PKLMT
10
IPEAKx
+
To Gate-Drive
Shut-down
CSx
Current
Synthesizer
DI
To Current
Amplifier
3V Average Current-sense
Signal Range, plus Ripple
Figure 20. Externally Programmable Peak Current Limit
Copyright © 2010–2011, Texas Instruments Incorporated
21
UCC28070-Q1
SLUSA71A – JULY 2010 – REVISED JUNE 2011
www.ti.com
Linear Multiplier
The multiplier of the UCC28070 generates a reference current which represents the desired wave shape and
proportional amplitude of the ac input current. This current is converted to a reference voltage signal by the RIMO
resistor, which is scaled in value to match the voltage of the current-sense signals. The instantaneous multiplier
current is dependent upon the rectified, scaled input voltage VVINAC and the voltage-error amplifier output VVAO.
The VVINAC signal conveys three pieces of information to the multiplier:
1. The overall wave-shape of the input voltage (typically sinusoidal),
2. The instantaneous input voltage magnitude at any point in the line cycle,
3. The rms level of the input voltage.
The VVAO signal represents the total output power of the PFC pre-regulator.
A major innovation in the UCC28070 multiplier architecture is the internal quantized VRMS feed-forward (QVFF)
circuitry, which eliminates the requirement for external filtering of the VINAC signal and the subsequent slow
response to transient line variations. A unique circuit algorithm detects the transition of the peak of VVINAC
through seven thresholds and generates an equivalent VFF level centered within the eight QVFF ranges. The
boundaries of the ranges expand with increasing VIN to maintain an approximately equal-percentage delta
between levels. These eight QVFF levels are spaced to accommodate the full "universal" line range of 85 V-265
VRMS.
A great benefit of the QVFF architecture is that the fixed kVFF factors eliminate any contribution to distortion of the
multiplier output, unlike an externally-filtered VINAC signal which unavoidably contains 2nd-harmonic distortion
components. Furthermore, the QVFF algorithm allows for rapid response to both increasing and decreasing
changes in input rms voltage so that disturbances transmitted to the PFC output are minimized. 5% hysteresis in
the level thresholds help avoid "chattering" between QVFF levels for VVINAC voltage peaks near a particular
threshold or containing mild ringing or distortion. The QVFF architecture requires that the input voltage be largely
sinusoidal, and relies on detecting zero-crossings to adjust QVFF downward on decreasing input voltage.
Zero-crossings are defined as VVINAC falling below 0.7 V for at least 50 μs typically.
Table 1 reflects the relationship between the various VINAC peak voltages and the corresponding kVFF terms for
the multiplier equation.
Table 1. VINAC Peak Voltages
(1)
22
LEVEL
VVINAC PEAK VOLTAGE
kVFF (V2)
8
2.60 V ≤ VVINAC(pk)
3.857
> 345 V
7
2.25 V ≤ VVINAC(pk) < 2.60 V
2.922
300 V to 345 V
6
1.95 V ≤ VVINAC(pk) < 2.25 V
2.199
260 V to 300 V
5
1.65 V ≤ VVINAC(pk) < 1.95 V
1.604
220 V to 260 V
4
1.40 V ≤ VVINAC(pk) < 1.65 V
1.156
187 V to 220 V
3
1.20 V ≤ VVINAC(pk) < 1.40 V
0.839
160 V to 187 V
2
1.00 V ≤ VVINAC(pk) < 1.20 V
0.600
133 V to 160 V
1
VVINAC(pk) ≤ 1.00 V
0.398
< 133 V
VIN PEAK VOLTAGE
(1)
The VIN peak voltage boundary values listed above are calculated based on a 400-V PFC output voltage and the use of a matched
resistor-divider network (kR = 3 V/400 V = 0.0075) on VINAC and VSENSE (as required for current synthesis). When VOUT is designed
to be higher or lower than 400 V, kR = 3 V/VOUT, and the VIN peak voltage boundary values for each QVFF level adjust to VVINAC(pk)/kR.
Copyright © 2010–2011, Texas Instruments Incorporated
UCC28070-Q1
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The multiplier output current IIMO for any line and load condition can thus be determined by the equation
I IMO =
17 m A ´ (VVINAC )´ (VVAO - 1)
kVFF
(13)
2
Because the kVFF value represents the scaled VRMS at the center of a level, VVAO will adjust slightly upwards or
downwards when VINACpk is either lower or higher than the center of the QVFF voltage range to compensate for
the difference. This is automatically accomplished by the voltage loop control when VIN varies, both within a level
and after a transition between levels.
The output of the voltage-error amplifier VAO is clamped at 5.0 V, which represents the maximum PFC output
power. This value is used to calculate the maximum reference current at the IMO pin, and sets a limit for the
maximum input power allowed (and, as a consequence, limits maximum output power).
Unlike a continuous VFF situation, where maximum input power is a fixed power at any VRMS input, the discrete
QVFF levels permit a variation in maximum input power within limited boundaries as the input VRMS varies within
each level.
The lowest maximum power limit occurs at the VINAC voltage of 0.76 V, while the highest maximum power limit
occurs at the increasing threshold from level-1 to level-2. This pattern repeats at every level transition threshold,
keeping in mind that decreasing thresholds are 95% of the increasing threshold values. Below VINAC = 0.76 V,
PIN is always less than PIN(max), falling linearly to zero with decreasing input voltage.
For example, to design for the lowest maximum power allowable, determine the maximum steady-state (average)
output power required of the PFC pre-regulator and add some additional percentage to account for line drop-out
recovery power (to recharge COUT while full load power is drawn) such as 10% or 20% of POUT(max). Then apply
the expected efficiency factor to find the lowest maximum input power allowable:
PIN (max) =
1.10 ´ POUT (max)
h
(14)
At the PIN(max) design threshold, VVINAC = 0.76 V, hence QVFF = 0.398 and input VAC = 73 VRMS (accounting for
2-V bridge-rectifier drop) for a nominal 400-V output system.
Thus I IN ( rms ) =
PIN (max)
73VRMS
, and I IN ( pk ) = 1.414 ´ I IN ( rms )
Copyright © 2010–2011, Texas Instruments Incorporated
(15)
23
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This IIN(pk) value represents the combined average current through the boost inductors at the peak of the line
voltage. Each inductor current is detected and scaled by a current-sense transformer (CT). Assuming equal
currents through each interleaved phase, the signal voltage at each current sense input pin (CSA and CSB) is
developed across a sense resistor selected to generate ~3 V based on (1/2) x IIN(pk) x RS/NCT, where RS is the
current sense resistor and NCT is the CT turns-ratio.
IIMO is then calculated at that same lowest maximum-power point, as
I IMO(max) = 17 m A ´
(0.76V )(5V - 1V ) = 130m A
0.398
(16)
RIMO is selected such that:
R
æ1ö
RIMO ´ I IMO(max) = ç ÷ ´ I IN ( pk ) ´ S
N CT
è2ø
(17)
Therefore:
RIMO
ææ 1 ö
ö
ç ç 2 ÷ ´ I IN ( pk ) ´ RS ÷
è ø
ø
=è
(NCT ´ I IMO(max) )
(18)
At the increasing side of the level-1 to level-2 threshold, it should be noted that the IMO current would allow
higher input currents at low-line:
I IMO( L1- L 2 ) = 17 m A ´
(1.0V )(5V - 1V ) = 171m A
0.398
(19)
However, this current may easily be limited by the programmable peak current limiting (PKLMT) feature of the
UCC28070 if required by the power stage design.
The same procedure can be used to find the lowest and highest input power limits at each of the QVFF level
transition thresholds. At higher line voltages, where the average current with inductor ripple is traditionally below
the PKLMT threshold, the full variation of maximum input power will be seen, but the input currents will inherently
be below the maximum acceptable current levels of the power stage.
The performance of the multiplier in the UCC28070 has been significantly enhanced when compared to previous
generation PFC controllers, with high linearity and accuracy over most of the input ranges. The accuracy is at its
worst as VVAO approaches 1 V because the error of the (VVAO-1) subtraction increases and begins to distort the
IMO reference current to a greater degree.
24
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Enhanced Transient Response (VA Slew-Rate Correction)
Due to the low voltage loop bandwidth required to maintain proper PFC and ignore the slight 120-Hz ripple on
the output, the response of ordinary controllers to input voltage and load transients will also be slow. However,
the QVFF function effectively handles the line transient response with the exception of any minor adjustments
needed within a QVFF level. Load transients on the other hand can only be handled by the voltage loop, therefore,
the UCC28070 has been designed to improve its transient response by pulling up on the output of the voltage
amplifier (VAO) with an additional 100 μA of current in the event the VSENSE voltage drops below 93% of
regulation (2.79 V). During a soft-start cycle, when VSENSE is ramping up from the 0.75-V PFC Enable
threshold, the 100-μA correction current source is disabled to ensure the gradual and controlled ramping of
output voltage and current during a soft start.
Voltage Biasing (VCC and VREF)
The UCC28070 operates within a VCC bias supply range of 10 V to 21 V. An Under-Voltage Lock-Out (UVLO)
threshold prevents the PFC from activating until VCC > 10.2 V, and 1 V of hysteresis assures reliable start-up
from a possibly low-compliance bias source. An internal 25-V zener-like clamp on VCC is intended only to protect
the device from brief energy-limited surges from the bias supply, and should NOT be used as a regulator with a
current-limited source.
At minimum, a 0.1-μF ceramic bypass capacitor must be applied from VCC to GND close to the device pins to
provide local filtering of the bias supply. Larger values may be required depending on ICC peak current
magnitudes and durations to minimize ripple voltage on VCC.
In order to provide a smooth transition out of UVLO and to make the 6-V voltage reference available as early as
possible, the VREF output is enabled when VCC exceeds 8 V typically.
The VREF circuitry is designed to provide the biasing of all internal control circuits and for limited use externally.
At minimum, a 22-nF ceramic bypass capacitor must be applied from VREF to GND close to the device pins to
ensure stability of the circuit. External load current on VREF should be limited to less than 2 mA, or degraded
regulation may result.
PFC Enable and Disable
The UCC28070 contains two independent circuits dedicated to disabling the GDx outputs based on the biasing
conditions of the VSENSE or SS pins. The first circuit which monitors the VVSENSE, is the traditional PFC Enable
that holds off soft-start and the overall PFC function until the output has pre-charged to ~25%. Prior to VVSENSE
reaching 0.75 V, almost all of the internal circuitry is disabled. Once VVSENSE reaches 0.75 V and VAO < 0.75 V,
the oscillator, multiplier, and current synthesizer are enabled and the SS circuitry begins to ramp up the voltage
on the SS pin. The second circuit provides an external interface to emulate an internal fault condition to disable
the GDx output without fully disabling the voltage loop and multiplier. By externally pulling the SS pin below 0.6
V, the GDx outputs are immediately disabled and held low. Assuming no other fault conditions are present,
normal PWM operation resumes when the external SS pulldown is released. It must be noted that the external
pulldown needs to be sized large enough to override the internal 1.5-mA adaptive SS pullup once the SS voltage
falls below the disable threshold. It is recommended that a MOSFET with less than 100-Ω RDS(on) resistance be
used to ensure the SS pin is held adequately below the disable threshold.
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Adaptive Soft Start
In order to maintain a controlled power up, the UCC28070 has been designed with an adaptive soft-start function
that overrides the internal reference voltage with a controlled voltage ramp during power up. On initial power up,
once VVSENSE exceeds the 0.75-V enable threshold (VEN), the internal pull down on the SS pin is released, and
the 1.5-mA adaptive soft-start current source is activated. This 1.5-mA pullup almost immediately pulls the SS pin
to 0.75 V (VVSENSE) to bypass the initial 25% of dead time during a traditional 0 V to Vregulation SS ramp. Once
the SS pin has reached the voltage on VSENSE, the 10-μA soft-start current (ISS) takes over. Thus, through the
selection of the soft-start capacitor (CSS), the effective soft-start time (tSS) may be easily programmed based on
the equation below.
æ 2.25V ö
tSS = CSS ´ ç
÷
è 10 m A ø
(20)
Often, a system restart is desired following a brief shut-down. In such a case, VSENSE may still have substantial
voltage if VOUT has not fully discharged or if high line has peak charged COUT. To eliminate the delay caused by
charging CSS from 0 V up to the pre-charged VVSENSE with only the 10-μA current source and minimize any
further output voltage sag, the adaptive soft start uses a 1.5-mA current source to rapidly charge CSS to VVSENSE,
after which time the 10-μA source controls the VSS accent to the desired soft-start ramp rate. In such a case, tSS
is estimated as follows:
æ 3V - VVSENSE 0 ö
tSS = CSS ´ ç
÷
è 10 m A
ø
(21)
where VVSENSE0 is the voltage at VSENSE at the moment a soft start or restart is initiated.
NOTE
For soft start to be effective and avoid overshoot on VOUT, the SS ramp must be slower
than the voltage-loop control response. Choose CSS ≥ CVZ to ensure this.
(V)
VSS
VVSENSE
VSS if no adaptive current
Time (s)
PFC externally
disabled due to
AC-line drop-out
Reduced delay to regulation
AC-Line recovers
and SS pin released
Figure 21. Soft-Start Ramp Rate
26
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UCC28070-Q1
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SLUSA71A – JULY 2010 – REVISED JUNE 2011
PFC Start-Up Hold Off
An additional feature designed into the UCC28070 is the "Start-Up Hold Off" logic that prevents the device from
initiating a soft-start cycle until the VAO is below the zero-power threshold (0.75 V). This feature ensures that the
SS cycle will initiate from zero-power and zero duty-cycle while preventing the potential for any significant inrush
currents due to stored charge in the VAO compensation network.
Output Over-Voltage Protection (OVP)
Because of the high voltage output and a limited design margin on the output capacitor, output over-voltage
protection is essential for PFC circuits. The UCC28070 implements OVP through the continuous monitoring of
the VSENSE voltage. In the event VVSENSE rises above 106% of regulation (3.18 V), the GDx outputs are
immediately disabled to prevent the output voltage from reaching excessive levels. Meanwhile the CAOx outputs
are pulled low in order to ensure a controlled recovery starting from 0% duty-cycle after an OVP fault is released.
Once the VVSENSE voltage has dropped below 3.08 V, the PWM operation resumes normal operation.
Zero-Power Detection
In order to prevent undesired performance under no-load and near no-load conditions, the UCC28070
zero-power detection comparator is designed to disable both GDA and GDB output in the event the VAO voltage
falls below 0.75 V. The 150 mV of hysteresis ensures that the output remains disabled until the VAO has nearly
risen back into the linear range of the multiplier (VAO ≥ 0.9 V).
Thermal Shutdown
In order to protect the power supplies from silicon failures at excessive temperatures, the UCC28070 has an
internal temperature-sensing comparator that shuts down nearly all of the internal circuitry, and disables the GDA
and GDB outputs, if the die temperature rises above 160°C. Once the die temperature falls below 140°C, the
device brings the outputs up through a typical soft start.
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Current Loop Compensation
The UCC28070 incorporates two identical and independent transconductance-type current-error amplifiers (one
for each phase) with which to control the shaping of the PFC input current waveform. The current-error amplifier
(CA) forms the heart of the embedded current control loop of the boost PFC pre-regulator, and is compensated
for loop stability using familiar principles [4, 5]. The output of the CA for phase-A is CAOA, and that for phase-B
is CAOB. Since the design considerations are the same for both, they are collectively referred to as CAOx,
where the "x" may be "A" or "B".
In a boost PFC pre-regulator, the current control loop comprises the boost power plant stage, the current sensing
circuitry, the wave-shape reference, the PWM stage, and the CA with compensation components. The CA
compares the average boost inductor current sensed with the wave-shape reference from the multiplier stage
and generates an output current proportional to the difference.
This CA output current flows through the impedance of the compensation network generating an output voltage,
VCAO, which is then compared with a periodic voltage ramp to generate the PWM signal necessary to achieve
PFC.
IMO
+
CAOx
CAx
CSx
Current
Synthesizer
CZC
gmc = 100µS
C PC
R ZC
Figure 22. Current Error Amplifier With Type II Compensation
For frequencies above boost LC resonance and below fPWM, the small-signal model of the boost stage, which
includes current sensing, can be simplified to:
R
Vout ´ S
vRS
N CT
=
vCA DVRMP ´ kSYNC ´ s ´ LB
(22)
where LB = mid-value boost inductance, RS = CT sense resistor, NCT = CT turns ratio, VOUT = average output
voltage, ∆VRMP = 4Vpk-pk amplitude of the PWM voltage ramp, kSYNC = ramp reduction factor (if PWM frequency is
synchronized to an external oscillator; kSYNC = 1 otherwise), s = Laplace complex variable
An RZCCZC network is introduced on CAOx to obtain high gain for the low-frequency content of the inductor
current signal, but reduced flat gain above the zero frequency out to fPWM to attenuate the high-frequency
switching ripple content of the signal (thus averaging it).
28
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The switching ripple voltage should be attenuated to less than 1/10 of the ΔVRMP amplitude so as to be
considered "negligible" ripple.
Thus, CAOx gain at fPWM is:
DVRMP ´ k SYNC
g mc Rzc £
DI LB ´
RS
10
N CT
(23)
where ∆ILB is the maximum peak-to-peak ripple current in the boost inductor, and gmc is the transconductance of
the CA, 100 μS.
Rzc £
4V ´ N CT
10 ´100 m S ´ DI LB ´ RS
(24)
The current-loop cross-over frequency is then found by equating the open loop gain to 1 and solving for fCXO:
Vout ´
fCXO =
RS
N CT
´ g mc Rzc
DVRMP ´ kSYNC ´ 2p ´ LB
(25)
CCZ is then determined by setting fZC = fCXO = 1/(2πxRZCxCZC) and solving for CZC. At fZC = fCXO, a phase margin
of 45° is obtained at fCXO. Greater phase margin may be had by placing fZC < fCXO.
An additional high-frequency pole is generally added at fPWM to further attenuate ripple and noise at fPWM and
higher. This is done by adding a small-value capacitor, Cpc, across the RzcCzc network.
Cpc =
1
2p ´ f PWM ´ Rzc
(26)
The procedure above is valid for fixed-value inductors.
NOTE
If a "swinging-choke" boost inductor (inductance decreases with increasing current) is
used, fCXO varies with inductance, so CZC should be determined at maximum inductance.
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Voltage Loop Compensation
The outer voltage control loop of the dual-phase PFC controller functions the same as with a single-phase
controller, and compensation techniques for loop stability are standard [4]. The bandwidth of the voltage-loop
must be considerably lower than the twice-line ripple frequency (f2LF) on the output capacitor, to avoid
distortion-causing correction to the output voltage. The output of the voltage-error amplifier (VA) is an input to the
multiplier, to adjust the input current amplitude relative to the required output power. Variations on VAO within the
bandwidth of the current loops will influence the wave-shape of the input current. Since the low-frequency ripple
on COUT is a function of input power only, its peak-to-peak amplitude is the same at high-line as at low-line. Any
response of the voltage-loop to this ripple will have a greater distorting effect on high-line current than on low-line
current. Therefore, the allowable percentage of 3rd-harmonic distortion on the input current contributed by VAO
should be determined using high-line conditions.
Because the voltage-error amplifier (VA) is a transconductance type of amplifier, the impedance on its input has
no bearing on the amplifier gain, which is determined solely by the product of its transconductance (gmv) with its
output impedance (ZOV). Thus the VSENSE input divider-network values are determined separately, based on
criteria discussed in the VINAC section. Its output is the VAO pin.
3V
+
VAO
VA
C ZV
VSENSE
gmv = 70µS
CPV
RZV
Figure 23. Voltage Error Amplifier With Type II Compensation
The twice-line ripple voltage component of VSENSE must be sufficiently attenuated and phase-shifted at VAO to
achieve the desired level of 3rd-harmonic distortion of the input current wave-shape [4]. For every 1% of
3rd-harmonic input distortion allowable, the small-signal gain GVEA = VVAOpk / vSENSEpk = gmvxZOV at the twice-line
frequency should allow no more than 2% ripple over the full VAO voltage range. In the UCC28070, VVAO can
range from 1 V at zero load power to ~4.2 V(see note below) at full load power for a ΔVVAO = 3.2 V, so 2% of 3.2
V is 64-mV peak ripple.
NOTE
Although the maximum VAO voltage is clamped at 5 V, at full load VVAO may vary around
an approximate center point of 4.2 V to compensate for the effects of the quantized
feed-forward voltage in the multiplier stage (see Multiplier Section for details). Therefore,
4.2 V is the proper voltage to use to represent maximum output power when performing
voltage-loop gain calculations.
30
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The output capacitor maximum low-frequency zero-to-peak ripple voltage is closely approximated by:
v0 pk =
Pinavg ´ X Cout
Voutavg
=
Pinavg
Voutavg ´ 2p ´ f 2 LF ´ Cout
(27)
where PIN(avg) is the total maximum input power of the interleaved-PFC pre-regulator, VOUT(avg) is the average
output voltage and COUT is the output capacitance.
VSENSEpk = vopkxkR, where kR is the gain of the resistor-divider network on VSENSE.
Thus, for k3rd% of allowable 3rd-harmonic distortion on the input current attributable to the VAO ripple,
Z OV ( f2 LF ) =
k3rd ´ 64mV ´ Voutavg ´ 2p f 2 LF ´ Cout
g mv ´ k R ´ Pinavg
(28)
This impedance on VAO is set by a capacitor (Cpv), where CPV = 1/( 2πf2LFxZOV(f2LF)) therefore,
Cpv =
g mv ´ k R ´ Pinavg
k3rd ´ 64mV ´ Voutavg ´ ( 2p f 2 LF )2 ´ Cout
(29)
The voltage-loop unity-gain cross-over frequency (fVXO) may now be solved by setting the open-loop gain equal
to 1:
æ Pinavg ´ X Cout
Tv( fVXO ) = GBST ´ GVEA ´ k R = ç
ç DVVAO ´ Voutavg
è
fVXO 2 =
so,
ö
÷÷ ´ (g mv ´ X Cpv )´ k R = 1
ø
(30)
g mv ´ k R ´ Pinavg
2
DVVAO ´ Voutavg ´ (2p ) ´ Cpv ´ Cout
(31)
The "zero-resistor" (RZV) from the zero-placement network of the compensation may now be calculated. Together
with CPV, RZV sets a pole right at fVXO to obtain 45° phase margin at the cross-over.
Rzv =
Thus,
1
2p fVXO ´ Cpv
(32)
Finally, a zero is placed at or below fVXO/6 with capacitor CZV to provide high gain at dc but with a breakpoint far
enough below fVXO so as not to significantly reduce the phase margin. Choosing fVXO/10 allows one to
approximate the parallel combination value of CZV and CPV as CZV, and solve for CZV simply as:
Czv =
10
» 10 ´ Cpv
2p fVXO ´ Rzv
(33)
By using a spreadsheet or math program, CZV, RZV, and CPV may be manipulated to observe their effects on fVXO
and phase margin and %-contribution to 3rd-harmonic distortion (see note below). Also, phase margin may be
checked as PIN(avg) level and system parameter tolerances vary.
NOTE
The percent of 3rd-harmonic distortion calculated in this section represents the
contribution from the f2LF voltage ripple on COUT only. Other sources of distortion, such as
the current-sense transformer, the current synthesizer stage, even distorted VIN, etc., can
contribute additional 3rd and higher harmonic distortion.
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Advanced Design Techniques
Current Loop Feedback Configuration
(Sizing of the Current Transformer Turns Ratio and Sense Resistor (RS)
A current-sense transformer (CT) is typically used in high-power applications to sense inductor current while
avoiding significant losses in the sensing resistor. For average current-mode control, the entire inductor current
waveform is required; however low-frequency CTs are obviously impracticable. Normally, two high-frequency
CTs are used, one in the switching leg to obtain the up-slope current and one in the diode leg to obtain the
down-slope current. These two current signals are summed together to form the entire inductor current, but this
is not the case for the UCC28070.
A major advantage of the UCC28070 design is the current synthesis function, which internally recreates the
inductor current down-slope during the switching period off-time. This eliminates the need for the diode-leg CT in
each phase, significantly reducing space, cost and complexity. A single resistor programs the synthesizer down
slope, as previously discussed in the Current Synthesizer section.
A number of trade-offs must be made in the selection of the CT. Various internal and external factors influence
the size, cost, performance, and distortion contribution of the CT.
These factors include, but are not limited to:
• Turns-ratio (NCT)
• Magnetizing inductance (LM)
• Leakage inductance (LLK)
• Volt-microsecond product (Vμs)
• Distributed capacitance (Cd)
• Series resistance (RSER)
• External diode drop (VD)
• External current sense resistor (RS)
• External reset network
Traditionally, the turns-ratio and the current sense resistor are selected first. Some iterations may be needed to
refine the selection once the other considerations are included.
32
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In general, 50 ≤ NCT ≤ 200 is a reasonable range from which to choose. If NCT is too low, there may be high
power loss in RS and insufficient LM. If too high, there could be excessive LLK and Cd. (A one-turn primary
winding is assumed.)
LLK
IDS
1
NCT
LM
iM
CSx
RSER
D
Cd
Reset
Network
RS
Figure 24. Current Sense Transformer Equivalent Circuit
A major contributor to distortion of the input current is the effect of magnetizing current on the CT output signal
(iRS). A higher turns-ratio results in a higher LM for a given core size. LM should be high enough that the
magnetizing current (iM) generated is a very small percentage of the total transformed current. This is an
impossible criterion to maintain over the entire current range, because iM unavoidably becomes a larger fraction
of iRS as the input current decreases toward zero. The effect of iM is to "steal" some of the signal current away
from RS, reducing the CSx voltage and effectively understating the actual current being sensed. At low currents,
this understatement can be significant and CAOx increases the current-loop duty-cycle in an attempt to correct
the CSx input(s) to match the IMO reference voltage. This unwanted correction results in overstated current on
the input wave shape in the regions where the CT understatement is significant, such as near the ac line zero
crossings. It can affect the entire waveform to some degree under the high line, light-load conditions.
The sense resistor RS is chosen, in conjunction with NCT, to establish the sense voltage at CSx to be about 3 V
at the center of the reflected inductor ripple current under maximum load. The goal is to maximize the average
signal within the common-mode input range VCMCAO of the CAOx current-error amplifiers, while leaving room for
the peaks of the ripple current within VCMCAO. The design condition should be at the lowest maximum input power
limit as determined in the Multiplier Section. If the inductor ripple current is so high as to cause VCSx to exceed
VCMCAO, then RS or NCT or both must be adjusted to reduce peak VCSx, which could reduce the average sense
voltage center below 3 V. There is nothing wrong with this situation; but be aware that the signal is more
compressed between full- and no-load, with potentially more distortion at light loads.
The matter of volt-second balancing is important, especially with the widely varying duty-cycles in the PFC stage.
Ideally, the CT is reset once each switching period; that is, the off-time Vμs product equals the on-time Vμs
product. (Because a switching period is usually measured in microseconds, it is convenient to convert the
volt-second product to volt-microseconds to avoid sub-decimal numbers.) On-time Vμs is the time-integral of the
voltage across LM generated by the series elements RSER, LLK, D, and RS. Off-time Vμs is the time-integral of the
voltage across the reset network during the off-time. With passive reset, Vμs-off is unlikely to exceed Vμs-on.
Sustained unbalance in the on or off Vμs products will lead to core saturation and a total loss of the
current-sense signal. Loss of VCSx causes VCAOx to quickly rise to its maximum, programming a maximum
duty-cycle at any line condition. This, in turn causes the boost inductor current to increase without control, until
the system fuse or some component failure interrupts the input current.
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It is vital that the CT has plenty of Vμs design-margin to accommodate various special situations where there to
be several consecutive maximum duty-cycle periods at maximum input current, such as during peak current
limiting.
Maximum Vμs(on) can be estimated by:
Vm (on )max = tON (max ) ´ (VRS + VD + VRSER + VLK )
(34)
where all factors are maximized to account for worst-case transient conditions and tON(max) occurs during the
lowest dither frequency when frequency dithering is enabled. For design margin, a CT rating of ~5*Vμs(on)max
or higher is suggested. The contribution of VRS varies directly with the line current. However, VD may have a
significant voltage even at near-zero current, so substantial Vμs(on) may accrue at the zero-crossings where the
duty-cycle is maximum. VRSER is the least contributor, and often can be neglected if RSER
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