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UCC28180
SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
UCC28180 Programmable Frequency, Continuous Conduction Mode (CCM), Boost Power
Factor Correction (PFC) Controller
1 Features
3 Description
•
•
The UCC28180 is a flexible and easy-to-use, 8-pin,
active Power-Factor Correction (PFC) controller that
operates under Continuous Conduction Mode (CCM)
to achieve high Power Factor, low current distortion
and excellent voltage regulation of boost preregulators in AC - DC front-ends. The controller is
suitable for universal AC input systems operating in
100-W to few-kW range with the switching frequency
programmable between 18 kHz to 250 kHz, to
conveniently support both power MOSFET and IGBT
switches. An integrated 1.5-A and 2-A (SRC-SNK)
peak gate drive output, clamped internally at 15.2 V
(typical), enables fast turn-on, turn-off, and easy
management of the external power switch without the
need for buffer circuits.
1
•
•
•
•
•
•
•
•
•
•
•
8-Pin Solution (No AC Line Sensing Needed)
Wide Range Programmable Switching Frequency
(18 kHz to 250 kHz for MOSFET and IGBT-based
PFC Converters)
Trimmed Current Loop Circuits for Low iTHD
Reduced Current Sense Threshold (Minimizes
Power Dissipation in Shunt)
Average Current-Mode Control
Soft Over Current and Cycle-by-Cycle Peak
Current Limit Protection
Output Overvoltage Protection With Hysteresis
Recovery
Audible Noise Minimization Circuitry
Open Loop Detection
Enhance Dynamic Response During Output
Overvoltage and Undervoltage Conditions
Maximum Duty Cycle of 96% (Typical)
Burst Mode for No Load Regulation
VCC UVLO, Low ICC Start-Up (< 75 µA)
2 Applications
•
•
•
•
•
Universal AC Input, CCM Boost PFC Converters
in 100-W to Few-kW Range
Server and Desktop Power Supplies
White Good Appliances (Air Conditioners,
Refrigerators)
Industrial Power Supplies (DIN Rail)
Flat Panel (PDP, LCD, and LED) TVs
Low-distortion wave shaping of the input current
using average current mode control is achieved
without input line sensing, reducing the external
component count. In addition, the controller features
reduced current sense thresholds to facilitate the use
of small-value shunt resistors for reduced power
dissipation, especially important in high-power
systems. To enable low current distortion, the
controller also features trimmed internal current loop
regulation circuits for eliminating associated
inaccuracies.
Device Information(1)
PART NUMBER
UCC28180
PACKAGE
BODY SIZE (NOM)
SOIC (8)
4.90 mm × 3.91 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
space
space
space
Typical Application Schematic
VOUT
EMI Filter
LINE
INPUT
±
Bridge
Rectifier
+
1
GND
2
ICOMP
3
ISENSE
4
FREQ
GATE
8
VCC
7
VSENSE
6
VCOMP
5
Auxilary
Supply
Rload
Copyright © 2016, Texas Instruments Incorporated
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
UCC28180
SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
7
8
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Description (Continued) ........................................
Pin Configuration and Functions .........................
Specifications.........................................................
1
1
1
2
3
4
5
7.1
7.2
7.3
7.4
7.5
7.6
5
5
5
5
6
8
Absolute Maximum Ratings ......................................
ESD Ratings ............................................................
Recommended Operating Conditions.......................
Thermal Information ..................................................
Electrical Characteristics...........................................
Typical Characteristics ..............................................
Detailed Description ............................................ 12
8.1 Overview ................................................................. 12
8.2 Functional Block Diagram ....................................... 13
8.3 Feature Description................................................. 14
8.4 Device Functional Modes........................................ 20
9
Application and Implementation ........................ 21
9.1 Application Information............................................ 21
9.2 Typical Application .................................................. 22
10 Power Supply Recommendations ..................... 36
10.1 Bias Supply ........................................................... 36
11 Layout................................................................... 36
11.1 Layout Guidelines ................................................. 36
11.2 Layout Example .................................................... 38
12 Device and Documentation Support . . ............. 39
12.1
12.2
12.3
12.4
12.5
12.6
Documentation Support .......................................
Receiving Notification of Documentation Updates
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
39
39
39
39
39
39
13 Mechanical, Packaging, and Orderable
Information ........................................................... 39
4 Revision History
Changes from Revision C (April 2016) to Revision D
Page
•
Changed the correct page number in the C Revision History of the diode addition to the Functional Block Diagram. ......... 2
•
Changed text value of 0.538 to 0.366 to align with Equation 85. Updated change was implemented in the C revision
and recorded in the D revision. ............................................................................................................................................ 31
•
Added D4 to Table 2. Updated change was implemented in the C revision and recorded in the D revision. ..................... 37
•
Added Receiving Notification of Documentation Updates.................................................................................................... 39
•
Added Community Resources.............................................................................................................................................. 39
Changes from Revision B (December 2014) to Revision C
Page
•
Added a diode to the Typical Application Schematic image. ................................................................................................. 1
•
Changed ICC Standby current MAX rate from 2.95 mA to 3.47 mA...................................................................................... 6
•
Changed ISENSE threshold, soft over current (SOC) TYP value from –0.295 V to –0.285 V. ............................................. 6
•
Changed Maximum current under EDR operation MAX rating from –241 µA to –275 µA. ................................................... 6
•
Added a diode to the Functional Block Diagram. ................................................................................................................. 13
•
Added Diode to Soft Overcurrent/Peak-Current Limit image. .............................................................................................. 17
•
Added ISENSE Pin section. ................................................................................................................................................ 18
•
Added diode to the Design Example Schematic image. ..................................................................................................... 22
•
Changed Equation 101 3kHz to 5kHz. ................................................................................................................................. 32
•
Changed Recommended Layout for UCC28180 image....................................................................................................... 38
Changes from Revision A (November 2013) to Revision B
•
2
Page
Added ESD Rating table, Feature Description section, Device Functional Modes, Application and Implementation
section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and
Mechanical, Packaging, and Orderable Information section ................................................................................................. 1
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SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
5 Description (Continued)
Simple external networks allow for flexible compensation of the current and voltage control loops. In addition,
UCC28180 offers an enhanced dynamic response circuit that is based on the voltage feedback signal to deliver
improved response under fast load transients, both for output overvoltage and undervoltage conditions. An
unique VCOMP discharge circuit provided in UCC28180 is activated whenever the voltage feedback signal
exceeds VOVP_L thus allowing a chance for the control loop to stabilize quickly and avoid encountering the
overvoltage protection function when PWM shutoff can often cause audible noise. Controlled soft start gradually
regulates the input current during start-up and reduces stress on the power switches. Numerous system-level
protection features available in the controller include VCC UVLO, peak current limit, soft overcurrent, output
open-loop detection, output overvoltage protection and open-pin detection (VISNS). A trimmed internal reference
provides accurate protection thresholds and regulation set-point. The user can control low-power standby mode
by pulling the VSENSE pin below 0.82 V.
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UCC28180
SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
www.ti.com
6 Pin Configuration and Functions
8-Pin SOIC
D Package
(TOP VIEW)
1 GND
2 ICOMP
3 ISENSE
4 FREQ
GATE 8
VCC 7
VSENSE 6
VCOMP 5
Pin Functions
PIN
NAME
NO.
GATE
8
GND
1
ICOMP
2
O
DESCRIPTION
Gate Drive: Integrated push-pull gate driver for one or more external power MOSFETs. Typical 2.0-A sink
and 1.5-A source capability. Output voltage is typically clamped at 15.2 V (typical).
Ground: device ground reference.
O
Current Loop Compensation: Transconductance current amplifier output. A capacitor connected to GND
provides compensation and averaging of the current sense signal in the current control loop. The controller is
disabled if the voltage on ICOMP is less than 0.2 V, (ICOMPP protection function).
I
Inductor Current Sense: Input for the voltage across the external current sense resistor, which represents
the instantaneous current through the PFC boost inductor. This voltage is averaged by the current amplifier to
eliminate the effects of ripple and noise. Soft Over Current (SOC) limits the average inductor current. Cycleby-cycle peak current limit (PCL) immediately shuts off the GATE drive if the peak-limit voltage is exceeded.
An internal 2.3-µA current source pulls ISENSE above 0.085 V to shut down PFC operation if this pin
becomes open-circuited, (ISOP protection function). Use a 220-Ω resistor between this pin and the current
sense resistor to limit inrush-surge currents into this pin.
ISENSE
3
VCC
7
Device Supply: External bias supply input. Under-Voltage Lockout (UVLO) disables the controller until VCC
exceeds a turn-on threshold of 11.5 V. Operation continues until VCC falls below the turn-off (UVLO)
threshold of 9.5 V. A ceramic by-pass capacitor of 0.1 µF minimum value should be connected from VCC to
GND as close to the device as possible for high-frequency filtering of the VCC voltage.
VCOMP
5
O
Voltage Loop Compensation: Transconductance voltage error amplifier output. A resistor-capacitor network
connected from this pin to GND provides compensation. VCOMP is held at GND until VCC, and VSENSE
exceed their threshold voltages. Once these conditions are satisfied, VCOMP is charged until the VSENSE
voltage reaches its nominal regulation level. When Enhanced Dynamic Response (EDR) is engaged, a higher
transconductance is applied to VCOMP to reduce the charge or discharge time for faster transient response.
Soft Start is programmed by the capacitance on this pin. VCOMP is pulled low when VCC UVLO,
OLP/Standby, ICOMPP and ISOP functions are activated.
FREQ
4
O
Switching Frequency Setting: This pin allows the setting of the operating switching frequency by connecting
a resistor to ground. The programmable frequency range is from 18 kHz to 250 kHz.
I
Output Voltage Sense: An external resistor-divider network connected from this pin to the PFC output
voltage provides feedback sensing for regulation to the internal 5-V reference voltage. A small capacitor from
this pin to GND filters high-frequency noise. Standby disables the controller and discharges VCOMP when
the voltage at VSENSE drops below the Open-Loop Protection (OLP) threshold of 16.5%VREF (0.82 V). An
internal 100-nA current source pulls VSENSE to GND during pin disconnection. Enhanced Dynamic
Response (EDR) rapidly returns the output voltage to its normal regulation level when a system line or load
step causes VSENSE to rise above 105% or fall below 95% of the reference voltage. Two level Output OverVoltage Protection (OVP): a 4-kΩ resistor connects VCOMP to ground to rapidly discharge VCOMP when
VSENSE exceeds 107% (VOVP_L) of the reference voltage. If VSENSE exceeds 109% (VOVP_H) of the
reference voltage, GATE output will be disabled until VSENSE drops below 102% of the reference voltage.
VSENSE
4
I/O
6
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SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
7 Specifications
7.1 Absolute Maximum Ratings (1)
Over operating free-air temperature range, all voltages are with respect to GND (unless otherwise noted). Currents are
positive into and negative out of the specified terminal.
MIN
MAX
UNIT
VCC, GATE
–0.3
22
V
FREQ, VSENSE, VCOMP, ICOMP
–0.3
7
ISENSE
–24
7
Input current range
VSENSE, ISENSE
–1
1
mA
Junction temperature, TJ
Operating
–55
150
°C
Lead temperature, TSOL
Soldering, 10 s
300
°C
150
°C
Input voltage range
Storage temperature, Tstg
(1)
–65
Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other condition beyond those included under “recommended operating
conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods of time may affect device reliability.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
±2000
Charged-device model (CDM), per JEDEC specification JESD22C101 (2)
±500
UNIT
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
VCC input voltage from a low-impedance source
Operating junction temperature, TJ
Operating frequency
MIN
MAX
VCCOFF + 1V
21
UNIT
V
–40
125
°C
18
250
kHz
7.4 Thermal Information
UCC28180
THERMAL METRIC
(1)
D
UNIT
8 PINS
RθJA
Junction-to-ambient thermal resistance (2)
(3)
116.1
RθJCtop
Junction-to-case (top) thermal resistance
RθJB
Junction-to-board thermal resistance (4)
56.4
ψJT
Junction-to-top characterization parameter (5)
14.4
ψJB
Junction-to-board characterization parameter (6)
55.9
(1)
(2)
(3)
(4)
(5)
(6)
62.2
°C/W
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report (SPRA953).
The junction-to-ambient thermal resistance under natural convection is obtained in a simulation on a JEDEC-standard, high-K board, as
specified in JESD51-7, in an environment described in JESD51-2a.
The junction-to-case (top) thermal resistance is obtained by simulating a cold plate test on the package top. No specific JEDECstandard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
The junction-to-board thermal resistance is obtained by simulating in an environment with a ring cold plate fixture to control the PCB
temperature, as described in JESD51-8.
The junction-to-top characterization parameter, ψJT, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining RθJA, using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-board characterization parameter, ψJB, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining RθJA, using a procedure described in JESD51-2a (sections 6 and 7).
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UCC28180
SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
www.ti.com
7.5 Electrical Characteristics
Unless otherwise noted, VCC=15Vdc, 0.1µF from VCC to GND, –40°C ≤ TJ = TA ≤ +125°C. All voltages are with respect to
GND. Currents are positive into and negative out of the specified terminal.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
VCC BIAS SUPPLY
ICCPRESTART
ICC Pre-start current
VCC = VCCOFF – 0.2 V
75
µA
ICCSTBY
ICC Standby current
VSENSE = 0.5 V
1.8
2.4
3.47
mA
ICCON_load
ICC Operating current
VSENSE = 4.0 V, CGATE = 4.7 nF
5.8
7
8.8
mA
UNDER VOLTAGE LOCKOUT (UVLO)
VCCON
VCC Turn on threshold
10.8
11.5
12.1
V
VCCOFF
VCC Turn off threshold
9.1
9.5
10.3
V
1.6
1.7
2
V
UVLO Hysteresis
VARIABLE FREQUENCY
Minimum switching frequency
RFREQ = 130 kΩ
16.3
18
19.8
kHz
Typical switching frequency
RFREQ = 32.7 kΩ
61.75
65
68.25
kHz
Maximum switching frequency
RFREQ = 8.2 kΩ
225
250
275
kHz
Voltage at FREQ pin
TA = 25°C
1.43
1.5
1.56
V
DMIN
Minimum duty cycle
VSENSE = 5.1 V, ISENSE = –0.25 V
DMAX
Maximum duty cycle
VSENSE = 4.0 V, RFREQ = 32.7 Ω
tOFF(min)
Minimum off time
VSENSE = 3 V, ICOMP = 0.72 V
fSW
VFREQ
PWM
0%
94.8%
96.5%
98%
450
570
690
ns
SYSTEM PROTECTION
VSOC
ISENSE threshold, soft over current (SOC)
–0.259
–0.285
–0.312
V
VPCL
ISENSE threshold, peak current limit (PCL)
–0.345
–0.4
–0.438
V
IISOP
ISENSE bias current, ISENSE open-pin protection
(ISOP)
ISENSE = 0 V
–2.3
–2.95
µA
VISOP
ISENSE threshold, ISENSE open-pin protection
(ISOP)
ISENSE = open pin
0.085
0.14
V
VOLP
VSENSE threshold, open loop protection (OLP)
ICOMP = 1 V, ISENSE = 0 V
16.5
17.6
%VREF
Open loop protection (OLP) Internal pull-down
current
VSENSE = 0.5 V
100
325
nA
95
97
%VREF
15.6
VUVD
VSENSE threshold, output under-voltage detection
(UVD) used for enhanced dynamic response (1)
VOVD
VSENSE threshold, output over-voltage detection
(OVD) used for Enhanced dynamic response (1)
103
105 106.75
%VREF
VOVP_L
Output over-voltage protection low threshold,
VCOMP is discharged by a 4kΩ resistor when
VSENSE > VOVP_L
105
107
109
%VREF
VOVP_H
Output over-voltage protection high threshold, PWM
shuts off when VSENSE > VOVP_H
107
109
111
%VREF
VOVP_H(RST)
Output over-voltage protection (VOVP_H) reset
threshold, PWM turns on when VSENSE <
VOVP_H(RST)
100
102
104
%VREF
0.2
0.25
%VREF
0.95
1.1
mS
93.25
ICOMP threshold, external overload protection
CURRENT LOOP
gmi
Transconductance gain
Output linear range
0.75
(1)
±50
ICOMP voltage during OLP
VSENSE = 0 V
µA
2.7
3
3.3
V
TA = 25°C
4.93
5
5.07
V
–40°C ≤ TA ≤ +125°C
4.87
5
5.15
V
–40
–56
–70
µS
VOLTAGE LOOP
VREF
Reference voltage
gmv
Transconductance gain without EDR
gmv-EDR
Transconductance gain under EDR
(1)
6
–230
–280
–340
µS
Maximum sink current under normal operation
VSENSE = 5 V, VCOMP = 4 V
23
40
57
µA
Source current under soft start
VSENSE = 4 V, VCOMP = 4 V
–29
–40
–52
µA
Maximum current under EDR operation
VSENSE = 4 V, VCOMP = 2.5 V
–200
–275
µA
Not production tested. Characterized by design
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Electrical Characteristics (continued)
Unless otherwise noted, VCC=15Vdc, 0.1µF from VCC to GND, –40°C ≤ TJ = TA ≤ +125°C. All voltages are with respect to
GND. Currents are positive into and negative out of the specified terminal.
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
20
100
250
nA
0
0.04
0.10
VSENSE input bias current
VSENSE = 5 V
VCOMP voltage during OLP
VSENSE = 0.5 V, IVCOMP= 0.5 mA
VCOMP rapid discharge current
VCOMP = 2 V, VCC = floating
VPRECHARGE
VCOMP precharge voltage
IVCOMP = –100 µA, VSENSE = 4 V
1.5
V
IPRECHARGE
VCOMP precharge current
VCOMP = 0 V
–1
mA
VSENSE threshold, end-of-soft-start
Initial Start-up
98
%VREF
GATE current, peak, sinking (1)
CGATE = 4.7 nF
2
A
GATE current, peak, sourcing (1)
CGATE = 4.7 nF
–1.5
GATE rise time
CGATE = 4.7 nF, GATE = 2 V to 8 V
8
40
GATE fall time
CGATE = 4.7 nF, GATE = 8 V to 2 V
8
GATE low voltage, no load
IGATE = 0 A
GATE low voltage, sinking
IGATE = 20 mA
GATE low voltage, sourcing
IGATE = -20 mA
GATE low voltage, sinking, OFF
VCC = 5 V, IGATE = 5 mA
0.1
GATE low voltage, sinking, OFF
VCC = 5 V, IGATE = 20 mA
GATE high voltage
V
0.37
mA
GATE DRIVER
A
60
ns
25
40
ns
0
0.01
V
0.04
0.06
V
–0.04
–0.06
V
0.2
0.31
V
0.4
0.8
1.4
V
VCC = 20 V, CGATE = 4.7 nF
14.5
15.2
16.1
V
GATE high voltage
VCC = 12.2 V, CGATE = 4.7 nF
10.8
11.2
12
V
GATE high voltage
VCC = VCCOFF + 0.2 V,
CGATE = 4.7 nF
8.2
9
10.1
V
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7.6 Typical Characteristics
0.99
0.97
DMAX ± Maximum Duty Cycle
FSW ± Switching Frequency (kHz)
265
215
165
115
0.95
0.93
0.91
0.89
65
0.87
15
0.85
0
20
40
60
80
100
120
140
15
35
55
75
RFREQ (KŸ)
95 115 135 155 175 195 215 235 255
FSW ± Switching Frequency (kHz)
C001
C002
VCC = 15 V
Figure 1. Switching Frequency vs. Resistor
Figure 2. Maximum Duty Cycle vs. Switching Frequency
12.0
3.5
VCC Turn ON
3.0
11.0
ICC ± Supply Current (mA)
VCCON/VCCOFF ± UVLO Threshold (V)
11.5
10.5
10.0
VCC Turn OFF
9.5
9.0
2.5
2.0
ICC Turn ON
ICC Turn OFF
1.5
1.0
0.5
8.5
8.0
±40
10
60
110
0.0
0
5
TJ ± Temperature (ƒC)
10
15
20
25
VCC ± Bias Supply Voltage (V)
C003
C004
TJ = 25 °C
No Gate Load
Figure 3. UVLO Threshold vs. Temperature
Figure 4. Supply Current vs. Bias Supply Voltage
9
70
8
65
7
60
ICC ± Supply Current (µA)
Operating, GATE Load = 4.7 nF
ICC ± Supply Current (mA)
VSENSE= 3 V
FSW = 65 kHz
6
5
4
3
55
50
Pre-Start
45
40
2
Standby
35
1
0
30
±40
10
60
110
±40
TJ ± Temperature (ƒC)
10
60
C005
C006
VCC = 15 V
VCC = VCCON – 0.2 V
Figure 5. Supply Current vs. Temperature
8
110
TJ ± Temperature (ƒC)
Figure 6. Pre-Start Supply Current vs. Temperature
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75
75
73
73
71
71
fSW ± Switching Frequency (kHz)
fSW ± Switching Frequency (kHz)
Typical Characteristics (continued)
69
67
65
Switching Frequency
63
61
69
67
65
Switching Frequency
63
61
59
59
57
57
55
55
10
±40
60
110
9
11
TJ ± Temperature (ƒC)
13
15
17
19
21
C007
VCC = 15 V
C008
FSW = 65 kHz
TJ = 25 °C
20.0
250.0
19.5
249.5
19.0
18.5
18.0
17.5
17.0
16.5
16.0
FSW = 65 kHz
Figure 8. Oscillator Frequency (65 kHz) vs. Bias Supply
Voltage
Switching Frequency (kHz)
Oscillator Frequency (kHz)
Figure 7. Oscillator Frequency (65 kHz) vs. Temperature
15.5
249.0
248.5
248.0
247.5
247.0
246.5
246.0
245.5
15.0
245.0
±50
±25
0
25
50
75
100
Temperature (ƒC)
125
±50
±25
0
25
50
75
100
Temperature (ƒC)
C006
VCC = 15 V
125
C002
VCC = 15 V
Figure 9. Oscillator Frequency (18 kHz) vs. Temperature
Figure 10. Oscillator Frequency (250 kHz) vs. Temperature
20.0
250.0
19.5
249.5
Oscillator Frequency (kHz)
Oscillator Frequency (kHz)
23
VCC ± Bias Supply Voltage (V)
19.0
18.5
18.0
17.5
17.0
16.5
16.0
15.5
249.0
248.5
248.0
247.5
247.0
246.5
246.0
245.5
15.0
245.0
9
11
13
15
17
Bias Supply Voltage (V)
19
21
9
11
13
15
17
19
Bias Supply Voltage (V)
C003
TJ = 25 °C
21
C004
TJ = 25 °C
Figure 11. Oscillator Frequency (18 kHz) vs. Bias Voltage
Figure 12. Oscillator Frequency (250 kHz) vs. Bias Voltage
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2.0
±45
1.8
±47
1.6
±49
1.4
±51
gmv ± Gain (µA/V)
gmi ± Gain (mA/V)
Typical Characteristics (continued)
1.2
1.0
0.8
Gain
±53
±55
Gain, No EDR
±57
0.6
±59
0.4
±61
0.2
±63
0.0
±65
10
±40
60
110
±40
10
TJ ± Temperature (ƒC)
60
110
TJ t Temperature (ƒC)
C009
C010
VCC = 15 V
VCC = 15 V
Figure 14. Voltage Loop Gain vs. Temperature
5.5
0.00
5.4
±0.05
5.3
±0.10
VSOC ± ISENSE Threshold (V)
VREF ± Reference Voltage (V)
Figure 13. Current Loop Gain vs. Temperature
5.2
5.1
5.0
Reference Voltage
4.9
4.8
±0.15
±0.20
±0.25
±0.30
4.7
±0.40
4.6
±0.45
4.5
Soft Over-Current Protection (SOC)
±0.35
±0.50
10
±40
60
110
±40
10
TJ ± Temperature (ƒC)
60
110
TJ ± Temperature (ƒC)
C011
C012
VCC = 15 V
VCC = 15 V
Figure 16. ISENSE Threshold Soft Over Current (SOC) vs.
Temperature
115
2.0
1.8
VOVP_H
110
1.6
VOLP ± VSENSE Threshold (V)
VOVP_H/VOVP_L/VOVD/VOVP_H(RST)/VUVD t VSENSE Threshold (% of VREF)
Figure 15. Reference Voltage vs. Temperature
VOVP_L
VOVD
105
VOVP_H(RST)
100
VUVD
95
1.4
1.2
1.0
VOLP
0.8
0.6
0.4
0.2
0.0
90
±40
±15
10
35
60
85
110
±40
10
60
C013
C014
VCC = 15 V
VCC = 15 V
Figure 17. VSENSE Threshold vs. Temperature
10
110
TJ ± Temperature (ƒC)
TJ t Temperature (ƒC)
Figure 18. VSENSE Threshold Open Loop vs. Temperature
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650
50
630
45
610
40
590
35
570
30
t ± Time (ns)
t ± Time (ns)
Typical Characteristics (continued)
tOFF(min)
550
530
Rise Time
25
20
Fall Time
510
15
490
10
470
5
450
0
±40
±15
10
35
60
85
110
±40
±15
10
TJ ± Temperature (ƒC)
35
60
85
110
TJ ± Temperature (ƒC)
C015
ICOMP = 0.72 V
VSENSE = 3 V
FSW = 65 kHz
Figure 19. Minimum Off Time vs. Temperature
1.8
40
1.6
VGATE ± Gate Low Voltage (V)
2.0
45
30
Rise Time
25
20
Fall Time
15
VGATE = 2 V-8 V
1.4
1.2
1.0
0.8
VGATE
0.6
10
0.4
5
0.2
0
CGATE = 4.7 nF
Figure 20. Gate Drive Rise/Fall Time vs. Temperature
50
35
t ± Time (ns)
C016
VCC = 15 V
0.0
10
12
14
16
18
20
22
±40
±15
10
35
60
C017
TJ = 25 °C
CGATE = 4.7 nF
85
110
TJ ± Temperature (ƒC)
VCC ± Bias Supply Voltage (V)
VGATE = 2 V-8 V
Figure 21. Gate Drive Rise/Fall Time vs. Bias Supply Voltage
C018
VCC = 15 V
IGATE = 20 mA
Figure 22. Gate Low Voltage vs. Temperature
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8 Detailed Description
8.1 Overview
The UCC28180 is a boost controller for power factor correction operating at a fixed frequency in continuous
conduction mode. The UCC28180 requires few external components to operate as an active PFC pre-regulator.
UCC28180 employs two control loops. An internal error amplifier and 5-V reference provide a slow outer loop to
control output voltage. External compensation of this outer loop is applied by means of the VCOMP pin. The
inner current loop shapes the average input current to match the sinusoidal input voltage. The inner current loop
avoids the need to sense input voltage by exploiting the relationship between input voltage and boost duty-cycle.
External compensation of the inner current loop is applied by means of the ICOMP pin.
The operating switching frequency can be programmed from 18 kHz to 250 kHz simply by connecting the FREQ
pin to ground through a resistor.
UCC28180 includes a number of protection functions designed to ensure it is reliable, and will provide safe
operation under all conditions, including abnormal or fault conditions.
12
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8.2 Functional Block Diagram
EMI Filter
LBST
LINE
INPUT
–
Bridge
Rectifier
DBST
VOUT
+
RFB1
QBST
CIN
RGATE
COUT
RLOAD
RFB2
RSENSE
Auxiliary Supply
UCC28180 Block Diagram
ICOMP Protection
+
0.2V
ICOMP
+
Under voltage lockout
ICOMPP
Q
S
Q
R
7
VCCON
11.5V
CVCC
VCCOFF
9.5V
+
UVLO
VCC
1
GND
2
Current
Amplifier
CICOMP
PWM
Comparator
KPC(s)
+
Gate Driver
gmi
S
Q
R
Q
+
3V
PWM
RAMP
M2
GAIN
M1, K1
OI O
V SL
P OP
| P
H
FAULT
OVP_H
Min Off Time
Oscillator
M2
8
PCL
S
Q
Clock
R
Q
GATE
Pre-Drive and
Clamp Circuit
VCOMP
M1
OVP_H
RISENSEfilter
Q
S
Q
R
+
5.45V
4k
ISENSE
+
Peak Current Limit (PCL)
3
300ns
Leading Edge
Blanking
1V
CISENSEfilter
+
5.10V
Over voltage protection
PCL
OVP_L
+
+
5.35V
SOC
-2.5X
ISENSE
Open-pin
Protection
+
EDR
Soft Over Current (SOC)
0.72V
ISOP
5.25V
Over voltage detector
+
4.75V
SOC
EDR
+
Under voltage detector
+
0.82V
OLP/STANDBY
FREQ
4
Oscillator
RFREQ
ICOMPP
ISOP
UVLO
OLP
Voltage Error
Amplifier
FAULT
100 nA
+
5V
gmv
6
gmv Enhancement
CVSENSE
End of soft start detector
+
END OF SS
4.9V
5
UVLO
Rapid Discharge
when
VCC < VCC OFF
Q
S
END OF SS
VPRECHARGE
FAULT
VCOMP
RCV
EDR
SS
VSENSE
Q
R
FAULT
CCV2
CCV1
FAULT
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8.3 Feature Description
8.3.1 Soft Start
Soft-Start controls the rate of rise of VCOMP in order to obtain a linear control of the increasing duty cycle as a
function of time. VCOMP, the output of the voltage loop transconductance amplifier, is pulled low during UVLO,
ICOMPP, ISOP and OLP (Open-Loop Protection)/STANDBY. Once the fault condition is released, an initial precharge source rapidly charges VCOMP to 1.5 V. After that point, a constant 40 µA of current is sourced into the
compensation components causing the voltage on this pin to ramp linearly until the output voltage reaches 85%
of its final value. At this point, the sourcing current decreases until the output voltage reaches its final rated
voltage. The soft-start time is controlled by the voltage error amplifier compensation capacitor values selected,
and is user programmable based on desired loop crossover frequency. Once the output voltage exceeds 98% of
rated voltage, soft start is over, the initial pre-charge source is disconnected, and EDR is no longer inhibited.
Soft-Start
+
–
VCOMP
5V
gmv
VSENSE
FAULT
VCOMP
ISS = –40 uA
for VSENSE < 4.25 V
during Soft-Start
FAULT
END OF SS
(LATCHED)
+
–
1.5 V source for
rapid pre-charge
of VCOMP prior
to Soft-Start
Figure 23. Soft Start
8.3.2 System Protection
System-level protection features help keep the system within safe operating limits.
8.3.3 VCC Undervoltage LockOut (UVLO)
UVLO
VCC
Auxiliary Supply
+
VCC ON 11.5 V
C DECOUPLE
Q
R
Q
UVLO
GND
VCC OFF 9.5 V
S
+
Figure 24. UVLO
14
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Feature Description (continued)
During startup, Under-Voltage LockOut (UVLO) keeps the device in the off state until VCC rises above the 11.5V enable threshold, VCCON. With a typical 1.7 V of hysteresis on UVLO to increase noise immunity, the device
turns off when VCC drops to the 9.5-V disable threshold, VCCOFF.
If, during a brief AC-line dropout, the VCC voltage falls below the level necessary to bias the internal FAULT
circuitry, the UVLO condition enables a special rapid discharge circuit which continues to discharge the VCOMP
capacitors through a low impedance despite a complete lack of VCC. This helps to avoid an excessive current
surge should the AC-line return while there is still substantial voltage stored on the VCOMP capacitors. Typically,
these capacitors can be discharged to less than 1 V within 150 ms of loss of VCC.
8.3.4 Output Overvoltage Protection (OVP)
There are two levels of OVP: When VSENSE exceeds 107% (VOVP_L) of the reference voltage, a 4-kΩ resistor
connects VCOMP to ground to rapidly discharge VCOMP. If VSENSE exceeds 109% (VOVP_H) of the reference
voltage, GATE output is disabled until VSENSE drops below 102% of the reference voltage.
8.3.5 Open Loop Protection/Standby (OLP/Standby)
If the output voltage feedback components were to fail and disconnect (open loop) the signal from the VSENSE
input, then it is likely that the voltage error amp would increase the GATE output to maximum duty cycle. To
prevent this, an internal pull-down forces VSENSE low. If the output voltage falls below 16.5% of its rated
voltage, causing VSENSE to fall below 0.82 V, the device is put in standby, a state where the PWM switching is
halted and the device is still on but draws standby current below 2.95 mA. This shutdown feature also gives the
designer the option of pulling VSENSE low with an external switch (standby function).
8.3.6 ISENSE Open-Pin Protection (ISOP)
If the current feedback components were to fail and disconnect (open loop) the signal to the ISENSE input, then
it is likely that the PWM stage would increase the GATE output to maximum duty cycle. To prevent this, an
internal pull-up source drives ISENSE above 0.085 V so that a detector forces a state where the PWM switching
is halted and the device is still on but draws standby current below 2.95 mA. This shutdown feature avoids
continual operation in OVP and severely distorted input current.
8.3.7 ICOMP Open-Pin Protection (ICOMPP)
If the ICOMP pin shorts to ground, then the GATE output increases to maximum duty cycle. To prevent this,
once ICOMP pin voltage falls below 0.2 V, the PWM switching is halted and the device is still on but draws
standby current below 2.95 mA .
8.3.8 FAULT Protection
VCC UVLO, OLP/Standby, ISOP and ICOMPP funtions constitute the fault protection feature in the UCC28180.
Under fault protection, VCOMP pin is pulled low and the device is in standby.
8.3.9 Output Overvoltage Detection (OVD), Undervoltage Detection (UVD) and Enhanced Dynamic
Response (EDR)
During normal operation, small perturbations on the PFC output voltage rarely exceed ±5% deviation and the
normal voltage control loop gain drives the output back into regulation. For large changes in line or load, if the
output voltage perturbation exceeds ±5%, an output over-voltage (OVD) or under-voltage (UVD) is detected and
Enhanced Dynamic Response (EDR) acts to speed up the slow response of the low-bandwidth voltage loop.
During EDR, the transconductance of the voltage error amplifier is increased approximately five times to speed
charging or discharging the voltage-loop compensation capacitors to the level required for regulation. EDR is
disabled when 5.25 V > VSENSE > 4.75 V. The EDR feature is not activated until soft start is completed. The
UVD is disabled during soft over protection (SOC) condition (since UVD and SOC conflict with each other).
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Feature Description (continued)
Over Voltage Protection
Enhanced Dynamic Response
Open Loop Protection/ Standby
Soft-Start Complete
+
OVP_L
5.35 V
+
OVP_H
5.45 V
OVERVOLTAGE
PROTECTION
5.10 V
+
S
Q
R
Q
Output Voltage
RFB1
Standby
VSENSE
RFB2
Optional
OVERVOLTAGE
DETECTION
5.25 V
UNDERVOLTAGE
DETECTION
4.75 V
SOFT-START COMPLETE
+
EDR
+
EDR
4.9 V
END OF SS
+
OPEN LOOP
PROTECTION/STANDBY 0.82 V
+
OLP/STANDBY
Figure 25. OVP_H, OVP_L, EDR, OLP, Soft Start Complete
8.3.10 Overcurrent Protection
Inductor current is sensed by RISENSE, a low value resistor in the return path of input rectifier. The other side of
the resistor is tied to the system ground. The voltage is sensed on the rectifier side of the sense resistor and is
always negative. The voltage at ISENSE is buffered by a fixed gain of -2.5 to provide a positive internal signal to
the current functions. There are two overcurrent protection features; Soft Overcurrent (SOC) protects against an
overload on the output and Peak Current Limit (PCL) protects against inductor saturation.
16
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Feature Description (continued)
Soft Over Current (SOC)
LINE
INPUT
0.72V
ISENSE Open-Pin
Protection (ISOP)
±
+
IISOP
2µA
VOUT
SOC
+
VISOP
0.082V
RISENSE
ISOP
+
ISENSE
RISENSEfilter
CISENSEfilter
3
(Optional)
300ns
Leading Edge
Blanking
1V
+
PCL
+
-2.5x
Peak Current Limit (PCL)
Figure 26. Soft Overcurrent/Peak-Current Limit
8.3.11 Soft Overcurrent (SOC)
Soft Overcurrent (SOC) limits the input current. SOC is activated when the current sense voltage on ISENSE
reaches –0.285 V. This is a soft control as it does not directly switch off the gate driver. Instead a 4-kΩ resistor
connects VCOMP to ground to discharge VCOMP and the control loop is adjusted to reduce the PWM duty
cycle. The under-voltage detection (UVD) is disabled during SOC.
8.3.12 Peak Current Limit (PCL)
Peak Current Limit (PCL) operates on a cycle-by-cycle basis. When the current sense voltage on ISENSE
reaches –0.4 V, PCL is activated, immediately terminating the active switch cycle. PCL is leading-edge blanked
to improve noise immunity against false triggering.
8.3.13 Current Sense Resistor, RISENSE
The current sense resistor, RISENSE, is sized using the minimum threshold value of Soft Over Current (SOC),
VSOC(min) . To avoid triggering this threshold during normal operation, resulting in a decreased duty-cycle, the
resistor is sized for an overload current of 10% more than the peak inductor current,
VSOC(min)
RISENSE £
1.1 IL _ PEAK(max)
(1)
Since RISENSE “sees” the average input current, worst-case power dissipation occurs at input low-line when input
current is at its maximum. Power dissipated by the sense resistor is given by:
(
PRISENSE = IIN _ RMS(max)
2
) RISENSE
(2)
Peak current limit (PCL) protection turns off the output driver when the voltage across the sense resistor reaches
the PCL threshold, VPCL. The absolute maximum peak current, IPCL, is given by:
V
/ 2.5
IPCL = PCL
RISENSE
(3)
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Feature Description (continued)
8.3.14 ISENSE Pin
The voltage at the ISENSE pin should be limited between 0 V and –1.1 V. Inrush currents at start-up have the
potential to drive the ISENSE pin significantly more negative so a diode clamp should be used between ISENSE
and GND to prevent the ISENSE pin going more negative than 1.1 V, (see Figure 26). The diode Vf should be
greater than the maximum PCL threshold (–0.438 V) and less than –1.1 V across temperature and component
variations.
8.3.15 Gate Driver
The GATE output is designed with a current-optimized structure to directly drive large values of total
MOSFET/IGBT gate capacitance at high turn-on and turn-off speeds. An internal clamp limits voltage on the
MOSFET gate to 15.2 V (typical). When VCC voltage is below the UVLO level, the GATE output is held in the off
state. An external gate drive resistor, RGATE, can be used to limit the rise and fall times and dampen ringing
caused by parasitic inductances and capacitances of the gate drive circuit and to reduce EMI. The final value of
the resistor depends upon the parasitic elements associated with the layout and other considerations. A 10-kΩ
resistor close to the gate of the MOSFET/IGBT, between the gate and ground, discharges stray gate capacitance
and helps protect against inadvertent dv/dt-triggered turn-on.
Gate Driver
7
VCC
PWM
FAULT
OVP_H
8
PCL
S
Q
Clock
R
Q
1
Pre-Drive and
Clamp Circuit
GATE
GND
Figure 27. Gate Driver
8.3.16 Current Loop
The overall system current loop consists of the current averaging amplifier stage, the pulse width modulator
(PWM) stage, the external boost inductor stage and the external current sensing resistor.
8.3.17 ISENSE and ICOMP Functions
The negative polarity signal from the current sense resistor is buffered and inverted at the ISENSE input. The
internal positive signal is then averaged by the current amplifier (gmi), whose output is the ICOMP pin. The
voltage on ICOMP is proportional to the average inductor current. An external capacitor to GND is applied to the
ICOMP pin for current loop compensation and current ripple filtering. The gain of the averaging amplifier is
determined by the internal VCOMP voltage. This gain is non-linear to accommodate the world-wide AC-line
voltage range.
ICOMP is connected to 3-V internally whenever OVP_H, ISOP, or OLP is triggered.
8.3.18 Pulse Width Modulator
The PWM stage compares the ICOMP signal with a periodic ramp to generate a leading-edge-modulated output
signal which is high whenever the ramp voltage exceeds the ICOMP voltage. The slope of the ramp is defined by
a non-linear function of the internal VCOMP voltage.
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Feature Description (continued)
The PWM output signal always starts low at the beginning of the cycle, triggered by the internal clock. The output
stays low for a minimum off-time, tOFF_min, after which the ramp rises linearly to intersect the ICOMP voltage. The
ramp-ICOMP intersection determines tOFF, and hence DOFF. Since DOFF = VIN/VOUT by the boost-topology
equation, and since VIN is sinusoidal in wave-shape, and since ICOMP is proportional to the inductor current, it
follows that the control loop forces the inductor current to follow the input voltage wave-shape to maintain boost
regulation. Therefore, the average input current is also sinusoidal in wave-shape.
PWM cycle
V ICOMP
VRAMP = F(VCOMP)
tON
tOFF
Figure 28. PWM Generation
8.3.19 Control Logic
The output of the PWM comparator stage is conveyed to the GATE drive stage, subject to control by various
protection functions incorporated into the device. The GATE output duty-cycle may be as high as 98%, but
always has a minimum off-time tOFF_min. Normal duty-cycle operation can be interrupted directly by OVP_H and
PCL. UVLO, ISOP, ICOMMP and OLP/Standby also terminate the GATE output pulse, and further inhibit output
until the SS operation can begin.
8.3.20 Voltage Loop
The outer control loop of the PFC controller is the voltage loop. This loop consists of the PFC output sensing
stage, the voltage error amplifier stage, and the non-linear gain generation.
8.3.21 Output Sensing
A resistor-divider network from the PFC output voltage to GND forms the sensing block for the voltage control
loop. The resistor ratio is determined by the desired output voltage and the internal 5-V regulation reference
voltage.
The very low bias current at the VSENSE input allows the choice of the highest practicable resistor values for
lowest power dissipation and standby current. A small capacitor from VSENSE to GND serves to filter the signal
in a high-noise environment. This filter time constant should generally be less than 100 µs.
8.3.22 Voltage Error Amplifier
The transconductance error amplifier (gmv) generates an output current proportional to the difference between the
voltage feedback signal at VSENSE and the internal 5-V reference. This output current charges or discharges
the compensation network capacitors on the VCOMP pin to establish the proper VCOMP voltage for the system
operating conditions. Proper selection of the compensation network components leads to a stable PFC preregulator over the entire AC-line range and 0% to 100% load range. The total capacitance also determines the
rate-of-rise of the VCOMP voltage at Soft Start, as discussed earlier.
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Feature Description (continued)
The amplifier output VCOMP is pulled to GND during any fault or standby condition to discharge the
compensation capacitors to an initial zero state. Usually, the large capacitor has a series resistor which delays
complete discharge for their respective time constant (which may be several hundred milliseconds). If VCC bias
voltage is quickly removed after UVLO, the normal discharge transistor on VCOMP loses drive and the large
capacitor could be left with substantial voltage on it, negating the benefit of a subsequent Soft Start. The
UCC28180 incorporates a parallel discharge path which operates without VCC bias, to further discharge the
compensation network after VCC is removed.
If the output voltage perturbations exceed ±5%, and output over-voltage (OVD) or under-voltage (UVD) is
detected, the OVD or UVD function invokes EDR which immediately increases the voltage error amplifier
transconductance to about 280 µS. This higher gain facilitates faster charging or discharging the compensation
capacitors to the new operating level. When output voltage perturbations greater than 107%VREF appear at the
VSENSE input, a 4-kΩ resistor connects VCOMP to ground to quickly reduce VCOMP voltage. When output
voltage perturbations are greater than 109%VREF, the GATE output is shut off until VSENSE drops below 102%
of regulation.
8.3.23 Non-Linear Gain Generation
The voltage at VCOMP is used to set the current amplifier gain and the PWM ramp slope. This voltage is subject
to modification by the SOC function, as discussed earlier.
Together the current gain and the PWM slope adjust to the different system operating conditions (set by the ACline voltage and output load level) as VCOMP changes, to provide a low-distortion, high-power-factor, inputcurrent wave shape following that of the input voltage.
8.4 Device Functional Modes
This device has no functional modes.
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The UCC28180 is a switch-mode controller used in boost converters for power factor correction operating at a
fixed frequency in continuous conduction mode. The UCC28180 requires few external components to operate as
an active PFC pre-regulator. The operating switching frequency can be programmed from 18 kHz to 250 kHz
simply by connecting the FREQ pin to ground through a resistor.
The internal 5-V reference voltage provides for accurate output voltage regulation over the typical world-wide 85VAC to 265-VAC mains input range from zero to full output load. The usable system load ranges from 100 W to
few kW.
Regulation is accomplished in two loops. The inner current loop shapes the average input current to match the
sinusoidal input voltage under continuous inductor current conditions. Under light-load conditions, depending on
the boost inductor value, the inductor current may go discontinuous but still meet Class-A/D requirements of IEC
61000-3-2 despite the higher harmonics. The outer voltage loop regulates the PFC output voltage by generating
a voltage on VCOMP (dependent upon the line and load conditions) which determines the internal gain
parameters for maintaining a low-distortion, steady-state, input-current wave shape.
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J1
3
2
1
LINE
C1
0.47 µF
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Bias
12Vdc
+
-
J2
GND
COMPONENTS MAY GET HOT
WARNING! HIGH VOLTAGE
HS1 COMMON TO Q1, BR1, AND D3
OUTPUT VOLTAGE: 390 VDC nominal
MAXIMUM OUTPUT POWER: 360 W
MAXIMUM OUTPUT CURRENT: 0.923 A
NOTES:
LINE INPUT VOLTAGE: 85 VRMS - 265 VRMS, 47 Hz - 63 Hz
PEAK INPUT CURRENT: 7 A
EARTH
LINE
NEUTRAL
Vin = 85 VAC to 265 VAC, 47 Hz to 63 Hz
VAR1
S10K275E2
F1
250 VAC
8A
TP1
47 µF
C3
C2
2200 pF
L1
5 mH
TP2
GND_EARTH
1
C6
0603
1
R1
0603
C4
2200 pF
C5
0.47 µF
C7
2700pF
TP3
~
1 Do Not Populate
5 ohm
t°
RT1
+
~
R3
17.8k
BR1
GBU8J-BP
C8
1000 pF
R2
221
JP1
-
22
UCC28180D
1
FREQ
ISENSE
ICOMP
GND
U1
C9
4
3
2
1
R4
0.032
C10
0.33 µF
C11
1 µF
VCOMP
VSENSE
VCC
GATE
5
6
7
8
TP4
C12
0.1 µF
TP6
TP5
3.3
R5
C13
4.7µF
R6
22.6k
R7
10.0k
D1
MBR140SFT1G
L2
327 µH
D2
TP7
C14
0.47µF
TP8
R13
13.3k
R12
0
R11
340k
R10
332k
R9
332k
R8
49.9
GND
C17
0.1 µF
C18
0.1 µF
TP12
TP11
1
2
3
4
J3
VOUT RTN
+VOUT
OUTPUT: 390 VDC NOMINAL, 0.923 A MAX
Copyright © 2016, Texas Instruments Incorporated
TP10
TP9
C16
270 µF
C15
820 pF
Q1
SPP20N60C3
HS1
D3
C3D04060A
1N5406
UCC28180
SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
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9.2 Typical Application
Figure 29. Design Example Schematic
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UCC28180
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SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
Typical Application (continued)
9.2.1 Design Requirements
This example illustrates the design process and component selection for a continuous mode power factor
correction boost converter utilizing the UCC28180. The pertinent design equations are shown for a universal
input, 360-W PFC converter with an output voltage of 390 V.
Table 1. Design Goal Parameters
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
INPUT CHARACTERISTICS
VIN
Input voltage
85
265
VAC
fLINE
Input frequency
47
63
Hz
IIN(peak)
Peak input current
VIN = VIN(min),
IOUT = IOUT(max)
7
A
OUTPUT CHARACTERISTICS
VOUT
Output voltage
VIN(min) ≤ VIN ≤ VIN(max),
fLINE(min) ≤ fLINE≤ fLINE(max),
IOUT ≤ IOUT(max)
Line Regulation
VIN(min) ≤ VIN≤ VIN(max),
IOUT = IOUT(max)
5%
VIN = 115 VAC,
fLINE = 60 Hz,
IOUT(min) ≤ IOUT ≤ IOUT(max)
5%
VIN = 230 VAC,
fLINE = 60 Hz,
IOUT(min)≤ IOUT ≤ IOUT(max)
5%
Load Regulation
379
390
402
VDC
IOUT
Output Load Current
VIN(min) ≤ VIN ≤ VIN(max)
fLINE(min) ≤ fLINE ≤ fLINE(max)
0
0.923
A
POUT
Output Power
VIN(min) ≤ VIN ≤ VIN(max)
fLINE(min) ≤ fLINE ≤ fLINE(max)
0
360
W
VRIPPLE(SW)
High frequency
Output voltage ripple
VRIPPLE(f_LINE Line frequency
Output voltage ripple
)
VIN = 115 VAC,
fLINE = 60 Hz
IOUT = IOUT(max)
2.5
VIN = 230 VAC,
fLINE = 50 Hz
IOUT = IOUT(max)
2.5
3.9
VIN = 115 VAC,
fLINE = 60 Hz,
IOUT = IOUT(max)
11.6
19.5
VIN = 230 VAC,
fLINE = 50 Hz,
IOUT = IOUT(max)
13.3
3.9
VP-P
VP-P
VOUT(OVP)
Output overvoltage
protection
425
VOUT(UVP)
Output undervoltage
protection
370
19.5
V
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Typical Application (continued)
Table 1. Design Goal Parameters (continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
CONTROL LOOP CHARACTERISTICS
fSW
f(CO)
PF
THD
η
Switching frequency
TJ = 25°C
Voltage Loop Bandwidth
VIN = 162 VDC,
IOUT = 0.466 A
8
Hz
Voltage Loop Phase
Margin
VIN = 162 VDC,
IOUT = 0.466 A
68
°
Power Factor
VIN = 115 VAC,
IOUT = IOUT(max)
0.99
VIN = 115 VAC,
fLINE = 60 Hz,
IOUT = IOUT(max)
4.3%
10%
VIN = 230 VAC,
fLINE = 50 Hz
IOUT = IOUT(max)
4%
10%
VIN = 115 VAC,
fLINE = 60 Hz,
IOUT = IOUT(max)
94%
Total harmonic distortion
Full load efficiency
Ambient temperature
114
120
25
126
kHz
°C
9.2.2 Detailed Design Procedure
9.2.2.1 Current Calculations
The input fuse, bridge rectifier, and input capacitor are selected based upon the input current calculations. First,
determine the maximum average output current, IOUT(max):
POUT(max)
IOUT(max) =
VOUT
(4)
IOUT(max) =
360 W
@ 0.923 A
390 V
(5)
The maximum input RMS line current, IIN_RMS(max), is calculated using the parameters from Table 1 and the
efficiency and power factor initial assumptions:
POUT(max)
I IN _ RMS(max) =
hVIN(min)PF
(6)
360 W
I IN _ RMS(max) =
= 4.551A
0.94 ´ 85 V ´ 0.99
(7)
Based upon the calculated RMS value, the maximum input current, IIN (max), and the maximum average input
current, IIN_AVG(max), assuming the waveform is sinusoidal, can be determined.
IIN(max) = 2IIN _ RMS(max)
(8)
IIN(max) = 2 ´ 4.551A = 6.436 A
(9)
IIN _ AVG(max) =
IIN _ AVG(max)
24
2IIN(max)
(10)
p
2 ´ 6.436 A
=
= 4.097 A
p
(11)
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9.2.2.2 Switching Frequency
The UCC28180 switching frequency is user programmable with a single resistor on the FREQ pin to ground. For
this design, the switching frequency, fSW, was chosen to be 120 kHz. Figure 30 (same as Figure 1) could be
used to select the suitable resistor to program the switching frequency or the value can be calculated using
constant scaling values of fTYP and RTYP. In all cases, fTYP is a constant that is equal to 65 kHz, RINT is a constant
that is equal to 1 MΩ, and RTYP is a constant that is equal to 32.7 kΩ. Simply applying the calculation below
yields the appropriate resistor that should be placed between FREQ and GND:
fTYP ´ R TYP ´ RINT
RFREQ =
(fSW ´ RINT ) + (R TYP ´ fSW ) - (R TYP ´ fTYP )
(12)
RFREQ =
65kHz ´ 32.7kW ´ 1MW
= 17.451kW
(120kHz ´ 1MW) + (32.7kW ´ 120kHz) - (32.7kW ´ 65kHz)
(13)
A typical value of 17.8 kΩ for the FREQ resistor results in a switching frequency of 118 kHz.
FSW ± Switching Frequency (kHz)
265
215
165
115
65
15
0
20
40
60
80
100
120
140
RFREQ (KŸ)
C001
Figure 30. Frequency vs. RFREQ
9.2.2.3 Bridge Rectifier
The input bridge rectifier must have an average current capability that exceeds the input average current.
Assuming a forward voltage drop, VF_BRIDGE, of 1 V across the rectifier diodes, BR1, the power loss in the input
bridge, PBRIDGE, can be calculated:
PBRIDGE = 2 VF _ BRIDGEIIN _ AVG(max)
(14)
PBRIDGE = 2 ´ 1V ´ 4.097 A = 8.195 W
(15)
Heat sinking will be required to maintain operation within the bridge rectifier’s safe operating area.
9.2.2.4 Inductor Ripple Current
The UCC28180 is a Continuous Conduction Mode (CCM) controller but if the chosen inductor allows relatively
high-ripple current, the converter will be forced to operate in Discontinuous Mode (DCM) at light loads and at the
higher input voltage range. High-inductor ripple current has an impact on the CCM/DCM boundary and results in
higher light-load THD, and also affects the choices for the input capacitor, RSENSE and CICOMP values. Allowing an
inductor ripple current, ΔIRIPPLE, of 20% or less will result in CCM operation over the majority of the operating
range but requires a boost inductor that has a higher inductance value and the inductor itself will be physically
large. As with all converter designs, decisions must be made at the onset in order to optimize performance with
size and cost. In this design example, the inductor is sized in such a way as to allow a greater amount of ripple
current in order to minimize space with the understanding that the converter operates in DCM at the higher input
voltages and at light loads but optimized for a nominal input voltage of 115 VAC at full load. Although specifically
defined as a CCM controller, the UCC28180 is shown in this application to meet the overall performance goals
while transitioning into DCM at high-line voltage, at a higher load level.
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9.2.2.5 Input Capacitor
The input capacitor must be selected based upon the input ripple current and an acceptable high frequency input
voltage ripple. Allowing an inductor ripple current, ΔIRIPPLE, of 40% and a high frequency voltage ripple factor,
ΔVRIPPLE_IN, of 7%, the maximum input capacitor value, CIN, is calculated by first determining the input ripple
current, IRIPPLE, and the input voltage ripple, VIN_RIPPLE:
I RIPPLE = DIRIPPLEIIN(max)
(16)
DIRIPPLE = 0.4
(17)
I RIPPLE = 0.4 ´ 6.436 A = 2.575 A
(18)
VIN _ RIPPLE = DVRIPPLE _ IN VIN _ RECTIFIED(min)
(19)
DVRIPPLE _ IN = 0.07
(20)
VIN _ RECTIFIED = 2VIN
(21)
VIN _ RECTIFIED = 2 ´ 85 V = 120 V
(22)
VIN _ RIPPLE = 0.07 ´ 120 V = 8.415 V
(23)
The recommended value for the input x-capacitor can now be calculated:
IRIPPLE
CIN =
8fSW VIN _ RIPPLE
CIN =
2.575 A
= 0.324 mF
8 ´ 118kHz ´ 8.415 V
(24)
(25)
A standard value 0.33-µF Y2/X2 film capacitor is used.
9.2.2.6 Boost Inductor
Based upon the allowable inductor ripple current discussed above, the boost inductor, LBST, is selected after
determining the maximum inductor peak current, IL_PEAK:
I
IL _ PEAK(max) = IIN(max) + RIPPLE
2
(26)
2.575 A
IL _ PEAK(max) = 6.436 A +
= 7.724 A
2
(27)
The minimum value of the boost inductor is calculated based upon the acceptable ripple current, IRIPPLE, at a
worst case duty cycle of 0.5:
V
D(1 - D)
LBST(min) ³ OUT
fSWIRIPPLE
(28)
LBST(min) ³
390 V ´ 0.5(1 - 0.5)
³ 321mH
118kHz ´ 2.575 A
(29)
The recommended minimum value for the boost inductor assuming a 40% ripple current is 321 µH; the actual
value of the boost inductor that will be used is 327 µH. With this actual value used, the actual resultant inductor
current ripple will be:
LBST = 327 mH
(30)
IRIPPLE(actual) =
VOUTD(1 - D)
fSW LBST
(31)
IRIPPLE(actual) =
390 V ´ 0.5(1 - 0.5)
= 2.527 A
118kHz ´ 327 mH
(32)
IL _ PEAK(max)
26
2.527 A
= 6.436 A +
= 7.7 A
2
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The duty cycle is a function of the rectified input voltage and will be continuously changing over the half line
cycle. The duty cycle, DUTY(max), can be calculated at the peak of the minimum input voltage:
VOUT - VIN _ RECTIFIED(min)
DUTY(max) =
VOUT
(34)
VIN _ RECTIFIED(min) = 2 ´ 85 V = 120 V
DUTY(max)
(35)
390 V - 120 V
=
= 0.692
390 V
(36)
9.2.2.7 Boost Diode
The diode losses are estimated based upon the forward voltage drop, VF, at 125°C and the reverse recovery
charge, QRR, of the diode. Using a silicon carbide Schottky diode, although more expensive, will essentially
eliminate the reverse recovery losses and result in less power dissipation:
PDIODE = VF _ 125CIOUT(max) + 0.5fSW VOUT QRR
(37)
VF _ 125°C = 1V
(38)
QRR = 0nC
(39)
PDIODE = (1V ´ 0.923 A ) + (0.5 ´ 119kHz ´ 390 V ´ 0nC ) = 0.923 W
(40)
This output diode should have a blocking voltage that exceeds the output over voltage of the converter and be
attached to an appropriately sized heat sink.
9.2.2.8 Switching Element
The MOSFET/IGBT switch will be driven by a GATE output that is clamped at 15.2 V for VCC bias voltages
greater than 15.2 V. An external gate drive resistor is recommended to limit the rise time and to dampen any
ringing caused by the parasitic inductances and capacitances of the gate drive circuit; this will also help in
meeting any EMI requirements of the converter. The design example uses a 3.3-Ω resistor; the final value of any
design is dependent upon the parasitic elements associated with the layout of the design. To facilitate a fast turn
off, a standard 40-V, 1-A Schottky diode is placed anti-parallel with the gate drive resistor. A 10-kΩ resistor is
placed between the gate of the MOSFET/IGBT and ground to discharge the gate capacitance and protect from
inadvertent dv/dt triggered turn-on.
The conduction losses of the switch MOSFET, in this design are estimated using the RDS(on) at 125°C, found in
the device data sheet, and the calculated drain to source RMS current, IDS_RMS:
2
PCOND = IDS
_ RMSRDS(on)125°C
(41)
RDS(on)125°C = 0.35 W
IDS _ RMS =
IDS _ RMS
POUT(max)
VIN _ RECTIFIED(min)
(42)
2-
16VIN _ RECTIFIED(min)
3pVOUT
(43)
360 W
16 ´ 120 V
=
= 3.639 A
2120 V
3p ´ 390 V
(44)
2
PCOND = 3.639 A ´ 0.35 W = 4.636 W
(45)
The switching losses are estimated using the rise time, tr, and fall time, tf, of the MOSFET gate, and the output
capacitance losses.
tr = 5ns
t f = 4.5ns
tr=5 nsCOSS = 780pF
PSW = fSW é0.5VOUTIIN(max) (tr +
ë
(46)
2
ù
t f ) + 0.5COSS VOUT
û
(47)
2ù
P = 118kHz é0.5 ´ 390 V ´ 6.436A(5ns + 4.5ns) + 0.5 ´ 780pF ´ 390 V = 8.407 W
ë
û
tr=5 ns SW
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Total FET losses
PCOND + PSW = 4.636 W + 8.407 W = 13.042 W
(49)
The MOSFET requires an appropriately sized heat sink.
9.2.2.9 Sense Resistor
To accommodate the gain of the non-linear power limit, the sense resistor, RSENSE, is sized such that it triggers
the soft over current at 10% higher than the maximum peak inductor current using the minimum soft over current
threshold of the ISENSE pin, VSOC, of ISENSE equal to 0.265 V.
VSOC(min)
RSENSE =
IL _ PEAK(max) ´ 1.1
(50)
RSENSE =
0.259 V
= 0.032 W
7.7 A ´ 1.1
(51)
The power dissipated across the sense resistor, PRSENSE, must be calculated:
2
PRSENSE = IIN
_ RMS(max)RSENSE
(52)
2
PRSENSE = 4.551A ´ 0.032 W = 0.663 W
(53)
The peak current limit, PCL, protection feature is triggered when current through the sense resistor results in the
voltage across RSENSE to be equal to the VPCL threshold. For a worst case analysis, the maximum VPCL threshold
is used:
VPCL(max)
IPCL =
RSENSE
(54)
IPCL =
0.438 V
= 13.688 A
0.032 W
(55)
To protect the device from inrush current, a standard 220-Ω resistor, RISENSE, is placed in series with the ISENSE
pin. A 1000-pF capacitor is placed close to the device to improve noise immunity on the ISENSE pin.
9.2.2.10 Output Capacitor
The output capacitor, COUT, is sized to meet holdup requirements of the converter. Assuming the downstream
converters require the output of the PFC stage to never fall below 300 V, VOUT_HOLDUP(min), during one line cycle,
tHOLDUP = 1/fLINE(min), the minimum calculated value for the capacitor is:
2POUT(max) tHOLDUP
COUT(min) ³ 2
2
VOUT - VOUT
_ HOLDUP(min)
(56)
COUT(min) ³
2 ´ 360 W ´ 21.28ms
390 V 2 - 300 V 2
³ 247 mF
(57)
It is advisable to de-rate this capacitor value by 10%; the actual capacitor used is 270 µF.
Verifying that the maximum peak-to-peak output ripple voltage will be less than 5% of the output voltage ensures
that the ripple voltage will not trigger the output over-voltage or output under-voltage protection features of the
controller. If the output ripple voltage is greater than 5% of the regulated output voltage, a larger output capacitor
is required. The maximum peak-to-peak ripple voltage, occurring at twice the line frequency, and the ripple
current of the output capacitor is calculated:
VOUT _ RIPPLE(pp) < 0.05 VOUT
(58)
VOUT _ RIPPLE(pp) < 0.05 ´ 390 V = 19.5 VPP
VOUT _ RIPPLE(pp) =
VOUT _ RIPPLE(pp)
28
(59)
IOUT
2p(2fLINE(min) )COUT
(60)
0.923A
=
= 5.789 V
2p(2 ´ 47Hz) ´ 270 mF
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The required ripple current rating at twice the line frequency is equal to:
IOUT(max)
ICOUT _ 2fline =
2
0.923 A
ICOUT _ 2fline =
= 0.653 A
2
(62)
(63)
There is a high frequency ripple current through the output capacitor:
ICOUT _ HF = IOUT(max)
ICOUT _ HF = 0.923 A
16 VOUT
- 1.5
3pVIN _ RECTIFIED(min)
(64)
16 ´ 390 V
- 1.5 = 1.848 A
3p ´ 120 V
(65)
The total ripple current in the output capacitor is the combination of both and the output capacitor must be
selected accordingly:
2
2
I COUT _ RMS(total) = ICOUT
_ 2fline + ICOUT _ HF
2
(66)
2
I COUT _ RMS(total) = 0.653 A + 1.848 A = 1.96 A
(67)
9.2.2.11 Output Voltage Set Point
For low power dissipation and minimal contribution to the voltage set point, it is recommended to use 1 MΩ for
the top voltage feedback divider resistor, RFB1. Multiple resistors in series are used due to the maximum
allowable voltage across each. Using the internal 5-V reference, VREF, the bottom divider resistor, RFB2, is
selected to meet the output voltage design goals.
VREFRFB1
RFB2 =
VOUT - VREF
(68)
RFB2 =
5 V ´ 1MW
= 13.04kW
390 V - 5 V
(69)
A standard value 13-kΩ resistor for RFB2 results in a nominal output voltage set point of 391 V.
An output over voltage is detected when the output voltage exceeds its nominal set-point level by 5%, as
measured when the voltage at VSENSE is 105% of the reference voltage, VREF. At this threshold, the enhanced
dynamic response (EDR) is triggered and the non-linear gain to the voltage error amplifier will increase the
transconductance to VCOMP and quickly return the output to its normal regulated value. This EDR threshold
occurs when the output voltage reaches the VOUT(ovd) level:
VOVD = 1.05 VREF = 1.05 ´ 5 V = 5.25 V
(70)
æR
+ RFB2 ö
VOUT(ovd) = VOVD ç FB1
÷
RFB2
è
ø
VOUT(ovd)
(71)
æ 1MW + 13kW ö
= 5.25 V ´ ç
÷ = 410.7 V
13kW
è
ø
(72)
In the event of an extreme output over voltage event, the GATE output will be disabled if the output voltage
exceeds its nominal set-point value by 9%. The output voltage, VOUT(ovp), at which this protection feature is
triggered is calculated as follows:
æR
+ RFB2 ö
VOUT(ovp) = 1.09 ´ VREF ç FB1
÷ = 426.4 V
R
FB2
è
ø
(73)
An output under voltage is detected when the output voltage falls below 5% below its nominal set-point as
measured when the voltage at VSENSE is 95% of the reference voltage, VREF:
VUVD = 0.95 VREF = 0.95 ´ 5 V = 4.75 V
(74)
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æR
+ RFB2 ö
VOUT(uvp) = VUVD ç FB1
÷
RFB2
è
ø
VOUT(uvp)
(75)
æ 1MW + 13kW ö
= 4.75 V ´ ç
÷ = 371.6 V
13kW
è
ø
(76)
A small capacitor on VSENSE must be added to filter out noise. Limit the value of the filter capacitor such that
the RC time constant is limited to approximately 10 µs so as not to significantly reduce the control response time
to output voltage deviations.
10 ms
C VSENSE =
= 769pF
RFB2
(77)
The closest standard value of 820 pF was used on VSENSE for a time constant of 10.66 µs.
9.2.2.12 Loop Compensation
The current loop is compensated first by determining the product of the internal loop variables, M1M2, using the
internal controller constants K1 and KFQ. Compensation is optimized maximum load and nominal input voltage,
115 VAC is used for the nominal line voltage for this design:
M1M2 =
KFQ =
KFQ =
2
IOUT(max) VOUT
2.5RSENSEK1
2
hVIN
_ RMSKFQ
(78)
1
fSW
1
= 8.475 ms
118kHz
K1 = 7
M1M2 =
(79)
0.923 A ´ 390 V 2 ´ 2.5 ´ 0.032 W ´ 7
0.92 ´ 115 V 2 ´ 8.475 ms
V
= 0.751
ms
(80)
The VCOMP operating point is found on the following chart, M1M2 vs. VCOMP. Once the M1M2 result is
calculated above, find the resultant VCOMP voltage at that operating point to calculate the individual M1 and M2
components.
4.0
3.5
3.0
M1M2
2.5
2.0
1.5
1.0
0.5
0.0
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
VCOMP (V)
4.5
5.0
C007
Figure 31. M1M2 vs. VCOMP
For the given M1M2 of 0.751 V/µs, the VCOMP approximately equal to 3 V, as shown in Figure 31.
The individual loop factors, M1 which is the current loop gain factor, and M2 which is the voltage loop PWM ramp
slope, are calculated using the following conditions:
The M1 non-linear current loop gain factor follows the following identities:
30
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M1 = 0.068 if VCOMP < 1 V
(81)
M1 = 0.156 ´ VCOMP - 0.088 if 1 V < VCOMP < 2 V
(82)
M1 = 0.313 ´ VCOMP - 0.401 if 2 V < VCOMP < 4.5 V
(83)
M1 = 1.007 if 4.5 V < VCOMP < 5 V
(84)
In this example, according to the chart in Figure 31, VCOMP is approximately equal to 3 V, so M1 is calculated
to be approximately equal to 0.366:
M1 = 0.313 ´ 2.45 - 0.401 = 0.366
(85)
The M2 non-linear PWM ramp slope will obey the following relationships:
V
M2 = 0
ms if VCOMP ≤ 0.5 V
M2 =
(86)
fSW
V
´ 0.1223 ´ (VCOMP - 0.5)2
65kHz
ms if 0.5 V ≤ VCOMP ≤ 4.6 V
(87)
f
V
M2 = SW ´ 2.056
65kHz
ms if 4.6 V ≤ VCOMP ≤ 5 V
(88)
In this example, with VCOMP approximately equal to 3 V, M2 equals 1.388 V/µs:
118kHz
V
V
M2 =
´ 0.1223 ´ (3 - 0.5)2
= 1.388
65kHz
ms
ms
(89)
Verify that the product of the individual gain factors, M1 and M2, is approximately equal to the M1M2 factor
determined above, if not, iterate the VCOMP value and recalculate M1M2
V
V
M1 ´ M2 = 0.538 ´ 1.388
= 0.747
ms
ms
(90)
The product of M1 and M2 is within 1% of the M1M2 factor previously calculated:
M1 ´ M2 @ M1M2
(91)
V
V
0.747
@ 0.751
ms
ms
(92)
If more accuracy was desired, iteration results in a VCOMP value of 3.004 V where M1M2 and M1 x M2 are both
equal to 0.751 V/µs.
The non-linear gain variable, M3, can now be calculated:
M3 = 0 if VCOMP < 5 V
(93)
f
V
M3 = SW ´
´ (0.0166 ´ VCOMP - 0.0083)
65kHz ms
if 0.5 V < VCOMP < 1 V
(94)
f
V
M3 = SW ´
´ (0.0572 ´ VCOMP2 - 0.0597 ´ VCOMP + 0.0155)
65kHz ms
if 1 V < VCOMP < 2 V
fSW
V
2
M3 =
´
´ (0.1148 ´ VCOMP - 0.1746 ´ VCOMP + 0.0586)
65kHz ms
if 2 V < VCOMP < 4.5 V
f
V
M3 = SW ´
´ (0.1148 ´ VCOMP2 - 0.1746 ´ VCOMP + 0.0586)
65kHz ms
if 4.5 V < VCOMP < 4.6 V
M3 = 0 if 4.6 V < VCOMP < 5 V
In this example, using 3.004 V for VCOMP for a more precise calculation, M3 calculates to 1.035 V/µs:
118kHz V
V
M3 =
´
´ (0.1148 ´ 3.0042 - 0.1746 ´ 3.004 + 0.0586) = 1.035
65kHz ms
ms
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(95)
(96)
(97)
(98)
(99)
31
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For designs that allow a high inductor ripple current, the current averaging pole, which functions to flatten out the
ripple current on the input of the PWM comparator, should be at least decade before the converter switching
frequency. Analysis on the completed converter may be needed to determine the ideal compensation pole for the
current averaging circuit as too large of a capacitor on ICOMP will add phase lag and increase iTHD where as too
small of an ICOMP capacitor will result in not enough averaging and an unstable current averaging loop. The
frequency of the current averaging pole, fIAVG, is chosen to be at approximately 5 kHz for this design as the
current ripple factor, ∆IRIPPLE, was chosen at the onset of the design process to be 40%, which is large enough to
force DCM operation and result in relatively high inductor ripple current. The required capacitor on ICOMP,
CICOMP, for this is determined using the transconductance gain, gmi, of the internal current amplifier:
g ´ M1
CICOMP = mi
K12pfIAVG
(100)
CICOMP =
0.95mS ´ 0.538
= 2330pF
7 ´ 2 ´ p ´ 5kHz
(101)
A standard value 2700-pF capacitor for CICOMP results in a current averaging pole frequency of 4.314 kHz.
gmi ´ M1
= 4.314kHz
fIAVG =
K1 ´ 2 ´ p ´ 2700pF
(102)
The transfer function of the current loop can be plotted:
K 2.5RSENSE VOUT
1
GCL (f) = 1
´
KFQM1M2LBST
s(f)2 K1CICOMP
s(f) +
gmi ´ M1
(103)
)
(104)
100
Gain (dB)
±80
Gain
80
±90
Phase
60
±100
40
±110
20
±120
0
±130
±20
±140
±40
±150
±60
±160
±80
±170
±100
Phase (ƒ)
(
GCLdB (f) = 20log GCL (f)
±180
10
100
1k
10k
100k
Frequency (Hz)
1M
C005
Figure 32. Bode Plot of the Current Averaging Circuit
The voltage transfer function, GVL(f) contains the product of the voltage feedback gain, GFB, and the gain from the
pulse width modulator to the power stage, GPWM_PS, which includes the pulse width modulator to power stage
pole, fPWM_PS. The plotted result is shown in Figure 32.
RFB2
GFB =
RFB1 + RFB2
GFB =
32
13kW
= 0.013
1MW + 13kW
(105)
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1
fPWM _ PS =
2p
fPWM _ PS =
3
K12.5RSENSE VOUT
COUT
2
KFQM1M2 VIN(nom)
1
7 ´ 2.5 ´ 0.032W ´ 390V 3 ´ 270 mF
2p
V
8.475 ms ´ 0.539 ´ 1.392 ´ 115 V 2
ms
= 1.479Hz
(106)
M3 VOUT
M1M2 ´ 1V
GPWM _ PS (f) =
s(f)
1+
2pfPWM _ PS
(107)
GVL (f) = GFBGPWM _ PS (f)
)
(108)
100
PWM to Power Stage Gain
Total Open Loop Gain
Total Open Loop Phase
80
Gain (dB)
60
0
±10
±20
40
±30
20
±40
0
±50
±20
±60
±40
±70
±60
±80
±80
±90
±100
0.01
0.1
1
10
100
Frequency (Hz)
Phase (ƒ)
(
GVLdB (f) = 20log GVL (f)
±100
1000
C008
Figure 33. Bode Plot of the Open Voltage Loop without Error Amplifier
The voltage error amplifier is compensated with a zero, fZERO, at the fPWM_PS pole and a pole, fPOLE, placed at 20
Hz to reject high frequency noise and roll off the gain amplitude. The overall voltage loop crossover, fV, is desired
to be at 10 Hz. The compensation components of the voltage error amplifier are selected accordingly.
1
fZERO =
2pR VCOMPC VCOMP
(109)
1
fPOLE =
2p
R VCOMPC VCOMPC VCOMP _ P
C VCOMP + C VCOMP _ P
(110)
é
ù
ê
ú
ê
ú
1 + s(f)R VCOMPC VCOMP
GEA (f) = gmv ê
ú
é
æ R VCOMPC VCOMPC VCOMP _ P ö ù ú
ê
÷ú ú
ê C VCOMP + C VCOMP _ P s(f) ê1 + s(f) çç
÷ú
êë
êë
è C VCOMP + C VCOMP _ P
ø û úû
(
)
(111)
From Figure 33, the gain of the voltage transfer function at 10 Hz is approximately 0.081 dB. Estimating that the
parallel capacitor, CVCOMP_P, is much smaller than the series capacitor, CVCOMP, the unity gain will be at fV, and
the zero will be at fPWM_PS, the series compensation capacitor is determined:
fV = 10Hz
(112)
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UCC28180
SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
gmv
C VCOMP =
10
fV
fPWM _ PS
0 - GVLdB (f )
20
´ 2pfV
(113)
10Hz
56 ms ´
1.479Hz
C VCOMP =
10
www.ti.com
0 - 0.081dB
20
= 6.08 mF
´ 2 ´ p ´ 10Hz
(114)
The capacitor for VCOMP must have a voltage rating that is greater than the absolute maximum voltage rating of
the VCOMP pin, which is 7 V. The readily available standard value capacitor that is rated for at least 10 V in the
package size that would fit the application was 4.7 µF and this is the value used for CVCOMP in this design
example.
RVCOMP is calculated using the actual CVCOMP capacitor value.
C VCOMP = 4.7 mF
R VCOMP =
R VCOMP =
(115)
1
2pfZEROC VCOMP
(116)
1
= 22.89kW
2 ´ p ´ 1.479Hz ´ 4.7 mF
(117)
A 22.6-kΩ resistor is used for RVCOMP.
C VCOMP
C VCOMP _ P =
2pfPOLER VCOMPC VCOMP - 1
C VCOMP _ P
(118)
4.7 mF
=
= 0.381mF
2 ´ p ´ 20Hz ´ 22.6k kW ´ 4.7 mF - 1
(119)
A 0.47-µF capacitor is used for CVCOMP_P.
The total closed loop transfer function, GVL_total, contains the combined stages and is plotted in Figure 34.
GVL _ total (f) = GFB (f)GPWM _ PS (f)GEA (f)
Gain (dB)
)
(121)
100
100
50
80
0
60
±50
40
±100
±150
0.01
20
EA Gain
Total Closed Loop Gain
Total Closed Loop Phase Margin
0.1
1
10
Phase (ƒ)
(
GVL _ totaldB (f) = 20log GVL _ total (f)
(120)
100
0
1000
Frequency (Hz)
C001
Figure 34. Closed Loop Voltage Bode Plot
34
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SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
EFFICIENCY
9.2.3 Application Curve
1.00
0.99
0.98
0.97
0.96
0.95
0.94
0.93
0.92
0.91
0.90
0.89
0.88
0.87
0.86
0.85
0.84
0.83
0.82
0.81
0.80
85 VAC, 60 Hz
115 VAC, 60 Hz
230 VAC, 50 HZ
265 VAC, 50 Hz
0.0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
LOAD (A)
1.0
C001
Figure 35. UCC28180EVM-573 Efficiency
(As a Function of Line Voltage and Load Current)
Figure 36. UCC28180EVM-573 Power Factor
(As a Function of Line Voltage and Load Current)
0.14
PWR 573
AMPLITUDE (A)
0.12
EN61000-3-2 Class D max
230 VAC, 50 Hz, Full Load
0.10
0.08
0.06
0.04
0.02
0.00
3 5 7 9 11 13 15 17 19 21 23 25 27 29 31 33 35 37 39
HARMONIC NUMBER
Figure 37. UCC28180EVM-573 Total Harmonic Distortion
(As a Function of Line Voltage and Load Current)
C004
Figure 38. UCC28180EVM-573 Current Harmonics,
(230-VAC, 50-Hz Input, Full Load, Without the Fundamental)
0.14
PWR 573
AMPLITUDE (A)
0.12
0.10
EN61000-3-2 Class D max
115 VAC, 60 Hz, Full Load
0.08
0.06
0.04
0.02
0.00
3 5 7 9 11 13 15 17 19 21 23 25 27 29 31 33 35 37 39
HARMONIC NUMBER
C005
Figure 39. UCC28180EVM-573 Current Harmonics,
(115-VAC, 60-Hz Input, Full Load, Without the Fundamental)
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10 Power Supply Recommendations
10.1 Bias Supply
The UCC28180 operates from an external bias supply. It is recommended that the device be powered from a
regulated auxiliary supply. (This device is not intended to be used from a bootstrap bias supply. A bootstrap bias
supply is fed from the input high voltage through a resistor with sufficient capacitance on VCC to hold up the
voltage on VCC until current can be supplied from a bias winding on the boost inductor. For that reason, the
minimal hysteresis on VCC would require an unreasonable value of hold-up capacitance.)
During normal operation, when the output is regulated, current drawn by the device includes the nominal run
current plus the current supplied to the gate of the external boost switch. Decoupling of the bias supply must take
switching current into account in order to keep ripple voltage on VCC to a minimum. A ceramic capacitor of 0.1µF minimum value from VCC to GND with short, wide traces is recommended.
VCC
VCC(ON) 11.5 V
VCC(OFF) 9.5 V
ICC
ICC(ON)
ICC(stby) < 2.95 mA
ICC(prestart) < 75 µA
Controller
State
PWM
State
UVLO
Soft-Start
Run
OFF
Ramp
Regulated
Fault/standby
OFF
SoftStart
Run
Ramp Regulated
UVLO
OFF
Figure 40. Device Supply States
The device's bias operates in several states. During startup, VCC Under-Voltage LockOut (UVLO) sets the
minimum operational DC input voltage of the controller. There are two UVLO thresholds. When the UVLO turn-on
threshold is exceeded, the PFC controller turns ON. If the VCC voltage falls below the UVLO turn-off threshold,
the PFC controller turns off. During UVLO, current drawn by the device is minimal. After the device turns on, Soft
Start (SS) is initiated and the boost inductor current is ramped up in a controlled manner to reduce the stress on
the external components and avoids output voltage overshoot. During soft start and after the output is in
regulation, the device draws its normal run current. If any of several fault conditions are encountered or if the
device is put in standby with an external signal, the device draws a reduced standby current.
11 Layout
11.1 Layout Guidelines
As with all PWM controllers, the effectiveness of the filter capacitors on the signal pins depends upon the
integrity of the ground return. Separating the high di/dt induced noise on the power ground from the low current
quiet signal ground is required for adequate noise immunity. Even with a signal layer PCB design, the pin out of
the UCC28180 is ideally suited to minimize noise on the small signal traces. As shown in Figure 41, the
capacitors on VSENSE, VCOMP, ISENSE, ICOMP, and FREQ (if used) must be all be returned directly to the
portion of the ground plane that is the quiet signal GND and not in high-current return path of the converter,
shown as power GND. The trace from the FREQ pin to the frequency programming resistor should be as short
as possible. It is recommended that the compensation components on ICOMP and VCOMP are located as close
as possible to the UCC28180. Placement of these components should take precedence, paying close attention
to keeping their traces away from high noise areas. The bypass capacitors on VCC must be located physically
close the VCC and GND pins of the UCC28180 but should not be in the immediate path of the signal return.
36
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Layout Guidelines (continued)
Other layout considerations should include keeping the switch node as short as possible, with a wide trace to
reduce induced ringing caused by parasitic inductance. Every effort should be made to avoid noise from the
switch node from corrupting the small signal traces with adequate clearance and ground shielding. As some
compromises must be made due to limitation of PCB layers or space constraints, traces that must be made long,
such as the signal from the current sense resistor shown in Figure 41, should be as wide as possible, avoid long
narrow traces.
Table 2. Layout Component Description for Figure 41
LAYOUT COMPONENTS
REFERENCE DESIGNATOR
FUNCTION
U1
Controller, UCC28180
Q1
Main switch
D2
Boost diode
R5
RGATE
R7
Pull-down resistor on GATE
D1
Turn-off diode on GATE
D4
ISENSE pin diode
C11, C12
VCC bypass capacitors
C7
ICOMP compensation, CICOMP
R1, C6
Placeholders for additional ICOMP compensation, if needed
C8
ISENSE filter, CISENSE
R2
ISENSE inrush current limiting resistor, RISENSE
R3
Frequency programming resistor, RFREQ
C9
Placeholder for FREQ filter, if needed
R6, C13, C14
VCOMP compensation components, RVCOMP, CVCOMP_P, CVCOMP
C15
VSENSE filter, CVSENSE
R11, R12
RFB1 on VSENSE
R13
RFB2 on VSENSE
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11.2 Layout Example
Figure 41. Recommended Layout for UCC28180
38
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SLUSBQ5D – NOVEMBER 2013 – REVISED JULY 2016
12 Device and Documentation Support
12.1 Documentation Support
12.1.1 Related Documentation
These references, additional design tools, and links to additional references, including design software and
models may be found on the web at http://www.power.ti.com under Technical Documents.
• User Guide, Using the UCC28180EVM-573, 360-W Power Factor Correction, SLUUAT3
• Design Spreadsheet, UCC28180 Design Calculator, SLUC506
12.2 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. In the upper
right corner, click on Alert me to register and receive a weekly digest of any product information that has
changed. For change details, review the revision history included in any revised document.
12.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.4 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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39
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UCC28180D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
U28180
UCC28180DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
U28180
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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