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UCC28881
SLUSC36B – NOVEMBER 2015 – REVISED JANUARY 2016
UCC28881 700-V, 225-mA Low Quiescent Current Off-Line Converter
1 Features
3 Description
•
•
The UCC28881 integrates the controller and a 14-Ω,
700-V power MOSFET into one monolithic device.
The device also integrates a high-voltage current
source, enabling start up and operation directly from
the rectified mains voltage. UCC28881 is the same
family device of UCC28880, with higher current
handling capability.
1
•
•
•
•
•
•
•
Integrated 14-Ω, 700-V Power MOSFET
Integrated High-Voltage Current Source for
Internal Device Bias Power
Integrated Current Sense
Internal Soft Start
Self-Biased Switcher (Start Up and Operation
Directly from Rectified Mains Voltage)
Supports Buck, Buck-Boost and Flyback
Topologies
VFB_TH)
58
100
µA
IFL
Internal supply current, full
load
FB = 0.75 V (> VFB_TH)
86
120
µA
ICH0
Charging VDD Cap current
VVDD = 0 V,
ICH1
Charging VDD Cap current
VVDD = 4.4V, VFB = 1.25 V
VVDD
Internally regulated low
Voltage supply (supplied from
HVIN pin)
VFB_TH
FB pin reference threshold
VVDD(on)
VDD turn-on threshold
VDD low-to-high
VVDD(off)
VDD turn-off threshold
VDD high-to-low
ΔVVDD(uvlo)
VDD UVLO Hysteresis
VDD high-to-low
DMAX
Maximum Duty Cycle
FB = 0.75 V
45%
ILIMIT
Current Limit
–3.8
–1.6
–0.4
mA
–3.40
–1.30
–0.25
mA
4.5
5.0
5.5
V
0.96
1.03
1.105
V
3.55
3.92
4.28
V
3.28
3.62
3.89
V
0.27
0.345
0.39
V
55%
Static, TA = –40°C
Static, TA = 25°C
330
Static, TA = 125°C
315
440
630
mA
570
mA
mA
TJ(stop)
Thermal Shutdown
Temperature
Internal junction temperature
138.5
150
°C
TJ(hyst)
Thermal Shutdown Hysteresis
Internal junction temperature
37.
45
°C
BV
Power MOSFET Breakdown
Voltage
TJ = 25°C
Power MOSFET OnResistance (includes internal
sense-resistor)
ID = 60 mA, TJ = 25°C
14
18
Ω
RDS(on)
ID = 60 mA, TJ = 125°C
24
30
Ω
DRAIN_ILEAKAGE
Power MOSFET off state
leakage current
VDRAIN = 700V, TJ = 25°C
HVIN_IOFF
VVDD(clamp)
HVIN off state current
VDD clamp voltage
700
V
VDRAIN = 400 V, TJ = 125°C
VHVIN = 700 V, TJ = 25°C, VVDD =
5.8 V
4.0
7.5
VHVIN = 400 V, TJ = 125°C, VVDD =
5.8 V
IVDD = 250 µA
5
µA
20
µA
36.0
µA
20
µA
V
6.0
6.7
7.5
7.6 Switching Characteristics
over operating free-air temperature range (unless otherwise noted)
MIN
TYP
MAX
UNIT
fSW(max)
Maximum switching frequency
52
62
75
kHz
tON_MAX
Maximum switch on time (current limiter not triggered), FB = 0.75 V
6.5
8.3
9.7
µs
tOFF_MIN
Minimum switch off time follows every tON time, FB = 0.75 V
6.5
8.3
9.7
µs
tMIN
Minimum on time
0.17
0.27
0.30
µs
tOFF(ovl)
Max off time (OL condition), tOFF(ovl) = tSW – tON(max)
130
200
270
µs
tON_TO
Inductor current run away protection time threshold
6
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450
ns
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7.7 Typical Characteristics
1.6
Current Limit (normalized to 50 mA/Ps)
Current Limit (normalized to 25oC)
1.2
1.15
1.1
1.05
1
0.95
0.9
0.85
0.8
-40
-20
0
20
40
60
80
Temperature (oC)
100
120
1.5
1.4
1.3
1.2
1.1
1
0.9
0.8
0.7
0.6
0.5
0.4
50
140
150
ICH0 and ICH1 (normalized to 25°C)
IFL and INL (normalized to 25oC)
1.15
1.1
1.05
1
0.95
0.9
IFL
INL
-20
0
20
40
60
80
Temperature (oC)
100
120
D002
ICH0
ICH1
1.4
1.3
1.2
1.1
1
0.9
0.8
0.7
0.6
0.5
-40
140
-20
0
D303
20
40
60
80
Temperature (°C)
100
120
140
D001
Figure 4. ICH0 and ICH1 vs Temperature
900
1.01
VVDD(on)
VVDD Hysteresis
800
Drain to Source Current (mA)
1.005
1
0.995
0.99
0.985
700
600
500
400
300
200
IDRAIN 25oC
IDRAIN 125oC
100
0.98
-40
750
1.5
Figure 3. INL and IFL vs Temperature
VVDD(on) and Hysteresis (normalized to 25oC)
650
Figure 2. ILIMIT vs Drain Current Slope
Figure 1. ILIMIT vs Temperature
1.2
0.85
-40
250
350
450
550
Drain Current Slope (mA/Ps)
D301
0
-20
0
20
40
60
80
Temperature (oC)
100
120
140
0
5
D305
Figure 5. VVDD(on) and VVDD Hysteresis vs Temperature
10
15
20 25 30 35 40 45
Drain to Source Voltage (V)
50
55
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D316
Figure 6. IDS vs VDS at 25°C and 125°C
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Typical Characteristics (continued)
1.02
Feedback Voltage (normalized to 25oC)
FSW(max) (normalized to 25°C)
1.02
1.01
1
0.99
0.98
0.97
0.96
-50
0
50
Temperature (°C)
100
1.01
1
0.99
0.98
0.97
0.96
-40
150
-20
0
Figure 7. Maximum Switching Frequency vs Temperature
1.045
1.04
1.035
1.03
1.025
1.02
1.015
1.01
1.005
1
TOFF(min)
TON(max)
0.995
-20
0
20
40
60
80
Temperature (°C)
100
120
140
D001
Figure 9. tON(max) and tOFF(min) vs Temperature
40
60
80
Temperature (oC)
100
120
140
D308
Figure 8. VFB_TH vs Temperature
DRAIN Breakdown Voltage (normalized to 25oC)
tON(max) and tOFF(min) (normalized to 25°C)
1.05
0.99
-40
20
D012
1.12
1.1
1.08
1.06
1.04
1.02
1
0.98
0.96
0.94
0.92
0.9
-40
-20
0
20
40
60
80
Temperature (oC)
100
120
140
D310
Figure 10. DRAIN breakdown voltage vs Temperature
140
136
132
RTJA (oC/W)
128
124
120
116
112
108
104
100
0
200
400
600
800
1000
Copper Area (mm2)
1200
1400
D001
D201
Figure 11. RTHJA vs Copper Area
8
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SLUSC36B – NOVEMBER 2015 – REVISED JANUARY 2016
8 Detailed Description
8.1 Overview
The UCC28881 integrates a controller and a 700-V power MOSFET into one monolithic device. The device also
integrates a high-voltage current source, enabling start up and operation directly from the rectified mains voltage.
UCC28881 is the same family device as UCC28880 and it provides higher power handling capability.
The low-quiescent current of the device enables excellent efficiency. The device is suitable for non-isolated ACto-DC low-side buck and buck-boost configurations with level-shifted direct feedback, but also more traditional
high-side buck, buck boost and low-power flyback converters with low standby power can be built using a
minimum number of external components.
The device generates its own internal low-voltage supply (5 V referenced to the device’s ground, GND) from the
integrated high-voltage current source. The PWM signal generation is based on a maximum constant on-time,
minimum off-time concept, with the triggering of the on-pulse depending on the feedback voltage level. Each onpulse is followed by a minimum off-time to ensure that the power MOSFET is not continuously driven in an onstate. The PWM signal is AND-gated with the signal from a current limit circuit. No internal clock is required, as
the switching of the power MOSFET is load dependent. A special protection mechanism is included to avoid
runaway of the inductor current when the converter operates with the output shorted or in other abnormal
conditions that can lead to an uncontrolled increase of the inductor current. This special protection feature keeps
the MOSFET current at a safe operating level. The device is also protected from other fault conditions with
thermal shutdown, under-voltage lockout and soft-start features.
8.2 Functional Block Diagram
HVIN
5
High Voltage
Current Source
8
Thermal
Shutdown
VDD
4
Gate
LDO
UVLO
DRAIN
S
Q
R
Q
Current
Limit
Control and
Reference
VREF_TH = 1 V
+
FB
3
PWM Controller
and Output Short
Circuit Protection
Leading Edge
Blanking Time
LEB
1, 2
GND
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8.3 Feature Description
The UCC28881 integrates a 700-V rated power MOSFET switch, a PWM controller, a high-voltage current
source to supply a low-voltage power supply regulator. A bias and reference block, thermal shutdown and the
following protection features, current limiter, under voltage lockout (UVLO) and overload protection for situations
like short circuit at the output are also integrated inside UCC28881. UCC28881 is the same family device as
UCC28880 and it provides higher power handling capability.
The positive high-voltage input of the converter node (VIN+) functions as a system reference ground for the
output voltage in low-side topologies. In the low-side buck topology the output voltage is negative with respect to
the positive high-voltage input (VIN+), and in low-side buck-boost topology the output voltage is positive with
respect to the positive high-voltage input (VIN+).
In low-side buck and buck-boost topologies, the external level-shifted direct feedback circuit can be implemented
by two resistors and a high-voltage PNP transistor.
In high-side buck configuration, as well as in non-isolated flyback configuration, the output voltage is positive with
respect to the negative high-voltage input (VIN–), which is the system reference ground.
UCC28881 can operate under continuous conduction mode (CCM) or discontinuous conduction mode (DCM)
mode. CCM operation allows the converter to deliver more output power because of less current ripple. However,
it requires a higher inductor value and generally results in lower efficiency due to the added losses associated
with diode reverse recovery current. DCM mode operation uses a smaller inductor and achieves better efficiency,
but the output current handling capability is reduced because of increased current ripple. The table below shows
the current handling capability in DCM and CCM operation for the UCC28880, UCC28881 family.
Table 1. Current Handling Capability for UCC28880 and UCC28881
DEVICE
CURRENT HANDLING MODE
230 VAC ±15%
85 V ~ 265 VAC
UCC28881
Discontinuous Conduction Mode (DCM)
150 mA
150 mA
UCC28881
Continuous Conduction Mode (CCM)
225 mA
225 mA
UCC28880
Discontinuous Conduction Mode (DCM)
70 mA
70 mA
UCC28880
Continuous Conduction Mode (CCM)
100 mA
100 mA
When the bus voltage is higher than 400 V, it is recommended to limit operation to DCM mode only to avoid the
diode reverse recovery current and the associated internal MOSFET stress. Due to the higher switching loss and
device stresses at higher bus voltage, it is recommended to keep the converter input voltage less than 560 V for
the buck applications.
UCC28881 power handling capability is reduced at higher ambient temperature, as a function of the power
dissipation of the device, the device's recommended maximum operating junction temperature and the thermal
dissipation capability of the total system. De-rating of the output current according to maximum ambient
temperature can be estimated from Figure 12. The de-rating estimation assumes CCM operation, 10 µJ of
switching loss and 135°C/W RθJA and 30-kHz, full-load switching frequency. This is a conservative estimation.
The thermal handling capability can be more accurately determined through experimental measurement. For
more information on thermal evaluation methods see the IC Package and Thermal Metrics application report:
SPRA953.
10
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1.2
Output Current Derating Factor
1.1
1
0.9
0.8
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
-40
-20
0
20
40
60
80
100
Ambient Temperature (oC)
120
140
D012
Figure 12. Output Current (De-Rating Factor) vs. Temperature
It is required to use fast recovery diode for the buck freewheeling diode. When designed in CCM, the diode
reverse recovery time should be less than 35 ns to keep low reverse recovery current and the switching loss. For
example, STTH1R06A provides 25-ns reverse recovery time. When designed in DCM, a slower diode can be
used, but the reverse-recovery time should be kept less than 75 ns. MURS160 can fit the requirement.
The device has a low-standby power consumption (no-load condition), only 18 mW (typical) when connected to a
230-VAC mains and 9 mW when connected to an 115-VAC mains.
The standby power does not include the power dissipated in the external feedback path, the power dissipated in
the external pre-load, the inductor in the freewheeling diode and the converter input stage (rectifiers and filter).
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8.4 Device Functional Modes
8.4.1 Start-Up Operation
The device includes a high-voltage current source connected between the HVIN pin and the internal supply for
the regulator. When the voltage on the HVIN pin rises, the current source is activated and starts to supply current
to the internal 5-V regulator. The 5-V regulator charges the external capacitor connected between VDD pin and
GND pin. When the VDD voltage exceeds the VDD turn-on threshold, VVDD(on), device starts operations. The
minimum voltage across HVIN and GND pins, to ensure enough current to charge the capacitance on VDD pin,
is VHVIN(min). At the First switching cycle the minimum MOSFET off time is set to be >100 μs and cycle-by-cycle is
progressively reduced up to tOFF(min) providing soft start.
8.4.2 Feedback and Voltage Control Loop
The feedback circuit consists of a voltage comparator with the positive input connected to an internal reference
voltage (referenced to GND) and the negative input connected to FB pin. When the feedback voltage at the FB
pin is below the reference voltage VFB_TH logic high is generated at the comparator output. This logic high
triggers the PWM controller, which generates the PWM signal turning on the MOSFET. When the feedback
voltage at the FB pin is above the reference voltage, it indicates that the output voltage of the converter is above
the targeted output voltage set by the external feedback circuitry and PWM is stopped.
12
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Device Functional Modes (continued)
8.4.3 PWM Controller
UCC28881 operates under on/off control. When the FB pin voltage is below internal reference 1 V, the converter
is switching and sending power to the load. When the FB pin voltage is above internal reference 1 V, the
converter shuts off and stops delivering power to the load.
The PWM controller does not need a clock signal. The PWM signal’s frequency is set to fSW(max) = (1/(tON(max) +
tOFF(min))) which occurs when the voltage on the FB pin is continuously below VFB_TH.
PWM duty cycle is determined by both the peak current and maximum on time. At each switching cycle, after
turn on, the MOSFET is turned off if its current reaches the fixed peak-current threshold or its on time reaches
the maximum value of on-time pulse tON(max).
Normally the converter would operate under frequency control, which means the converter is only enabled one
switching cycle and then disabled. Next switching cycle starts when output voltage decays and the feedback
enable the converter again. This way, the converter appears to operate under variable switching frequency
control.
The user might observe the converter operates in burst mode that converter is enabled for multiple switching
cycles and then stopped for multiple switching cycles. This causes larger output voltage ripple. However, due to
the infrequent switching it actually helps on the standby power at no load.
VFB
VFB_TH
t
FB_COMP_OUT
t
PWM
t
CURRENT LIMIT
t
RSTN
t
GATE
t
tON(max)
tOFF(min)
tON(max)
tOFF(min)
tON(max)
tOFF(min)
tON(max)
tOFF(min)
Figure 13. UCC28881 Timing Diagram
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Device Functional Modes (continued)
8.4.4 Current Limit
The current limit circuit senses the current through the power MOSFET. The sensing circuit is located between
the source of the power MOSFET and the GND pin. When the current in the power MOSFET exceeds the
threshold ILIMIT, the internal current limit signal goes high, which sets the internal RSTN signal low. This disables
the power MOSFET by driving its gate low. The current limit signal is set back low after the falling edge of the
PWM signal. After the rising edge of the GATE signal, there is a blanking time. During this blanking time, the
current limit signal cannot go high. This blanking time and the internal propagation delay result in minimum on
time, tMIN.
8.4.5 Inductor Current Runaway Protection
To protect the device from overload conditions, including a short circuit at the output, the PWM controller
incorporates a protection feature which prevents the inductor current from runaway. When the output is shorted
the inductor demagnetization is very slow, with low di/dt, and when the next switching cycle starts energy stored
in the inductance is still high. After the MOSFET switches on, the current starts to rise from pre-existing DC value
and reaches the current-limit value in a short duration of time. Because of the intrinsic minimum on-time of the
device the MOSFET on-time cannot be lower than tMIN, in an overload or output short circuit the energy
inductance is not discharged sufficiently during MOSFET off-time, it is possible to lose control of the current
leading to a runaway of the inductor current. To avoid this, if the on-time is less than tON_TO (tON_TO is a device
internal time out), the controller increases the MOSFET off-time (tOFF). If the MOSFET on-time is longer than
tON_TO then tOFF is decreased. The controller increases tOFF, cycle-by-cycle, through discrete steps until the ontime continues to stay below tON_TO. The tOFF is increased up to tOFF(ovl) after that, if the on-time is still below
tON_TO the off-time is kept equal to tOFF(ovl). The controller decreases tOFF cycle-by-cycle until the on-time
continues to stay above tON_TO up to tOFF(min). This mechanism prevents control loss of the inductor current and
prevents over stress of the MOSFET (see typical waveforms in Figure 14 and Figure 15). At start up, the tOFF is
set to tOFF(ovl) and reduced cycle-by-cycle (if the on-time is longer than tON_TO) up to tOFF(min) providing a soft start
for the power stage.
I LIMIT
Inductor Current
Drain Current
tON_ MAX
t
tOFF
PWM
t
Current Limit
t
LEB
~200 ns
~200 ns
t
tON_TO
tON_TO
tON_TO
t
Increase tOFF
( Decrease fSW )
Decrease tOFF
(Increase fSW )
CNT_IN
t
Gate
tON
tON
t
Figure 14. Current Runaway Protection Logic Timing Diagram
14
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Device Functional Modes (continued)
Output Shorted Here
VFB
VFB_TH
ILIMIT
INDUCTOR
CURRENT
DRAIN
CURRENT
t
tON_TO
t
GATE
tON_TO
tON < tON_TO
tON_TO
tON > tON_TO
tON_TO
tON < tON_TO
tON_TO
t
tON < tON_TO
Figure 15. Current Runaway Protection, Inductor and MOSFET Current
A minimal value needs to be imposed on the inductance value to avoid nuisance tripping of the protection feature
that prevents the loss of control of the inductor current. Inadvertent operation of the protection feature limits the
output-power capability of the converter. This condition depends on the converter's maximum input operating
voltage and temperature. Use Equation 1 to calculate your minimum inductance value.
VIN max
L!
u tON _ TO
ILIMIT
(1)
In this equation, VIN(max) is the maximum voltage on the DC input. If the input is a rectifed AC voltage, it should
be the peak value of the maximum AC line. For a DC input case, it should be the maximum DC input voltage.
If the inductance value is too low, such that the MOSFET on-time is always less than tON_TO timeout and the
device progressively increases the MOSFET off-time up to tOFF(ovl), the output power is reduced and the
converter fails to supply the load.
8.4.6 Over Temperature Protection
If the junction temperature rises above TJ(stop), the over temperature protection is triggered. This disables the
power MOSFET switching. To re-enable the switching of the MOSFET the junction temperature has to fall by
TJ(hyst) below the TJ(stop) where the device moves out of over temperature protection.
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9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The UCC28881 can be used in various application topologies with direct or isolated feedback. The device can be
used in low-side buck, where the output voltage is negative, or as a low-side buck-boost configuration, where the
output voltage is positive. In both configurations the common reference node is the positive input node (VIN+).
The device can also be configured as a LED driver in either of the above mentioned configurations. If the
application requires the AC-to-DC power supply output to be referenced to the negative input node (VIN–), the
UCC28881 can also be configured as a traditional high-side buck as shown in Figure 16. In this configuration,
the voltage feedback is sampling the output voltage VOUT, making the DC regulation less accurate and load
dependent than in low-side buck configuration, where the feedback is always tracking the VOUT. However, highconversion efficiency can still be obtained.
9.2 Typical Application
9.2.1 13-V, 225-mA High-Side Buck Converter
Figure 16 shows a typical application example of a non-isolated power supply, where the UCC28881 is
connected in a high-side buck configuration having an output voltage that is positive with respect to the negative
high-voltage input (VIN–). The output voltage is set to 13 V in this example, but can easily be changed by
changing the value of RFB1. This application can be used for a wide variety of household appliances and
automation, or any other applications where mains isolation is not required.
RF
L2
D2
HVIN
VDD
UCC28881
CVDD
C1
+
DRAIN
RFB1
FB
VIN
C2
-
GND
CFB
RFB2
D4
L1
D1
D3
CL
RL
+
VOUT
-
Figure 16. Universal Input, 12-V, 225-mA Output High-Side Buck
16
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Typical Application (continued)
9.2.1.1 Design Requirements
Table 2. Design Specification
DESCRIPTION
MIN
TYP
MAX
UNIT
DESIGN INPUT
VIN
AC input voltage
85
265
fLINE
Line frequency
47
63
VRMS
Hz
IOUT
Output current
0
225
mA
DESIGN REQUIREMENTS
PNL
No-load input power
VOUT
Output voltage
ΔVOUT
Output voltage ripple
η
Converter full-load efficiency
12.5
13
500
mW
17.5
V
350
mV
70%
9.2.1.2 Detailed Design Procedure
9.2.1.2.1 Input Stage (RF, D2, D3, C1, C2, L2)
•
•
•
Resistor RF is a flame-proof fusible resistor. RF limits the inrush current, and also provide protection in case
any component failure causes a short circuit. Value for its resistance is generally selected between 4.7 Ω to
15 Ω.
A half-wave rectifier is chosen and implemented by diode D2 (1N4937). It is a general purpose 1-A, 600-V
rated diode. It has a fast reverse recovery time (200 ns) for improved differential-mode-conducted EMI noise
performance. Diode D3 (1N4007) is a general purpose 1-A, 1-kV rated diode with standard reverse recovery
time (>500 ns), and is added for improved common-mode-conducted EMI noise performance. D3 can be
removed and replaced by a short if not needed.
EMI filtering is implemented by using a single differential-stage filter (C1-L2-C2).
Capacitors C1 and C2 in the EMI filter also acts as storage capacitors for the high-voltage input DC voltage
(VIN). The required input capacitor size can be calculated according Equation 2.
2 u PIN
fLINE min
CBULK
min
ª
1
u«
« RCT
¬«
§ V
BULK min
1
u arccos ¨
¨ 2uV
2uS
IN min
©
2
2
2 u VIN
VBULK
min
min
·º
¸»
¸»
¹ ¼»
where
•
•
•
•
•
•
CBULK(min) is minimum value for the total input capacitor value (C1 + C2 in the schematic of Figure 16).
RCT = 1 in case of half wave rectifier and RCT = 2 in case of full-wave rectifier .
PIN is the converter input power.
VIN(min) is the minimum RMS value of the AC input voltage.
VBULK(min) is the minimum allowed voltage value across bulk capacitor during converter operation.
fLINE(min) is the minimum line frequency when the line voltage is VIN(min).
The converter input power can be easily calculated as follow:
•
•
The converter maximum output power is: POUT = IOUT x VOUT = 0.225 A x 13 V = 2.925 W
Assuming the efficiency η = 68% the input power is PIN = POUT/η = 4.178 W
Using the following values for the other parameters
•
•
•
VBULK(min) = 80 V
VIN(min) = 85 VRMS (from design specification table)
fLINE(min) = 57 Hz
(2)
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CBULK(min) = 15.6 μF. Considering that electrolytic capacitors, generally used as bulk capacitor, have 20% of
tolerance in value, the minimum nominal value required for CBULK is:
CBULK(min)
CBULKn(min) !
19.5 PF
1 TOLCBULK
(3)
Select C1 and C2 to be 10 μF each (CBULK = 10 μF + 10 μF = 20 μF > CBULKn(min)).
By using a full-wave rectifier allows a smaller capacitor for C1 and C2, almost 50% smaller.
9.2.1.2.2 Regulator Capacitor (CVDD)
Capacitor CVDD acts as the decoupling capacitor and storage capacitor for the internal regulator. A 100-nF, 10-V
rated ceramic capacitor is enough for proper operation of the device's internal LDO.
9.2.1.2.3 Freewheeling Diode (D1)
The freewheeling diode has to be rated for high-voltage with as short as possible reverse-recovery time (trr).
The maximum reverse voltage that the diode should experience in the application, during normal operation, is
given by Equation 4.
VD1(max)
2 u VIN(max)
2 u 265V 375V
(4)
A margin of 20% is generally considered.
The use of a fast recovery diode is required for the buck-freewheeling rectifier. When designed in CCM, the
diode reverse recovery time should be less than 35 ns to keep low reverse recovery current and the switching
loss. For example, STTH1R06A provides 25-ns reverse recovery time. When designed in DCM, slower diode can
be used, but the reverse recovery time should be kept less than 75 ns. MURS160 can fit the requirement.
9.2.1.2.4 Output Capacitor (CL)
The value of the output capacitor impacts the output ripple. Depending on the combination of capacitor value and
equivalent series resistor (RESR). A larger capacitor value also has an impact on the start-up time. For a typical
application, the capacitor value can start from 47 μF, to hundreds of μF. A guide for sizing the capacitor value
can be calculated by the following equations:
ILIMIT IOUT
440mA 225mA
CL ! 20 u
20 u
200 PF
fSW max u 'VOUT
62kHz u 350mV
(5)
RESR
'VOUT
ILIMIT
0.8 :
(6)
Take into account that both CL and RESR contribute to output voltage ripple. A first pass capacitance value can be
selected and the contribution of CL and RESR to the output voltage ripple can be evaluated. If the total ripple is
too high the capacitance value has to increase or RESR value must be reduced. In the application example CL
was selected (330 µF) and it has an RESR of 0.03 Ω. So the RESR contributes for 4% of the total ripple. The
formula that calculates CL is based on the assumption that the converter operates in burst of twenty switching
cycles. The number of bursts per cycle could be different, the formula for CL is a first approximation.
9.2.1.2.5 Pre-Load Resistor (RL)
The pre-load resistor connected at the output is required for the high-side buck topology. Unlike low side buck
topology, the output voltage is directly sensed, in high-side buck topology the output is sampled and estimated.
At no-load condition, because the feedback loop runs with its own time constant, the buck converter operates
with a fixed minimum switching frequency. Select the pre-load resistor or using a zener diode to prevent output
voltage goes too high at no-load condition.
A simple zener diode would be a good choice without going through the calculation. Besides the simplifying the
calculation, zener diode doesn't consumes power at heavy load condition, which helps to improve the converter
heavy-load efficiency.
18
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A simple resistor can also be used to limit the output voltage at no load condition. However, this resistor
connects to the output all the time and it reduces the full-load efficiency. The pre-load resistor can be calculated
based on below equation or based on experiments. In this equation, the VMAX is allowed maximum output
voltage, and VREG is the regulated output voltage.
RL
4 u VMAX 2 u VMAX
VMAX
VREG
VREG
u
CFB u RFB1 RFB2
L1 u ILIMIT 2
(7)
9.2.1.2.6 Inductor (L1)
Initial calculations:
Half of the peak-to-peak ripple current at full load:
'IL 2 u ILIMIT IOUT
(8)
When operating in DCM, the peak-to-peak current ripple is the peak current of the device.
Average MOSFET conduction minimum duty cycle at continuous conduction mode is:
VOUT Vd
DMIN
VIN(max) Vd
(9)
If the converter operates in discontinuous conduction mode:
I
VOUT Vd
DMIN 2 u OUT
ILIMIT VIN(max) Vd
(10)
Maximum allowed switching frequency at VIN(max) and full load:
DMIN
FSW _ VIN(max)
tON _ TO
(11)
Switching frequency has a maximum value limit of fSW(max).
The worst case ILIMIT = 315 mA, but assuming ΔIL = 180 mA.
The converter works in continuous conduction mode (ΔIL < ILIMIT) so based on VOUT = 13 V, Vd = 0.5 V and
VIN(max) = 375 V
VOUT Vd
DMIN
3.61%
VIN(max) Vd
(12)
The maximum allowed switching frequency is 62 kHz because the calculated value exceeds it.
DMIN
FSW _ VIN(max)
80kHz ! fSW(max) 62kHz
tON _ TO
(13)
The duty cycle does not force the MOSFET on time to go below tON_TO. If DMIN/TON_TO < fSW(max), the switching
frequency is reduced by current runaway protection and the maximum average switching frequency is lower than
fSW(max), the converter can't support full load.
The minimum inductance value satisfies both the following conditions:
VOUT Vd
L!
1 mH
'IL u FSW _ VIN max
L!
VIN max
ILIMIT
u tON _ TO
(14)
375 V
u 450ns # 536 PH
315mA
(15)
In the application example, 1 mH is selected as the minimum standard value that satisfy Equation 14 and
Equation 15.
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9.2.1.2.7 Feedback Path (CFB, RFB1 and RFB2) and Load Resistor (RL)
In low-side buck converter the output voltage is always sensed by the FB pin and UCC28881 internal controller
can turn on the MOSFET on VOUT. In high-side buck converter applications the information on the output
voltage value is stored on CFB capacitor. This information is not updated in real time. The information on CFB
capacitor is updated just after MOSFET turn-off event. When the MOSFET is turned off, the inductor current
forces the freewheeling diode (D1 in Figure 16) to turn on and the GND pin of UCC28881 goes negative at -Vd1
(where Vd1 is the forward drop voltage of diode D1) with respect to the negative terminal of bulk capacitor (C1 in
Figure 16). When D1 is on, through diode D4, the CFB capacitor is charged at VOUT – Vd4 + Vd1. Set the output
voltage regulation level using Equation 16.
RFB1 VOUT(T) Vd4 Vd1 VFB _ TH VOUT(T) VFB _ TH
#
RFB2
VFB _ TH
VFB _ TH
where
•
•
•
•
•
WFB
VFB_TH is the FB pin reference voltage.
VOUT_T is the target output voltage.
RFB1, RFB2 is the resistance of the resistor divider connected with FB pin (see Figure 16)
The capacitor CFB after D1 is discharged with a time constant that is τFB = CFB x (RFB1 + RFB2 ).
Select the time constant τFB, given in Equation 17
CFB u RFB1 RFB2
0.1u CL u RLOAD
(16)
(17)
In this equation, RLOAD is the full load resistor value.
The time constant selection leads to a slight output-voltage increase in no-load or light-load conditions. In order
to reduce the output-voltage increase, increase τFB. The drawback of increasing τFB is t in high-load conditions
VOUT could drop.
Because of the nature of sample and hold of output voltage feedback, the feedback loop components need some
adjustment after the initial design. The larger time constant of the feedback components leads to lower no-load
switching frequency. As the results, the no-load standby power and light-load voltage regulation are improved.
Because of larger time constant, at heavier load, the load regulation start to get worse. On the contrast,
decreasing the time constant makes the heavy load regulation better but increases the no-load standby power
and makes the light-load voltage regulation worse. Some tradeoff is required to make the regulation and standby
power. Below table summarizes the relationship between the feedback loop time constant and the load
regulation. This can be used for an easy guideline for fine tuning the circuit performance.
Table 3. Feedback Loop Time Constant Adjustment
FEEDBACK LOOP TIME
CONSTANT (τFB)
20
NO-LOAD REGULATION
HEAVY-LOAD OUTPUT
VOLTAGE RIPPLE
Increase
Better
Worse
Better
Decrease
Worse
Better
Worse
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9.2.1.3 Application Curves
Figure 17 shows the output voltage vs output current. Different curves correspond to different converter AC input
voltages. Figure 18 shows efficiency changes vs output power. Different curves correspond to different converter
AC input voltages.
18
100
115V AC
230V AC
16
90
80
70
12
Efficiency (%)
Output Voltage (V)
14
10
8
6
60
50
40
30
4
20
2
10
0
115V AC
230V AC
0
0
0.05
0.1
0.15
Output Current (A)
0.2
0.25
0
D001
Figure 17. Output IV Characteristic
0.3 0.6 0.9 1.2 1.5 1.8 2.1 2.4 2.7
Output Power (W)
3
D001
Figure 18. Efficiency vs POUT
Table 4. Converter Efficiency
VIN_AC (VRMS)
LOAD (mA)
EFFICIENCY (%)
AVERAGE EFFICIENCY (%)
115
50
82
81
100
81
150
81
225
80
230
50
80
100
81
150
81
225
81
80.8
Table 5. Key Component List for 13-V 225-mA High-Side Buck Converter
DES
PART NUMBER
MANUFACTURER (1)
Bulk capacitor, 10 μF, 450 V
EEUED2W100
Panasonic
CFB
Feedback capacitor, ceramic, 0.015 μF, 50 V
C0805C153K5RACTU
Kemet
CL
Output capacitor, 330 μF, 35 V
EEU-FM1V331L
Panasonic
D1
Buck freewheeling diode, ultrafast, 600 V, 1 A
STTH1R06A
ST Microelectronics
Rectifier diodes, standard recovery rectifier, 1000
V, 1 A
1N4007
Fairchild semiconductor
C1, C2
D2, D3
(1)
DESCRIPTION
D4
Feedback diode, standard recovery rectifier, 600 V, 1N4006-T
1A
Diodes Inc.
L1
Buck inductor, 1 mH, 0.4 A,
7447471102
Wurth Elektronik
L2
Filter inductor, 1 mH, 0.2 A
5800-102-RC
Bourns
RFB1
Upper feedback resistor 121 kΩ, 1%
ERJ-6ENF1213V
Panasonic
RFB2
Lower feedback resistor, 10.0 kΩ, 1%
ERJ-6ENF1002V
Panasonic
See Third-Party Products Discalimer
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9.2.2 Additional UCC28881 Application Topologies
9.2.2.1 Low-Side Buck and LED Driver – Direct Feedback (Level Shifted)
Features include:
• Output Referenced to Input
• Negative Output (VOUT) with Respect to VIN+
• Step Down: VOUT < VIN
• Direct Level-Shifted Feedback
RFB1
D1
+
CL
+
VDD
VIN
-
Q1
L1
HVIN
VOUT
DRAIN
UCC28881
FB
GND
RFB2
Figure 19. Low-Side Buck, Direct Feedback (Level Shifted)
RSENSE
C1
R2
D1
RFB1
CL
R1
Q2
+
Current Feedback
VIN
-
L1
HVIN
VDD
VOUT
DRAIN
Q1
UCC28881
FB
GND
RFB1
Figure 20. Low-Side Buck LED Driver, Direct Feedback (Level Shifted)
22
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9.2.2.2 High-Side Buck Converter
Features include:
• Output Referenced to Input
• Positive Output (VOUT) with Respect to VIN–
• Step Down (VOUT < VIN)
HVIN
VDD
+
DRAIN
UCC28881
VIN
FB
-
10
GND
D2
CFB
RFB2
L1
+
CL
D1
VOUT
-
Figure 21. High-Side Buck Converter Schematic
9.2.2.3 Non-Isolated, Low-Side Buck-Boost Converter
Features Include:
• Output Referenced to Input
• Positive Output (VOUT) with Respect to VIN+
• Step Up, Step Down: VOUT > VIN or VOUT < VIN
• Direct Level-Shifted Feedback
CL
VOUT
D1
VDD
+
VIN
-
-
L1
HVIN
+
DRAIN
UCC28881
FB
GND
RFB2
Figure 22. Low-Side Buck-Boost Converter
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9.2.2.4 Non-Isolated, High-Side Buck-Boost Converter
Features include:
• Output Referenced to Input
• Positive Output (VOUT) with Respect to VIN–
• Step Up, Step Down: VOUT > VIN or VOUT < VIN
HVIN
VDD
DRAIN
UCC28881
+
FB
VIN
RFB1
GND
-
CFB
RFB2
D1
D2
+
VOUT
-
CL
L1
Figure 23. High-Side Buck-Boost Converter
9.2.2.5 Non-Isolated Flyback Converter
Features include:
• Output Referenced to Input
• Positive Output (VOUT) with Respect VIN–
• Direct Feedback
• Higher Output Current Capability, 4.5 W for 85 VAC ~ 265 VAC Input and 6 W for 176 VAC ~ 265 VAC Input
RFB1
CL
HVIN
+
VDD
VIN
-
+
VOUT
-
DRAIN
UCC28881
CVDD
FB
GND
RFB2
Figure 24. Non-Isolated Flyback Configuration
24
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9.2.2.6 Isolated Flyback Converter
Features include:
• Output Isolated from Input
• Direct Feedback
• Higher Output Current Capability, 4.5 W for 85 VAC ~ 265 VAC Input and 6 W for 176 VAC ~ 265VAC Input
CL
HVIN
+
VDD
VIN
-
+
VOUT
-
DRAIN
UCC28881
CVDD
FB
GND
RFB
Figure 25. Isolated Flyback Converter
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10 Power Supply Recommendations
The VDD capacitor recommended value is 100 nF to ensure high-phase margin of the internal 5-V regulator and
it should be placed close to VDD pin and GND pins to minimize the series resistance and inductance.
The VDD pin provides a regulated 5-V output but it is not intended as a supply for external load. Do not supply
VDD pin with external voltage source (for example the auxiliary winding of flyback converter).
Always keep GND pin 1 and GND pin 2 connected together with the shortest possible connection.
11 Layout
11.1 Layout Guidelines
•
•
•
•
In both buck and buck-boost low-side configurations, the copper area of the switching node DRAIN should be
minimized to reduce EMI.
Similarly, the copper area of the FB pin should be minimized to reduce coupling to feedback path. Loop CL,
Q1, RFB1 should be minimized to reduce coupling to feedback path.
In high-side buck and buck boost the GND, VDD and FB pins are all part of the switching node so the copper
area connected with these pins should be optimized. Large copper area allows better thermal management,
but it causes more common mode EMI noise. Use the minimum copper area that is required to handle the
thermal dissipation.
Minimum distance between 700-V coated traces is 1.41 mm (60 mils).
11.2 Layout Example
Figure 26 shows and example PCB layout for UCC28881 in low-side buck configuration.
L2
D2
Rf
C1
C2
xx
xx
D3
60 mils
AC
INPUT
DRAIN
GND_IC
L1
GND_IC
xxx
xxx
xx
xxxx
xx
D1
NC
FB
HVIN
VDD
Rfb2
= top layer
= bottom layer
= connect top and bot
Rfb1
Q1
CL
RL
DC
OUTPUT
Figure 26. UCC28881 Layout Example
26
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12 Device and Documentation Support
12.1 Device Support
12.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
12.2 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
12.3 Trademarks
E2E is a trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.5 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
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PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UCC28881D
ACTIVE
SOIC
D
7
75
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
U28881
UCC28881DR
ACTIVE
SOIC
D
7
2500
RoHS & Green
NIPDAU
Level-2-260C-1 YEAR
-40 to 125
U28881
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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