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UCC28950
SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
UCC28950 Green Phase-Shifted Full-Bridge Controller With Synchronous Rectification
1 Features
3 Description
•
The UCC28950 enhanced phase-shifted controller
builds upon Texas Instrument’s industry standard
UCx895 phase-shifted controller family with
enhancements that offer best in class efficiency in
today’s high performance power systems. The
UCC28950 implements advanced control of the fullbridge along with active control of the synchronous
rectifier output stage.
1
•
•
•
•
•
•
•
•
•
•
Enhanced Wide Range Resonant Zero Voltage
Switching (ZVS) Capability
Direct Synchronous Rectifier (SR) Control
Light-Load Efficiency Management Including
– Burst Mode Operation
– Discontinuous Conduction Mode (DCM),
Dynamic SR On and Off Control with
Programmable Threshold
– Programmable Adaptive Delay
Average or Peak Current Mode Control With
Programmable Slope Compensation and Voltage
Mode Control
Naturally Handles Pre-Biased Start Up With DCM
Mode
Closed Loop Soft Start and Enable Function
Programmable Switching Frequency up to 1 MHz
with Bi-Directional Synchronization
(±3%) Cycle-by-Cycle Current Limit Protection
With Hiccup Mode Support
150-µA Start-Up Current
VDD Undervoltage Lockout
Wide Temperature Range –40°C to +125°C
The primary-side signals allow programmable delays
to ensure ZVS operation over wide-load current and
input voltage range, while the load current naturally
tunes the secondary-side synchronous rectifiers
switching delays, maximizing overall system
efficiency.
The UCC28950 also offers multiple light-load
management features including burst mode and
dynamic SR on/off control when transitioning in and
out of Discontinuous Current Mode (DCM) operation,
ensuring ZVS operation is extended down to much
lighter loads.
In addition, the UCC28950 includes support for
current or voltage mode control. Programmable
switching frequency up to 1 MHz and a wide set of
protection features including cycle-by-cycle current
limit, UVLO and thermal shutdown. A 90-degree
phase-shifted interleaved synchronized operation can
be easily arranged between two converters.
2 Applications
•
•
•
•
•
•
The UCC28950 is available in TSSOP-24 package.
Phase-Shifted Full-Bridge Converters
Datacom, Telecom, and Wireless Base-Station
Power
Server, Power Supplies
Industrial Power Systems
High-Density Power Architectures
Solar Inverters, and Electric Vehicles
Device Information(1)
PART NUMBER
UCC28950
PACKAGE
BODY SIZE (NOM)
TSSOP (24)
7.80 mm x 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the datasheet.
UCC28950 Typical Application
+
CT
VBIAS
VIN
CIN
UCC28950
CREF
R1
R2
1
VREF
GND 24
2
EA+
VDD 23
3
EA-
OUTA 22
A
4
COMP
OUTB 21
B
-
R3
C1
RLF2
VDD
CVDD
VDD
VDD
R5
VSENSE
R6 R4
C2
QA
QC
A
C
C3
CSS
5
SS/EN
OUTC 20
C
RAB
6
DELAB
OUTD 19
D
RCD
7
DELCD
OUTE 18
E
T1
ENABLE
VDD
VDD
LOUT
QB
REF
8
DELEF
OUTF 17
F
RTMIN
9
TMIN
SYNC 16
SYNC
RAHI
RT
VREF
10 RT
B
D
VOUT
CS 15
+
RSUM
11 RSUM
UCC27324
ADEL 14
RAEFHI
RDCMHI
VREF
QD
12 DCM
UCC27324
QE
QF
E
F
ADELEF 13
RA
RLF1
DA
R7
RCS
RDCM
CLF
COUT
-
RAEF
VSENSE
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
UCC28950
SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
www.ti.com
Table of Contents
1
2
3
4
5
6
Features ..................................................................
Applications ...........................................................
Description .............................................................
Revision History.....................................................
Pin Configuration and Functions .........................
Specifications.........................................................
6.1
6.2
6.3
6.4
6.5
6.6
6.7
6.8
7
1
1
1
2
4
5
Absolute Maximum Ratings ...................................... 5
ESD Ratings.............................................................. 5
Recommended Operating Conditions....................... 5
Thermal Information .................................................. 6
Electrical Characteristics........................................... 6
Timing Requirements ................................................ 8
Dissipation Ratings ................................................... 8
Typical Characteristics ............................................ 11
Detailed Description ............................................ 15
7.1 Overview ................................................................. 15
7.2 Functional Block Diagram ....................................... 16
7.3 Feature Description................................................. 17
7.4 Device Functional Modes........................................ 34
8
Application and Implementation ........................ 35
8.1 Application Information............................................ 35
8.2 Typical Application .................................................. 38
9 Power Supply Recommendations...................... 66
10 Layout................................................................... 66
10.1 Layout Guidelines ................................................. 66
10.2 Layout Example .................................................... 67
11 Device and Documentation Support ................. 68
11.1
11.2
11.3
11.4
11.5
11.6
Device Support......................................................
Documentation Support ........................................
Community Resources..........................................
Trademarks ...........................................................
Electrostatic Discharge Caution ............................
Glossary ................................................................
68
68
68
68
68
68
12 Mechanical, Packaging, and Orderable
Information ........................................................... 68
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision C (November 2015) to Revision D
Page
•
Changed Pin Functions table to be alphabetized. ................................................................................................................. 4
•
Added text to UCC28950 Startup Timing Diagram note, "Narrower pulse widths (less than 50% duty cycle) may be
observed in the first OUTD pulse of a burst. The user must design the bootstrap capacitor charging circuit of the
gate driver device so that the first OUTC pulse is transmitted to the MOSFET gate in all cases. Transformer based
gate driver circuits are not affected. This behavior is described in more detail in the application note, Gate Driver
Design Considerations". ......................................................................................................................................................... 9
Changes from Revision B (October 2011) to Revision C
Page
•
Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device
Functional Modes, Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ................................................................................................................................................................................... 1
•
Moved Standard Temperature Range from ESD table to Absolute Maximum Ratings table ............................................... 5
•
Changed Figure 6 Startup Current value from mA to µA. .................................................................................................... 11
•
Changed Figure 11 Nominal Switching Frequency value from Hz to kHz. .......................................................................... 12
•
Changed Figure 12 Maximum Switching Frequency value from Hz to kHz. ....................................................................... 12
•
Updated Adaptive Delay section. ......................................................................................................................................... 19
•
Changed values in Equation 3 ............................................................................................................................................ 19
•
Changed values in Equation 4 ............................................................................................................................................ 19
•
Changed line in Figure 34 to stop at RTMIN= 10 kΩ and TMIN = 800 ns. ............................................................................. 23
•
Changed content in Slope Compensation (RSUM) section. ................................................................................................... 25
•
Added TMIN setting to Figure 39............................................................................................................................................ 27
•
Updated Synchronization (SYNC) section............................................................................................................................ 31
•
Changed Detailed Design Procedure in the Typical Application section. ............................................................................ 39
•
Deleted Vg vs. Qg for QE and QF FETs graph from Select FETs QE and QF section. ....................................................... 48
•
Added Daughter Board Schematic. ..................................................................................................................................... 62
2
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•
SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
Added Power Stage Schematic. .......................................................................................................................................... 63
Changes from Revision A (July 2010) to Revision B
Page
•
Added Naturally Handles Pre-Biased Start Up with DCM Mode bullet .................................................................................. 1
•
Added Datacom, Telecom, and Wireless Base-Station Power .............................................................................................. 1
•
Changed Server, Telecom Power Supplies bullet to Server, Power Supplies ....................................................................... 1
Changes from Original (March 2010) to Revision A
Page
•
Changed UCC28950 Typical Application Diagram................................................................................................................. 1
•
Changed Converter switching frequency from 1400 kHz to 1000 kHz................................................................................... 5
•
Changed Functional Block Diagram ..................................................................................................................................... 16
•
Added Figure 30 ................................................................................................................................................................... 20
•
Changed Equation ................................................................................................................................................................ 21
•
Added Typical Application Diagram...................................................................................................................................... 21
•
Added always deliver even number of Power cycles to Power transformer. ....................................................................... 23
•
Deleted deliver either one or two power delivery cycle pulses. If controller delivers a power delivery cycle for OUTB
and OUTC, then it stops. If it starts delivering to OUTA and OUTD, then it continues with another power delivery
cycle to OUTB and OUTC, and then it stops. ...................................................................................................................... 23
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UCC28950
SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
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5 Pin Configuration and Functions
PW Package
24-Pin TSSOP
Top View
UCC28950
1
VREF
2
EA+
GND 24
VDD 23
3
EA-
OUTA 22
4
COMP
OUTB 21
5
SS/EN
OUTC 20
6
DELAB
OUTD 19
7
DELCD
OUTE 18
8
DELEF
OUTF 17
9
TMIN
SYNC 16
10 RT
11 RSUM
12 DCM
CS 15
ADEL 14
ADELEF 13
Pin Functions
PIN
4
I/O
DESCRIPTION
ADEL
I
Dead-time programming for the primary switches over CS voltage range, TABSET and
TCDSET.
13
ADELEF
I
Delay-time programming between primary side and secondary side switches, TAFSET
and TBESET.
4
COMP
I/O
15
CS
I
Current sense for cycle-by-cycle over-current protection and adaptive delay functions.
12
DCM
I
DCM threshold setting.
6
DELAB
I
Dead-time delay programming between OUTA and OUTB.
7
DELCD
I
Dead-time delay programming between OUTC and OUTD.
8
DELEF
I
Delay-time programming between OUTA to OUTF, and OUTB to OUTE.
2
EA+
I
Error amplifier non-inverting input.
3
EA–
I
Error amplifier inverting input.
24
GND
—
Ground. All signals are referenced to this node.
22
OUTA
O
0.2-A sink/source primary switching output.
21
OUTB
O
0.2-A sink/source primary switching output.
20
OUTC
O
0.2-A sink/source primary switching output.
19
OUTD
O
0.2-A sink/source primary switching output.
18
OUTE
O
0.2-A sink/source synchronous switching output.
11
RSUM
I
Slope compensation programming. Voltage mode or peak current mode setting.
10
RT
I
Oscillator frequency set. Master or slave mode setting.
5
SS/EN
I
Soft-start programming, device enable and hiccup mode protection circuit.
16
SYNC
I/O
Synchronization out from Master controller to input of slave controller.
17
OUTF
O
0.2-A sink/source synchronous switching output.
9
TMIN
I
Minimum duty cycle programming in burst mode.
23
VDD
I
Bias supply input.
1
VREF
O
5-V, ±1.5%, 20-mA reference voltage output.
NO.
NAME
14
Error amplifier output and input to the PWM comparator.
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SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
6 Specifications
6.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted)
(1) (2)
MIN
MAX
UNIT
–0.4
20
V
OUTA, OUTB, OUTC, OUTD, OUTE, OUTF
–0.4
VDD + 0.4
V
Input voltage on DELAB, DELCD, DELEF, SS/EN, DCM, TMIN, RT, SYNC, RSUM, EA+, EA-,
COMP, CS, ADEL, ADELEF
–0.4
VREF + 0.4
V
Output voltage on VREF
–0.4
5.6
V
Input supply voltage, VDD
(3)
Continuous total power dissipation
See Dissipation Ratings
Operating virtual junction temperature, TJ
–40
+150
°C
Operating ambient temperature, TA
–40
+125
°C
+300
°C
+150
°C
Lead temperature (soldering, 10 sec.)
Storage temperature, Tstg
(1)
(2)
(3)
–65
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under Recommended Operating
Conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
These devices are sensitive to electrostatic discharge; follow proper device handling procedures.
All voltages are with respect to GND unless otherwise noted. Currents are positive into, negative out of the specified terminal. See
Packaging Section of the datasheet for thermal limitations and considerations of packages.
6.2 ESD Ratings
V(ESD)
(1)
(2)
Electrostatic discharge
VALUE
UNIT
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001 (1)
2000
V
Charged device model (CDM), per JEDEC specification JESD22C101 (2)
500
V
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
Supply voltage range, VDD
MIN
NOM
8
12
MAX
UNIT
17
V
–40
125
°C
Converter switching frequency setting range, FSW(nom)
50
1000
kHz
Programmable delay range between OUTA, OUTB and OUTC, OUTD set by
resistors DELAB and DELCD and parameter KA (1)
30
1000
ns
Programmable delay range between OUTA, OUTF and OUTB, OUTE set by
resistor DELEF, and parameter KEF (1)
30
1400
ns
Programmable DCM range as percentage of voltage at CS (1)
5%
30%
Programmable TMIN range
100
800
Operating junction temperature range
(1)
ns
Verified during characterization only.
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6.4 Thermal Information
UCC28950
THERMAL METRIC (1)
PW (TSSOP)
UNIT
24 PINS
RθJA
Junction-to-ambient thermal resistance
93.3
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
24.2
°C/W
RθJB
Junction-to-board thermal resistance
47.9
°C/W
ψJT
Junction-to-top characterization parameter
0.7
°C/W
ψJB
Junction-to-board characterization parameter
47.4
°C/W
RθJC(bot)
Junction-to-case (bottom) thermal resistance
n/a
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report, SPRA953.
6.5 Electrical Characteristics
VDD = 12 V, TA = TJ = –40°C to +125°C, CVDD = 1 µF, CREF = 1 µF, RAB = 22.6 kΩ, RCD = 22.6 kΩ , REF = 13.3 kΩ, RSUM = 124
kΩ, RTMIN = 88.7 kΩ, RT = 59 kΩ connected between RT pin and 5-V voltage supply to set FSW = 100 kHz (FOSC = 200 kHz)
(unless otherwise noted). All component designations are from Figure 48.
PARAMETER
TEST CONDITIONS
MIN
TYP (1)
MAX
UNIT
UNDERVOLTAGE LOCKOUT (UVLO)
UVLO_RTH
Start threshold
6.75
7.3
7.9
V
UVLO_FTH
Minimum operating voltage
after start
6.15
6.7
7.2
V
UVLO_HYST
Hysteresis
0.53
0.6
0.75
V
150
270
µA
5
10
mA
5
5.075
SUPPLY CURRENTS
IDD(off)
Startup current
IDD
Operating supply current
VDD is 5.2 V
VREF OUTPUT VOLTAGE
VREF
VREF total output range
0 ≤ IR ≤ 20 mA; VDD = from 8 V to 17 V
ISCC
Short circuit current
VREF = 0 V
4.925
–53
V
–23
mA
100
108
kHz
95%
97%
SWITCHING FREQUENCY (½ OF INTERNAL OSCILLATOR FREQUENCY FOSC)
FSW(nom)
Total range
DMAX
Maximum duty cycle
92
SYNCHRONIZATION
PHSYNC
Total range
RT = 59 kΩ between RT and GND; Input pulses
200 kHz, D = 0.5 at SYNC
85
90
95
°PH
FSYNC
Total range
RT = 59 kΩ between RT and 5 V; –40 °C ≤ TJ ≤
+125°C
180
200
220
kHz
TPW
Pulse width
2.2
2.5
2.8
µs
(1)
6
Typical values for TA = 25°C
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Electrical Characteristics (continued)
VDD = 12 V, TA = TJ = –40°C to +125°C, CVDD = 1 µF, CREF = 1 µF, RAB = 22.6 kΩ, RCD = 22.6 kΩ , REF = 13.3 kΩ, RSUM = 124
kΩ, RTMIN = 88.7 kΩ, RT = 59 kΩ connected between RT pin and 5-V voltage supply to set FSW = 100 kHz (FOSC = 200 kHz)
(unless otherwise noted). All component designations are from Figure 48.
PARAMETER
TYP (1)
TEST CONDITIONS
MIN
MAX
UNIT
VICM range ensures parameters, the functionality
ensured for 3.6 V < VICM < VREF + 0.4 V, and
–0.4 V < VICM < 0.5 V
0.5
3.6
–7
7
mV
1
µA
ERROR AMPLIFIER
VICM
Common mode input
voltage range
VIO
Offset voltage
IBIAS
Input bias current
EAHIGH
High-level output voltage
(EA+) – (EA–) = 500 mV, IEAOUT = –0.5 mA
EALOW
Low-level output voltage
(EA+) – (EA–) = –500 mV, IEAOUT = 0.5 mA
ISOURCE
Error amplifier source
current
–8
ISINK
Error amplifier sink current
2.7
IVOL
Open-loop dc gain
GBW
Unity gain bandwidth
–1
3.9
4.25
V
0.25
0.35
V
–3.75
–0.5
mA
4.6
5.75
mA
100
(2)
V
dB
3
MHz
CYCLE-BY-CYCLE CURRENT LIMIT
VCS_LIM
CS pin cycle-by-cycle
threshold
1.94
2
2.06
V
15
20
25
µA
3.2
3.6
4.2
V
1.90
2.55
3.2
µA
20
25
30
µA
0.25
0.50
0.70
V
INTERNAL HICCUP MODE SETTINGS
IDS
Discharge current to set
cycle-by-cycle current limit
duration
VHCC
Hiccup OFF Time threshold
IHCC
Discharge current to set
Hiccup Mode OFF Time
CS = 2.5 V, VSS = 4 V
SOFT START/ENABLE
ISS
Charge current
VSS_STD
Shutdown/restart/reset
threshold
VSS_PU
Pull up threshold
VSS_CL
Clamp voltage
VSS = 0 V
3.3
3.7
4.3
V
4.20
4.65
4.95
V
LIGHT-LOAD EFFICIENCY CIRCUIT
VDCM
IDCM_SRC
DCM threshold, T = 25°C
VDCM = 0.4 V, Sweep CS confirm there are
OUTE and OUTF pulses
0.37
0.39
0.41
V
DCM threshold, T = 0°C to
+85°C (3)
VDCM = 0.4 V, Sweep CS, confirm there are
OUTE and OUTF pulses
0.364
0.390
0.416
V
DCM threshold, T= –40°C
to +125°C (3)
VDCM = 0.4 V, Sweep CS, confirm there are
OUTE and OUTF pulses
0.35
0.39
0.43
V
DCM Sourcing Current
CS < DCM threshold
14
20
26
µA
OUTPUTS OUTA, OUTB, OUTC, OUTD, OUTE, OUTF
ISINK/SRC
Sink/Source peak current (3)
RSRC
Output source resistance
IOUT = 20 mA
10
20
35
Ω
RSINK
Output sink resistance
IOUT = 20 mA
5
10
30
Ω
0.2
A
THERMAL SHUTDOWN
Rising threshold (3)
160
°C
Falling threshold (3)
140
°C
20
°C
Hysteresis
(2)
(3)
Verified during characterization only.
Verified during characterization only.
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6.6 Timing Requirements
MIN
NOM
MAX
UNIT
CYCLE-BY-CYCLE CURRENT LIMIT
Propagation delay from CS to OUTC and OUTD outputs
Input pulse between CS and GND from zero to 2.5 V
TCS
100
ns
PROGRAMMABLE DELAY TIME SET ACCURACY AND RANGE (1) (2) (3) (4) (5)
TABSET1
Short delay time set accuracy between OUTA and OUTB
CS = ADEL = ADELEF = 1.8 V
32
45
56
ns
TABSET2
Long delay time set accuracy between OUTA and OUTB
CS = ADEL = ADELEF = 0.2 V
216
270
325
ns
TCDSET1
Short delay time set accuracy between OUTC and OUTD
CS = ADEL = ADELEF = 1.8 V
32
45
56
ns
TCDSET2
Long delay time set accuracy between OUTC and OUTD
CS = ADEL = ADELEF = 0.2 V
216
270
325
ns
TAFSET1
Short delay time set accuracy between falling OUTA, OUTF
CS = ADEL = ADELEF = 0.2 V
22
35
48
ns
TAFSET2
Long delay time set accuracy between falling OUTA, OUTF
CS = ADEL = ADELEF = 1.8 V
190
240
290
ns
TBESET1
Short delay time set accuracy between falling OUTB, OUTE
CS = ADEL = ADELEF = 0.2 V
22
35
48
ns
TBESET2
Long delay time set accuracy between falling OUTB, OUTE
CS = ADEL = ADELEF = 1.8 V
190
240
290
ns
ΔTADBC
Pulse matching between OUTA rise, OUTD fall and OUTB rise,
OUTC fall
CS = ADEL = ADELEF = 1.8 V, COMP = 2 V
–50
0
50
ns
ΔTABBA
Half cycle matching between OUTA rise, OUTB rise and OUTB rise,
OUTA rise
CS = ADEL = ADELEF = 1.8 V, COMP = 2 V
–50
0
50
ns
ΔTEEFF
Pulse matching between OUTE fall, OUTE rise and OUTF fall, OUTF
rise
CS = ADEL = ADELEF = 0.2 V, COMP = 2 V
–60
0
60
ns
ΔTEFFE
Pulse matching between OUTE fall, OUTF rise and OUTF fall, OUTE
rise
CS = ADEL = ADELEF = 0.2 V, COMP = 2 V
–60
0
60
ns
425
525
625
ns
LIGHT-LOAD EFFICIENCY CIRCUIT
TMIN
Total range, RTMIN = 88.7 kΩ
OUTPUTS OUTA, OUTB, OUTC, OUTD, OUTE, OUTF
TR
Rise time, CLOAD = 100 pF
9
25
ns
TF
Fall time, CLOAD = 100 pF
7
25
ns
(1)
(2)
(3)
(4)
(5)
See Figure 28 for timing diagram and TABSET1, TABSET2, TCDSET1, TCDSET2 definitions.
See Figure 31 for timing diagram and TAFSET1, TAFSET2, TBESET1, TBESET2 definitions.
Pair of outputs OUTC, OUTE and OUTD, OUTF always going high simultaneously.
Outputs A or B are never allowed to go high if both outputs OUTE and OUTF are high.
All delay settings are measured relative to 50% of pulse amplitude.
6.7 Dissipation Ratings
over operating free-air temperature range (unless otherwise noted)
PACKAGE
PW
8
DERATING FACTOR
POWER RATING
ABOVE TA = 25°C
TA < 25°C
TA = 70°C
TA = 85°C
10.7 mW/°C
1.07 W
0.59 W
0.429 W
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VDD 7.3-V rise, 6.7-V fall
VDD_GOOD
VREF
VREF_GOOD
4.8-V rise, 4.6-V fall
SS > 0.5 V, then release COMP, DCM, CS , Outputs A,B,C,D,E and F
CLK
TMIN
TMIN
COMP
RAMP
PWM
Add 0.85 V offset to RAMP
No PWM pulses shorter than TMIN
except during cycle-by-cycle current limit
PWM
TMIN
2 VP-P
A
B
C
D
E
F
Burst Mode at the beginning of
start up until PWM> TMIN pulses
No output delay shown, COMP-to-RAMP offset not included.
There is no pulse on OUTE during burst mode at startup. Two falling edge PWM pulses are required before enabling
the synchronous rectifier outputs. Narrower pulse widths (less than 50% duty cycle) may be observed in the first
OUTD pulse of a burst. The user must design the bootstrap capacitor charging circuit of the gate driver device so that
the first OUTC pulse is transmitted to the MOSFET gate in all cases. Transformer based gate driver circuits are not
affected. This behavior is described in more detail in the application note, Gate Driver Design Considerations.
Figure 1. UCC28950 Startup Timing Diagram
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VDD failed and VDD_GOOD goes low,
Everything is shutdown
7.3V rise, 6.7V fall
VDD
VDD_GOOD
4.8V rise, 4.6V fall
VREF
VREF_GOOD
TMIN
CLK
TMIN
Add 0.85V offset to RAMP
COMP
2Vp-p
RAMP
PWM
No PWM pulses shorter than TMIN except
during cycle-by-cycle current limit
A
B
C
D
E
F
No output delay shown, COMP-to-RAMP offset not included.
Figure 2. UCC28950 Steady State/Shutdown Timing Diagram
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6.8 Typical Characteristics
640
UVLO - Under Voltage Lockout Hysteresis - mV
UVLO - Under Voltage Lockout Thresholds - V
7.6
UVLO_RTH
7.4
7.2
7.0
UVLO_FTH
6.8
6.6
6.4
620
UVLO_HYST
610
600
590
580
6.2
-40
-40
125
25
125
TJ - Temperature - °C
Figure 3. UVLO Thresholds vs Temperature
Figure 4. UVLO Hysteresis vs Temperature
250
3.8
IDD - Startup Current - PA
200
3.7
3.6
150
100
3.5
3.4
50
-40
125
25
-40
TJ - Temperature - °C
25
125
TJ - Temperature - °C
Figure 6. Startup Current vs Temperature
Figure 5. Supply Current vs Temperature
5.010
5.001
ILOAD = 10µA
5.000
4.999
VREF - Line Voltage Regulation - V
5.005
VREF - Voltage Reference - V
25
TJ - Temperature - °C
3.9
IDD - Operating Supply Current - mA
630
ILOAD = 1 mA
4.995
ILOAD = 10 mA
4.990
4.985
ILOAD = 20 mA
4.980
VREF _ 10 mA _ 12 VDD
4.997
4.995
VREF _ 10 mA _ 10 VDD
4.993
4.991
VREF _ 10 mA _ 8 VDD
4.989
4.975
4.987
-40
25
125
4.985 -40
TJ - Temperature - °C
25
125
TJ - Temperature - °C
Figure 7. Voltage Reference (VDD = 12 V) vs Temperature
Figure 8. Line Voltage Regulation (ILOAD = 10 mA) vs
Temperature
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Typical Characteristics (continued)
38.5
95.4
95.2
DMAX - Maximum Duty Cycle - %
Short Circuit Current - mA
38.0
37.5
37.0
36.5
36.0
95.0
94.8
94.6
94.4
94.2
94.0
35.5
93.8
35.0
93.6
-40
25
125
-40
125
TJ - Temperature - °C
Figure 9. Short Circuit Current vs Temperature
Figure 10. Maximum Duty Cycle vs Temperature
1079
FSW(max) - Maximum Switching Frequency - kHz
FSW(nom) - Nominal Switching Frequency - kHz
95.4
95.0
94.6
94.0
1059
1039
1019
999
93.6
-40
25
125
-40
TJ - Temperature - °C
125
Figure 12. Maximum Switching Frequency vs Temperature
125
0.00
-0.05
120
AVOL - Voltage Error Amplifier - dB
Error Amplifier OFFSET voltage - mV
25
TJ - Temperature - °C
Figure 11. Nominal Switching Frequency vs Temperature
-0.10
-0.15
-0.20
VIO = 500 mV
-0.25
VIO = 3.6 V
-0.30
-0.35
VIO = 2.5 V
-0.40
115
110
105
100
95
90
-0.45
85
-0.50
-40
25
-40
125
25
125
TJ - Temperature - °C
TJ - Temperature - °C
Figure 13. Error Amplifier Offset Voltage vs Temperature
12
25
TJ - Temperature - °C
Figure 14. Voltage Error Amplifier (Open Loop Gain) vs
Temperature
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Typical Characteristics (continued)
0.60
VSS(std) - Shutdown/Restart/Reset Threshold - V
26.0
ISS - Charge Current – µA
25.5
25.0
24.5
24.0
23.5
0.55
0.50
0.45
0.40
0.35
0.30
-40
25
125
-40
TJ - Temperature - °C
25
125
TJ - Temperature - °C
Figure 15. ISS Charge Current vs Temperature
Figure 16. Shutdown/Restart/Reset Threshold vs
Temperature
3.715
4.692
4.688
3.710
VSS(CL) - SS Clamp Voltage - V
VSS(pu) - SS Pullup Threshold - V
4.690
3.705
4.686
4.684
4.682
4.680
3.700
4.678
4.676
3.695
4.674
-40
25
125
-40
TJ - Temperature - °C
Figure 17. SS Pull-Up Threshold vs Temperature
125
Figure 18. SS Clamp Voltage vs Temperature
110
TCS(prop) - Current Sense Propagation Delay - ns
1.996
VCS(lim) - Current Sense Cycle-By-Cycle Limit - V
25
TJ - Temperature - °C
1.994
1.992
1.990
1.988
1.986
1.984
107
104
101
98
95
-40
25
125
-40
TJ - Temperature - °C
25
125
TJ - Temperature - °C
Figure 19. Current Sense Cycle-by-Cycle Limit vs
Temperature
Figure 20. Current Sense Propagation Delay vs Temperature
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Typical Characteristics (continued)
17.5
RSINK - Outputs Sink Resistance – Ω
RSINK - Outputs Sink Resistance – Ω
17.5
RSINK_OUTF
RSINK_OUTD
15.5
RSINK_OUTA
13.5
11.5
9.5
RSINK_OUTE
RSINK_OUTC
15.5
RSINK_OUTB
13.5
11.5
9.5
7.5
7.5
-40
25
125
-40
25
TJ - Temperature - °C
Figure 21. Outputs Sink Resistance vs Temperature
Figure 22. Outputs Sink Resistance vs Temperature
25
RSRC - Outputs Source Resistance – Ω
25
RSRC - Outputs Source Resistance – Ω
125
TJ - Temperature - °C
RSRC_OUTF
RSRC_OUTC
23
RSRC_OUTA
21
19
17
RSRC_OUTE
RSRC_OUTD
23
RSRC_OUTB
21
19
17
15
15
-40
25
-40
125
25
125
TJ - Temperature - °C
TJ - Temperature - °C
Figure 23. Outputs Source Resistance vs Temperature
Figure 24. Outputs Source Resistance vs Temperature
50
280
TCDSET2
270
TOFFTIME - Dead Time Delay - ns
TOFFTIME - Dead Time Delay - ns
TCDSET1
TABSET1
45
40
TAFSET1
35
250
TAFSET2
240
TBESET2
230
TBESET1
30
220
-40
14
TABSET2
260
25
125
-40
25
125
TJ - Temperature - °C
TJ - Temperature - °C
Figure 25. Dead Time Delay vs Temperature
Figure 26. Dead Time Delay vs Temperature
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Typical Characteristics (continued)
0.405
0.400
DCM Threshold - V
0.395
0.390
0.385
0.380
0.380
0.375
-40
25
125
TJ - Temperature - °C
Figure 27. DCM Threshold vs Temperature
7 Detailed Description
7.1 Overview
The UCC28950 device combines all the functions necessary to control a phase-shifted full bridge power stage in
a 24-pin TSSOP package. The device includes two Synchronous-Rectifier (SR), gate-drive outputs as well as the
outputs needed to drive all four switches in the full-bridge circuit. The dead times between the upper and lower
switches in the full bridge may be set using the DELAB and DELCD inputs. Further, this dead time may be
dynamically adjusted according to the load level using the ADEL pin. This allows the user to optimize the dead
time for their particular power circuit and to achieve ZVS over the entire operating range. In a similar manner, the
dead times between the full bridge switches and the secondary SRs may be optimized using the DELEF input.
This dead time may also be dynamically adjusted according to the load, using the ADELEF input to the
controller. A DCM (Discontinuous Conduction Mode) option disables the SRs at a user settable light load in order
to improve power circuit efficiency. The device enters a light-load-burst mode if the feedback loop demands a
conduction time less than a user settable level (TMIN).
At higher-power levels, two or more UCC28950 devices may be easily synchronized in a Master/Slave
configuration. A SS/EN input may be used to set the length of the soft start process and to turn the controller on
and off. The controller may be configured for Voltage mode or Current mode control. Cycle-by-cycle current
limiting is provided in Voltage mode and Peak Current mode. The switching frequency may be set over a wide
range making this device suited to both IGBT and MOSFET based designs.
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7.2 Functional Block Diagram
ADEL
14
VDD
UVLO
COMP
VDD
Thermal
Shutdown
VDD 23
EN
Reference
Generator
+
+
VREF
1
COMP
4
EA-
3
EA+
2
22 OUTA
VDD
ON/OFF
5V LDO
+
+
20 OUTC
+
Programmable
Delay CD
CLK
7
DELCD
19 OUTD
Oscillator
RAMP
2.8 V
0.8 V
13 ADELEF
Ramp
Summing
18 OUTE
+
CS
Cycle-by-Cycle
ILIM
CS 15
Synchronization
Block
+
-
Programmable
Delay EF
8
DELEF
CS
Light-Load
Efficiency Block
2V
16
DELAB
21 OUTB
PWM
COMP
Lower "+" Input
is Dominant
RSUM 11
6
7.3 V Rise
6.7 V Fall
Logic Block
RT 10
Programmable
Delay AB
Soft Start and Enable
with 0.55 V Threshold
16
24
12
9
5
SYNC
GND
DCM
TMIN
SS/EN
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7.3 Feature Description
7.3.1 Start-Up Protection Logic
Before the UCC28950 controller will start up, the following conditions must be met:
• VDD voltage exceeds rising UVLO threshold 7.3-V typical.
• The 5-V reference voltage is available.
• Junction temperature is below the thermal shutdown threshold of 140°C.
• The voltage on the soft-start capacitor is not below 0.55-V typical.
If all those conditions are met, an internal enable signal EN is generated that initiates the soft start process. The
duty cycle during the soft start is defined by the voltage at the SS pin, and cannot be lower than the duty cycle
set by TMIN, or by cycle-by-cycle current limit circuit depending on load conditions.
7.3.2 Voltage Reference (VREF)
The accurate (±1.5%) 5-V reference voltage regulator with a short circuit protection circuit supplies internal
circuitry and provides up to 20-mA external output current. Place a low ESR and ESL, preferably ceramic
decoupling capacitor CREF in 1-µF to 2.2-µF range from this pin to GND as close to the related pins as possible
for best performance. The only condition where the reference regulator is shut down internally is during under
voltage lockout.
7.3.3 Error Amplifier (EA+, EA–, COMP)
The error amplifier has two uncommitted inputs, EA+ and EA-, with a 3-MHz unity gain bandwidth, which allows
flexibility in closing the feedback loop. The EA+ is a non-inverting input, the EA– is an inverting input and the
COMP is the output of the error amplifier. The input voltage common mode range, where the parameters of the
error amplifier are guaranteed, is from 0.5 V to 3.6 V. The output of the error amplifier is connected internally to
the non-inverting input of the PWM comparator. The range of the error amplifier output of 0.25 V to 4.25 V far
exceeds the PWM comparator input ramp-signal range, which is from 0.8 V to 2.8 V. The soft-start signal serves
as an additional non-inverting input of the error amplifier. The lower of the two non-inverting inputs of the error
amplifier is the dominant input and sets the duty cycle where the output signal of the error amplifier is compared
with the internal ramp at the inputs of the PWM comparator.
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Feature Description (continued)
7.3.4 Soft Start and Enable (SS/EN)
The soft-start pin SS/EN is a multi-function pin used for the following operations:
• Closed loop soft start with the gradual duty cycle increase from the minimum set by TMIN up to the steady
state duty cycle required by the regulated output voltage.
• Setting hiccup mode conditions during cycle-by-cycle overcurrent limit.
• On/off control for the converter.
During soft start, one of the voltages at the SS/EN or EA+ pins, whichever is lower (SS/EN – 0.55 V) or EA+
voltage (see Block Diagram), sets the reference voltage for a closed feedback loop. Both SS/EN and EA+ signals
are non-inverting inputs of the error amplifier with the COMP pin being its output. Thus the soft start always goes
under the closed feedback loop and the voltage at COMP pin sets the duty cycle. The duty cycle defined by the
COMP pin voltage can not be shorter than TMIN pulse width set by the user. However, if the shortest duty cycle
is set by the cycle-by-cycle current limit circuit, then it becomes dominant over the duty cycle defined by the
COMP pin voltage or by the TMIN block.
The soft-start duration is defined by an external capacitor CSS, connected between the SS/EN pin and ground,
and the internal charge current that has a typical value of 25 µA. Pulling the soft-start pin externally below 0.55 V
shuts down the controller. The release of the soft-start pin enables the controller to start, and if there is no
current limit condition, the duty cycle applied to the output inductor gradually increases until it reaches the steady
state duty cycle defined by the regulated output voltage of the converter. This happens when the voltage at the
SS/EN pin reaches and then exceeds by 0.55 V, the voltage at the EA+ pin. Thus for the given soft-start time
TSS, the CSS value can be defined by Equation 1 or Equation 2:
CSS(master ) =
CSS(slave) =
TSS ´ 25 mA
(0.55 + EA+)
(1)
TSS
æ
ö
20.6
825k ´ Ln ç
÷
è 20.6 – 0.55 – EA+ ø
(2)
For example, in Equation 1, if the soft-start time TSS is selected to be 10 ms, and the EA+ pin is 2.5 V, then the
soft-start capacitor CSS is equal to 82 nF.
NOTE
If the converter is configured in Slave Mode, place an 825-kΩ resistor from SS pin to
ground.
7.3.5 Light-Load Power Saving Features
The UCC28950 offers four different light-load management techniques for improving the efficiency of a power
converter over a wide load current range.
1. Adaptive Delay,
(a) ADEL, which sets and optimizes the dead-time control for the primary switches over a wide load current
range.
(b) ADELEF, which sets and optimizes the delay-time control between the primary side switches and the
secondary side switches.
2. TMIN, sets the minimum pulse width as long as the part is not in current limit mode.
3. Dynamic synchronous rectifier on/off control in DCM Mode, For increased efficiency at light loads. The DCM
Mode starts when the voltage at CS pin is lower than the threshold set by the user. In DCM Mode, the
synchronous output drive signals OUTE and OUTF are brought down low.
4. Burst Mode, for maximum efficiency at very light loads or no load. Burst Mode has an even number of PWM
TMIN pulses followed by off time. Transition to the Burst Mode is defined by the TMIN duration set by the
user.
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Feature Description (continued)
7.3.6 Adaptive Delay, (Delay between OUTA and OUTB, OUTC and OUTD (DELAB, DELCD, ADEL))
The resistor RAB from the DELAB pin, DELAB to GND, along with the resistor divider RAHI from CS pin to ADEL
pin and RA from ADEL pin to GND sets the delay TABSET between one of outputs OUTA or OUTB going low and
the other output going high Figure 28. The total resistance of this resistor divider should be in the range between
10 kΩ and 20 kΩ
OUTA
(OUTC)
TABSET2
TABSET2
TCDSET2
TCDSET2
TABSET1
TABSET1
TCDSET1
TCDSET1
OUTB
(OUTD)
Figure 28. Delay Definitions Between OUTA and OUTB, OUTC and OUTD
This delay gradually increases as a function of the CS signal from TABSET1, which is measured at VCS = 1.8 V, to
TABSET2, which is measured at the VCS = 0.2 V. This approach ensures there will be no shoot-through current
during the high-side and low-side MOSFET switching and optimizes the delay for acheiving ZVS condition over a
wide load current range. The ratio between the longest and shortest delays is set by the resistor divider RAHI and
RA. The max ratio is achieved by tying the CS and ADEL pins together. If ADEL is connected to GND, then the
delay is fixed, defined only by the resistor RAB from DELAB to GND. The delay TCDSET1 and TCDSET2 settings and
their behaviour for outputs OUTC and OUTD are very similar to the one described for OUTA and OUTB. The
difference is that resistor RCD connected between DELCD pin and GND sets the delay TCDSET. The ratio between
the longest and shortest delays is set by the resistor divider RAHI and RA.
The delay time TABSET is defined by the following Equation 3.
æ
ö
5 ´ R AB
TABSET = ç
÷ ns
è 0.26 V + CS ´ K A ´ 1.3 ø
(3)
The same equation is used to define the delay time TCDSET in another leg except RAB is replaced by RCD.
æ
ö
5 ´ RCD
TCDSET = ç
÷ ns
è 0.26 V + CS ´ K A ´ 1.3 ø
(4)
In these equations RAB and RCD are in kΩ and CS, the voltage at pin CS, is in volts and KA is a numerical
coefficient in the range from 0 to 1. The delay time TABSET and TCDSET are in ns, and is measured at the IC pins.
These equations are empirical and they are approximated from measured data. Thus, there is no unit agreement
in the equations. As an example, assume RAB = 15 kΩ, CS = 1 V and KA = 0.5. Then the TABSET will be
approximately 90 ns. In both Equation 3 and Equation 4, KA is the same and is defined as:
KA =
RA
R A + R AHI
(5)
KA sets how the delay varies with the CS pin voltage as shown in Figure 29 and Figure 30.
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Feature Description (continued)
It is recommended to start by setting KA = 0 and set TABSET and TCDSET relatively large using equations or plots in
this data sheet to avoid hard switching or even shoot through current. The delay between outputs A, B and C, D
set by resistors RAB and RCS accordingly. Program the optimal delays at light load first. Then by changing KA set
the optimal delay for the outputs A, B at maximum current. KA for outputs C, D is the same as for A,D. Usually
outputs C, D always have ZVS if sufficient delay is provided.
NOTE
The allowed resistor range on DELAB and DELCD, RAB and RCD is 13 kΩ to 90 kΩ.
RA and RAHI define the portion of voltage at pin CS applied to the pin ADEL (See Figure 48). KA defines how
significantly the delay time depends on CS voltage. KA varies from 0, where ADEL pin is shorted to ground (RA =
0) and the delay does not depend on CS voltage, to 1, where ADEL is tied to CS (RAHI = 0). Setting KA, RAB and
RCD provides the ability to maintain optimal ZVS conditions of primary switches over load current because the
voltage at CS pin includes the load current reflected to the primary side through the current sensing circuit. The
plots in Figure 29 and Figure 30 show the delay time settings as a function of CS voltage and KA for two different
conditions: RAB = RCD = 13 kΩ (Figure 29) and RAB = RCD = 90 kΩ (Figure 30).
350
2000
KA = 0
KA = 0.1
KA = 0.25
KA = 0.5
KA = 0.75
KA = 1
250
1600
TABSET, TCDSET - Time Delay - ns
TABSET, TCDSET - Time Delay - ns
300
KA = 0
KA = 0.1
KA = 0.25
KA = 0.5
KA = 0.75
KA = 1
1800
200
150
100
1400
1200
1000
800
600
400
50
200
0
0
0
20
0.2
0.4
0.6
0.8
1
1.2
CS Voltage - V
1.4
1.6
1.8
2
0
0.5
1
CS Voltage - V
1.5
2
G001
G001
Figure 29. Delay Time Set TABSET and TCDSET
(Over CS voltage variation and selected KA for RAB and
RCD equal 13 kΩ)
Figure 30. Delay time set TABSET and TCDSET
(Over CS voltage variation and selected KA for RAB and
RCD equal 90 kΩ)
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Feature Description (continued)
7.3.7 Adaptive Delay (Delay between OUTA and OUTF, OUTB and OUTE (DELEF, ADELEF))
The resistor REF from the DELEF pin to GND along with the resistor divider RAEFHI from CS pin to ADELEF pin
and RAEF from ADELEF pin to GND sets equal delays TAFSET and TBESET between outputs OUTA or OUTB going
low and related output OUTF or OUTE going low Figure 31. The total resistance of this resistor divider should be
in the range between 10kΩ and 20kΩ.
OUTA
(OUTB)
OUTD
(OUTC)
TAFSET1
TBESET1
OUTF
(OUTE)
TAFSET2
TBESET2
Figure 31. Delay Definitions Between OUTA and OUTF, OUTB and OUTE
These delays gradually increase as function of the CS signal from TAFSET1, which is measured at VCS = 0.2 V, to
TAFSET2, which is measured at VCS = 1.8 V. This is opposite to the DELAB and DELCD behavior and this delay is
longest (TAFSET2) when the signal at CS pin is maximized and shortest (TAFSET1) when the CS signal is
minimized. This approach will reduce the synchronous rectifier MOSFET body diode conduction time over a wide
load current range thus improving efficiency . The ratio between the longest and shortest delays is set by the
resistor divider RAEFHI and RAEF. If CS and ADELEF are tied, the ratio is maximized. If ADELEF is connected to
GND, then the delay is fixed, defined only by resistor REF from DELEF to GND.
The delay time TAFSET is defined by the following Equation 6. Equation 6 also defines the delay time TBESET.
ææ
ö
ö
5 ´ REF
TAFSET = ç ç
ns + 4ns ÷
÷
ç
÷
è è 2.65 V - CS ´ K EF ´ 1.32 ø
ø
(6)
In this equation REF is in kΩ, the CS, which is the voltage at pin CS, is in volts and KEF is a numerical gain factor
of CS voltage from 0 to 1. The delay time TAFSET is in ns, and is measured at the IC pins. This equation is an
empirical approximation of measured data, thus, there is no unit agreement in it. As an example, assume REF =
15 kΩ, CS = 1 V and KEF = 0.5. Then the TAFSET is going to be 41.7 ns. KEF is defined as:
K EF =
R AEF
R AEF + R AEF(hi)
(7)
RAEF and RAEFHI define the portion of voltage at pin CS applied to the pin ADELEF (See Figure 48). KEF defines
how significantly the delay time depends on CS voltage. KEF varies from 0, where ADELEF pin is shorted to
ground (RAEF = 0) and the delay does not depend on CS voltage, to 1, where ADELEF is tied to CS (RAEFHI = 0).
NOTE
The allowed resistor range on DELEF, REF is 13 kΩ to 90 kΩ.
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Feature Description (continued)
The plots in Figure 32 and Figure 33 show delay time settings as function of CS voltage and KEF for two different
conditions: REF = 13kΩ (Figure 32) and REF = 90kΩ (Figure 33)
350
2000
1800
300
KA = 0.00
KA = 0.25
200
KA = 0.50
KA = 0.75
150
KA = 0.90
KA = 1.00
100
TAFSET, TBESET - Time Delay - ns
TAFSET, TBESET - Time Delay - ns
1600
250
1400
KA = 0.0
KA = 0.4
1200
KA = 0.5
1000
KA = 0.8
800
KA = 0.9
KA = 1.0
600
400
50
200
5
5
0.0
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
0.0
0.2
CS Voltage - V
0.6
0.8
1.0
1.2
1.4
1.6
1.8
CS Voltage - V
Figure 32. Delay Time TAFSET and TBESET
(Over CS Voltage and Selected KEF for REF equal 13 kΩ)
22
0.4
Figure 33. Delay Time TAFSET and TBESET
(Over CS Voltage and Selected KEF for REF equal 90 kΩ)
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Feature Description (continued)
7.3.8 Minimum Pulse (TMIN)
The resistor RTMIN from the TMIN pin to GND sets a fixed minimum pulse width. This pulse is applied to the
transformer and enables ZVS at light load. If the output PWM pulse demanded by the feedback loop is shorter
than TMIN, then the controller proceeds to burst mode operation where an even number of TMIN pulses are
followed by the off time dictated by the feedback loop. The proper selection of the TMIN duration is dictated by
the time it takes to raise sufficient magnetizing current in the power transformer to maintain ZVS. The TMIN
pulse is measured from the rising edge of OUTA to the falling edge of OUTD – or from the rising edge of OUTB
to the falling edge of OUTC. The minimum pulse TMIN is then defined by Equation 8.
TMIN = (5.92 ´ RTMIN ) ns
(8)
Where TMIN is in ns and RTMIN is in kΩ The pulse width measured at the transformer will be modified (usually
increased) by various propagation and response time delays in the power circuit. Because of the propagation
and response time delays in the power circuit, selecting the correct TMIN setting will be an iterative process.
NOTE
The minimum allowed resistor on TMIN, RTMIN is 10 kΩ.
The related plot is shown in Figure 34.
800
700
600
TMIN (ns)
500
400
300
200
100
0
0
20
40
60
80
RTMIN (k:)
100
120
140
D001
Figure 34. Minimum Time TMIN Over Setting Resistor RTMIN
The value of minimum duty cycle DMIN is determined by Equation 9.
(
)
DMIN = TMIN ´ FSW (osc ) ´ 10-4 %
(9)
Here, FSW(osc) is oscillator frequency in kHz, TMIN is the minimum pulse in ns and DMIN is in percent.
7.3.9 Burst Mode
If the converter is commanding a duty cycle lower than TMIN, then the controller will go into Burst Mode. The
controller will always deliver an even number of Power cycles to the Power transformer. The controller always
stops its bursts with an OUTB and an OUTC power delivery cycle. If the controller is still demanding a duty cycle
less than TMIN, then the controller goes into shut down mode. Then it waits until the converter is demanding a
duty cycle equal or higher than TMIN before the controller puts out TMIN or a PWM duty cycle as dictated by
COMP voltage pin.
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Feature Description (continued)
7.3.10 Switching Frequency Setting
Connecting an external resistor RT between the RT pin and VREF pins sets the fixed frequency operation and
configures the controller as a master providing synchronization output pulses at SYNC pin with 0.5 duty cycle
and frequency equal to the internal oscillator. To set the converter in slave mode, connect the external resistor
RT between the RT pin to GND and place an 825-kΩ resistor from the SS pin to GND in parallel with the SS_EN
capacitor. This configures the controller as a slave. The slave controller operates with 90° phase shift relative to
the master converter if their SYNC pins are tied together. The switching frequency of the converter is equal to the
frequency of output pulses. The following Equation 10 defines the nominal switching frequency of the converter
configured as a master (resistor RT between the RT pin and VREF). On the UCC28950 there is an internal clock
oscillator frequency which is twice as that of the controller's output frequency.
FSW (nom)
æ
ö
ç
÷
2.5 ´ 103
ç
÷ kHz
=
çæ
ö
RT
kW ÷
+ 1´
çç ç
÷ ÷÷
V
2.5
V
V
REF
è
øø
è
(10)
In this equation RT is in kΩ, VREF is in volts and FSW(nom) is in kHz. This is also an empirical approximation and
thus, there is no unit agreement. Assume for example, VREF = 5 V, RT = 65 kΩ. Then the switching frequency
FSW(nom) is going to be 92.6 kHz.
Equation 11 defines the nominal switching frequency of converter if the converter configured as a slave and the
resistor RT is connected between the RT pin and GND.
FSW (nom)
æ
ö
ç
÷
3
2.5 ´ 10
÷ kHz
=ç
ç æ RT
kW ö ÷
+ 1´
çç
÷
V ÷ø ø
è è 2.5 V
(11)
In this equation the RT is in kΩ, and FSW(nom) is in kHz. Notice that for VREF = 5 V, Equation 10 and Equation 11
yield the same results.
The plot in Figure 35 shows how FSW(nom) depends on the resistor RT value when the VREF = 5 V. As it is seen
from Equation 10 and Equation 11, the switching frequency FSW(nom) is set to the same value for either master or
slave configuration provided the same resistor value RT is used.
1000
FSW(nom) - Switching Frequency - kHz
900
800
700
600
500
400
300
200
100
0
5
15 25 35 45 55 65 75 85 95 105 115 125
RT - Resistor - kΩ
Figure 35. Converter Switching Frequency FSW(nom) Over resistor RT Value
24
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Feature Description (continued)
7.3.11 Slope Compensation (RSUM)
Slope compensation is needed to prevent a sub-harmonic oscillation in a controller operating in peak current
mode (PCM) control or during cycle-by-cycle current limit at duty cycles above 50% (some publications suggest it
may happen at D < 50%). Slope compensation in the UCC28950 adds an additional ramp signal to the CS signal
and is applied:
• To the PWM comparator in the case of peak current mode control.
• To the input of the cycle-by-cycle comparator.
At low duty cycles and light loads the slope compensation ramp reduces the noise sensitivity of Peak Current
Mode control.
Placing a resistor from the RSUM pin to ground allows the controller to operate in PCM control. Connecting a
resistor from RSUM to VREF switches the controller to voltage mode control (VMC) with the internal PWM ramp.
In VMC the resistor at RSUM provides CS signal slope compensation for operation in cycle-by-cycle current limit.
That is, in VMC, the slope compensation is applied only to the cycle-by-cycle comparator while in PCM the slope
compensation is applied to both the PWM and cycle-by-cycle current limit comparators. The operation logic of
the slope compensation circuit is shown in Figure 36.
COMP
4
+
+
Oscillator
VREF
VCM
0.85 V
CLK
PCM
Ramp
Generator
VMC
RAMP
Cycle-by-Cycle ILIM
RSUM
11
Two Direction
Current Sense
Ramp
Summing
CS_SLOPECOMP
+
CS 15
2V
+
-
Mode Select
GND
PCM
7
GND
Figure 36. The Operation Logic of Slope Compensation Circuit
Too much slope compensation reduces the benefits of PCM control. In the case of cycle-by-cycle current limit,
the average current limit becomes lower and this might reduce the start-up capability into large output
capacitances.
The optimum compensation ramp varies, depending on duty cycle, LOUT and LMAG. A good starting point in
selecting the amount of slope compensation is to set the slope compensation ramp to be half the inductor current
ramp downslope (inductor current ramp during the off time). The inductor current ramp downslope – as seen at
the CS pin input, and neglecting the effects of any filtering at the CS pin, will be:
m0 =
VOUT
RS
LOUT a1´ CTRAT
(12)
Where, VOUT is the converter’s output voltage of the converter, LOUT is the output inductor value, a1 is the
transformer turns ratio (Np/Ns), CTRAT is the current transformer ratio (Ip/Is, typically 100:1). Selection of LOUT, a1
and CTRAT are described elsewhere in this document. The total slope compensation is 0.5 m0. Part of this ramp
will be due to magnetizing current in the transformer, the rest is added by an appropriately chosen resistor from
RSUM to ground.
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Feature Description (continued)
The slope of the additional ramp, me, added to the CS signal by placing a resistor from RSUM to ground is
defined by Equation 13.
æ
öV
2.5
me = ç
÷
è 0.5 ´ RSUM ø ms
(13)
If the resistor from the RSUM pin is connected to the VREF pin, then the controller operates in voltage mode
control, still having the slope compensation ramp added to the CS signal used for cycle-by-cycle current limit. In
this case the slope is defined by Equation 14.
æ (V - 2.5 V) ö V
me = ç REF
÷
è 0.5 ´ RSUM ø ms
(14)
In Equation 13 and Equation 14, VREF is in volts, RSUM is in kΩ and me is in V/μs. These are empirically derived
equations without units agreement. As an example, substituting VREF = 5V and RSUM = 40 kΩ, yields the result
0.125 V/μs. The related plot of me as a function of RSUM is shown in Figure 37, Because VREF = 5V, the plots
generated from Equation 13 and Equation 14 coincide.
0.50
0.45
0.40
Slope - V/µs
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0
5
20
40
60
80
100 120 140 160 180 200
Rsum - Resistor - kΩ
Figure 37. Slope of the Added Ramp Over Resistor RSUM
NOTE
The recommended resistor range for RSUM is 10 kΩ to 1 MΩ.
26
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Feature Description (continued)
7.3.12 Dynamic SR ON/OFF Control (DCM Mode)
The voltage at the DCM pin provided by the resistor divider RDCMHI between VREF pin and DCM, and RDCM from
DCM pin to GND, sets the percentage of 2-V current limit threshold for the Current Sense pin, (CS). If the CS pin
voltage falls below the DCM pin threshold voltage, then the controller initiates the light load power saving mode,
and shuts down the synchronous rectifiers, OUTE and OUTF. If the CS pin voltage is higher than the DCM pin
threshold voltage, then the controller runs in CCM mode. Connecting the DCM pin to VREF makes the controller
run in DCM mode and shuts both Outputs OUTE and OUTF. Shorting the DCM pin to GND disables the DCM
feature and the controller runs in CCM mode under all conditions.
VREF
1
20 mA
RDCM(hi)
CS
DCM
R = 77 kW
PWM
DCM_COMP
15
2-Cycle
Counter
+
R = 77 kW
0 = DCM
1 = CCM
12
C = 6.5 pF
RDCM
C = 6.5 pF
Other Blocks
Figure 38. DCM Functional Block
Moving into
DCM Mode
0.8
VS(max)
0.6
Duty Cycle - %
VS(min)
TMIN Setting
0.4
0.2
Burst Mode
Area
0
0
1
2
3
4
5
6
7
8
9
10
Load Current - A
Figure 39. Duty Cycle Change Over Load Current Change
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Feature Description (continued)
A nominal 20-µA switched current source is used to create hysteresis. The current source is active only when the
system is in DCM Mode. Otherwise, it is inactive and does not affect the node voltage. Therefore, when in the
DCM region, the DCM threshold is the voltage divider plus ΔV explained in Equation 15. When in the CCM
region, the threshold is the voltage set by the resistor divider. When the CS pin reaches the threshold set on the
DCM pin, the system waits to see two consecutive falling edge PWM cycles before switching from CCM to DCM
and vice-versa. The magnitude of the hysteresis is a function of the external resistor divider impedance. The
hysteresis can be calculated using the following Equation 15:
DV = 2 ´ 10 -5
RDCMHI ´ RDCM
RDCMHI + RDCM
(15)
PWM
DCM Threshold
+ Hysteresis
CS
E
F
Figure 40. Moving from DCM to CCM Mode
PWM
DCM Threshold
+ Hysteresis
CS
E
F
Figure 41. Moving from CCM to DCM Mode
DCM must be used in order to prevent reverse current in the output inductor which could cause the synchronous
FETS to fail.
The controller must switch to DCM mode at a level where the output inductor current is positive. If the output
inductor current is negative when the controller switches to DCM mode then the synchronous FETs will see a
large VDS spike and may fail.
28
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Feature Description (continued)
7.3.13 Current Sensing (CS)
The signal from the current sense pin is used for cycle-by-cycle current limit, peak-current mode control, lightload efficiency management and setting the delay time for outputs OUTA, OUTB, OUTC, OUTD and delay time
for outputs OUTE, OUTF. Connect the current sense resistor RCS between CS and GND. Depending on layout,
to prevent a potential electrical noise interference, it is recommended to put a small R-C filter between the RCS
resistor and the CS pin.
7.3.14 Cycle-by-Cycle Current Limit Current Protection and Hiccup Mode
The cycle-by-cycle current limit provides peak current limiting on the primary side of the converter when the load
current exceeds its predetermined threshold. For peak current mode control, a certain leading edge blanking time
is needed to prevent the controller from false tripping due to switching noise. An internal 30-ns filter at the CS
input is provided. The total propagation delay TCS from CS pin to outputs is 100 ns. An external RC filter is still
needed if the power stage requires more blanking time. The 2.0-V ±3% cycle-by-cycle current limit threshold is
optimized for efficient current transformer based sensing. The duration when a converter operates at cycle-bycycle current limit depends on the value of soft-start capacitor and how severe the overcurrent condition is. This
is achieved by the internal discharge current IDS Equation 16 and Equation 17 at SS pin.
IDS(master ) = (-25 ´ (1 - D ) + 5 )mA
(16)
IDS(slave) = (-25 ´ (1 - D ))mA
(17)
The soft-start capacitor value also determines the so called hiccup mode off-time duration. The behavior of the
converter during different modes of operation, along with related soft start capacitor charge/discharge currents
are shown in Figure 42.
SS Pin (V)
SS Clamp Voltage
4.65
Pull Up Threshold
3.70
3.60
Soft Start
Cycle-by-Cycle ILIM
Normal
.
Operation
OFF Time Before Restart
25 mA
Soft Restart
Fast Pull Up
by 1 kW Switch IDS = (-25 x (1-D)+5) mA
Output Enable
Threshold
0.55
0.00
ISS=25 mA
IHCC = 2.5 mA
Output Pulses (D)
Figure 42. Timing Diagram of Soft-Start Voltage VSS
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Feature Description (continued)
The largest discharge current of 20 µA is when the duty cycle is close to zero. This current sets the shortest
operation time during the cycle-by-cycle current limit which is defined as:
TCL(on _ master ) =
TCL(on _ slave) =
CSS ´ (4.65 V - 3.7 V )
20 mA
(18)
CSS ´ (4.65 V - 3.7 V )
25 mA
(19)
Thus, if the soft-start capacitor CSS = 100 nF is selected, then the TCL(on) time will be 5 ms.
To calculate the hiccup off time TCL(off) before the restart, the following Equation 20 or Equation 21 needs to be
used:
TCL(off _ master ) =
TCL(off _ slave) =
C SS ´ (3.6 V - 0.55 V )
2.5 mA
(20)
CSS ´ (3.6 V - 0.55 V )
4.9 mA
(21)
With the same soft start capacitor value 100 nF, the off time before the restart is going to be 122 ms. Notice, that
if the overcurrent condition happens before the soft start capacitor voltage reaches the 3.7-V threshold during
start up, the controller limits the current but the soft start capacitor continues to be charged. As soon as the 3.7-V
threshold is reached, the soft-start voltage is quickly pulled up to the 4.65-V threshold by an internal 1-kΩ RDS(on)
switch and the cycle-by-cycle current limit duration timing starts by discharging the soft start capacitor.
Depending on specific design requirements, the user can override this default behavior by applying external
charge or discharge currents to the soft start capacitor. The whole cycle-by-cycle current limit and hiccup
operation is shown in Figure 42. In this example the cycle-by-cycle current limit lasts about 5 ms followed by 122
ms of off time.
Similarly to the overcurrent condition, the hiccup mode with the restart can be overridden by the user if a pullup
resistor is connected between the SS and VREF pins. If the pullup current provided by the resistor exceeds 2.5
µA, then the controller remains in the latch off mode. In this case, an external soft-start capacitor value should be
calculated with the additional pull-up current taken into account. The latch off mode can be reset externally if the
soft-start capacitor is forcibly discharged below 0.55 V or the VDD voltage is lowered below the UVLO threshold.
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Feature Description (continued)
7.3.15 Synchronization (SYNC)
The UCC28950 allows flexible configuration of converters operating in synchronized mode by connecting all
SYNC pins together and by configuration of the controllers as master and/or slaves. The controller configured as
master (resistor between RT and VREF) provides synchronization pulses at the SYNC pin with the frequency
equal to 2X the converter frequency FSW(nom) and 0.5 duty cycle. The controller configured as a slave (resistor
between RT and GND and 825-kΩ resistor between SS_EN pin to GND) does not generate the synchronization
pulses. The Slave controller synchronizes its own clock to the falling edge of the synchronization signal thus
operating 90° phase shifted versus the master converter’s frequency FSW(nom).
The output inductor in a full bridge converter sees a switching frequency which is twice that seen by the
transformer. In the case of the UCC28950 this means that the output inductor operates at 2 x FSW(nom). This
means that the 90° phase shift between master and slave controllers gives a 180° phase shift between the
currents in the output inductors and hence maximum ripple cancellation. For more information about
synchronizing more than two UCC28950 devices, see Synchronizing Three or More UCC28950 Phase-Shifted,
Full-Bridge Controllers, SLUA609.
If the synchronization feature is not used then the SYNC pin may be left floating, but connecting the SYNC pin to
GND via a 10-kΩ resistor will reduce noise pickup and switching frequency jitter.
• If any converter is configured as a slave, the SYNC frequency must be greater than or equal to 1.8 times the
converter frequency.
• Slave converter does not start until at least one synchronization pulse has been received.
• If any or all converters are configured as slaves, then each converter operates at its own frequency without
synchronization after receiving at least one synchronization pulse. Thus, If there is an interruption of
synchronization pulses at the slave converter, then the controller uses its own internal clock pulses to
maintain operation based on the RT value that is connected to GND in the slave converter.
• In master mode, SYNC pulses start after SS pin passes its enable threshold which is 0.55 V.
• Slave starts generating SS/EN voltage even though synchronization pulses have not been received.
• It is recommended that the SS on the master controller starts before the SS on the slave controller; therefore
SS/EN pin on master converter must reach its enable threshold voltage before SS/EN on the slave converter
starts for proper operation. On the same note, it’s recommended that TMIN resistors on both master and slave
are set at the same value.
CLK
SYNC_OUT
A
B
Figure 43. SYNC_OUT (Master Mode) Timing Diagram
SYNC_IN
CLK
A
B
Figure 44. SYNC_IN (Slave Mode) Timing Diagram
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Feature Description (continued)
7.3.16 Outputs (OUTA, OUTB, OUTC, OUTD, OUTE, OUTF)
• All MOSFET control outputs have 0.2-A drive capability.
• The control outputs are configured as P-MOS and N-MOS totem poles with typical RDS(on) 20 Ω and 10 Ω
accordingly.
• The control outputs are capable of charging 100-pF capacitor within 12 ns and discharge within 8 ns.
• The amplitude of output control pulses is equal to VDD.
• Control outputs are designed to be used with external gate MOSFET/IGBT drivers.
• The design is optimized to prevent the latch up of outputs and verified by extensive tests.
The UCC28950 device has outputs OUTA, OUTB driving the active leg, initiating the duty cycle leg of power
MOSFETs in a phase-shifted full bridge power stage, and outputs OUTC, OUTD driving the passive leg,
completing the duty cycle leg, as it is shown in the typical timing diagram in Figure 46. Outputs OUTE and OUTF
are optimized to drive the synchronous rectifier MOSFETs (Figure 48). These outputs have 200-mA peak-current
capabilities and are designed to drive relatively small capacitive loads like inputs of external MOSFET or IGBT
drivers. Recommended load capacitance should not exceed 100 pF. The amplitude of the output signal is equal
to the VDD voltage.
7.3.17 Supply Voltage (VDD)
Connect this pin to a bias supply in the range from 8 V to 17 V. Place high quality, low ESR and ESL, at least 1µF ceramic bypass capacitor CVDD from this pin to GND. It is recommended to use a 10-Ω resistor in series from
the bias supply to the VDD pin to form an RC filter with the CVDD capacitor.
7.3.18 Ground (GND)
All signals are referenced to this node. It is recommended to have a separate quiet analog plane connected in
one place to the power plane. The analog plane connects the components related to the pins VREF, EA+, EA-,
COMP, SS/EN, DELAB, DELCD, DELEF, TMIN, RT, RSUM. The power plane connects the components related
to the pins DCM, ADELEF, ADEL, CS, SYNC, OUTF, OUTE, OUTD, OUTC, OUTB, OUTA, and VDD. An
example of layout and ground planes connection is shown in Figure 45.
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Feature Description (continued)
UCC28950
CREF
R1
R2
CVDD
1
VREF
GND 24
2
EA+
VDD 23
VDD
C1
R3
VSENSE
3
EA-
OUTA 22
A
4
COMP
OUTB 21
B
CSS
5
SS/EN
OUTC 20
C
RAB
6
DELAB
OUTD 19
D
RCD
7
DELCD
OUTE 18
E
EF
8
DELEF
OUTF 17
F
RT(min)
9
TMIN
SYNC 16
SYNC
R5
R4
R6
C3
C2
ENABLE
Power Plane
Analog Plane R
10 RT
RT
CS 15
RA(hi)
RSUM
11 RSUM
ADEL 14
RA
RDCM(hi)
12 DCM
VREF
ADELEF 13
RAEF(hi)
Current Sense
R7
RCS
RDCM
RAEF
Figure 45. Layout Recommendation for Analog and Power Planes
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7.4 Device Functional Modes
The UCC28950 has a number of operational modes. These modes are described in detail in Feature Description.
• Current mode (1). The UCC28950 device will operate in current mode control if the RSUM pin is connected to
GND through a resistor (RSUM) . The resistor sets the amount of slope compensation.
• Voltage mode (1). The UCC28950 device will operate in voltage mode control if the RSUM pin is connected to
VREF through a resistor (RSUM). The resistor value is chosen to give the correct amount of slope
compensation for operation in current limit mode (cycle-by-cycle current limit).
• DCM mode. The UCC28950 device enters DCM mode if the signal at the CS pin falls below the level set by
the resistor at the DCM pin. The SR drives (OUTE and OUTF) are turned off and secondary rectification is
through the body diodes of the SRs.
• Burst mode. The UCC28950 device enters burst mode if the pulse width demanded by the feedback signal
falls below the width set by the resistor at the TMIN pin.
• Master mode. This is the default operation mode of the UCC28950 device and is the mode used if there is
only one UCC28950 device in the system. Connect the timing resistor (RT) from the RT pin to VREF. In a
system with more than one UCC28950, one will be configured as the master and the others as slaves (1).
• Slave mode. The slave controller will operate with a 90° phase shift relative to the Master (providing their
SYNC pins are tied together). Connect the timing resistor (RT) from the RT pin to GND and connect an 825
kΩ resistor from the SS/EN pin to GND (1).
• Synchronized mode. If a UC28950 is configured as a slave then its SYNC pin is used as an input. The slave
will synchronize its internal oscillator at 90° to the signal at its SYNC pin. App note slua609 discusses how
multiple Slave controllers may be synchronized to a single master oscillator.
• Hiccup mode. This mode provides overload protection to the power circuit. The UCC28950 device stops
switching after a certain time in current limit. It starts again (soft start) after a delay time. The user can control
the time spent in current limit before switching is stopped and the delay time before the soft start happens.
• Current-limit mode. The UCC28950 device will provide cycle-by-cycle current limiting if the signal at the CS
pin reaches 2V.
• Latch-off mode. Connect a resistor between the SS pin and VREF. The UCC28950 will then latch off if the
controller enters Current Limit mode. (1)
(1)
34
Current mode control and voltage mode control are mutually exclusive as are master and slave modes.
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The high efficiency of a phase-shifted full-bridge DC/DC converter using the UCC28950 is achieved by using
synchronous rectification, a control algorithm providing ZVS condition over the entire load current range, accurate
adaptive timing of the control signals between primary and secondary FETs and special operating modes at light
load. A simplified electrical diagram of this converter is shown in Figure 48. The controller device is located on
the secondary side of converter, although it could be located on primary side as well. The location on secondary
side allows easy power system level communication and better handling of some transient conditions that require
fast direct control of the synchronous rectifier MOSFETs. The power stage includes primary side MOSFETs, QA,
QB, QC, QD and secondary side synchronous rectifier MOSFETs, QE and QF. For example, for the 12-V output
converters in server power supplies use of the center-tapped rectifier scheme with L-C output filter is a popular
choice.
To maintain high efficiency at different output power conditions, the converter operates in synchronous
rectification mode at mid and high output power levels, transitioning to diode rectifier mode at light load and then
into burst mode as the output power becomes even lower. All these transitions are based on current sensing on
the primary side using a current sense transformer in this specific case.
The major waveforms of the phase-shifted converter during normal operation are shown in Figure 46. The upper
six waveforms in Figure 46 show the output drive signals of the controller. In normal mode, the outputs OUTE
and OUTF overlap during the part of the switching cycle when both rectifier MOSFETs are conducting and the
windings of the power transformer are shorted. Current, IPR, is the current flowing through the primary winding of
the power transformer. The bottom four waveforms show the drain-source voltages of rectifier MOSFETs, VDS_QE
and VDS_QF, the voltage at the output inductor, V LOUT, and the current through the output inductor, I LOUT. Proper
timing between the primary switches and synchronous rectifier MOSFETs is critical to achieve highest efficiency
and reliable operation in this mode. The controller device adjusts the turn OFF timing of the rectifier MOSFETs
as a function of load current to ensure minimum conduction time and reverse recovery losses of their internal
body diodes.
ZVS is an important feature of relatively high input voltage converters in reducing switching losses associated
with the internal parasitic capacitances of power switches and transformers. The controller ensures ZVS
conditions over the entire load current range by adjusting the delay time between the primary MOSFETs
switching in the same leg in accordance to the load variation. The controller also limits the minimum ON-time
pulse applied to the power transformer at light load, allowing the storage of sufficient energy in the inductive
components of the power stage for the ZVS transition.
As the load current reduces from full load down to the no-load condition, the controller selects the most efficient
power saving mode by moving from the normal operation mode to the discontinuous-current diode-rectification
mode and, eventually, at very light-load and at no-load condition, to the burst mode. These modes and related
output signals, OUTE, OUTF, driving the rectifier MOSFETs, are shown in Figure 47.
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Application Information (continued)
TSW(nom)
TABSET2
OUTA
TABSET1
OUTB
TCDSET2
TSW(osc)
OUTC
TCDSET1
OUTD
IPR
VOUTx(1-D) /D
VLOUT
VOUT
ILOUT
IOUT
Figure 46. Major Waveforms of Phase-Shifted Converter
36
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Application Information (continued)
OUTE
(CCM Mode)
OUTF
(CCM Mode)
OUTE
OUTE and OUTF are disabled if VCS < VDCM
OUTF
OUTE and OUTF are disabled if VCS < VDCM
Burst Mode at light load with TMIN maintaining ZVS
(The time scale is different versus above diagram)
Transformer
Winding
Magnetizing
Current
Figure 47. Major Waveforms During Transitions Between Different Operating Modes
It is necessary to prevent the reverse current flow through the synchronous rectifier MOSFETs and output
inductor at light load, during parallel operation and at some transient conditions. Such reverse current results in
circulating of some extra energy between the input voltage source and the load and, therefore, causes increased
losses and reduced efficiency. Another negative effect of such reverse current is the loss of ZVS condition. The
suggested control algorithm prevents reverse current flow, still maintaining most of the benefits of synchronous
rectification by switching off the drive signals of rectifier MOSFETs in a predetermined way. At some predetermined load current threshold, the controller disables outputs OUTE and OUTF by bringing them down to
zero.
Synchronous rectification using MOSFETs requires some electrical energy to drive the MOSFETs. There is a
condition below some light-load threshold when the MOSFET drive related losses exceed the saving provided by
the synchronous rectification. At such light load, it is best to disable the drive circuit and use the internal body
diodes of rectifier MOSFETs, or external diodes in parallel with the MOSFETs, for more efficient rectification. In
most practical cases, the drive circuit needs to be disabled close to DCM mode. This mode of operation is called
discontinuous-current diode-rectification mode.
At very light-load and no-load condition, the duty cycle, demanded by the closed-feedback-loop control circuit for
output voltage regulation, can be very low. This could lead to the loss of ZVS condition and increased switching
losses. To avoid the loss of ZVS, the control circuit limits the minimum ON-time pulse applied to the power
transformer using resistor from TMIN pin to GND. Therefore, the only way to maintain regulation at very light load
and at no-load condition is to skip some pulses. The controller skips pulses in a controllable manner to avoid
saturation of the power transformer. Such operation is called burst mode. In Burst Mode there are always an
even number of pulses applied to the power transformer before the skipping off time. Thus, the flux in the core of
the power transformer always starts from the same point during the start of every burst of pulses.
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8.2 Typical Application
A typical application for the UCC28950 device is a controller for a phase-shifted full-bridge converter that
converts a 390-VDC input to a regulated 12-V output using synchronous rectifiers to achieve high efficiency.
+
CT
VBIAS
VIN
CIN
VREF
CREF
R1
UCC28950
R2
1
VREF
GND 24
2
EA+
VDD 23
3
EA-
OUTA 22
A
4
COMP
OUTB 21
B
CSS
5
SS/EN
OUTC 20
C
-
R3
C1
RLF2
VDD
CVDD
VDD
VDD
R5
VSENSE
R6 R4
C2
QA
QC
DB
A
C
C3
RAB
6
DELAB
OUTD 19
D
RCD
7
DELCD
OUTE 18
E
REF
8
DELEF
OUTF 17
F
RTMIN
9
TMIN
SYNC 16
SYNC
T1
LS
DC
ENABLE
VDD
VDD
LOUT
QB
B
QD
D
VOUT
RT
VREF
10 RT
CS 15
RAHI
RSUM
11 RSUM
+
VREF
UCC27324
ADEL 14
UCC27324
QE
RDCMHI
QF
E
VREF
12 DCM
F
ADELEF 13
COUT
RAEFHI
RLF1
DA
CLF
RDCM
RCS
R7
RAEF
-
RA
VSENSE
Figure 48. UCC28950 Typical Application
8.2.1 Design Requirements
Table 1 lists the requirements for this application.
Table 1. UCC28950 Typical Application Design Requirements
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
370
390
410
V
2
A
12.6
V
INPUT CHARACTERISTICS
VIN
DC input voltage range
IIN(max)
Maximum input current
VIN= 370 VDC to 410 VDC
OUTPUT CHARACTERISTICS
VOUT
Output voltage
VIN= 370 VDC to 410 VDC
IOUT
Output current
VIN= 370 VDC to 410 VDC
Output voltage transient
90% load step
Continuous output power
VIN= 370 VDC to 410 VDC
600
W
Load regulation
VIN = 370 VDC to 410 VDC, IOUT= 5 A to 50 A
140
mV
Line regulation
VIN = 370 VDC to 410 VDC, IOUT= 5 A to 50 A
140
mV
Output ripple voltage
VIN = 370 VDC to 410 VDC, IOUT= 5 A to 50 A
200
mV
POUT
11.4
12
50
600
A
mV
SYSTEM
FSW
Switching Frequency
Full-load efficiency
38
100
VIN= 370 VDC to 410 VDC, POUT= 500 W
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94%
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8.2.2 Detailed Design Procedure
In high-power server applications to meet high-efficiency and green standards some power-supply designers
have found it easier to use a phase-shifted, full-bridge converter. This is because the phase-shifted, full-bridge
converter can obtain zero-voltage switching on the primary side of the converter, reducing switching losses, and
EMI and increasing overall efficiency.
This is a review of the design of a 600-W, phase-shifted, full-bridge converter for one of these power systems
using TI's UCC28950 device, which is based on typical values. In a production design, the values may need to
be modified for worst-case conditions. TI has provided a MathCAD Design Tool and an Excel Design Tool to
support the system designer. Both tools can be accessed in the Tools and Software tab of the UCC28950
product folder on TI.com, or can be downloaded through the following links: MathCAD Design Tool, Excel Design
Tool.
NOTE
FSW refers to the switching frequency applied to the power transformer. The output
inductor experiences a switching frequency which is 2 x FSW .
8.2.2.1 Power Loss Budget
To meet the efficiency goal a power loss budget needs to be set.
æ 1- h ö
PBUDGET = POUT ´ ç
÷ » 45.2 W
è h ø
(22)
8.2.2.2 Preliminary Transformer Calculations (T1)
Transformer turns ratio (a1):
a1 =
NP
NS
(23)
Estimated FET voltage drop (VRDSON):
VRDSON = 0.3 V
(24)
Select transformer turns based on 70% duty cycle (DMAX) at minimum specified input voltage. This will give some
room for dropout if a PFC front end is used.
a1 =
a1 =
NP
NS
(25)
(VINMIN - 2 ´ VRDSON )´ DMAX
VOUT + VRDSON
» 21
(26)
Turns ratio rounded to the nearest whole turn.
a1 = 21
(27)
Calculated typical duty cycle (DTYP) based on average input voltage.
DTYP =
(VOUT + VRDSON )´ a1 » 0.66
(VIN - 2 ´ VRDSON )
(28)
Output inductor peak-to-peak ripple current is set to 20% of the output current.
DILOUT =
POUT ´ 0.2
= 10 A
VOUT
(29)
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Care must be taken in selecting the correct amount of magnetizing inductance (LMAG). The following equations
calculate the minimum magnetizing inductance of the primary of the transformer (T1) to ensure the converter
operates in current-mode control. As LMAG reduces, the increasing magnetizing current becomes an increasing
proportion of the signal at the CS pin. If the magnetizing current increases enough it can swamp out the current
sense signal across RCS and the converter will operate increasingly as if it were in voltage mode control rather
than current mode.
LMAG ³
VIN ´ (1 - DTYP )
» 2.78mH
DILOUT ´ 0.5
´ 2 ´ FSW
a1
(30)
Figure 49 shows T1 primary current (IPRIMARY) and synchronous rectifiers QE (IQE) and QF (IQF) currents with
respect to the synchronous rectifier gate drive currents. Note that IQE and IQF are the same as the secondary
winding currents of T1. Variable D is the duty cycle of the converter.
IPP
IMP2 » IPP - DILOUT / (2 ´ a1)
IMP2
IPRIMARY
IMP
0A
D
On
QEg
Off
On
QFg
Off
IQE
0A
IQF
IPS
IMS2
IMS
IMS2 » IPS - DILOUT /2
0A
DILOUT /2
Figure 49. T1 Primary and QE and QF FET Currents
40
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Calculate T1 secondary RMS current (ISRMS):
IPS =
POUT DILOUT
+
» 55 A
VOUT
2
(31)
POUT DILOUT
» 45 A
VOUT
2
(32)
ΔI
- LOUT » 50 A
2
(33)
IMS =
IMS2 = IPS
Secondary RMS current (ISRMS1) when energy is being delivered to the secondary:
2
IPS - IMS ) ù
(
æ DMAX ö é
ú » 29.6 A
ISRMS1 = ç
÷ êIPS ´ IMS +
3
è 2 ø êë
úû
(34)
Secondary RMS current (ISRMS2) when current is circulating through the transformer when QE and QF are both
on.
(I - I )
æ 1 - DMAX ö é
êIPS ´ IMS2 + PS MS2
= ç
÷
2
3
è
ø êë
2
ISRMS2
ù
ú » 20.3 A
úû
(35)
Secondary RMS current (ISRMS3) caused by the negative current in the opposing winding during freewheeling
period, please refer to Figure 49.
ISRMS3 =
DILOUT æ 1 - DMAX ö
ç 2 ´ 3 ÷ » 1.1A
2
è
ø
(36)
Total secondary RMS current (ISRMS):
ISRMS = ISRMS12 + ISRMS22 + ISRMS3 2 » 36.0 A
(37)
Calculate T1 Primary RMS Current (IPRMS):
D ILMAG =
VINMIN ´ DMAX
» 0.47 A
LMAG ´ 2 ´ FSW
(38)
æ P
DI
IPP = ç OUT + LOUT
2
è VOUT ´ h
ö1
÷ + DILMAG » 3.3 A
ø a1
(39)
æ P
DI
IMP = ç OUT - LOUT
2
è VOUT ´ h
ö1
÷ + DILMAG » 2.8A
ø a1
(40)
é
(IPP - IMP )2 ùú
ê
IPRMS1 = (DMAX ) IPP ´ IMP +
» 2.5 A
ê
ú
3
ë
û
(41)
æ DI
ö1
IMP2 = IPP - ç LOUT ÷ » 3.0 A
è 2 ø a1
(42)
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T1 Primary RMS (IPRMS1) current when energy is being delivered to the secondary.
é
(IPP - IMP )2 ùú
ê
IPRMS1 = (DMAX ) IPP ´ IMP +
» 2.5 A
ê
ú
3
ë
û
(43)
T1 Primary RMS (IPRMS2) current when the converter is free wheeling.
IPRMS2
2
é
IPP - IMP2 ) ù
(
ú » 1.7 A
= (1 - DMAX )êIPP ´ IMP2 +
3
êë
úû
(44)
Total T1 primary RMS current (IPRMS):
IPRMS = IPRMS12 + IPRMS22 » 3.1A
(45)
For this design a Vitec™ transformer was selected part number 75PR8107 that had the following specifications.
a1= 21
LMAG = 2.8mH
(46)
(47)
Measure leakage inductance on the Primary:
LLK = 4 mH
(48)
Transformer Primary DC resistance:
DCRP = 0.215 W
(49)
Transformer Secondary DC resistance:
DCRS = 0.58 W
(50)
Estimated transformer core losses (PT1) are twice the copper loss.
NOTE
This is just an estimate and the total losses may vary based on magnetic design.
(
)
PT1 » 2 ´ IPRMS 2 ´ DCRP + 2 ´ ISRMS 2 ´ DCRS » 7.0 W
(51)
Calculate remaining power budget:
PBUDGET = PBUDGET - PT1 » 38.1W
42
(52)
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8.2.2.3 QA, QB, QC, QD FET Selection
In this design to meet efficiency and voltage requirements 20 A, 650 V, CoolMOS FETs from Infineon are chosen
for QA..QD.
FET drain to source on resistance:
Rds(on)QA = 0.220 W
(53)
FET Specified COSS:
COSS _ QA _ SPEC = 780pF
(54)
Voltage across drain-to-source (VdsQA) where COSS was measured, data sheet parameter:
VdsQA = 25 V
(55)
Calculate average Coss [2]:
COSS _ QA _ AVG = COSS _ QA _ SPEC
VdsQA
» 193pF
VINMAX
(56)
QA FET gate charge:
QA g = 15nC
(57)
Voltage applied to FET gate to activate FET:
Vg = 12 V
(58)
Calculate QA losses (PQA) based on Rds(on)QA and gate charge (QAg):
PQA
IPRMS2 u RDS(on)QA
2 u QAg u Vg u fSW | 2.1W
(59)
Recalculate power budget:
PBUDGET = PBUDGET - 4 ´ PQA » 29.7 W
(60)
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8.2.2.4 Selecting LS
Calculating the value of the shim inductor (LS) is based on the amount of energy required to achieve zero voltage
switching. This inductor needs to able to deplete the energy from the parasitic capacitance at the switch node.
The following equation selects LS to achieve ZVS at 100% load down to 50% load based on the primary FET’s
average total COSS at the switch node.
NOTE
The actual parasitic capacitance at the switched node may differ from the estimate and LS
may have to be adjusted accordingly.
VINMAX 2
- LLK » 26 mH
LS ³ (2 ´ COSS _ QA _ AVG )
2
æ IPP DILOUT ö
ç 2 - 2 ´ a1 ÷
è
ø
(61)
For this design a 26-µH Vitec inductor was chosen for LS, part number 60PR964. The shim inductor had the
following specifications.
LS = 26 mH
(62)
LS DC Resistance:
DCRLS = 27mW
(63)
Estimate LS power loss (PLS) and readjust remaining power budget:
PLS = 2 ´ IPRMS 2 ´ DCRLS » 0.5 W
(64)
PBUDGET = PBUDGET - PLS » 29.2 W
(65)
8.2.2.5 Selecting Diodes DB and DC
There is a potential for high voltage ringing on the secondary rectifiers, caused by the difference in current
between the transformer and the shim inductor when the transformer comes out of freewheeling. Diodes DB and
DC provide a path for this current and prevent any ringing by clamping the transformer primary to the primary
side power rails. Normally these diodes do not dissipate much power but they should be sized to carry the full
primary current. The worse case power dissipated in these diodes is:
P = 0.5 ´ LS ´ I2PRMS ´ FSW
(66)
The diodes should be ultra-fast types and rated for the input voltage of the converter – VIN (410 VDC in this
case).
A MURS360 part is suitable at this power level.
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8.2.2.6 Output Inductor Selection (LOUT)
Inductor LOUT is designed for 20% inductor ripple current (∆ILOUT):
DILOUT =
LOUT
POUT ´ 0.2 600 W ´ 0.2
=
» 10 A
VOUT
12 V
(67)
VOUT u (1 DTYP )
| 2 PH
'ILOUT u 2 u fSW
(68)
Calculate output inductor RMS current (ILOUT_RMS):
2
ILOUT _ RMS
2
æ P ö æ DI
ö
= ç OUT ÷ + ç LOUT ÷ = 50.3 A
è VOUT ø è 3 ø
(69)
A 2-µH inductor from Vitec Electronics Corporation, part number 75PR8108, is suitable for this design. The
inductor has the following specifications.
LOUT = 2 mH
(70)
Output inductor DC resistance:
DCRLOUT = 750 mW
(71)
Estimate output inductor losses (PLOUT) and recalculate power budget. Note PLOUT is an estimate of inductor
losses that is twice the copper loss. Note this may vary based on magnetic manufactures. It is advisable to
double check the magnetic loss with the magnetic manufacture.
PLOUT = 2 ´ ILOUT _ RMS 2 ´ DCRLOUT » 3.8 W
(72)
PBUDGET = PBUDGET - PLOUT » 25.4 W
(73)
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8.2.2.7 Output Capacitance (COUT)
The output capacitor is selected based on holdup and transient (VTRAN) load requirements.
Time it takes LOUT to change 90% of its full load current:
tHU
LOUT ´ POUT ´ 0.9
VOUT
=
= 7.5 ms
VOUT
(74)
During load transients most of the current will immediately go through the capacitors equivalent series resistance
(ESRCOUT). The following equations are used to select ESRCOUT and COUT based on a 90% load step in current.
The ESR is selected for 90% of the allowable transient voltage (VTRAN), while the output capacitance (COUT) is
selected for 10% of VTRAN.
ESRCOUT £
COUT
VTRAN ´ 0.9
= 12mW
POUT ´ 0.9
VOUT
(75)
POUT ´ 0.9 ´ tHU
VOUT
³
» 5.6mF
VTRAN ´ 0.1
(76)
Before selecting the output capacitor, the output capacitor RMS current (ICOUT_RMS) must be calculated.
ICOUT _ RMS =
DILOUT
3
» 5.8 A
(77)
To meet the design requirements five 1500-µF, aluminum electrolytic capacitors are chosen for the design from
United Chemi-Con™, part number EKY-160ELL152MJ30S. These capacitors have an ESR of 31 mΩ.
Number of output capacitors:
n=5
(78)
Total output capacitance:
COUT = 1500 mF ´ n » 7500 mF
(79)
Effective output capacitance ESR:
ESRCOUT =
31mW
= 6.2mW
n
(80)
Calculate output capacitor loss (PCOUT):
PCOUT = ICOUT _ RMS 2 ´ ESRCOUT » 0.21W
(81)
Recalculate remaining Power Budget:
PBUDGET = PBUDGET - PCOUT » 25.2 W
46
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8.2.2.8 Select FETs QE and QF
Selecting FETs for a design is an iterative process. To meet the power requirements of this design, we select 75V, 120-A FETs, from Fairchild, part number FDP032N08. These FETs have the following characteristics.
QEg = 152nC
(83)
Rds(on)QE = 3.2mW
(84)
Calculate average FET COSS (COSS_QE_AVG) based on the data sheet parameters for COSS (COSS_SPEC), and drain
to source voltage where COSS_SPEC was measured (Vds_spec), and the maximum drain to source voltage in the
design (VdsQE) that will be applied to the FET in the application.
Voltage across FET QE and QF when they are off:
VdsQE =
VINMAX
» 19.5 V
a1
(85)
Voltage where FET COSS is specified and tested in the FET data sheet:
Vds _ spec = 25 V
(86)
Specified output capacitance from FET data sheet:
COSS _ SPEC = 1810pF
(87)
Average QE and QF COSS [2]:
COSS _ QE _ AVG = COSS _ SPEC
VdsQE
» 1.6nF
Vds _ spec
(88)
QE and QF RMS current:
IQE _ RMS = ISRMS = 36.0 A
(89)
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To estimate FET switching loss the Vg vs. Qg curve from the FET data sheet needs to be studied. First the gate
charge at the beginning of the miller plateau needs to be determined (QEMILLER_MIN) and the gate charge at the
end of the miller plateau (QEMILLER_MAX) for the given VDS.
Maximum gate charge at the end of the miller plateau:
QEMILLER _ MAX » 100nC
(90)
Minimum gate charge at the beginning of the miller plateau:
QEMILLER _ MIN » 52nC
(91)
NOTE
The FETs in this design are driven with a UCC27324 Gate Driver IC, setup to drive 4-A
(IP) of gate drive current.
IP » 4 A
(92)
Estimated FET Vds rise and fall time:
tr » t f =
100nC - 52nC 48nC
=
» 24ns
IP
4A
2
2
(93)
Estimate QE and QF FET Losses (PQE):
PQE
IQE _RMS2 u Rds(on)QE
POUT
u VdsQE tr
VOUT
t f fSW
2 u COSS _ QE _ AVG u VdsQE2fSW
2 u QgQE u VgQEfSW
(94)
PQE » 9.3 W
(95)
Recalculate the power budget.
PBUDGET = PBUDGET - 2 ´ PQE » 6.5 W
48
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8.2.2.9 Input Capacitance (CIN)
The input voltage in this design is 390 VDC, which is generally fed by the output of a PFC boost pre-regulator.
The input capacitance is generally selected based on holdup and ripple requirements.
NOTE
The delay time needed to achieve ZVS can act as a duty cycle clamp (DCLAMP).
Calculate tank frequency:
fR =
1
2p LS ´ (2 ´ COSS _ QA _ AVG )
(97)
Estimated delay time:
tDELAY =
2
» 314ns
f R ´4
(98)
Effective duty cycle clamp (DCLAMP):
DCLAMP
§ 1
¨
© 2 u fSW
·
tDELAY ¸ u 2 u fSW
¹
94%
(99)
VDROP is the minimum input voltage where the converter can still maintain output regulation. The converter’s input
voltage would only drop down this low during a brownout or line-drop condition if this converter was following a
PFC pre-regulator.
æ 2 ´ DCLAMP ´ VRDSON + a1´ (VOUT + VRDSON ) ö
VDROP = ç
÷ = 276.2 V
DCLAMP
è
ø
(100)
CIN was calculated based on one line cycle of holdup:
2 ´ POUT ´
CIN ³
(V
2
IN
1
60Hz
- VDROP 2
)
» 364 mF
(101)
Calculate high frequency input capacitor RMS current (ICINRMS).
2
ICINRMS = I
2
PRMS1
æ POUT ö
-ç
÷ = 1.8 A
è VINMIN ´ a1 ø
(102)
To meet the input capacitance and RMS current requirements for this design a 330-µF capacitor was chosen
from Panasonic part number EETHC2W331EA.
CIN = 330 mF
(103)
This capacitor has a high frequency (ESRCIN) of 150 mΩ, measured with an impedance analyzer at 200 kHz.
ESRCIN = 0.150 W
(104)
Estimate CIN power dissipation (PCIN):
PCIN = ICINRMS 2 ´ ESRCIN = 0.5 W
(105)
Recalculate remaining power budget:
PBUDGET = PBUDGET - PCIN » 6.0 W
(106)
There is roughly 6.0 W left in the power budget left for the current sensing network, and biasing the control
device and all resistors supporting the control device.
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8.2.2.10 Current Sense Network (CT, RCS, R7, DA)
The CT chosen for this design has a turns ratio (CTRAT) of 100:1
CTRAT =
IP
= 100
IS
(107)
Calculate nominal peak current (IP1) at VINMIN:
Peak primary current:
æ P
DI
IP1 = ç OUT + LOUT
2
è VOUT ´ h
ö1
VINMIN ´ DMAX
» 3.3 A
÷ +
ø a1 LMAG ´ 2 ´ FSW
(108)
The CS pin voltage where peak current limit will trip.
VP = 2 V
(109)
Calculate current sense resistor (RCS) and leave 300mV for slope compensation. Include a 1.1 factor for margin:
RCS =
VP - 0.3 V
» 47 W
IP1
´ 1.1
CTRAT
(110)
Select a standard resistor for RCS:
RCS = 47 W
(111)
Estimate power loss for RCS:
2
PRCS
æI
ö
= ç PRMS1 ÷ ´ RCS » 0.03 W
è CTRAT ø
(112)
Calculate maximum reverse voltage (VDA) on DA:
VDA = VP
DCLAMP
» 29.8 V
1 - DCLAMP
(113)
Estimate DA power loss (PDA):
PDA =
POUT ´ 0.6 V
» 0.01W
VINMIN ´ h ´ CTRAT
(114)
Calculate reset resistor R7:
Resistor R7 is used to reset the current sense transformer CT.
50
R7 = 100 ´ RCS = 4.7kW
(115)
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Resistor RLF1 and capacitor CLF form a low pass filter for the current sense signal (Pin 15). For this design we
chose the following values. This filter has a low frequency pole (fLFP) at 482 kHz. This should work for most
applications but may be adjusted to suit individual layouts and EMI present in the design.
RLF1 = 1kW
(116)
CLF = 330pF
(117)
fLFP =
1
= 482kHz
2pf ´ RLF1 ´ CLF
(118)
The UCC28950 VREF output (Pin 1) needs a high frequency bypass capacitor to filter out high frequency noise.
This pin needs at least 1 µF of high frequency bypass capacitance (CREF).
CREF = 1 mF
(119)
The voltage amplifier reference voltage (Pin 2, EA +) can be set with a voltage divider (R1, R2), for this design
example, the error amplifier reference voltage (V1) will be set to 2.5 V. Select a standard resistor value for R1
and then calculate resistor value R2.
UCC28950 reference voltage:
VREF = 5 V
(120)
Set voltage amplifier reference voltage:
V1 = 2.5 V
R1 = 2.37kW
R1´ (VREF - V1)
R2 =
= 2.37kW
V1
(121)
(122)
(123)
Voltage divider formed by resistor R3 and R4 are chosen to set the DC output voltage (VOUT) at Pin 3 (EA-).
Select a standard resistor for R3:
R3 = 2.37kW
(124)
Calculate R4:
R4 =
R3 ´ (VOUT - V1)
V1
» 9kW
(125)
Then choose a standard resistor for R4:
R4 =
R3 ´ (VOUT - V1)
V1
» 9.09kW
(126)
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8.2.2.10.1 Voltage Loop Compensation Recommendation
For best results in the voltage loop, TI recommends using a Type 2 or Type 3 compensation network (Figure 50).
A Type 2 compensation network does not require passive components CZ2 and RZ2. Type 1 compensation is not
versatile enough for a phase shifted full bridge. When evaluating the COMP for best results, TI recommends
placing a 1-kΩ resistor between the socpe probe and the COMP pin of the UCC28950.
VOUT
VREF
CZ2
EA+
RI
+
EA1 k:
RD
RZ2
R
CZ1
RZ1
CP1
R
When evaluating COMP, for best results put
a 1-k: resistor between COMP and probe.
Figure 50. Type 3 Compensation Evaluation
Compensating the feedback loop can be accomplished by properly selecting the feedback components (R5, C1
and C2). These components are placed as close as possible to pin 3 and 4 of the UCC28950. A Type 2
compensation network is designed in this example.
Calculate load impedance at 10% load (RLOAD):
RLOAD =
VOUT 2
= 2.4 W
POUT ´ 0.1
(127)
Approximation of control to output transfer function (GCO(f)) as a function of frequency:
GCO (f ) »
æ 1 + 2pj ´ f ´ ESRCOUT ´ COUT
R
DVOUT
= a1´ CTRAT ´ LOAD ´ ç
RCS è 1 + 2pj ´ f ´ RLOAD ´ COUT
DVC
ö
÷´
ø
1
æ S(f ) ö
S(f )
1+
+ç
÷
2p ´ fPP è 2p ´ fPP ø
2
(128)
Double pole frequency of GCO(f):
fPP »
FSW
= 50kHz
2
(129)
Angular velocity:
S(f ) = 2p ´ j ´ f
52
(130)
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Compensate the voltage loop with type 2 feedback network. The following transfer function is the compensation
gain as a function of frequency (GC(f)).
GC (f ) =
DVC
=
DVOUT
2pj ´ f ´ R5 ´ C2 + 1
æ 2pj ´ f ´ C2 ´ C1´ R5 ö
+ 1÷
2pj ´ f ´ (C2 + C1)R4 ç
C2 + C1
è
ø
(131)
th
Calculate voltage loop feedback resistor (R5) based on crossing the voltage loop (fC) over at a 10 of the double
pole frequency (fPP).
fPP
= 5kHz
10
R4
» 27.9kW
R5 =
æ fPP ö
GCO ç
÷
è 10 ø
fC =
(132)
(133)
Select a standard resistor for R5.
R5 » 27.4kW
(134)
Calculate the feedback capacitor (C2) to give added phase at crossover.
C2 =
1
f
2 ´ p ´ R5 ´ C
5
» 5.8nF
(135)
Select a standard capacitance value for the design.
C2 = 5.6nF
(136)
Put a pole at two times fC.
C1 =
1
» 580pF
2 ´ p ´ R5 ´ fC ´ 2
(137)
Select a standard capacitance value for the design.
C1 = 560pF
(138)
Loop gain as a function of frequency (TV(f)) in dB.
TV dB(f ) = 20log (GC (f ) ´ GCO (f ) )
(139)
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80
180
60
135
40
90
20
45
0
0
-20
-45
-40
-90
TvdB(f)
-60
Phase in Degrees
Gain in dB
Plot theoretical loop gain and phase to graphically check for loop stability (Figure 51). The theoretical loop gain
crosses over at roughly 3.7 kHz with a phase margin of greater than 90 degrees.
-135
&Tv(f)
-80
100
1000
-180
100000
10000
Frequency in Hz
Figure 51. Loop Gain and Phase vs Frequency
NOTE
TI recommends checking your loop stability of your final design with transient testing
and/or a network analyzer and adjust the compensation (GC(f)) feedback as necessary.
LMAG ³
VIN ´ (1 - DTYP )
» 2.78mH
DILOUT ´ 0.5
´ 2 ´ FSW
a1
where
•
54
Loop Gain (TVdB(f)), Loop Phase (ΦTV(f))
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To limit over shoot during power up the UCC28950 has a soft-start function (SS, Pin 5) which in this application
was set for a soft start time of 15 ms (tSS).
t ss = 15ms
CSS =
(141)
t SS ´ 25 mA
» 123nF
V1 + 0.55
(142)
Select a standard capacitor for the design.
CSS = 150nF
(143)
This application note presents a fixed delay approach to achieving ZVS from 100% load down to 50% load.
Adaptive delays can be generated by connecting the ADEL and ADELEF pins to the CS pin as shown in
Figure 52.
RAHI
UCC28950
CS 15
ADEL 14
RAEFHI
ADELEF 13
RA
RAEF
Figure 52. UCC28950 Adaptive Delays
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When the converter is operating below 50% load the converter will be operating in valley switching. In order to
achieve zero voltage switching on switch node of QBd, the turn-on (tABSET) delays of FETs QA and QB need to
be initially set based on the interaction of LS and the theoretical switch node capacitance. The following
equations are used to set tABSET initially.
Equate shim inductance to two times COSS capacitance:
2p ´ fRLS =
1
2p ´ fR ´ (2 ´ COSS _ QA _ AVG )
(144)
Calculate tank frequency:
fR =
1
2p LS ´ (2 ´ COSS _ QA _ AVG )
(145)
Set initial tABSET delay time and adjust as necessary.
NOTE
The 2.25 factor of the tABSET equation was derived from empirical test data and may vary
based on individual design differences.
t ABSET =
2.25
» 346ns
f R ´4
(146)
The resistor divider formed by RA and RAHI programs the tABSET, tCDSET delay range of the UCC28950. Select a
standard resistor value for RAHI.
NOTE
tABSET can be programmed between 30 ns to 1000 ns.
R AHI = 8.25kW
(147)
The voltage at the ADEL input of the UCC28950 (VADEL) needs to be set with RA based on the following
conditions.
If tABSET > 155 ns set VADEL = 0.2 V, tABSET can be programmed between 155 ns and 1000 ns:
If tABSET ≤ 155 ns set VADEL = 1.8 V, tABSET can be programmed between 29 ns and 155 ns:
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Based on VADEL selection, calculate RA:
RA =
R AHI ´ VADEL
» 344 W
5V - VADEL
(148)
Select the closest standard resistor value for RA:
R A = 348 W
(149)
Recalculate VADEL based on resistor divider selection:
VADEL =
5 V ´ RA
= 0.202 V
R AHI + R A
(150)
Resistor RAB programs tABSET:
R AB =
TABSET
´ (0.26 + CS ´ K A ´ 1.3 ) » 30.6kW
5
(151)
Select a standard resistor value for the design:
R AB = 30.1kW
(152)
NOTE
Once you have a prototype up and running it is recommended you fine tune tABSET at light
load to the peak and valley of the resonance between LS and the switch node
capacitance. In this design the delay was set at 10% load. Please refer to Figure 53.
Set t
ABSET
at resonant tank Peak and Valley
t ABSET = t 1 - t 0
t ABSET = t 4 - t 3
QB d
QA g
Miller Plateau
tMILLER = t
QB
2
- t1
Miller Plateau
g
t MILLER = t 5 - t 4
t0 t1 t2
t3 t4 t5
Figure 53. tABSET to Achieve Valley Switching at Light Loads
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The QC and QD turnon delays (tCDSET) should be initially set for the same delay as the QA and QB turn on
delays (Pin 6). The following equations program the QC and QD turn-on delays (tCDSET) by properly selecting
resistor RDELCD (Pin 7).
t ABSET = t CDSET
(153)
Resistor RCD programs tCDSET:
RCD =
TCDSET
´ (0.26 + CS ´ K A ´ 1.3 ) » 30.6kW
5
(154)
Select a standard resistor for the design:
RCD = 30.1kW
(155)
NOTE
Once you have a prototype up and running it is recommended to fine tune tCDSET at light
load. In this design the CD node was set to valley switch at roughly 10% load. Please
refer to Figure 54. Obtaining ZVS at lighter loads with switch node QDd is easier due to
the reflected output current present in the primary of the transformer at FET QD and QC
turnoff/on. This is because there was more peak current available to energize LS before
this transition, compared to the QA and QB turnoff/on.
Set t
t CDSET
QD
QC
=t
1
CDSET
t CDSET
- t0
=t
4
- t3
d
g
Miller Plateau
t MILLER
QD
at resonant tank Peak and Valley
=t
2
-t1
Miller Plateau
g
t MILLER
t 0 t1 t 2
=t
5
-t4
t 3 t4 t 5
Figure 54. tCDSET to Achieve Valley Switching at Light Loads
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There is a programmable delay for the turnoff of FET QF after FET QA turnoff (tAFSET) and the turnoff of FET QE
after FET QB turnoff (tBESET). A good place to set these delays is 50% of tABSET. This will ensure that the
appropriate synchronous rectifier turns off before the AB ZVS transition. If this delay is too large it will cause
OUTE and OUTF not to overlap correctly and it will create excess body diode conduction on FETs QE and QF.
t AFSET = tBESET = t ABSET ´ 0.5
(156)
The resistor divider formed by RAEF and RAEFHI programs the tAFSET and tBESET delay range of the UCC28950.
Select a standard resistor value for RAEFHI.
NOTE
tEFSET and tBESET can be programmed between 32 ns to 1100 ns.
R AEFHI = 8.25kW
(157)
The voltage at the ADELEF pin of the UCC28950 (VADELEF) needs to be set with RAEF based on the following
conditions.
If tAFSET < 170 ns set VADEL = 0.2 V, tABSET can be programmed between 32 ns and 170 ns:
If tABSET > or = 170 ns set VADEL = 1.7 V, tABSET can be programmed between 170 ns and 1100 ns:
Based on VADELEF selection, calculate RAEF:
R AEF =
R AEFHI ´ VADELEF
» 4.25kW
5 V - VADELEF
(158)
Select the closest standard resistor value for RAEF:
R AEF = 4.22kW
(159)
Recalculate VADELEF based on resistor divider selection:
VADELEF =
5 V ´ R AEF
= 1.692 V
R AEFHI + R AEF
(160)
The following equation was used to program tAFSET and tBESET by properly selecting resistor REF.
REF
t AFSET ´ 0.5 - 4ns ) (2.65 V - VADELEF ´ 1.32 )´ 103
(
=
´
´
ns
5
1
» 14.1kW
1A
(161)
A standard resistor was chosen for the design.
REF = 14kW
(162)
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Resistor RTMIN programs the minimum on time (tMIN) that the UCC28950 (Pin 9) can demand before entering
burst mode. If the UCC28950 controller tries to demand a duty cycle on time of less than tMIN the power supply
will go into burst mode operation. For this design we set the minimum on time to 75 ns.
tMIN = 75ns
(163)
The minimum on time is set by selecting RTMIN with the following equation.
RTMIN =
tMIN
» 12.7kW
5.92
(164)
A standard resistor value is then chosen for the design.
RTMIN = 13kW
(165)
A resistor from the RT pin to ground sets the converter switching frequency.
æ
ö
6 WHz
ç 2.5 ´ 10 V
W÷
- ÷ ´ (VREF - 2.5 V )´ 2.5 ´ 103 » 60kW
RT = ç
FSW
V÷
ç
2
è
ø
(166)
Select a standard resistor for the design.
RT = 61.9kW
(167)
The UCC28950 provides slope compensation. The amount of slope compensation is set by the resistor RSUM. As
suggested earlier, we set the slope compensation ramp to be half the inductor current ramp downslope (inductor
current ramp during the off time), reflected through the main transformer and current sensing networks as
explained earlier in Slope Compensation (RSUM).
The required slope compensation ramp is
me
0.5 u
VOUT u RCS
LOUT u a1 u CTRAT
0.5 u
12 u 47
2 u 10 6 u 21 u 100
67
mV
Ps
(168)
The magnetizing current of the power transformer provides part of the compensating ramp and is calculated from
Equation 169.
´ RCS
V
400 ´ 47
mV
=
»7
mMAG = INMAX
ms
LMAG ´ CTRAT 2.76 x10-3 ´ 100
(169)
The required compensating ramp is
mV
mV
mSUM = me - mMAG = (67 - 7)
= 60
ms
ms
(170)
The value for the resistor, RSUM, may be found from the graph in Figure 37 or calculated from rearranged
versions of Equation 12 or Equation 13 depending on whether the controller is operating in Current or Voltage
Control Mode. In this case we are using Current Mode Control and Equation 12 is rearranged and evaluated as
follows
2.5
2.5
RSUM =
=
» 82k W
0.5 ´ mSUM 0.5 ´ 60 x10-3
(171)
60
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Check that the 300mV we allowed for the slope compensation ramp when choosing RCS in Equation 110 is
sufficient.
mV
´ 0.7
60
mSUM ´ DMAX
ms
DVSLOPE -COMP =
=
= 210mV
2 ´ FSW
2 ´ 100kHz
(172)
To increase efficiency at lighter loads the UCC28950 is programmed (Pin 12, DCM) under light load conditions to
disable the synchronous FETs on the secondary side of the converter (QE and QF). This threshold is
programmed with resistor divider formed by RDCMHI and RDCM. This DCM threshold needs to be set at a level
before the inductor current goes discontinuous. The following equation sets the level at which the synchronous
rectifiers are disabled at roughly 15% load current.
VRCS
æ POUT ´ 0.15 DILOUT ö
+
ç
÷ ´ RCS
VOUT
2 ø
è
=
= 0.29 V
a1´ CTRAT
(173)
Select a standard resistor value for RDCM.
RDCM = 1kW
(174)
Calculate resistor value RDCMHI.
RDCMHI =
RDCM (VREF - VRCS )
» 16.3kW
VRCS
(175)
Select a standard resistor value for this design
RDCMHI = 16.9kW
(176)
NOTE
It is recommended to use an RCD clamp to protect the output synchronous FETs from
overvoltage due to switch node ringing.
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Figure 55. Daughter Board Schematic
62
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+
+
+
+
+
+
+
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Figure 56. Power Stage Schematic
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8.2.3 Application Curves
Valley Switching
QB
QD
d
d
Valley Switching
QD g
QB g
QB = off
QB = on
QD = off
QD = on
QC = on
QC = off
0V
QA= on
QA/QB = off
t ABSET
t D Slight Delay after t
0V
t CDSET
ABSET
t D Slight Delay after t
CDSET
before Miller Plateau
before Miller Plateau
VIN = 390 V
IOUT = 5 A
VIN = 390 V
IOUT = 5 A
Figure 57. Full Bridge Gate Drives
and Primary Switch Nodes (QBd and QDd)
Valley Switching
QB d
Figure 58. Full Bridge Gate Drives
and Primary Switch Nodes (QDg QDd)
QD d
QB g
QD g
ZVS
QB = off
0V
QA = on
QA/QB = off
tABSET
tD
Slight Delay after t
before Miller Plateau
0V
QD = off
QB = on
QD = on
0V
QC = on
QC = off
t CDSET
ABSET
tD
Slight Delay after t
before Miller Plateau
CDSET
VIN = 390 V
IOUT = 10 A
VIN = 390 V
IOUT = 10 A
Figure 59. Full-Bridge Gate Drives
and Switch Nodes (QBg QBd)
Figure 60. Full-Bridge Gate Drives
and Switch Nodes (QDg QDd)
NOTE
Switch node QBd is valley switching and node QDd has achieved ZVS. Please refer to
Figure 59 and Figure 60. It is not uncommon for switch node QDd to obtain ZVS before
QBd. This is because during the QDd switch node voltage transition, the reflected output
current provides immediate energy for the LC tank at the switch node. Where at the QBd
switch node transition the primary has been shorted out by the high side or low side FETs
in the H bridge. This transition is dependent on the energy stored in LS and LLK to provide
energy for the LC tank at switch node QBd making it take longer to achieve ZVS.
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ZVS Achieved
QBd
QD
d
QD g
ZVS
QBg
QB = off
QD = off
QB = on
0V
QA = on
QA/QB = off
QD = on
0V
QC = on
QC= off
t CDSET
t ABSET
VIN = 390 V
IOUT = 25 A
VIN = 390 V
IOUT = 25 A
Figure 61. Full-Bridge Gate Drives
and Switch Nodes (QBg QBd)
Figure 62. Full-Bridge Gate Drives
and Switch Nodes (QDg QDd)
NOTE
When the converter is running at 25 A, both switch nodes are operating into zero voltage
switching (ZVS). It is also worth mentioning that there is no evidence of the gate miller
plateau during gate driver switching. This is because the voltage across the drains and
sources of FETs QA through QD have already transitioned before.
ZVS
QB d
QD
d
QD g
ZVS
QB g
QB = off
QD = off
QB = on
QD = on
0V
0V
QA = on
QC = on
QA/QB = off
QC = off
t ABSET
t CDSET
VIN = 390 V
IOUT = 50 A
VIN = 390 V
IOUT = 50 A
Figure 63. Full-Bridge Gate Drives
and Switch Nodes (QBg QBd)
Figure 64. Full-Bridge Gate Drives
and Switch Nodes (QDg QDd)
NOTE
ZVS maintained from 50% to 100% output power.
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9 Power Supply Recommendations
The UCC28950 device should be operated from a VDD rail within the limits given in the Recommended Operating
Conditions section of this datasheet. To avoid the possibility that the device might stop switching, VDD must not
be allowed to fall into the UVLO_FTH range. In order to minimize power dissipation in the device, VDD should not
be unnecessarily high. Keeping VDD at 12 V is a good compromise between these competing constraints. The
gate drive outputs from the UCC28950 device deliver large-current pulses into their loads. This indicates the
need for a low-ESR decoupling capacitor to be connected as directly as possible between the VDD and GND
terminals.
TI recommends ceramic capacitors with stable dielectric characteristics over temperature, such as X7R. Avoid
capacitors which have a large drop in capacitance with applied DC voltage bias. For example, use a part that has
a low-voltage co-efficient of capacitance. The recommended decoupling capacitance is 1 μF, X7R, with at least a
25-V rating with a 0.1-µF NPO capacitor in parallel.
10 Layout
10.1 Layout Guidelines
In order to increase the reliability and robustness of the design, TI recommends the following layout guidelines.
• VREF pin. Decouple this pin to GND with a good quality ceramic capacitor. A 1-uF, X7R, 25V capacitor is
recommended. Keep VREF PCB tracks as far away as possible from sources of switching noise.
• EA+ pin. This is the non-inverting input to the error amplifier. It is a high impedance pin and is susceptible to
noise pickup. Keep tracks from this pin as short as possible.
• EA– pin. This is the inverting input to the error amplifier. It is a high impedance pin and is susceptible to noise
pickup. Keep tracks from this pin as short as possible.
• COMP pin. The error amplifier compensation network is normally connected to this pin. Keep tracks from this
pin as short as possible.
• SS/EN pin. Keep tracks from this pin as short as possible. If the Enable signal is coming from a remote
source then avoid running it close to any source of high dv/dt (MOSFET Drain connections for example) and
add a simple RC filter at the SS/EN pin.
• DELAB, DELCD, DELEF, TMIN, RT, RSUM, DCM, ADELEF and ADEL pins. The components connected to
these pins are used to set important operating parameters. Keep these components close to the IC and
provide short, low impedance return connections to the GND pin.
• CS pin. This connection is arguably the most important single connection in the entire PSU system. Avoid
running the CS signal traces near to sources of high dv/dt. Provide a simple RC filter as close to the pin as
possible to help filter out leading edge noise spikes which will occur at the beginning of each switching cycle.
• SYNC pin. This pin is essentially a digital I/O port. If it is unused, then it may be left open circuit or tied to
ground via a 1-kΩ resistor. If Synchronisation is used, then route the incoming Synchronisation signal as far
away from noise sensitive input pins as possible.
• OUTA, OUTB, OUTC, OUTD, OUTE and OUTF pins. These are the gate drive output pins and will have a
high dv/dt rate associated with their rising and falling edges. Keep the tracks from these pins as far away from
noise sensitive input pins as possible. Ensure that the return currents from these outputs do not cause
voltage changes in the analog ground connections to noise sensitive input pins. Follow the layout
recommendation for Analog and Power ground Planes in Figure 45.
• VDD pin. This pin must be decoupled to GND using ceramic capacitors as detailed in the 'Power Supply
Recommendations' section. Keep this capacitor as close to the VDD and GND pins as possible.
• GND pin. This pin provides the ground reference to the controller. Use a Ground Plane to minimize the
impedance of the ground connection and to reduce noise pickup.
66
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UCC28950
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SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
10.2 Layout Example
VREF pin decoupled to GND (C1), close to the device
R28
R6
C8
VDD decoupling as close to the
device as possible. (C6, C5)
Top Side
R2
R7
C1
C3 R9
C6
C5 R15
U1
C4
R5
R1
OUTA through OUTE signals
routed as far as possible from
signal pins. (pins 17 through 22)
R12
R11
R13
R24
R16
C7
R14
R27
R25 R17
R26
R28
RC filter close to CS pin.
(C7, R27, pin 15)
R22
R23
Short tracks at EA+, EA-, COMP, SS/EN, DELAB, DELCD,
TMIN, RT, RSUM, DCM, ADELEF, and ADEL pins.
(pins 1, 2, 3, 4, 5, 6, 7, 8, 9, 10, 11, 12, 13 and 14)
Figure 65. UCC28950 Layout Example (Top Side)
Bottom Side
R29
C2
J1
R3
R4
R8
R20
R10
Figure 66. UCC28950 Layout Example (Bottom Side)
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Product Folder Links: UCC28950
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UCC28950
SLUSA16D – MARCH 2010 – REVISED NOVEMBER 2016
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
11.1.2 Development Support
For the MathCAD Design Tool, see SLUC210.
For the Excel Design Tool, see SLUC222.
11.2 Documentation Support
11.2.1 Related Documentation
Synchronizing Three or More UCC28950 Phase-Shifted, Full-Bridge Controllers, SLUA609.
11.3 Community Resources
The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective
contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of
Use.
TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration
among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help
solve problems with fellow engineers.
Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and
contact information for technical support.
11.4 Trademarks
E2E is a trademark of Texas Instruments.
United Chemi-Con is a trademark of United Chemi-Con.
Vitec is a trademark of Vitec Electronics Corporation.
All other trademarks are the property of their respective owners.
11.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.6 Glossary
SLYZ022 — TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
68
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Product Folder Links: UCC28950
PACKAGE OPTION ADDENDUM
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10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UCC28950PW
ACTIVE
TSSOP
PW
24
60
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
UCC28950
UCC28950PWR
ACTIVE
TSSOP
PW
24
2000
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
UCC28950
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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