UCC28C40-Q1, UCC28C41-Q1, UCC28C42-Q1, UCC28C43-Q1, UCC28C44-Q1, UCC28C45-Q1
UCC28C40-Q1, UCC28C41-Q1, UCC28C42-Q1, UCC28C43-Q1,
UCC28C44-Q1,
UCC28C45-Q1
SLUSA12F – DECEMBER
2009 – REVISED
OCTOBER 2020
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SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
UCC28C4x-Q1 Automotive BiCMOS Low-Power Current-Mode PWM Controllers
1 Features
3 Description
•
The UCC28C4x-Q1 family devices are
performance current-mode PWM controllers.
•
•
•
•
•
•
•
•
•
•
•
AEC-Q100 qualified with the following results:
– Device temperature grade 1:-40⁰C to 125⁰C
– Device HBM classification level 2: ±2 kV
– Device CDM classification level C4B: 750 V
Functional Safety-Capable
– Documentation available to aid functional safety
system design
1-MHz operation
50-μA standby current, 100-μA maximum
Low operating current of 2.3 mA at 52 kHz
Fast 35-ns cycle-by-cycle over-current limiting
±1-A peak output current
Rail-to-rail output swings with 25-ns rise and 20-ns
fall times
±1% initial trimmed 2.5-V error amplifier reference
Trimmed oscillator discharge current
New undervoltage lockout versions
Create a custom design using the UCC28C4x-Q1
with the WEBENCH® Power Designer
The UCC28C4x-Q1 family supports a fixed-frequency,
peak current-mode flyback implementation with UVLO
and maximum duty-cycle limit options. UCC28C4XQ1 can be configured with either primary-side
regulation using an auxiliary winding or secondaryside regulation with the use of an opto-coupler. This
family of devices features high frequency operation up
to 1 MHz with low start up and operating currents,
thus minimizing start up loss and operating power
consumption for improved efficiency. The devices also
feature a fast current sense to output delay time of 35
ns, and a ±1-A peak output current capability for
driving large external MOSFETs directly.
The UCC28C4x-Q1 family is offered in the 8-pin SOIC
(D) package.
Device Information (1)
PART NUMBER
•
•
•
PACKAGE
BODY SIZE (NOM)
UCC28C40-Q1
2 Applications
•
high
UCC28C41-Q1
OBC and DC/DC converter isolated bias power
supply
Traction inverter HV to LV backup supply
HVAC compressor HV isolated power supply
AC and DC EV charging equipment isolated power
supply
UCC28C42-Q1
SOIC (8)
UCC28C43-Q1
4.90 mm × 3.91 mm
UCC28C44-Q1
UCC28C45-Q1
1. For all available packages, see the orderable
addendum at the end of the datasheet.
D CLAMP
CSNUB
RSNUB
DOUT
VIN
CIN
CSS
NP
RSTART
DBIAS
NS
VOUT
COUT
RVDD
NA
RSS
CVDD
UCC28C42-Q1
R COMPp
C COMPp
RRT
1
COMP
VREF
8
2 FB
VDD
7
3 CS
OUT
6
4 RT/CT
GND
5
RG
QSW
DZ
CVDDbp
CVREF
R BLEEDER
RCS
CCT
CRAMP
RCSF
R DIS
RRAMP
RFBU
CCSF
RFBB
Copyright © 2016, Texas Instruments Incorporated
Typical Automotive Application Example
An©IMPORTANT
NOTICEIncorporated
at the end of this data sheet addresses availability, warranty, changes, use in
safety-critical
applications,
Copyright
2020 Texas Instruments
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UCC28C45-Q1
1
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SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
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Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Device Comparison Table...............................................3
6 Pin Configuration and Functions...................................3
7 Specifications.................................................................. 4
7.1 Absolute Maximum Ratings........................................ 4
7.2 ESD Ratings............................................................... 4
7.3 Recommended Operating Conditions.........................4
7.4 Thermal Information....................................................5
7.5 Electrical Characteristics.............................................5
7.6 Typical Characteristics................................................ 7
8 Detailed Description...................................................... 11
8.1 Overview................................................................... 11
8.2 Functional Block Diagram......................................... 12
8.3 Feature Description...................................................12
8.4 Device Functional Modes..........................................21
9 Application and Implementation.................................. 22
9.1 Application Information............................................. 22
9.2 Typical Application.................................................... 22
10 Power Supply Recommendations..............................36
11 Layout........................................................................... 37
11.1 Layout Guidelines................................................... 37
11.2 Layout Example...................................................... 39
12 Device and Documentation Support..........................40
12.1 Device Support....................................................... 40
12.2 Documentation Support.......................................... 40
12.3 Related Links.......................................................... 40
12.4 Support Resources................................................. 40
12.5 Trademarks............................................................. 40
12.6 Electrostatic Discharge Caution..............................41
12.7 Glossary..................................................................41
13 Mechanical, Packaging, and Orderable
Information.................................................................... 41
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision E (June, 2020) to Revision F (October, 2020)
Page
• Added Functional Safety-Capable information................................................................................................... 1
• Added Thermal Information table....................................................................................................................... 5
Changes from Revision D (October 2019) to Revision E (June 2020)
Page
• Updated Available Options Table heading from TA to TA = TJ ........................................................................... 3
• Updated Operating Junction Temperature in the Recommended Operating Conditions Table, from –55 to 150
to –40 to 105.......................................................................................................................................................4
Changes from Revision C (August 2019) to Revision D (October 2019)
Page
• Changed 3 V to 2.5 V....................................................................................................................................... 16
Changes from Revision B (October 2017) to Revision C (August 2019)
Page
• Changed Updated Applications section..............................................................................................................1
• Changed Updated description............................................................................................................................ 1
• Changed the marketing status Advance Information to Initial Release for the UCC28C41-Q1 , UCC28C42Q1, UCC28C44-Q1 and UCC28C45-Q1 devices............................................................................................... 1
• Changed AGain TA = 25°C MIN from 2.75 V to 2.85 V........................................................................................ 5
• Changed AGain TA = Full Range, MIN from 2.85 V to 2.75 V.............................................................................. 5
• Added Link to Technical Support ..................................................................................................................... 40
2
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SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
5 Device Comparison Table
UVLO
TURN ON AT 14.5 V
TURN OFF AT 9 V
SUITABLE FOR OFF-LINE
APPLICATIONS
TURN ON AT 8.4 V
TURN OFF AT 7.6 V
SUITABLE FOR DC/DC
APPLICATIONS
TURN ON AT 7 V
TURN OFF AT 6.6 V
SUITABLE FOR BATTERY
APPLICATIONS
MAXIMUM
DUTY
CYCLE
UCC28C42QDRQ1
UCC28C43QDRQ1
UCC28C40QDRQ1
100%
UCC28C44QDRQ1
UCC28C45QDRQ1
UCC28C41QDRQ1
50%
TEMPERATURE (TA)
–40°C to 125°C
6 Pin Configuration and Functions
COMP
1
8
VREF
FB
2
7
VDD
CS
3
6
OUT
RT/CT
4
5
GND
Not to scale
Figure 6-1. D Package 8-Pin SOIC Top View
Table 6-1. Pin Functions
PIN
NAME
NO.
COMP
1
I/O
DESCRIPTION
O
This pin provides the output of the error amplifier for compensation. In addition, the COMP pin is frequently
used as a control port, by utilizing a secondary-side error amplifier to send an error signal across the
secondary-primary isolation boundary through an opto-isolator. The error amplifier is internally current limited
so the user can command zero duty cycle by externally forcing COMP to GND.
CS
3
I
Primary-side current sense pin. The current sense pin is the noninverting input to the PWM comparator.
Connect to current sensing resistor. This signal is compared to a signal proportional to the error amplifier
output voltage. The PWM uses this to terminate the OUT switch conduction. A voltage ramp can be applied to
this pin to run the device with a voltage mode control configuration.
FB
2
I
This pin is the inverting input to the error amplifier. FB is used to control the power converter voltage-feedback
loop for stability. The noninverting input to the error amplifier is internally trimmed to 2.5 V ±1%.
GND
5
—
Ground return pin for the output driver stage and the logic level controller section.
O
The output of the on-chip drive stage. OUT is intended to directly drive a MOSFET. The OUT pin in the
UCC28C40-Q1, UCC28C42-Q1, and UCC28C43-Q1 is the same frequency as the oscillator, and can operate
near 100% duty cycle. In the UCC28C41-Q1, UCC28C44-Q1, and UCC28C45-Q1, the frequency of OUT is
one-half that of the oscillator due to an internal T flipflop. This limits the maximum duty cycle to < 50%. Peak
currents of up to 1 A are sourced and sunk by this pin. OUT is actively held low when VDD is below the turnon
threshold.
I/O
Fixed frequency oscillator set point. Connect timing resistor (RRT) to VREF and timing capacitor (CCT) to GND
from this pin to set the switching frequency. For best performance, keep the timing capacitor lead to the device
GND as short and direct as possible. If possible, use separate ground traces for the timing capacitor and all
other functions. The switching frequency (fSW) of the UCC28C40-Q1, UCC28C42-Q1, and UCC28C43-Q1
gate drive is equal to fOSC; the switching frequency of the UCC28C41-Q1, UCC28C44-Q1, and UCC28C45Q1 is equal to half of the fOSC.
I
Analog controller bias input that provides power to the device. Total VDD current is the sum of the quiescent
VDD current and the average OUT current. A bypass capacitor, typically 0.1 µF, connected directly to GND
with minimal trace length, is required on this pin. Additional capacitance at least 10 times greater than the gate
capacitance of the main switching FET used in the design is also required on VDD.
O
5-V reference voltage. VREF is used to provide charging current to the oscillator timing capacitor through the
timing resistor. It is important for reference stability that VREF is bypassed to GND with a ceramic capacitor
connected as close to the pin as possible. A minimum value of 0.1 µF ceramic is required. Additional VREF
bypassing is required for external loads on VREF.
OUT
RT/CT
VDD
VREF
6
4
7
8
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1) (2)
MAX
UNIT
Input voltage
VDD
MIN
20
V
Input current
IVDD
30
mA
±1
A
5
µJ
Output drive current (peak)
Output energy (capacitive load), EOUT
Analog input voltage
COMP, CS, FB, RT/CT
–0.3
6.3
Output driver voltage
OUT
–0.3
20
Reference voltage
VREF
7
Error amplifier output sink current
COMP
10
mA
Total power dissipation at TA = 25°C
D package
50
°C/W
300
°C
Lead temperature (soldering, 10 s), TLEAD
V
Operating junction temperature, TJ
–40
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Section 7.3.
Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to GND pin. Currents are positive into and negative out of the specified terminals.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
Electrostatic discharge
Human-body model (HBM), per AEC
Q100-002(1)
UNIT
±2000
Charged-device model (CDM), per AEC Q100-011 Section 7.2
V
±750
AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
7.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted)
MIN
VVDD
VOUT
Output driver voltage
IOUT
Average output driver current(1)
IOUT(VREF)
Reference output current(1)
TJ
Operating junction temperature(1)
(1)
4
Input voltage
MAX
UNIT
18
UCC28C4x-Q1
–40
V
18
V
200
mA
–20
mA
125
°C
TI recommends against operating the device under conditions beyond those specified in this table for extended periods of time.
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SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
7.4 Thermal Information
UCC28C4x-Q1
THERMAL METRIC(1)
UNIT
D (SOIC) 8 PINS
RθJA
Junction-to-ambient thermal resistance
115.0
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
57.1
°C/W
RθJB
Junction-to-board thermal resistance
58.5
°C/W
ψJT
Junction-to-top characterization parameter
13.0
°C/W
ψJB
Junction-to-board characterization parameter
57.8
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor anddevicePackage Thermal Metrics
application report.
7.5 Electrical Characteristics
VVDD = 15 V(1), RRT = 10 kΩ, CCT = 3.3 nF, CVDD = 0.1 µF and no load on the outputs, TA = –40°C to 125°C (unless otherwise
noted).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
5
5.1
V
0.2
20
mV
25
mV
REFERENCE
VVREF
IVREF
VREF voltage, initial accuracy
IOUT = 1 mA
Line regulation
VVDD = 12 V to 18 V
Load regulation
1 mA to 20 mA
Temperature stability
See (2)
Total output variation
See (2)
VREF noise voltage
10 Hz to 10 kHz, TA = 25°C, see (2)
Long term stability
1000 hours, TA = 125°C, see (2)
4.9
3
0.2
4.82
0.4 mV/°C
5.18
50
V
µV
5
25
mV
–30
–45
–55
mA
TA = 25°C, see (3)
50.5
53
55
kHz
TA = Full Range, see (3)
50.5
57
kHz
Output short circuit
OSCILLATOR
fOSC
Initial accuracy
Voltage stability
12 V ≤ VVDD ≤ 18 V
Temperature stability
TA(MIN) to TA(MAX), see (2)
Amplitude
RT/CT pin peak-to-peak voltage
Discharge current
0.2%
1%
1%
2.5%
1.9
V
TA = 25°C, VRT/CT = 2 V, see (4)
7.7
8.4
9
TA = Full Range, VRT/CT = 2 V, see (4)
7.2
8.4
9.5
2.475
2.5
2.525
V
2.45
2.5
2.55
V
–0.1
–2
µA
mA
ERROR AMPLIFIER
VFB
Feedback input voltage, initial accuracy TA = 25°C, VCOMP = 2.5 V
Feedback input voltage, total variation
TA = Full Range, VCOMP = 2.5 V
IFB
Input bias current
VFB = 5 V
AVOL
Open-loop voltage gain
2 V ≤ VOUT ≤ 4 V
(2)
Unity gain bandwidth
See
Power supply rejection ratio
12 V ≤ VVDD ≤ 18 V
Output sink current
VFB = 2.7 V, VCOMP = 1.1 V
Output source current
VFB = 2.3 V, VCOMP = 5 V
VOH
High-level COMP voltage
VFB = 2.7 V, RCOMP = 15 kΩ COMP to GND
VOL
Low-level COMP voltage
VFB = 2.7 V, RCOMP = 15 kΩ COMP to VREF
PSRR
Copyright © 2020 Texas Instruments Incorporated
65
90
dB
1
1.5
MHz
60
dB
2
14
mA
–0.5
–1
mA
5
6.8
0.1
V
1.1
V
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SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
VVDD = 15 V(1), RRT = 10 kΩ, CCT = 3.3 nF, CVDD = 0.1 µF and no load on the outputs, TA = –40°C to 125°C (unless otherwise
noted).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
2.85
3
3.15
V/V
2.75
3
3.15
V/V
0.9
1
1.1
V
–0.1
–2
µA
35
70
ns
CURRENT SENSE
TA = 25°C, See (5) (6)
ACS
Gain
VCS
Maximum input signal
TA = Full Range, See
PSRR
Power supply rejection ratio
ICS
Input bias current
tD
CS to output delay
(5) (6)
VFB < 2.4 V
VVDD = 12 V to 18
V(2) (5)
70
dB
COMP to CS offset
VCS = 0 V
1.15
V
RDS(on) pulldown
ISINK = 200 mA
5.5
VOUT(high) RDS(on) pullup
ISOURCE = 200 mA
10
25
Ω
tRISE
Rise tIme
COUT = 1 nF
25
50
ns
tFALL
Fall tIme
COUT = 1 nF
20
40
ns
OUTPUT
VOUT(low)
15
Ω
UNDERVOLTAGE LOCKOUT
VDDON
VDDOFF
Start threshold
Minimum operating voltage
UCC28C42-Q1, UCC28C44-Q1
13.5
14.5
15.5
UCC28C43-Q1, UCC28C45-Q1
7.8
8.4
9
UCC28C40-Q1, UCC28C41-Q1
6.5
7
7.5
UCC28C42-Q1, UCC28C44-Q1
8
9
10
UCC28C43-Q1, UCC28C45-Q1
7
7.6
8.2
UCC28C40-Q1, UCC28C41-Q1
6.1
6.6
7.1
UCC28C42-Q1, UCC28C43-Q1, UCC28C40-Q1, V
FB < 2.4 V
94%
96%
UCC28C44-Q1, UCC28C45-Q1, UCC28C41-Q1, V
FB < 2.4 V
47%
48%
V
V
PWM
DMAX
Maximum duty cycle
DMIN
Minimum duty cycle
VFB > 2.6 V
0%
CURRENT SUPPLY
ISTART-UP
Start-up current
VVDD = VDDON – 0.5 V
50
100
µA
IVDD
Operating supply current
VFB = VCS = 0 V
2.3
3
mA
(1)
(2)
(3)
(4)
(5)
(6)
6
Adjust VVDD above the start threshold before setting at 15.5 V.
Ensured by design. Not production tested.
Output frequencies of the UCC28C41-Q1, UCC28C44-Q1, and the UCC28C45-Q1 are half the oscillator frequency.
Oscillator discharge current is measured with RRT = 10 kΩ to VREF.
Parameter measured at trip point of latch with VFB = 0 V.
Gain is defined as ACS = ΔVCOMP / ΔVCS , 0 V ≤ VCS ≤ 900 mV
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7.6 Typical Characteristics
9.5
IDISCH -- Oscillator Discharge Current -- mA
fOSC Oscillator Frequency (kHz)
1000
100
10
220 pF
470 pF
1 nF
2.2 nF
4.7 nF
1
9.0
8.5
8.0
7.5
7.0
1
10
RRT Timing Resistance (k:)
100
--50
--25
0
D001
Figure 7-1. Oscillator Frequency vs Timing
Resistance and Capacitance
50
75
100
125
Figure 7-2. Oscillator Discharge Current vs
Temperature
100
200
1.8
90
180
1.6
160
1.4
80
25
TJ -- Temperature -- °C
60
120
50
100
40
80
1.2
COMP to CS
140
Phase Margin -- (°)
Gain -- (dB)
GAIN
70
1.0
0.8
0.6
30
60
PHASE
MARGIN
20
0.4
40
10
0.2
20
0
1
10
100
1k
10 k
100 k
1M
0.0
0
10 M
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
f -- Frequency -- Hz
Figure 7-3. Error Amplifier Frequency Response
VCS = 0 V
Figure 7-4. COMP to CS Offset Voltage vs
Temperature
5.05
VEAREF -- Error Amplifier Reference Voltage -- V
2.55
VREF -- Reference Voltage -- V
5.04
5.03
5.02
5.01
5.00
4.99
4.98
4.97
4.96
4.95
--50
2.54
2.53
2.52
2.51
2.50
2.49
2.48
2.47
2.46
2.45
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-5. Reference Voltage vs Temperature
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--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-6. Error Amplifier Reference Voltage vs
Temperature
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200
--37
IBIAS -- Error Amplifier Input Bias Current -- nA
ISC -- Reference Short Circuit Current -- mA
--35
--39
--41
--43
--45
--47
--49
--51
--53
--55
--50
100
50
0
--50
--100
--150
--200
--25
0
25
50
75
TJ -- Temperature -- °C
100
125
Figure 7-7. Reference Short-Circuit Current vs
Temperature
--50
15
8.8
VUVLO -- UVLO Voltage -- V
9.0
13
UVLO
ON
12
UVLO
OFF
11
10
7.2
6
75
100
UVLO
ON
7.8
7.4
50
UVLO
OFF
7.0
--50
125
--25
0
TJ -- Temperature -- °C
UCC28C42-Q1 and UCC28C44-Q1
25
50
75
TJ -- Temperature -- °C
100
125
UCC28C43-Q1 and UCC28C45-Q1
Figure 7-9. Undervoltage Lockout vs Temperature
Figure 7-10. Undervoltage Lockout vs Temperature
7.3
25
7.2
IDD -- Supply Current -- mA
UVLO
ON
7.1
VUVLO -- UVLO Voltage -- V
125
8.0
7
25
100
8.2
7.6
0
25
50
75
TJ -- Temperature -- °C
8.4
8
--25
0
8.6
9
--50
--25
Figure 7-8. Error Amplifier Input Bias Current vs
Temperature
16
14
VUVLO -- UVLO Voltage -- V
150
7.0
6.9
6.8
6.7
6.6
6.5
1-nF LOAD
20
15
10
NO LOAD
5
UVLO
OFF
6.4
6.3
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
UCC28C40-Q1 and UCC28C41-Q1
0
0k
200 k
400 k
600 k
f -- Frequency -- Hz
800 k
1M
Figure 7-12. Supply Current vs Oscillator
Frequency
Figure 7-11. Undervoltage Lockout vs Temperature
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3.0
40
10% to 90%
VDD = 12 V
2.9
35
Output Rise and Fall TIme -- ns
IDD -- Supply Current -- mA
2.8
2.7
2.6
2.5
2.4
NO LOAD
2.3
2.2
tr
(1 nF)
30
tf
(1 nF)
25
20
15
2.1
2.0
10
--50
--25
0
25
50
75
100
125
--50
--25
TJ -- Temperature -- °C
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-13. Supply Current vs Temperature
Figure 7-14. Output Rise Time and Fall Time vs
Temperature
100
CT = 220 pF
Duty Cycle -- %
90
80
70
CT = 1 nF
60
50
0
500
1000
1500
2000
2500
f -- Frequency -- kHz
Figure 7-15. Maximum Duty Cycle vs Oscillator
Frequency
Figure 7-16. Maximum Duty Cycle vs Temperature
VCS_th -- Current Sense Threshold -- V
1.10
1.05
1.00
0.95
0.90
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-17. Maximum Duty Cycle vs Temperature
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Figure 7-18. Current Sense Threshold Voltage vs
Temperature
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70
tD -- CS to OUT Delay Time -- ns
65
60
55
50
45
40
35
30
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-19. Current Sense to Output Delay Time vs Temperature
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8 Detailed Description
8.1 Overview
The UCC28C4x-Q1 series of control integrated circuits provide the features necessary to implement AC-DC or
DC‑to-DC fixed-frequency current-mode control schemes with a minimum number of external components.
Protection circuitry includes undervoltage lockout (UVLO) and current limiting. Internally implemented circuits
include a start-up current of less than 100 µA, a precision reference trimmed for accuracy at the error amplifier
input, logic to ensure latched operation, a pulse-width modulation (PWM) comparator that also provides currentlimit control, and an output stage designed to source or sink high-peak current. The output stage, suitable for
driving N-channel MOSFETs, is low when it is in the OFF state. The oscillator contains a trimmed discharge
current that enables accurate programming of the maximum duty cycle and dead time limit, making this device
suitable for high-speed applications.
Major differences between members of this series are the UVLO thresholds, acceptable ambient temperature
range, and maximum duty cycle. Typical UVLO thresholds of 14.5 V (ON) and 9 V (OFF) on the UCC28C42-Q1
and UCC28C44-Q1 devices make them ideally suited to off-line AC-DC applications. The corresponding typical
thresholds for the UCC28C43-Q1 and UCC28C45-Q1 devices are 8.4 V (ON) and 7.6 V (OFF), making them
ideal for use with regulated input voltages used in DC-DC applications. The UCC28C40-Q1 and UCC28C41-Q1
feature a start-up threshold of 7 V and a turnoff threshold of 6.6 V (OFF), which makes them suitable for batterypowered applications. The UCC28C40-Q1, UCC28C42-Q1, and UCC28C43-Q1 devices operate to duty cycles
approaching 100%. The UCC28C41-Q1, UCC28C44-Q1, and UCC28C45-Q1 obtain a duty cycle from 0% to
50% by the addition of an internal toggle flip-flop, which blanks the output off every other clock cycle. The
UCC28C4x-Q1 series is specified for operation ambient temperature from –40°C to 125°C.
The UCC28C4x-Q1 series are an enhanced replacement with pin-to-pin compatibility to the bipolar UC284x,
UC384x, UC284xA, and UC384xA families. The new series offers improved performance when compared to
older bipolar devices and other competitive BiCMOS devices with similar functionality. These improvements
generally consist of tighter specification limits that are a subset of the older product ratings, maintaining drop-in
capability. In new designs, these improvements can reduce the component count or enhance circuit performance
when compared to the previously available devices.
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8.2 Functional Block Diagram
VDD
UVLO
EN
5V
VREF
VREF
VREF Good
Logic
RT/CT
Osc
( NOTE)
2. 5 V
S
2R
+
E/A
FB
OUT
T
PWM
Latch
R
R
1V
GND
PWM
Comparator
COMP
CS
Copyright © 2016, Texas Instruments Incorporated
Toggle flip-flop used only in UCC28C41-Q1, UCC28C44-Q1, and UCCx8C45-Q1
8.3 Feature Description
The BiCMOS design allows operation at high frequencies that were not feasible in the predecessor bipolar
devices. First, the output stage has been redesigned to drive the external power switch in approximately half the
time of the earlier devices. Second, the internal oscillator is more robust, with less variation as frequency
increases. This faster oscillator makes this device suitable for high speed applications and the trimmed
discharge current enables precise programming of the maximum duty cycle and dead-time limit. In addition, the
current sense to output delay has been reduced by a factor of three, to 45 ns (typical). The reduced delay times
in the current sense results in superior overload protection at the power switch. The reduced start-up current of
this device minimizes steady state power dissipation in the startup resistor, and the low operating current
maximizes efficiency while running, increasing the total circuit efficiency, whether operating off-line, DC input, or
battery operated circuits. These features combine to provide a device capable of reliable, high-frequency
operation.
Table 8-1. Key Parameters
PARAMETER
UCC28C4x-Q1
UCC28C4x
Supply current at 50 kHz
2.3 mA
2.3 mA
Start-up current
50 µA
50 µA
Overcurrent propagation delay
50 ns
50 ns
Reference voltage accuracy
± 1%
± 1%
Error amplifier reference voltage accuracy
± 25 mV
± 25mV
Maximum oscillator frequency
> 1 MHz
1 MHz
Output rise/fall times
25 ns
25 ns
UVLO turnon accuracy
±1V
±1V
Ambient Temperature
-40°C TO 125°C
-40°C TO 105°C
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8.3.1 Detailed Pin Description
8.3.1.1 COMP
The error amplifier in the UCC28C4x-Q1 family has a unity-gain bandwidth of 1.5 MHz. The COMP terminal can
both source and sink current. The error amplifier is internally current-limited, so that one can command zero duty
cycle by externally forcing COMP to GND.
8.3.1.2 FB
FB is the inverting input of the error amplifier. The noninverting input to the error amplifier is internally trimmed to
2.5 V ± 1%. FB is used to control the power converter voltage-feedback loop for stability. For best stability, keep
FB lead length as short as possible and FB stray capacitance as small as possible.
8.3.1.3 CS
The UCC28C4x-Q1 current sense input connects directly to the PWM comparator. Connect CS to the MOSFET
source current sense resistor. The PWM uses this signal to terminate the OUT switch conduction. A voltage
ramp can be applied to this pin to run the device with a voltage mode control configuration or to add slope
compensation. To prevent false triggering due to leading edge noises, an RC current sense filter may be
required. The gain of the current sense amplifier is typically 3 V/V.
8.3.1.4 RT/CT
RT/CT is the oscillator timing pin. For fixed frequency operation, set the timing capacitor charging current by
connecting a resistor from VREF to RT/CT. Set the frequency by connecting timing capacitor from RT/CT to
GND. For the best performance, keep the timing capacitor lead to GND as short and direct as possible. If
possible, use separate ground traces for the timing capacitor and all other functions.
The UCC28C4x-Q1’s oscillator allows for operation to 1 MHz. The device uses an external resistor to set the
charging current for the external capacitor, which determines the oscillator frequency. TI recommends timing
resistor values from 1 kΩ to 100 kΩ and timing capacitor values from 220 pF to 4.7 nF. The UCC28C4x-Q1
oscillator is true to the curves of the original bipolar devices at lower frequencies, yet extends the frequency
programmability range to at least 1 MHz. This allows the device to offer pin-to-pin capability where required, yet
capable of extending the operational range to the higher frequencies. See Figure 7-1 for component values for
setting the oscillator frequency.
8.3.1.5 GND
GND is the signal and power returning ground. TI recommends separating the signal return path and the high
current gate driver path so that the signal is not affected by the switching current.
8.3.1.6 OUT
The high-current output stage of the UCC28C4x-Q1 has been redesigned to drive the external power switch in
approximately half the time of the earlier devices. To drive a power MOSFET directly, the totem-pole OUT driver
sinks or source up to 1 A peak of current. The OUT of the UCC28C40-Q1, UCC28C42-Q1, and UCC28C43-Q1
devices switch at the same frequency as the oscillator and can operate near 100% duty cycle. In the
UCC28C41-Q1, UCC28C44-Q1, and UCC28C45-Q1, the switching frequency of OUT is one-half that of the
oscillator due to an internal T flip-flop. This limits the maximum duty cycle in the UCC28C41-Q1, UCC28C44-Q1,
and UCC28C45-Q1 to < 50%.
The UCC28C4x-Q1 family houses unique totem pole drivers exhibiting a 10-Ω impedance to the upper rail and a
5.5‑Ω impedance to ground, typically. This reduced impedance on the low-side switch helps minimize turnoff
losses at the power MOSFET, whereas the higher turnon impedance of the high-side is intended to better match
the reverse recovery characteristics of many high-speed output rectifiers. Transition times, rising and falling
edges, are typically 25 nanoseconds and 20 nanoseconds, respectively, for a 10% to 90% change in voltage.
A low impedance MOS structure in parallel with a bipolar transistor, or BiCMOS construction, comprises the
totem-pole output structure. This more efficient utilization of silicon delivers the high peak current required along
with sharp transitions and full rail-to-rail voltage swings. Furthermore, the output stage is self-biasing, active low
during under-voltage lockout type. With no VDD supply voltage present, the output actively pulls low if an
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attempt is made to pull the output high. This condition frequently occurs at initial power-up with a power
MOSFET as the driver load.
8.3.1.7 VDD
VDD is the power input connection for this device. In normal operation, power VDD through a current limiting
resistor. The absolute maximum supply voltage is 20 V, including any transients that may be present. If this
voltage is exceeded, device damage is likely. This is in contrast to the predecessor bipolar devices, which could
survive up to 30 V on the input bias pin. Also, because no internal clamp is included in the device, the VDD pin
must be protected from external sources which could exceed the 20 V level. If containing the start-up and
bootstrap supply voltage from the auxiliary winding N A below 20 V under all line and load conditions can not be
achieved, use a zener protection diode from VDD to GND. Depending on the impedance and arrangement of the
bootstrap supply, this may require adding a resistor, R VDD, in series with the auxiliary winding to limit the current
into the zener as shown in Figure 8-1. Insure that over all tolerances and temperatures, the minimum zener
voltage is higher than the highest UVLO upper turnon threshold. To ensure against noise related problems, filter
VDD with a ceramic bypass capacitor to GND. The VDD pin must be decoupled as close to the GND pin as
possible.
NP
NS
RSTART
To
Input
NA
DBIAS
RVDD
VDD
OUT
CVCC
DZCLAMP
CVDDbp
0.1 PF
GND
RCS
Figure 8-1. VDD Protection
Although quiescent VDD current is only 100 µA, the total supply current is higher, depending on the OUT
current. Total VDD current is the sum of quiescent VDD current and the average OUT current. Knowing the
operating frequency and the MOSFET gate charge (Q g), average OUT current can be calculated from Equation
1.
IOUT = Q g × fSW
(1)
8.3.1.8 VREF
VREF is the voltage reference for the error amplifier and also for many other internal circuits in the device. The
5-V reference tolerance is ±1% for the UCC28C4x-Q1 family. The high-speed switching logic uses VREF as the
logic power supply. The reference voltage is divided down internally to 2.5 V ±1% and connected to the error
amplifier's noninverting input for accurate output voltage regulation. The reference voltage sets the internal bias
currents and thresholds for functions such as the oscillator upper and lower thresholds along with the
overcurrent limiting threshold. The output short-circuit current is 55 mA (maximum). To avoid device over-heating
and damage, do not pull VREF to ground as a means to terminate switching. For reference stability and to
prevent noise problems with high-speed switching transients, bypass VREF to GND with a ceramic capacitor
close to thedevicepackage. A minimum of 0.1-µF ceramic capacitor is required. Additional VREF bypassing is
required for external loads on the reference. An electrolytic capacitor may also be used in addition to the ceramic
capacitor.
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8.3.2 Undervoltage Lockout
Three sets of UVLO thresholds are available with turnon and turnoff thresholds of: (14.5 V and 9 V), (8.4 V and
7.6 V), and (7 V and 6.6 V) respectively. The first set is primarily intended for off-line and 48-V distributed power
applications, where the wider hysteresis allows for lower frequency operation and longer soft-starting time of the
converter. The second group of UVLO options is ideal for high frequency DC-DC converters typically running
from a 12-VDC input. The third, and newest, set has been added to address battery powered and portable
applications. Table 8-2 shows the maximum duty cycle and UVLO thresholds by device.
Table 8-2. UVLO Options
MAXIMUM DUTY CYCLE
UVLO ON
UVLO OFF
PART NUMBER
100%
14.5 V
9V
UCC28C42-Q1
100%
8.4 V
7.6 V
UCC28C43-Q1
100%
7V
6.6 V
UCC28C40-Q1
50%
14.5 V
9V
UCC28C44-Q1
50%
8.4 V
7.6 V
UCC28C45-Q1
50%
7V
6.6 V
UCC28C41-Q1
During UVLO thedevicedraws less than 100 µA of supply current. Once crossing the turnon threshold
thedevicesupply current increases to a maximum of 3 mA, typically 2.3 mA. This low start-up current allows the
power supply designer to optimize the selection of the startup resistor value to provide a more efficient design. In
applications where low component cost overrides maximum efficiency, the low run current of 2.3 mA (typical)
allows the control device to run directly through the single resistor to (+) rail, rather than requiring a bootstrap
winding on the power transformer, along with a rectifier. The start and run resistor for this case must also pass
enough current to allow driving the primary switching MOSFET, which may be a few milliamps in small devices.
< 3 mA
IVDD
< 100 µA
VOFF
VON
VVDD
Figure 8-2. UVLO ON and OFF Profile
8.3.3 ±1% Internal Reference Voltage
The BiCMOS internal reference of 2.5 V has an enhanced design, and uses production trim to allow initial
accuracy of ±1% at room temperature and ±2% over the full temperature range. This can be used to eliminate an
external reference in applications that do not require the extreme accuracy afforded by the additional device.
This is useful for nonisolated DC-DC applications, where the control device is referenced to the same common
as the output. It is also applicable in off-line designs that regulate on the primary side of the isolation boundary
by looking at a primary bias winding, or from a winding on the output inductor of a buck-derived circuit.
8.3.4 Current Sense and Overcurrent Limit
An external series resistor (R CS) senses the current and converts this current into a voltage that becomes the
input to the CS pin. The CS pin is the noninverting input to the PWM comparator. The CS input is compared to a
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signal proportional to the error amplifier output voltage; the gain of the current sense amplifier is typically 3 V/V.
The peak ISENSE current is determined using Equation 2
ISENSE =
VCS
R CS
(2)
The typical value for VCS is 1 V. A small RC filter (RCSF and CCSF) may be required to suppress switch transients
caused by the reverse recovery of a secondary side diode or equivalent capacitive loading in addition to parasitic
circuit impedances. The time constant of this filter should be considerably less than the switching period of the
converter.
Error
Amplifier
COMP
2R
R
1V
ISENSE
RCSF
RCS
PWM
Comparator
CS
CCSF
GND
Copyright © 2016, Texas Instruments Incorporated
Figure 8-3. Current-Sense Circuit Schematic
Cycle-by-cycle pulse width modulation performed at the PWM comparator essentially compares the error
amplifier output to the current sense input. This is not a direct volt-to-volt comparison, as the error amplifier
output network incorporates two diodes in series with a resistive divider network before connecting to the PWM
comparator. The two-diode drop adds an offset voltage that enables zero duty cycle to be achieved with a low
amplifier output. The 2R/R resistive divider facilitates the use of a wider error amplifier output swing that can be
more symmetrically centered on the 2.5-V noninverting input voltage.
The 1-V zener diode associated with the PWM comparator’s input from the error amplifier is not an actual diode
in the device’s design, but an indication that the maximum current sense input amplitude is 1 V (typical). When
this threshold is reached, regardless of the error amplifier output voltage, cycle-by-cycle current limiting occurs,
and the output pulse width is terminated within 35 ns (typical). The minimum value for this current limit threshold
is 0.9 V with a 1.1-V maximum. In addition to the tolerance of this parameter, the accuracy of the current sense
resistor, or current sense circuitry, must be taken into account. It is advised to factor in the worst case of primary
and secondary currents when sizing the ratings and worst-case conditions in all power semiconductors and
magnetic components.
8.3.5 Reduced-Discharge Current Variation
The UCC28C4x-Q1 oscillator design incorporates a trimmed discharge current to accurately program maximum
duty cycle and operating frequency. In its basic operation, a timing capacitor (C CT) is charged by a current
source, formed by the timing resistor (R RT) connected to the device’s reference voltage (VREF). The oscillator
design incorporates comparators to monitor the amplitude of the timing capacitor’s voltage. The exponentially
shaped waveform charges up to a specific amplitude representing the oscillator’s upper threshold of 2.5 V. Once
reached, an internal current sink to ground is turned on and the capacitor begins discharging. This discharge
continues until the oscillator’s lower threshold has reached 0.7 V at which point the current sink is turned off.
Next, the timing capacitor starts charging again and a new switching cycle begins.
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VREF
VDDON
VDDOFF
RRT
CCT
CCT
RT/CT
tON
GND
8.4 mA
tOFF
tPERIOD
Copyright © 2016, Texas Instruments Incorporated
Figure 8-4. Oscillator Circuit
While the device is discharging the timing capacitor, resistor R RT is also still trying to charge C CT. It is the exact
ratio of these two currents, the discharging versus the charging current, which specifies the maximum duty cycle.
During the discharge time of CCT, the device’s output is always off. This represents an ensured minimum off time
of the switch, commonly referred to as dead-time. To program an accurate maximum duty cycle, use the
information provided in Figure 7-15 for maximum duty cycle versus oscillator frequency. Any number of
maximum duty cycles can be programmed for a given frequency by adjusting the values of RRT and CCT. Once R
RT is selected, the oscillator timing capacitor can be found using the curves in Figure 7-1. However, because
resistors are available in more precise increments, typically 1%, and capacitors are only available in 5%
accuracy, it might be more practical to select the closest capacitor value first and then calculate the timing
resistor value next.
8.3.6 Oscillator Synchronization
Synchronization is best achieved by forcing the timing capacitor voltage above the oscillator's internal upper
threshold. A small resistor is placed in series with C CT to GND. This resistor serves as the input for the sync
pulse which raises the CCT voltage above the oscillator’s internal upper threshold. The PWM is allowed to run at
the frequency set by RRT and CCT until the sync pulse appears. This scheme offers several advantages including
having the local ramp available for slope compensation. The UCC28C4x-Q1 oscillator must be set to a lower
frequency than the sync pulse stream, typically 20 percent with a 0.5-V pulse applied across the resistor.
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VREF
RRT
CCT + SYNC
CCT
RT/CT
SYNC
SYNC
50
GND
CCT
Copyright © 2016, Texas Instruments Incorporated
Figure 8-5. Oscillator Synchronization Circuit
8.3.7 Soft Start
Soft start is the technique to gradually power up the converter in a well-controlled fashion by slowly increasing
the effective duty cycle starting at zero and gradually rising. Following start-up of the PWM, the error amplifier
inverting input is low, commanding the error amplifier’s output to go high. The output stage of the amplifier can
source 1 mA typically, which is enough to drive most high impedance compensation networks, but not enough
for driving large loads quickly. Soft start is achieved by charging a fairly large value, >1-µF, capacitor (C SS)
connected to the error amplifier output through a PNP transistor as shown in Figure 8-6
VREF
RSS
COMP
ZF
+
2N2907
CSS
FB
ZI
To VOUT
Figure 8-6. Soft-Start Implementation
The limited charging current of the amplifier into the capacitor translates into a dv/dt limitation on the error
amplifier output. This directly corresponds to some maximum rate of change of primary current in a current mode
controlled system as one of the PWM comparator’s inputs gradually rises. The values of R SS and C SS must be
selected to bring the COMP pin up at a controlled rate, limiting the peak current supplied by the power stage.
After the soft-start interval is complete, the capacitor continues to charge to VREF, effectively removing the PNP
transistor from the circuit consideration. Soft start performs a different, frequently preferred function in current
mode controlled systems than it does in voltage mode control. In current mode, soft start controls the rising of
the peak switch current. In voltage mode control, soft start gradually widens the duty cycle, regardless of the
primary current or rate of ramp-up.
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The purpose of the resistor R SS and diode is to take the soft-start capacitor out of the error amplifier’s path
during normal operation, once soft start is complete and the capacitor is fully charged. The optional diode in
parallel with the resistor forces a soft start each time the PWM goes through UVLO condition that forces VREF to
go low. Without the diode, the capacitor remains charged during a brief loss of supply or brown-out, and no soft
start is enabled upon re-application of VDD.
8.3.8 Enable and Disable
There are a few ways to enable or disable the UCC28C4x-Q1 devices, depending on which type of restart is
required. The two basic techniques use external transistors to either pull the error amplifier output low (< 2 V BE)
or pull the current sense input high (> 1.1 V). Application of the disable signal causes the output of the PWM
comparator to be high. The PWM latch is reset dominant so that the output remains low until the next clock cycle
after the shutdown condition at the COMP or CS pin is removed. Another choice for restart without a soft start is
to pull the current sense input above the cycle-by-cycle current limiting threshold. A logic level P-channel FET
from the reference voltage to the current sense input can be used.
COMP
DISABLE
Figure 8-7. Disable Circuit
8.3.9 Slope Compensation
With current mode control, slope compensation is required to stabilize the overall loop with duty cycles
exceeding 50%. Although not required, slope compensation also improves stability in applications using below a
50% maximum duty cycle. Slope compensation is introduced by injecting a portion of the oscillator waveform to
the actual sensed primary current. The two signals are summed together at the current sense input (CS)
connection at the filter capacitor. To minimize loading on the oscillator, it is best to buffer the timing capacitor
waveform with a small transistor whose collector is connected to the reference voltage.
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VREF
0.1 µF
RRT
RT/CT
CCT
RRAMP
RCSF
ISENSE
CS
RCS
CCSF
Copyright © 2016, Texas Instruments Incorporated
Figure 8-8. Slope Compensation Circuit
8.3.10 Voltage Mode
In certain applications, voltage mode control may be a preferred control strategy for a variety of reasons. Voltage
mode control is easily executable with any current mode controller, especially the UCC28C4x-Q1 family
members. Implementation requires generating a 0-V to 0.9-V sawtooth shaped signal to input to the current
sense pin (CS) which is also one input to the PWM comparator. This is compared to the divided down error
amplifier output voltage at the other input of the PWM comparator. As the error amplifier output is varied, it
intersects the sawtooth waveform at different points in time, thereby generating different pulse widths. This is a
straightforward method of linearly generating a pulse whose width is proportional to the error voltage.
Implementation of voltage mode control is possible by using a fraction of the oscillator timing capacitor (C CT)
waveform. This can be divided down and fed to the current sense pin as shown in Figure 8-9. The oscillator
timing components must be selected to approximate as close to a linear sawtooth waveform as possible.
Although exponentially charged, large values of timing resistance and small values of timing capacitance help
approximate a more linear shaped waveform. A small transistor is used to buffer the oscillator timing
components from the loading of the resistive divider network.
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VREF
RRT
2N2222
RT/CT
CS
CCT
Figure 8-9. Current Mode PWM Used as a Voltage Mode PWM
8.4 Device Functional Modes
8.4.1 Normal Operation
During normal operating mode, the controller can be used in peak current mode or voltage mode control. When
the converter is operating in peak current mode, the controller regulates the converter's peak current and duty
cycle. When used in voltage mode control, the controller regulates the power converter's duty cycle. The
regulation of the system's peak current and duty cycle can be achieved with the use of the integrated error
amplifier and external feedback circuitry.
8.4.2 UVLO Mode
During the system start-up, VDD voltage starts to rise from 0 V. Before the VDD voltage reaches its
corresponding turnon threshold, thedeviceis operating in UVLO mode. In this mode, the VREF pin voltage is not
generated. When VDD is above 1 V and below the turnon threshold, the VREF pin is actively pulled low. This
way, VREF can be used as a logic signal to indicate UVLO mode. If the bias voltage to VDD drops below the
UVLO-OFF threshold, the PWM switching stops and VREF returns to 0 V. The device can be restarted by
applying a voltage greater than the UVLO-ON threshold to the VDD pin.
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9 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification, and TI
does not warrant its accuracy or completeness. TI’s customers are responsible for determining
suitability of components for their purposes. Customers should validate and test their design
implementation to confirm system functionality.
9.1 Application Information
The UCC28C4x-Q1 controllers are peak current mode pulse width modulators. These controllers have an
onboard amplifier and can be used in isolated and nonisolated power supply designs. There is an onboard totem
pole gate driver capable of delivering 1 A of peak current. This is a high-speed PWM capable of operating at
switching frequencies up to 1 MHz. Figure 9-1 shows a typical off-line application.
Figure 9-1. Typical Off-Line Application
9.2 Typical Application
A typical application for the UCC28C42-Q1 in an off-line flyback converter is shown in Figure 9-2. The
UCC28C42-Q1 uses an inner current control loop that contains a small current sense resistor which senses the
primary inductor current ramp. This current sense resistor transforms the inductor current waveform to a voltage
signal that is input directly into the primary side PWM comparator. This inner loop determines the response to
input voltage changes. An outer voltage control loop involves comparing a portion of the output voltage to a
reference voltage at the input of an error amplifier. When used in an off-line isolated application, the voltage
feedback of the isolated output is accomplished using a secondary-side error amplifier and adjustable voltage
reference, such as the TL431A-Q1 or TL431B-Q1. The error signal crosses the primary to secondary isolation
boundary using an opto-isolator whose collector is connected to the VREF pin and the emitter is connected to
FB. The outer voltage control loop determines the response to load changes.
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D CLAMP
~
VIN = 85 VAC
to 265 VAC
±
D BRIDGE
CSS
CSNUB
10 nF
RSNUB
50 k
DOUT
+
CIN
180µF
~
RSTART
420k
NP
DBIAS
NS
COUT
2200µF
RVDD
22
VOUT
12 V,
4A
NA
CVDD
120µF
RSS
LP =1. 5 mH
NP:NS = 10
NP:NA = 10
UCC28C42-Q1
R COMPp
10 k
C COMPp
10 nF
RRT
15. 4 k
COMP
VREF
8
2 FB
VDD
7
3 CS
OUT
6
4 RT/CT
GND
5
RG
10
QSW
DZ CVDDbp
18 V 0. 1 µF
CCT
1000pF
CRAMP
10 nF
RRAMP
24.9 k
1
CVREF
1 µF
R BLEEDER
10 k
RCS
0. 75
RLED
1.3 k
RCSF
3. 8 k
RTLbias
1k
R DIS
2.49 k
CCSF
100 pF
RP
Not Populated
OPTOCOUPLER
10 V
RFBG
4. 99 k
RFBU
9. 53 k
R COMPz C COMPz
88. 7 k
0. 01 µF
ROPTO
1k
TL431A-Q1
(TL431B-Q1)
RFBB
2. 49 k
Copyright © 2016, Texas Instruments Incorporated
Figure 9-2. Typical Application Design Schematic
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9.2.1 Design Requirements
Table 9-1 shows a typical set of performance requirements for an off-line flyback converter capable of providing
48 W at 12-V output voltage from a universal AC input. The design uses peak primary current control in a
continuous current mode PWM converter.
Table 9-1. Design Parameters
PARAMETER
VIN
Input Voltage
TEST CONDITIONS
fLINE
Line Frequency
VOUT
Output Voltage
IVOUT(min) ≤ IVOUT ≤ IVOUT(max)
IVOUT(min) ≤ IVOUT ≤ IVOUT(max)
VRIPPLE
Output Ripple Voltage
IVOUT
Output Current
fSW
Switching Frequency
η
Efficiency
MIN
NOM
MAX
UNIT
85
115/230
265
VRMS
47
50/60
63
Hz
11.75
12
12.25
V
0
4
100
110
mVpp
A
kHz
85%
9.2.2 Detailed Design Procedure
This procedure outlines the steps to design an off-line universal input continuous current mode (CCM) flyback
converter using the UCC28C42-Q1. See Figure 9-2 for component names referred to in the design procedure.
9.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the UCC28C4x-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
9.2.2.2 Input Bulk Capacitor and Minimum Bulk Voltage
Bulk capacitance may consist of one or more capacitors connected in parallel, often with some inductance
between them to suppress differential-mode conducted noise. The value of the input capacitor sets the minimum
bulk voltage; setting the bulk voltage lower by using minimal input capacitance results in higher peak primary
currents leading to more stress on the MOSFET switch, the transformer, and the output capacitors. Setting the
bulk voltage higher by using a larger input capacitor results in higher peak current from the input source and the
capacitor itself is physically larger. Compromising between size and component stresses determines the
acceptable minimum input voltage. The total required value for the primary-side bulk capacitance (C IN) is
selected based upon the power level of the converter (POUT), the efficiency target (η), the minimum input voltage
(VIN(min)), and is chosen to maintain an acceptable minimum bulk voltage level (VBULK(min)), using Equation 3.
CIN =
2 × PIN × F0.25 +
VBULK (min )
1
× arcsin F
GG
N
¾2 × VIN (min )
2
2
k2 × VIN
(min ) F VBULK (min ) o × fLINE (min )
(3)
where
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•
SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
VIN(min) is the RMS value of the minimum AC input voltage (85 VRMS) whose minimum line frequency is
denoted as fLINE(min), equal to 47 Hz
Based on Equation 3, to achieve a minimum bulk voltage of 75 V, assuming 85% converter efficiency, the bulk
capacitor must be larger than 126 µF; 180 µF was chosen for the design, taking into consideration component
tolerances and efficiency estimation.
9.2.2.3 Transformer Turns Ratio and Maximum Duty Cycle
The transformer design starts with selecting a suitable switching frequency for the given application. The
UCC28C42 is capable of switching up to 1 MHz but considerations such as overall converter size, switching
losses, core loss, system compatibility, and interference with communication frequency bands generally
determine an optimum frequency that should be used. For this off-line converter, the switching frequency (fSW) is
selected to be 110 kHz as a compromise to minimize the transformer size and the EMI filter size, and still have
acceptable losses.
The transformer primary to secondary turns ratio (N PS) can be selected based on the desired MOSFET voltage
rating and the secondary diode voltage rating. Because the maximum input voltage is 265 VRMS, the peak bulk
input voltage can be calculated as shown in Equation 4.
VBULK (max ) = ¾2 × VIN (max ) N 375 V
(4)
To minimize the cost of the system, a readily available 650V MOSFET is selected. Derating the maximum
voltage stress on the drain to 80% of its rated value and allowing for a leakage inductance voltage spike of up to
30% of the maximum bulk input voltage, the reflected output voltage must be less than 130 V as shown in
Equation 5.
VREFLECTED
0.8 u VDS(rated) 1.3 u VBULK(max)
130.2 V
(5)
The maximum primary to secondary transformer turns ratio (NPS) for a 12 V output can be selected as
NPS =
VREFLECTED
= 10.85
VOUT
(6)
A turns ratio of NPS = 10 is used in the design example.
The auxiliary winding is used to supply bias voltage to the UCC28C42-Q1. Maintaining the bias voltage above
the VDD minimum operating voltage after turnon is required for stable operation. The minimum VDD operating
voltage for the UCC28C42-Q1 version of the controller is 10 V. The auxiliary winding is selected to support a 12
V bias voltage so that it is above the minimum operating level but still keeps the losses low in the IC. The
primary to auxiliary turns ratio (NPA) can be calculated from Equation 7:
NPA = NPS ×
VOUT
= 10
VBIAS
(7)
The output diode experiences a voltage stress that is equal to the output voltage plus the reflected input voltage:
VDIODE =
VBULK :max ;
+ VOUT = 49.5 V
NPS
(8)
TI recommends a Schottky diode with a rated blocking voltage greater than 60 V to allow for voltage spikes due
to ringing. The forward voltage drop (VF) of this diode is estimated to be equal to 0.6 V
To avoid high peak currents, the flyback converter in this design operates in continuous conduction mode. Once
N PS is determined, the maximum duty cycle (D MAX) can be calculated using the transfer function for a CCM
flyback converter:
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DMAX
1
VOUT + VF
p×l
p
=l
NPS
1 F DMAX
VBULK :min ;
DMAX
NPS u VOUT
VF
VBULK(min) NPS u VOUT
VF
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(9)
0.627
(10)
Because the maximum duty cycle exceeds 50%, and the design is an off-line (AC-input) application, the
UCC28C42-Q1 is best suited for this application.
9.2.2.4 Transformer Inductance and Peak Currents
For this design example, the transformer magnetizing inductance is selected based upon the CCM condition. An
inductance value that allows the converter to stay in CCM over a wider operating range before transitioning into
discontinuous current mode is used to minimize losses due to otherwise high currents and also to decrease the
output ripple. The design of the transformer in this example sizes the inductance so the converter enters CCM
operation at approximately 10% load and minimum bulk voltage to minimize output ripple.
The inductor (LP) for a CCM flyback can be calculated using Equation 11.
2
LP =
1
×
2
2
NPS × VOUT
p
VBULK :min ; + NPS × VOUT
0.1 × PIN × fSW
kVBULK :min ; o × l
(11)
where
•
•
PIN is estimated by dividing the maximum output power (POUT) by the target efficiency (η)
fSW is the switching frequency of the converter
For the UCC28C42-Q1 the switching frequency is equal to the oscillator frequency and is set to 110 kHz.
Selecting f SW to be 110 kHz provides a good compromise between size of magnetics, switching losses, and
places the first harmonic below the 150-kHz lower limit of EN55022. Therefore, the transformer inductance must
be approximately 1.8 mH. A 1.5 mH inductance is chosen as the magnetizing inductance, L P, value for this
design.
Based on calculated inductor value and the switching frequency, the current stress of the MOSFET and output
diode can be calculated.
The peak current in the primary-side MOSFET of a CCM flyback can be calculated as shown in Equation 12.
IPK MOSFET
NPS × VOUT
VBULK (min ) VBULK :min ; + :NPS × VOUT ;
PIN
=
+n
×
r
NPS × VOUT
2 × Lm
fSW
VBULK :min ; ×
VBULK :min ; + :NPS × VOUT ;
(12)
The MOSFET peak current is 1.36 A. The RMS current of the MOSFET is calculated to be 0.97 A as shown in
Equation 13. Therefore, IRFB9N65A is selected to be used as the primary-side switch.
DMAX 2 × IPK MOSFET × VBULK (min )
VBULK (min ) 2
DMAX 3
p FF
IRM S MOSFET = ¨
×l
G + kDMAX × IPK MOSFET 2 o
LP × fSW
LP × fSW
3
(13)
The output diode peak current is equal to the MOSFET peak current reflected to the secondary side.
IPK DIODE = NPS × IPK MOSFET = 13.634 A
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The diode average current is equal to the total output current (4 A) combined with a required 60-V rating and
13.6-A peak current requirement, a 48CTQ060-1 is selected for the output diode.
9.2.2.5 Output Capacitor
The total output capacitance is selected based upon the output voltage ripple requirement. In this design, 0.1%
voltage ripple is assumed. Based on the 0.1% ripple requirement, the capacitor value can be selected using
Equation 15.
NPS × VOUT
VBULK :min ; + NPS × VOUT
= 1865 JF
0.001 × VOUT × fSW
IOUT ×
COUT R
(15)
To design for device tolerances, a 2200-µF capacitor was selected.
9.2.2.6 Current Sensing Network
The current sensing network consists of the primary-side current sensing resistor (R CS), filtering components R
and CCSF, and optional RP. Typically, the direct current sense signal contains a large amplitude leading edge
spike associated with the turnon of the main power MOSFET, reverse recovery of the output rectifier, and other
factors including charging and discharging of parasitic capacitances. Therefore, C CSF and R CSF form a low-pass
filter that provides immunity to suppress the leading edge spike. For this converter, CCSF is chosen to be 100 pF.
CSF
Without R P, R CS sets the maximum peak current in the transformer primary based on the maximum amplitude of
the CS pin, which is specified to be 1 V. To achieve 1.36-A primary side peak current, a 0.75-Ω resistor is chosen
for RCS.
The high current sense threshold of CS helps to provide better noise immunity to the system but also results in
higher losses in the current sense resistor. These current sense losses can be minimized by injecting an offset
voltage into the current sense signal using R P. R P and R CSF form a resistor divider network from the current
sense signal to the device’s reference voltage (V VREF) which adds an offset to the current sense voltage. This
technique still achieves current mode control with cycle-by-cycle over-current protection. To calculate required
offset value (VOFFSET), use Equation 16.
VOFFSET =
R CSF
× VREF
R CSF + R P
(16)
Once RP is added, adjust the RCS accordingly.
9.2.2.7 Gate Drive Resistor
R G is the gate driver resistor for the power switch (Q SW). The selection of this resistor value must be done in
conjunction with EMI compliance testing and efficiency testing. Using a larger resistor value for R G slows down
the turnon and turnoff of the MOSFET. A slower switching speed reduces EMI but also increases the switching
loss. A tradeoff between switching loss and EMI performance must be carefully performed. For this design, a
10‑Ω resistor was chosen for the gate drive resistor.
9.2.2.8 VREF Capacitor
A precision 5-V reference voltage performs several important functions. The reference voltage is divided down
internally to 2.5 V and connected to the error amplifier’s noninverting input for accurate output voltage regulation.
Other duties of the reference voltage are to set internal bias currents and thresholds for functions such as the
oscillator upper and lower thresholds. Therefore, the reference voltage must be bypassed with a ceramic
capacitor. A 1-µF, 16-V ceramic capacitor was selected for this converter. Placement of this capacitor on the
physical printed-circuit board layout must be as close as possible to the respective VREF and GND pins.
9.2.2.9 RT/CT
The internal oscillator uses a timing capacitor (C CT) and a timing resistor (R RT) to program the oscillator
frequency and maximum duty cycle. The operating frequency can be programmed based the curves in Figure
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7-1, where the timing resistor can be found once the timing capacitor is selected. It is best for the timing
capacitor to have a flat temperature coefficient, typical of most COG or NPO type capacitors. For this converter,
15.4 kΩ and 1000 pF were selected for RRT and CCT to operate at 110-kHz switching.
9.2.2.10 Start-Up Circuit
At start-up, thedevicegets its power directly from the high-voltage bulk, through a high-voltage resistor (R START).
The selection of the start-up resistor is the tradeoff between power loss and start-up time. The current flowing
through R START at the minimum input voltage must be higher than the VDD current under UVLO conditions (100
µA at its maximum value). A resistance of 420-kΩ was chosen for RSTART, providing 250 µA of start-up current at
low-line conditions. The start-up resistor is physically comprised of two 210-kΩ resistors in series to meet the
high voltage requirements and power rating at high-line.
After VDD is charged up above the UVLO-ON threshold, the UCC28C42-Q1 starts to consume full operating
current. The VDD capacitor is required to provide enough energy to prevent its voltage from dropping below the
UVLO-OFF threshold during start-up, before the output is able to reach its regulated level. A large bulk
capacitance would hold more energy but would result in slower start-up time. In this design, a 120-µF capacitor
is chosen to provide enough energy and maintain a start-up time of approximately 7 seconds. For faster start-up,
the bulk capacitor value may be decreased or the RSTART resistor modified to a lower value.
9.2.2.11 Voltage Feedback Compensation
Feedback compensation, also called closed-loop control, can reduce or eliminate steady state error, reduce the
sensitivity of the system to parametric changes, change the gain or phase of a system over some desired
frequency range, reduce the effects of small signal load disturbances and noise on system performance, and
create a stable system from an unstable system. A system is stable if its response to a perturbation is that the
perturbation eventually dies out. A peak current mode flyback uses an outer voltage feedback loop to stabilize
the converter. To adequately compensate the voltage loop, the open-loop parameters of the power stage must
be determined.
9.2.2.11.1 Power Stage Poles and Zeroes
The first step in compensating a fixed frequency flyback is to verify if the converter is continuous conduction
mode (CCM) or discontinuous conduction mode (DCM). If the primary inductance (L P) is greater than the
inductance for DCM or CCM boundary mode operation, called the critical inductance (L Pcrit), then the converter
operates in CCM:
LP > LPcrit , then CCM
LPcrit =
(17)
2
R OUT × :NPS ;2
VIN
p
×l
2 × fSW
VIN + VOUT × NPS
(18)
For the entire input voltage range, the selected inductor has a value larger than the critical inductor. Therefore,
the converter operates in CCM and the compensation loop requires design based on CCM flyback equations.
The current-to-voltage conversion is done externally with the ground-referenced R CS and the internal 2R/R
resistor divider which sets up the internal current sense gain, ACS = 3. The exact value of these internal resistors
is not critical but thedeviceprovides tight control of the resistor divider ratio, so regardless of the actual resistor
value variations their relative value to each other is maintained.
The DC open-loop gain (G O) of the fixed-frequency voltage control loop of a peak current mode control CCM
flyback converter shown in Equation 19 is approximated by first using the output load (R OUT), the primary to
secondary turns ratio (NPS), and the maximum duty cycle (D) as calculated in Equation 20.
GO =
28
R OUT × NPS
1
×
2
:1 F D;
R CS × ACS
+ :2 × M; + 1
RL
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In Equation 19, D is calculated with Equation 20, τ L is calculated with Equation 21, and M is calculated with
Equation 22.
D=
NPS × VOUT
VBULKmin + :NPS × VOUT ;
RL =
M=
(20)
2 × LP × fSW
R OUT × :NPS ;2
(21)
VOUT × NPS
VBULKmin
(22)
For this design, a converter with an output voltage (V OUT) of 12 V, and 48 W relates to an output load (R OUT)
equal to 3 Ω at full load. With a maximum duty cycle of 0.627, a current sense resistance of 0.75 Ω, and a
primary to secondary turns-ratio of 10, the open-loop gain calculates to 3.082 or 9.776 dB.
A CCM flyback has two zeroes that are of interest. The ESR and the output capacitance contribute a left-half
plane zero (ω ESRz) to the power stage, and the frequency of this zero (f ESRz), are calculated with Equation 23
and Equation 24.
XESRz =
R ESR
1
× COUT
(23)
1
fESRz =
2 × N × R ESR × COUT
(24)
The fESRz zero for an output capacitance of 2200 µF and a total ESR of 43 mΩ is located at 1.682 kHz.
CCM flyback converters have a zero in the right-half plane (RHP) in their transfer function. A RHP zero has the
same 20 dB per decade rising gain magnitude with increasing frequency just like a left-half plane zero, but it
adds a 90° phase lag instead of lead. This phase lag tends to limit the overall loop bandwidth. The frequency
location (fRHPz) of the RHP zero (ωRHPz) is a function of the output load, the duty cycle, the primary inductance (L
P), and the primary to secondary side turns ratio (NPS).
XRHPz =
fRHPz =
R OUT × :1 F D;2 × :NPS ;2
LP × D
(25)
R OUT × :1 F D;2 × :NPS ;2
2 × N × LP × D
(26)
The right-half plane zero frequency increases with higher input voltage and lighter load. Generally, the design
requires consideration of the worst case of the lowest right-half plane zero frequency and the converter must be
compensated at the minimum input and maximum load condition. With a primary inductance of 1.5 mH, at 75-V
DC input, the RHP zero frequency (fRHPz) is equal to 7.07 kHz at maximum duty cycle, full load.
The power stage has one dominate pole (ω P1) which is in the region of interest, located at a lower frequency (f
which is related to the duty cycle, the output load, and the output capacitance, and calculated with Equation
28. There is also a double pole placed at half the switching frequency of the converter (f P2) calculated with
Equation 30. For this example, pole fP1 is located at 40.37 Hz and fP2 is at 55 kHz.
P1);
XP1
:1 F D;3
+1+D
RL
=
R OUT × COUT
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fP1
:1 F D;3
+1+D
RL
=
2 × N × R OUT × COUT
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(28)
XP2 = N × fSW
(29)
fSW
2
(30)
fP2 =
9.2.2.11.2 Slope Compensation
Slope compensation is the large signal subharmonic instability that can occur with duty cycles that may extend
beyond 50% where the rising primary side inductor current slope may not match the falling secondary side
current slope. The subharmonic oscillation would result in an increase in the output voltage ripple and may even
limit the power handling capability of the converter.
The target of slope compensation is to achieve an ideal quality coefficient (QP), equal to 1 at half of the switching
frequency. The QP is calculated with Equation 31.
1
N × >MC × :1 F D; F 0.5?
QP =
(31)
where
•
•
D is the primary side switch duty cycle
MC is the slope compensation factor, which is defined with Equation 32
MC =
Se
+1
Sn
(32)
where
•
•
Se is the compensation ramp slope
Sn is the inductor rising slope
The optimal goal of the slope compensation is to achieve Q P = 1; upon rearranging Equation 32 the ideal value
of slope compensation factor is determined:
Mideal
1
+ 0.5
N
=
1FD
(33)
For this design to have adequate slope compensation, M C must be 2.193 when D reaches it maximum value of
0.627.
The inductor rising slope (Sn) at the CS pin is calculated with Equation 34.
Sn =
VINmin × R CS
V
= 0.038
Js
LP
(34)
The compensation slope (Se) is calculated with Equation 35.
Se = :MC F 1; × Sn = 44.74
30
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mV
Js
(35)
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The compensation slope is added into the system through R RAMP and R CSF. The C RAMP is an AC-coupling
capacitor that allows the voltage ramp of the oscillator to be used without adding an offset to the current sense;
select a value to approximate a high-frequency short circuit, such as 10 nF, as a starting point and make
adjustments if required. The R RAMP and R CSF resistors form a voltage divider from the oscillator charge slope
and this proportional ramp is injected into the CS pin to add slope compensation. Choose the value of R RAMP to
be much larger than the RRT resistor so that it does not load down the internal oscillator and result in a frequency
shift. The oscillator charge slope is calculated using the peak-to-peak voltage of the RT/CT sawtooth waveform
(VOSCpp) equal to 1.9 V, and the minimum ON time, as shown in Equation 37.
D
t ONmin =
SOSC =
fSW
(36)
VOSCpp
1.9 V
mV
=
= 333
5.7 Js
Js
t ONmin
(37)
To achieve a 44.74-mV/µs compensation slope, R CSF is calculated with Equation 38. In this design, R RAMP is
selected as 24.9 kΩ, a 3.8-kΩ resistor was selected for RCSF.
R CSF =
R RAMP
SOSC
F1
Se
(38)
It has to be noticed that due to the PN junction of the BJT transistor, it can only source current, which means the
capacitor C RAMP can only be charged, not discharged. Therefore, an extra discharge resistor R DIS is needed.
Choose RDIS to be 1/10 of the RRAMP.
9.2.2.11.3 Open-Loop Gain
Once the power stage poles and zeros are calculated and the slope compensation is determined, the power
stage open-loop gain and phase of the CCM flyback converter can be plotted as a function of frequency. The
power stage transfer function can be characterized with Equation 39.
HOPEN :s; = G0 ×
l1 +
s:f;
s:f;
p × l1 F
p
1
XESRz
XRHPz
×
s:f;
s:f;
s:f;2
1+
1
+
+
XP1
XP2 × Q P :XP2 ;2
(39)
The bode for the open-loop gain and phase can be plotted by using Equation 40.
GainOPEN :s; = 20 × log: HOPEN :s; ;
(40)
See Figure 9-3 and Figure 9-4.
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10
0
5
-45
Phase (q)
Gain (dB)
0
-5
-10
-15
-90
-135
-20
-180
-25
1
10
100
1000
frequency (Hz)
10000
100000
D001
Figure 9-3. Converter Open-Loop Bode Plot - Gain
1
10
100
1000
frequency (Hz)
10000
100000
D002
Figure 9-4. Converter Open-Loop Bode Plot Phase
9.2.2.11.4 Compensation Loop
The design of the compensation loop involves selecting the appropriate components so that the required gain,
poles, and zeros can be designed to result in a stable system over the entire operating range. There are three
distinct portions of the loop: the TL431A-Q1 (or TL431B-Q1), the opto-coupler, and the error amplifier. Each of
these stages combines with the power stage to result in a stable robust system.
For good transient response, the bandwidth of the finalized design must be as large as possible. The bandwidth
of a CCM flyback, fBW, is limited to ¼ of the RHP zero frequency, or approximately 1.77 kHz using Equation 41.
fBW =
fRHPz
4
(41)
The gain of the open-loop power stage at f BW can be calculated using Equation 40 or can be observed on the
Bode plot (Figure 9-3) and is equal to –19.55 dB and the phase at fBW is equal to –58°.
The secondary side portion of the compensation loop begins with establishing the regulated steady state output
voltage. To set the regulated output voltage, a TL431A-Q1 (or TL431B-Q1) adjustable precision shunt regulator
is ideally suited for use on the secondary side of isolated converters due to its accurate voltage reference and
internal op-amp. The resistors used in the divider from the output terminals of the converter to the TL431A-Q1
(or TL431B-Q1) REF pin are selected based upon the desired power consumption. Because the REF input
current for the TL431A-Q1 (or TL431B-Q1) is only 2 µA, selecting the resistors for a divider current (IFB_REF) of 1
mA results in minimal error. The top divider resistor (RFBU) is calculated:
R FBU =
VOUT F REFTL431
IFB _REF
(42)
The TL431A-Q1 (or TL431B-Q1) reference voltage (REFTL431) has a typical value of 2.495 V. A 9.53-kΩ resistor
is chosen for RFBU. To set the output voltage to 12 V, 2.49 kΩ is used for RFBB.
R FBB =
REFTL431
× R FBU
VOUT F REFTL431
(43)
For good phase margin, a compensator zero (f COMPz) is required and should be placed at 1/10th the desired
bandwidth:
fCOMPz =
32
fBW
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XCOMPz = 2 × N × fCOMPz
(45)
With this converter, f COMPz should be set at approximately 177 Hz. A series resistor (R COMPz) and capacitor (C
COMPz) placed across the TL431A-Q1 (or TL431B-Q1) cathode to REF sets the compensator zero location.
Setting CCOMPz to 0.01 µF, RCOMPz is calculated:
R COMPz =
XCOMPz
1
× CCOMPz
(46)
Using a standard value of 88.7 kΩ for RZ and a 0.01 µF for CZ results in a zero placed at 179 Hz.
In Figure 9-2, R TLbias provides cathode current to the TL431A-Q1 (or TL431B-Q1) from the regulated voltage
provided from the Zener diode (D REG). For robust performance, 10 mA is provided to bias the TL431A-Q1
(TL431B-Q1) by way of the 10-V Zener and a 1-kΩ resistor is used for RTLbias.
The gain of the TL431A-Q1 (or TL431B-Q1) portion of the compensation loop is calculated with Equation 47.
GTL431 :s; = lR COMPz +
1
1
p×
R FBU
s(f) × CZCOMPz
(47)
A compensation pole is required at the frequency of right half plane zero or the ESR zero, whichever is lowest.
Based previous the analysis, the right half plane zero (f RHPz) is located at 7.07 kHz and the ESR zero (f ESRz) is
at 1.68 kHz; therefore, for this design, the compensation pole must be put at 1.68 kHz. The opto-coupler
contains a parasitic pole that is difficult to characterize over frequency so the opto-coupler is set up with a pulldown resistor (R OPTO) equal to 1 kΩ, which moves the parasitic opto-coupler pole further out and beyond the
range of interest for this design.
The required compensation pole can be added to the primary side error amplifier using R COMPp and C COMPp.
Choosing RCOMPp as 10 kΩ, the required value of CCOMPp is determined using Equation 48.
CCOMPp =
1
2 × N × fESRz × R COMPp
= 9.46 nF
(48)
A 10-nF capacitor is used for CCOMPp setting the compensation pole at 1.59 kHz.
Adding a DC gain to the primary-side error amplifier may be required to obtain the required bandwidth and helps
to adjust the loop gain as needed. Using 4.99 kΩ for R FBG sets the DC gain on the error amplifier to 2. At this
point the gain transfer function of the error amplifier stage (G EA(s)) of the compensation loop can be
characterized using Equation 49.
GEA :s; = l
R COMPp
1
p×F
G
1 + s:f; × CCOMPp × R COMPp
R FBG
(49)
Using an opto-coupler whose current transfer ratio (CTR) is typically at 100% in the frequency range of interest
so that CTR = 1, the transfer function of the opto-coupler stage (GOPTO(s)) is found using Equation 50.
GOPTO (s) =
CTR × R OPTO
R LED
(50)
The bias resistor (R LED) to the internal diode of the opto-coupler and the pull-down resistor on the opto emitter
(R OPTO) sets the gain across the isolation boundary. R OPTO has already been set to 1 kΩ but the value of R LED
has not yet been determined.
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The total closed loop gain (G TOTAL(s)) is the combination of the open-loop power stage (H o(s)), the opto gain (G
OPTO(s)), the error amplifier gain (G EA(s)), and the gain of the TL431A-Q1 (or TL431B-Q1) stage (G TL431(s)), as
shown in Equation 51.
GTOTAL :s; = HOPEN :s; × GOPTO :s; × GEA :s; × GTL431 :s;
(51)
The required value for R LED can be selected to achieve the desired crossover frequency (f BW). By setting the
total loop gain equal to 1 at the desired crossover frequency and rearranging Equation 51, the optimal value for
RLED can be determined, as shown in Equation 52.
R LED Q HOPEN :s; × CTR × COPTO × GEA :s; × GTL431 :s;
(52)
A 1.3-kΩ resistor suits the requirement for RLED.
Based on the compensation loop structure, the entire compensation loop transfer function is written as Equation
53.
GCLOSED :s; = HOPEN :s; × l
×n
R COMPz
R COMPp
CTR × R OPTO
1
p×l
p×F
G
R LED
R FBG
1 + ks × CCOMPp × R COMPp o
1
A
+@
s × CCOMPz
r
R FBU
(53)
The final closed-loop bode plots are show in Figure 9-5 and Figure 9-6. The converter achieves a crossover
frequency of approximately 1.8 kHz and has a phase margin of approximately 67°.
TI recommends checking the loop stability across all the corner cases including component tolerances to ensure
system stability.
80
0
60
-45
Degrees (q)
Gain (dB)
40
20
-90
0
-135
-20
-180
-40
1
10
100
1000
frequency (Hz)
10000
100000
D003
Figure 9-5. Converter Closed-Loop Bode Plot –
Gain
34
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1
10
100
1000
frequency (Hz)
10000
100000
D001
D004
Figure 9-6. Converter Closed-Loop Bode Plot –
Phase
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9.2.3 Application Curves
Figure 9-7. Primary Side MOSFET Drain to Source
Voltage at 240-V AC Input (100 V/div)
CH1: Output Voltage AC Coupled, 200 mV/div
CH4: Output Current, 1 A/div
Figure 9-8. Primary Side MOSFET Drain to Source
Voltage at 120-V AC Input (100 V/div)
Figure 9-10. Output Voltage Ripple at Full Load
(100 mV/div)
Figure 9-9. Output Voltage During 0.9-A to 2.7-A
Load Transient
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Figure 9-11. Output Voltage Behavior at Full Load Start-up (5 V/div)
10 Power Supply Recommendations
The absolute maximum supply voltage is 20 V, including any transients that may be present. If this voltage is
exceeded, device damage is likely. This is in contrast to the predecessor bipolar devices, which could survive up
to 30 V. Thus, the supply pin must be decoupled as close to the GND pin as possible. Also, because no clamp is
included in the device, the supply pin must be protected from external sources which could exceed the 20-V
level.
To prevent false triggering due to leading edge noises, an RC current sense filter may be required on CS. Keep
the time constant of the RC filter well below the minimum on-time pulse width.
To prevent noise problems with high-speed switching transients, bypass VREF to ground with a ceramic
capacitor close to the device package. A minimum of 0.1-µF ceramic capacitor is required. Additional VREF
bypassing is required for external loads on the reference. An electrolytic capacitor may also be used in addition
to the ceramic capacitor.
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11 Layout
11.1 Layout Guidelines
11.1.1 Precautions
Careful layout of the printed board is a necessity for high-frequency power supplies. As the device-switching
speeds and operating frequencies increase, the layout of the converter becomes increasingly important.
This 8-pin device has only a single ground for the logic and power connections. This forces the gate-drive
current pulses to flow through the same ground that the control circuit uses for reference. Thus, the interconnect
inductance must be minimized as much as possible. One implication is to place the device (gate driver) circuitry
close to the MOSFET it is driving. This can conflict with the need for the error amplifier and the feedback path to
be away from the noise generating components.
The single most critical item in a PWM controlled printed-circuit board layout is the placement of the timing
capacitor. While both the supply and reference bypass capacitor locations are important, the timing capacitor
placement is far more critical. Any noise spikes on the C CT waveform due to lengthy printed circuit trace
inductance or pick-up noise from being in proximity to high power switching noise causes a variety of operational
problems. Dilemmas vary from incorrect operating frequency caused by pre-triggering the oscillator due to noise
spikes to frequency jumping with varying duty cycles, also caused by noise spikes. The placement of the timing
capacitor must be treated as the most important layout consideration. Keep PC traces as short as possible to
minimize added series inductance.
11.1.2 Feedback Traces
Try to run the feedback trace as far from the inductor and noisy power traces as possible. You would also like the
feedback trace to be as direct as possible and somewhat thick. These two sometimes involve a trade-off, but
keeping it away from EMI and other noise sources is the more critical of the two. If possible, run the feedback
trace on the side of the PCB opposite of the inductor with a ground plane separating the two.
11.1.3 Bypass Capacitors
When using a low value ceramic bypass capacitor, it must be placed as close to the VDD pin of the device as
possible. This eliminates as much trace inductance effects as possible and give the internal device rail a cleaner
voltage supply. Using surface mount capacitors also reduces lead length and lessens the chance of noise
coupling into the effective antenna created by through-hole components.
11.1.4 Compensation Components
For best stability, external compensation components must be placed close to the IC. Keep FB lead length as
short as possible and FB stray capacitance as small as possible. TI recommends surface mount components
here as well for the same reasons discussed for the filter capacitors. These must not be placed very close to
traces with high switching noise.
11.1.5 Traces and Ground Planes
Make all of the power (high current) traces as short, direct, and thick as possible. It is good practice on a
standard PCB board to make the traces an absolute minimum of 15 mils (0.381 mm) per ampere. The inductor,
output capacitors, and output diode must be as close to each other possible. This helps reduce the EMI radiated
by the power traces due to the high switching currents through them. This also reduces lead inductance and
resistance as well, which in turn reduces noise spikes, ringing, and resistive losses that produce voltage errors.
The grounds of the IC, input capacitors, output capacitors, and output diode, if applicable, must be connected
close together directly to a ground plane. It would also be a good idea to have a ground plane on both sides of
the PCB. This reduces noise as well by reducing ground loop errors as well as by absorbing more of the EMI
radiated by the inductor. For multi-layer boards with more than two layers, a ground plane can be used to
separate the power plane, where the power traces and components are, and the signal plane, where the
feedback and compensation and components are, for improved performance. On multi-layer boards the use of
vias is required to connect traces and different planes. It is good practice to use one standard via per 200 mA of
current if the trace conducts a significant amount of current from one plane to the other.
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Arrange the components so that the switching current loops curl in the same direction. Due to the way switching
regulators operate, there are two power states. One state when the switch is ON and one when the switch is
OFF. During each state there is a current loop made by the power components that are currently conducting.
Place the power components so that during each of the two states the current loop is conducting in the same
direction. This prevents magnetic field reversal caused by the traces between the two half-cycles and reduces
radiated EMI.
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11.2 Layout Example
Figure 11-1. UCC28C4x-Q1 Layout Example
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12 Device and Documentation Support
12.1 Device Support
12.1.1 Development Support
12.1.1.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the UCC28C4x-Q1 devices with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
12.2 Documentation Support
12.2.1 Related Documentation
(UCC28C4x-Q1 Technical Documents)
12.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 12-1. Related Links
PARTS
PRODUCT FOLDER
SAMPLE & BUY
TECHNICAL
DOCUMENTS
TOOLS &
SOFTWARE
SUPPORT &
COMMUNITY
UCC28C40-Q1
Click here
Click here
Click here
Click here
Click here
UCC28C41-Q1
Click here
Click here
Click here
Click here
Click here
UCC28C42-Q1
Click here
Click here
Click here
Click here
Click here
UCC28C43-Q1
Click here
Click here
Click here
Click here
Click here
UCC28C44-Q1
Click here
Click here
Click here
Click here
Click here
UCC28C45-Q1
Click here
Click here
Click here
Click here
Click here
12.4 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
12.5 Trademarks
TI E2E™ is a trademark of Texas Instruments.
WEBENCH® is a registered trademark of Texas Instruments.
All trademarks are the property of their respective owners.
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UCC28C45-Q1
UCC28C40-Q1, UCC28C41-Q1, UCC28C42-Q1, UCC28C43-Q1, UCC28C44-Q1, UCC28C45-Q1
www.ti.com
SLUSA12F – DECEMBER 2009 – REVISED OCTOBER 2020
12.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
12.7 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Copyright © 2020 Texas Instruments Incorporated
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Product Folder Links: UCC28C40-Q1 UCC28C41-Q1 UCC28C42-Q1 UCC28C43-Q1 UCC28C44-Q1
UCC28C45-Q1
41
PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UCC28C40QDRQ1
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
28C40Q
UCC28C41QDRQ1
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
28C41Q
UCC28C42QDRQ1
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
28C42Q
UCC28C43QDRQ1
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
28C43Q
UCC28C44QDRQ1
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
28C44Q
UCC28C45QDRQ1
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
28C45Q
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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