UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
UCCx8C4x BiCMOS Low-Power Current-Mode PWM Controller
In addition, lower startup voltage versions of 7
V are offered as UCCx8C40 and UCCx8C41 for
use in battery systems. The UCC28C4x series is
specified for operation from –40°C to 125°C, and the
UCC38C4x series is specified for operation from 0°C
to 85°C.
1 Features
•
•
•
•
•
•
•
•
•
•
Enhanced replacement for UCx84x and UCx84xA
family with pin-to-pin compatibility
Operating frequency: 1 MHz Maximum
50-μA Startup Current, 100-μA Maximum
Low operating current: 2.3 mA (at fOSC = 52 kHz)
Fast, cycle-by-cycle overcurrent limiting: 35-ns
Peak driving current: ±1-A. Rail-to-rail output:
– 25-ns rise time
– 20-ns fall time
±1% accurate 2.5-V error amplifier reference
Trimmed oscillator discharge current
Undervoltage lockout protections
VSSOP-8 package minimizes board space
Providing necessary features to control fixed
frequency, peak current-mode power supplies, this
family offers the following performance advantages.
The device offers high-frequency operation up to
1 MHz, suitable for high speed applications. The
trimmed discharge current enables more precise
programming of the maximum duty cycle and deadtime limit when compared to the UCCx8C4x family.
Reduced start-up and operating currents minimizes
start-up loss and low operating power consumption
for improved efficiency. The device also features a
fast current-sense-to-output delay time of 35 ns for
superior overload protection at the power switch, and
a ±1-A peak output current capability with improved
rise and fall times for driving large external MOSFETs
directly.
2 Applications
•
•
•
Switch-mode power supplies
General purpose single-ended DC-DC or off-line
isolated power converters
Board mount power modules
3 Description
The UCCx8C4x family is offered in 8-pin VSSOP
(DGK) and 8-pin SOIC (D) packages.
The UCCx8C4x family are high-performance,
current-mode PWM controllers. The UCCx8C4x
is an enhanced BiCMOS version with pin-for-pin
compatibility to the industry standard UCx84xA
family and UCx84x family of PWM controllers. The
BiCMOS technology offers lower power consumption
to improve efficiency as well as faster current sense
and oscillator frequency.
Device Information
PART NUMBER
UCC28C4x
UCC38C4x
(1)
PACKAGE(1)
BODY SIZE (NOM)
SOIC (8)
3.91 mm × 4.90 mm
VSSOP (8)
3.00 mm × 3.00 mm
For all available packages, see the orderable addendum at
the end of the data sheet.
VIN
VOUT
VDD
OUT
VREF
CS
UCC28C43
FB
RT/CT
GND
COMP
Copyright © 2016, Texas Instruments Incorporated
Simplified Application
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
www.ti.com
Table of Contents
1 Features............................................................................1
2 Applications..................................................................... 1
3 Description.......................................................................1
4 Revision History.............................................................. 2
5 Device Comparison Table...............................................3
6 Pin Configuration and Functions...................................4
7 Specifications.................................................................. 5
7.1 Absolute Maximum Ratings........................................ 5
7.2 ESD Ratings............................................................... 5
7.3 Recommended Operating Conditions.........................5
7.4 Thermal Information....................................................6
7.5 Electrical Characteristics.............................................6
7.6 Typical Characteristics................................................ 8
8 Detailed Description......................................................12
8.1 Overview................................................................... 12
8.2 Functional Block Diagram......................................... 13
8.3 Feature Description...................................................13
8.4 Device Functional Modes..........................................21
9 Application and Implementation.................................. 22
9.1 Application Information............................................. 22
9.2 Typical Application.................................................... 24
9.3 Power Supply Recommendations.............................36
9.4 Layout....................................................................... 37
10 Device and Documentation Support..........................40
10.1 Device Support....................................................... 40
10.2 Documentation Support.......................................... 40
10.3 Receiving Notification of Documentation Updates..40
10.4 Support Resources................................................. 40
10.5 Trademarks............................................................. 41
10.6 Electrostatic Discharge Caution..............................41
10.7 Glossary..................................................................41
11 Mechanical, Packaging, and Orderable
Information.................................................................... 41
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (January 2017) to Revision H (September 2022)
Page
• Changed -40°C to 105°C to -40°C to 125°C, and 0°C to 70°C to 0°C to 85°C...................................................1
• Removed PDIP package from Device Information............................................................................................. 1
• Updated TJ range in Device Comparison Table..................................................................................................3
• Removed PDIP package from Pin Configuration................................................................................................4
• Removed PDIP package from Absolute Maximum Table................................................................................... 5
• Updated Total Power Dissipation values in Absolute Maximum Table............................................................... 5
• Added VREF maximum continuous voltage from external circuitry in Recommended Operating Conditions......5
• Updated TJ max values in Recommended Operating Conditions Table.............................................................5
• Updated all Thermal Resistance Numbers in Thermal Information.................................................................... 6
• Updated Electrical Characteristics section ........................................................................................................ 6
• Corrected a drawing error of OUT pin high-side FET connection.....................................................................13
Changes from Revision F (August 2016) to Revision G (January 2017)
Page
• Changed VREFLECTED equation. ....................................................................................................................... 25
• Changed DMAX equation. ................................................................................................................................. 25
2
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
5 Device Comparison Table
UVLO
Junction
Temperature
(TJ) (°C)
Turn on at 14.5 V
Turn off at 9 V
for off-line applications
Turn on at 8.4 V
Turn off at 7.6 V
for dc/dc applications
Turn on at 7 V
Turn off at 6.6 V
for battery applications
UCC28C42
UCC28C43
UCC28C40
–40 to 125
UCC38C42
UCC38C43
UCC38C40
0 to 85
UCC28C44
UCC28C45
UCC28C41
–40 to 125
UCC38C44
UCC38C45
UCC38C41
0 to 85
Copyright © 2022 Texas Instruments Incorporated
Maximum
duty cycle
100%
50%
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3
UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
6 Pin Configuration and Functions
COMP
1
8
VREF
COMP
1
8
VREF
FB
2
7
VDD
FB
2
7
VDD
CS
3
6
OUT
CS
3
6
OUT
RT/CT
4
5
GND
RT/CT
4
5
GND
Not to scale
Not to scale
Figure 6-1. D Package 8-Pin SOIC (Top View)
Figure 6-2. DGK Package, 8-Pin VSSOP (Top View)
Table 6-1. Pin Functions
PIN
NAME
COMP
1
DESCRIPTION
(1)
O
This pin provides the output of the error amplifier for compensation. In addition, the COMP pin is frequently
used as a control port, by utilizing a secondary-side error amplifier to send an error signal across the
secondary-primary isolation boundary through an opto-isolator. The error amplifier is internally current limited
so the user can command zero duty cycle by externally forcing COMP to GND.
CS
3
I
Primary-side current sense pin. The current sense pin is the noninverting input to the PWM comparator.
Connect to current sensing resistor. This signal is compared to a signal proportional to the error amplifier
output voltage. The PWM uses this to terminate the OUT switch conduction. A voltage ramp can be applied to
this pin to run the device with a voltage mode control configuration.
FB
2
I
This pin is the inverting input to the error amplifier. FB is used to control the power converter voltage-feedback
loop for stability. The noninverting input to the error amplifier is internally trimmed to 2.5 V ± 1%.
GND
5
—
Ground return pin for the output driver stage and the logic level controller section.
O
The output of the on-chip drive stage. OUT is intended to directly drive a MOSFET. The OUT pin in the
UCCx8C40, UCCx8C42, and UCCx8C43 is the same frequency as the oscillator, and can operate near 100%
duty cycle. In the UCCx8C41, UCCx8C44, and UCCx8C45, the frequency of OUT is one-half that of the
oscillator due to an internal T flipflop. This limits the maximum duty cycle to < 50%. Peak currents of up to 1 A
are sourced and sunk by this pin. OUT is actively held low when VDD is below the turn-on threshold.
OUT
6
RT/CT
4
I/O
Fixed frequency oscillator set point. Connect timing resistor (RRT) to VREF and timing capacitor (CCT) to GND
from this pin to set the switching frequency. For best performance, keep the timing capacitor lead to the device
GND as short and direct as possible. If possible, use separate ground traces for the timing capacitor and all
other functions. The switching frequency (fSW) of the UCCx8C40, UCCx8C42, and UCCx8C43 gate drive is
equal to fOSC; the switching frequency of the UCCx8C41, UCCx8C44, and UCCx8C45 is equal to half of the
fOSC.
VDD
7
I
Analog controller bias input that provides power to the device. Total VDD current is the sum of the quiescent
VDD current and the average OUT current. A bypass capacitor, typically 0.1 µF, connected directly to GND
with minimal trace length, is required on this pin. Additional capacitance at least 10 times greater than the gate
capacitance of the main switching FET used in the design is also required on VDD.
O
5-V reference voltage. VREF is used to provide charging current to the oscillator timing capacitor through the
timing resistor. It is important for reference stability that VREF is bypassed to GND with a ceramic capacitor
connected as close to the pin as possible. A minimum value of 0.1 µF ceramic is required. Additional VREF
bypassing is required for external loads on VREF.
VREF
(1)
4
TYPE
NO.
8
I = input, O = output, G = ground
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
7 Specifications
7.1 Absolute Maximum Ratings
Over operating free-air temperature range (unless otherwise noted) (1) (2)
MAX
UNIT
Input voltage
VDD
MIN
20
V
Input current
IVDD
30
mA
±1
A
5
µJ
Output drive current (peak)
Output energy (capacitive load), EOUT
Analog input voltage
COMP, CS, FB, RT/CT
–0.3
6.3
Output driver voltage
OUT
–0.3
20
Reference voltage
VREF
Error amplifier output sink current
COMP
Total power dissipation at TA = 25°C
V
7
10
mA
D package
72.3
DGK package
98.1
Lead temperature (soldering, 10 s), TLEAD
°C/W
300
°C
Operating junction temperature, TJ
–55
150
°C
Storage temperature, Tstg
–65
150
°C
(1)
(2)
Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Section 7.3.
Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to GND pin. Currents are positive into and negative out of the specified terminals.
7.2 ESD Ratings
VALUE
V(ESD)
(1)
(2)
Electrostatic discharge
Human-body model (HBM), per ANSI/ESDA/JEDEC
JS-001(1)
UNIT
±2500
Charged-device model (CDM), per JEDEC specification JESD22-C101(2)
V
±1500
JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 Recommended Operating Conditions
Over operating free-air temperature range (unless otherwise noted)
MIN
MAX
UNIT
VVDD
Input voltage
18
V
VOUT
Output driver voltage
18
V
VREF
Maximum continuous voltage from external circuitry
5.5
V
IOUT
Average output driver current (source and sink) (1)
200
mA
IOUT(VREF)
Reference output current (source) (1)
20
mA
TJ
(1)
Operating junction temperature(1)
UCC28C4x
–40
125
UCC38C4x
0
85
°C
TI recommends against operating the device under conditions beyond those specified in this table for extended periods of time.
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5
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
7.4 Thermal Information
UCC28C4x, UCC38C4x
THERMAL METRIC(1)
D
(SOIC)
DGK
(VSSOP)
8 PINS
8 PINS
UNIT
RθJA
Junction-to-ambient thermal resistance
128.9
176.4
°C/W
RθJC(top)
Junction-to-case (top) thermal resistance
71.7
67.3
°C/W
RθJB
Junction-to-board thermal resistance
72.3
98.1
°C/W
ψJT
Junction-to-top characterization parameter
23.4
11.1
°C/W
ψJB
Junction-to-board characterization parameter
71.5
91.5
°C/W
(1)
For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application
report.
7.5 Electrical Characteristics
VVDD = 15 V (1), RRT = 10 kΩ, CCT = 3.3 nF, CVDD = 0.1 µF and no load on the outputs, TJ = –40°C to 125 °C for the
UCC28C4x and TJ = 0°C to 85 °C for the UCC38C4x (unless otherwise noted).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
4.9
5
5.1
V
0.2
20
mV
3
25
mV
REFERENCE
VVREF
IVREF
VREF voltage, initial accuracy
TJ = 25°C, IOUT = 1 mA
Line regulation
VVDD = 12 V to 18 V
Load regulation
1 mA to 20 mA
(2)
Temperature stability
See
Total output variation
See (2)
0.2
4.82
VREF noise voltage
10 Hz to 10 kHz, TJ = 25°C, see
Long term stability
1000 hours, TJ = 125°C, see (2)
(2)
Output short circuit (source current)
0.4 mV/°C
5.18
50
30
V
µV
5
25
mV
45
55
mA
kHz
OSCILLATOR
fOSC
Initial accuracy
TJ = 25°C, see (3)
Voltage stability
12 V ≤ VVDD ≤ 18 V
Temperature stability
Amplitude
Discharge current
53
55
0.2%
1%
TJ(MIN) to TJ(MAX), see (2)
1%
2.5%
RT/CT pin peak-to-peak voltage
1.9
TJ = 25°C, VRT/CT = 2 V, see
50.5
(4)
VRT/CT = 2 V, see (4)
V
7.7
8.4
9
7.2
8.4
9.5
2.475
2.5
2.525
2.45
2.5
2.55
V
0.1
2
µA
mA
ERROR AMPLIFIER
VFB
Feedback input voltage, initial accuracy VCOMP = 2.5 V, TJ = 25°C
Feedback input voltage, total variation
VCOMP = 2.5 V
IFB
Input bias current (source current)
VFB = 5 V
AVOL
Open-loop voltage gain
2 V ≤ VOUT ≤ 4 V
Unity gain bandwidth
See (2)
Power supply rejection ratio
12 V ≤ VVDD ≤ 18 V
Output sink current
VFB = 2.7 V, VCOMP = 1.1 V
Output source current
VFB = 2.3 V, VCOMP = 5 V
PSRR
6
VOH
High-level COMP voltage
VFB = 2.7 V, RCOMP = 15 kΩ COMP to GND
VOL
Low-level COMP voltage
VFB = 2.7 V, RCOMP = 15 kΩ COMP to VREF
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V
65
90
dB
1
1.5
MHz
2
14
mA
0.5
1
mA
60
dB
VREF 0.2
V
0.1
1.1
V
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
VVDD = 15 V (1), RRT = 10 kΩ, CCT = 3.3 nF, CVDD = 0.1 µF and no load on the outputs, TJ = –40°C to 125 °C for the
UCC28C4x and TJ = 0°C to 85 °C for the UCC38C4x (unless otherwise noted).
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
2.85
3
3.15
V/V
1
1.1
CURRENT SENSE
ACS
Gain
See (5) (6)
VCS
Maximum input signal
VFB < 2.4 V
PSRR
Power supply rejection ratio
VVDD = 12 V to 18 V(2) (5)
ICS
Input bias current (source current)
0.1
2
µA
tD
CS to output delay
35
70
ns
0.9
70
V
dB
COMP to CS offset
VCS = 0 V
1.15
V
VOUT(low)
RDS(on) pulldown
ISINK = 200 mA
5.5
15
Ω
VOUT(high)
RDS(on) pullup
ISOURCE = 200 mA
10
25
Ω
tRISE
Rise tIme
TJ = 25°C, COUT = 1 nF
25
50
ns
tFALL
Fall tIme
TJ = 25°C, COUT = 1 nF
20
40
ns
15.5
OUTPUT
UNDERVOLTAGE LOCKOUT
VDDON
VDDOFF
Start threshold
Minimum operating voltage
UCCx8C42, UCCx8C44
13.5
14.5
UCCx8C43, UCCx8C45
7.8
8.4
9
UCCx8C40, UCCx8C41
6.5
7
7.5
UCCx8C42, UCCx8C44
8
9
10
UCCx8C43, UCCx8C45
7
7.6
8.2
UCCx8C40, UCCx8C41
6.1
6.6
7.1
UCCx8C42, UCCx8C43, UCCx8C40, VFB < 2.4 V
94%
96%
UCCx8C44, UCCx8C45, UCCx8C41, VFB < 2.4 V
47%
48%
V
V
PWM
DMAX
Maximum duty cycle
DMIN
Minimum duty cycle
VFB > 2.6 V
0%
CURRENT SUPPLY
ISTART-UP
Start-up current
VVDD = VDDON – 0.5 V
50
100
µA
IVDD
Operating supply current
VFB = VCS = 0 V
2.3
3
mA
(1)
(2)
(3)
(4)
(5)
(6)
Adjust VVDD above the start threshold before setting at and 15.5 V.
Specified by design. Not production tested.
Output frequencies of the UCCx8C41, UCCx8C44, and the UCCx8C45 are half the oscillator frequency.
Oscillator discharge current is measured with RRT = 10 kΩ to VREF.
Parameter measured at trip point of latch with VFB = 0 V.
Gain is defined as ACS = ΔVCOMP / ΔVCS , 0 V ≤ VCS ≤ 900 mV
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7.6 Typical Characteristics
9.5
100
10
220 pF
470 pF
1 nF
2.2 nF
4.7 nF
1
1
10
RRT Timing Resistance (k:)
100
IDISCH -- Oscillator Discharge Current -- mA
fOSC Oscillator Frequency (kHz)
1000
8.5
8.0
7.5
7.0
D001
Figure 7-1. Oscillator Frequency vs Timing
Resistance and Capacitance
9.0
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-2. Oscillator Discharge Current vs
Temperature
100
200
1.8
90
180
1.6
80
160
1.4
140
60
120
50
100
40
80
1.2
COMP to CS
70
Phase Margin -- (°)
Gain -- (dB)
GAIN
1.0
0.8
0.6
30
60
PHASE
MARGIN
20
40
10
20
0
1
10
100
1k
10 k
100 k
1M
0
10 M
f -- Frequency -- Hz
Figure 7-3. Error Amplifier Frequency Response
0.4
0.2
0.0
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
VCS = 0 V
Figure 7-4. COMP to CS Offset Voltage vs
Temperature
8
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5.05
VEAREF -- Error Amplifier Reference Voltage -- V
2.55
VREF -- Reference Voltage -- V
5.04
5.03
5.02
5.01
5.00
4.99
4.98
4.97
4.96
4.95
--50
2.54
2.53
2.52
2.51
2.50
2.49
2.48
2.47
2.46
2.45
--25
0
25
50
75
100
125
--50
--25
0
TJ -- Temperature -- °C
Figure 7-5. Reference Voltage vs Temperature
75
100
125
200
--37
IBIAS -- Error Amplifier Input Bias Current -- nA
ISC -- Reference Short Circuit Current -- mA
50
Figure 7-6. Error Amplifier Reference Voltage vs
Temperature
--35
--39
--41
--43
--45
--47
--49
--51
--53
--55
--50
150
100
50
0
--50
--100
--150
--200
--25
0
25
50
75
TJ -- Temperature -- °C
100
125
Figure 7-7. Reference Short-Circuit Current vs
Temperature
--50
15
8.8
VUVLO -- UVLO Voltage -- V
9.0
13
UVLO
ON
12
UVLO
OFF
11
10
7.2
25
50
75
100
125
UCCx8C42 and UCCx8C44
Figure 7-9. Undervoltage Lockout vs Temperature
Copyright © 2022 Texas Instruments Incorporated
UVLO
ON
7.8
7
TJ -- Temperature -- °C
125
8.0
7.4
0
100
8.2
8
--25
25
50
75
TJ -- Temperature -- °C
8.4
7.6
--50
0
8.6
9
6
--25
Figure 7-8. Error Amplifier Input Bias Current vs
Temperature
16
14
VUVLO -- UVLO Voltage -- V
25
TJ -- Temperature -- °C
7.0
--50
UVLO
OFF
--25
0
25
50
75
TJ -- Temperature -- °C
100
125
UCCx8C43 and UCCx8C45
Figure 7-10. Undervoltage Lockout vs Temperature
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9
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7.3
25
7.2
IDD -- Supply Current -- mA
UVLO
ON
VUVLO -- UVLO Voltage -- V
7.1
7.0
6.9
6.8
6.7
6.6
1-nF LOAD
20
15
10
NO LOAD
6.5
5
UVLO
OFF
6.4
6.3
--50
--25
0
25
50
75
100
0
125
0k
200 k
400 k
600 k
f -- Frequency -- Hz
TJ -- Temperature -- °C
UCCx8C40 and UCCx8C41
Figure 7-11. Undervoltage Lockout vs Temperature
800 k
1M
Figure 7-12. Supply Current vs Oscillator
Frequency
3.0
40
10% to 90%
VDD = 12 V
2.9
35
Output Rise and Fall TIme -- ns
IDD -- Supply Current -- mA
2.8
2.7
2.6
2.5
2.4
NO LOAD
2.3
2.2
tr
(1 nF)
30
tf
(1 nF)
25
20
15
2.1
10
2.0
--50
--25
0
25
50
75
100
--50
125
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
TJ -- Temperature -- °C
Figure 7-13. Supply Current vs Temperature
Figure 7-14. Output Rise Time and Fall Time vs
Temperature
100
100
CT = 220 pF
98
Maximum Duty Cycle -- %
Duty Cycle -- %
90
80
70
CT = 1 nF
0
500
1000
1500
2000
2500
f -- Frequency -- kHz
Figure 7-15. Maximum Duty Cycle vs Oscillator
Frequency
10
94
92
60
50
96
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90
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-16. Maximum Duty Cycle vs Temperature
(for part numbers with maximum 100% duty cycle)
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50
VCS_th -- Current Sense Threshold -- V
1.10
Maximum Duty Cycle - %
49
48
47
46
45
1.05
1.00
0.95
0.90
--50
--25
0
25
50
75
100
125
--50
--25
TJ -- Temperature -- °C
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-17. Maximum Duty Cycle vs Temperature
(for part numbers with maximum 50% duty cycles)
Figure 7-18. Current Sense Threshold Voltage vs
Temperature
70
tD -- CS to OUT Delay Time -- ns
65
60
55
50
45
40
35
30
--50
--25
0
25
50
75
100
125
TJ -- Temperature -- °C
Figure 7-19. Current Sense to Output Delay Time vs Temperature
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8 Detailed Description
8.1 Overview
The UCCx8C4x series of control integrated circuits provide the features necessary to implement AC-DC or
DC‑to-DC fixed-frequency current-mode control schemes with a minimum number of external components.
Protection circuitry includes undervoltage lockout (UVLO) and current limiting. Internally implemented circuits
include a start-up current of less than 100 μA, a precision reference trimmed for accuracy at the error
amplifier input, logic to ensure latched operation, a pulse-width modulation (PWM) comparator that also provides
currentlimit control, and an output stage designed to source or sink high-peak current. The output stage, suitable
for driving N-channel MOSFETs, is low when it is in the OFF state. The oscillator contains a trimmed discharge
current that enables accurate programming of the maximum duty cycle and dead time limit, making this device
suitable for high-speed applications.
Major differences between members of this series are the UVLO thresholds, acceptable ambient temperature
range, and maximum duty cycle. Typical UVLO thresholds of 14.5 V (ON) and 9 V (OFF) on the UCCx8C42
and UCCx8C44 devices make them ideally suited to off-line AC-DC applications. The corresponding typical
thresholds for the UCCx8C43 and UCCx8C45 devices are 8.4 V (ON) and 7.6 V (OFF), making them ideal
for use with regulated input voltages used in DC-DC applications. The UCCx8C40 and UCCx8C41 feature a
start-up threshold of 7 V and a turnoff threshold of 6.6 V (OFF), which makes them suitable for battery-powered
applications. The UCCx8C40, UCCx8C42, and UCCx8C43 devices operate to duty cycles approaching 100%.
The UCCx8C41, UCCx8C44, and UCCx8C45 obtain a duty cycle from 0% to 50% by the addition of an internal
toggle flip-flop, which blanks the output off every other clock cycle. The UCC28C4x series is specified for
operation from –40°C to 125°C, and the UCC38C4x series is specified for operation from 0°C to 85°C.
The UCC28C4x and UCC38C4x series are an enhanced replacement with pin-to-pin compatibility to the bipolar
UC284x, UC384x, UC284xA, and UC384xA families. The new series offers improved performance when
compared to older bipolar devices and other competitive BiCMOS devices with similar functionality. These
improvements generally consist of tighter specification limits that are a subset of the older product ratings,
maintaining drop-in capability. In new designs, these improvements can reduce the component count or enhance
circuit performance when compared to the previously available devices.
12
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8.2 Functional Block Diagram
VDD
UVLO
EN
5V
VREF
VREF
VREF Good
Logic
RT/CT
Osc
( NOTE)
2. 5 V
S
2R
+
E/A
FB
OUT
T
PWM
Latch
R
R
1V
GND
PWM
Comparator
COMP
CS
Copyright © 2016, Texas Instruments Incorporated
Toggle flip-flop used only in UCCx8C41, UCCx8C44, and UCCx8C45
8.3 Feature Description
The BiCMOS design allows operation at high frequencies that were not feasible in the predecessor bipolar
devices. First, the output stage has been redesigned to drive the external power switch in approximately half
the time of the earlier devices. Second, the internal oscillator is more robust, with less variation as frequency
increases. This faster oscillator makes this device suitable for high speed applications and the trimmed
discharge current enables precise programming of the maximum duty cycle and dead-time limit. In addition,
the current sense to output delay is kept the same 45 ns (typical) . Such a delay time in the current sense results
in superior overload protection at the power switch. The reduced start-up current of this device minimizes steady
state power dissipation in the startup resistor, and the low operating current maximizes efficiency while running,
increasing the total circuit efficiency, whether operating off-line, DC input, or battery operated circuits. These
features combine to provide a device capable of reliable, high-frequency operation.
Table 8-1. Improved Key Parameters
PARAMETER
UCCx8C4x
UCx84x
Supply current at 50 kHz
2.3 mA
11 mA
Start-up current
50 µA
1 mA
Overcurrent propagation delay
50 ns
150 ns
Reference voltage accuracy
± 1%
± 2%
Error amplifier reference voltage accuracy
± 25 mV
± 80 mV
Maximum oscillator frequency
> 1 MHz
500 kHz
25 ns
50 ns
Output rise/fall times
UVLO turn-on accuracy
±1V
± 1.5 V
Smallest package option
VSSOP-8 (MSOP-8)
SOIC-8
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8.3.1 Detailed Pin Description
8.3.1.1 COMP
The error amplifier in the UCCx8C4x family has a unity-gain bandwidth of 1.5 MHz. The COMP terminal can
both source and sink current. The error amplifier is internally current-limited, so that one can command zero duty
cycle by externally forcing COMP to GND.
8.3.1.2 FB
FB is the inverting input of the error amplifier. The noninverting input to the error amplifier is internally trimmed to
2.5 V ± 1%. FB is used to control the power converter voltage-feedback loop for stability. For best stability, keep
FB lead length as short as possible and FB stray capacitance as small as possible.
8.3.1.3 CS
The UCCx8C4x current sense input connects directly to the PWM comparator. Connect CS to the MOSFET
source current sense resistor. The PWM uses this signal to terminate the OUT switch conduction. A voltage
ramp can be applied to this pin to run the device with a voltage mode control configuration or to add slope
compensation. To prevent false triggering due to leading edge noises, an RC current sense filter may be
required. The gain of the current sense amplifier is typically 3 V/V.
8.3.1.4 RT/CT
The internal oscillator uses a timing capacitor (CCT) and a timing resistor (RRT) to program the oscillator
frequency and maximum duty cycle. The operating frequency can be programmed based the curves in Figure
7-1, where the timing resistor can be found once the timing capacitor is selected. It is best for the timing
capacitor to have a flat temperature coefficient, typical of most COG or NPO type capacitors. For this converter,
15.4 kΩ and 1000 pF were selected for RRT and CCT to operate at 110-kHz switching.
8.3.1.5 GND
GND is the signal and power returning ground. TI recommends separating the signal return path and the high
current gate driver path so that the signal is not affected by the switching current.
8.3.1.6 OUT
The high-current output stage of the UCCx8C4x has been redesigned to drive the external power switch in
approximately half the time of the earlier devices. To drive a power MOSFET directly, the totem-pole OUT driver
sinks or source up to 1 A peak of current. The OUT of the UCCx8C40, UCCx8C42, and UCCx8C43 devices
switch at the same frequency as the oscillator and can operate near 100% duty cycle. In the UCCx8C41,
UCCx8C44, and UCCx8C45, the switching frequency of OUT is one-half that of the oscillator due to an internal T
flip-flop. This limits the maximum duty cycle in the UCCx8C41, UCCx8C44, and UCCx8C45 to < 50%.
The UCCx8C4x family houses unique totem pole drivers exhibiting a 10-Ω impedance to the upper rail and a
5.5‑Ω impedance to ground, typically. This reduced impedance on the low-side switch helps minimize turnoff
losses at the power MOSFET, whereas the higher turnon impedance of the high-side switch is intended to better
match the reverse recovery characteristics of many high-speed output rectifiers. Transition times, rising and
falling edges, are typically 25 nanoseconds and 20 nanoseconds, respectively, for a 10% to 90% change in
voltage.
A low impedance MOS structure in parallel with a bipolar transistor, or BiCMOS construction, comprises the
totem-pole output structure. This more efficient utilization of silicon delivers the high peak current required along
with sharp transitions and full rail-to-rail voltage swings. Furthermore, the output stage is self-biasing, active low
during undervoltage lockout type. With no VDD supply voltage present, the output actively pulls low if an attempt
is made to pull the output high. This condition frequently occurs at initial power-up with a power MOSFET as the
driver load.
14
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8.3.1.7 VDD
VDD is the power input connection for this device. In normal operation, power VDD through a current limiting
resistor. The absolute maximum supply voltage is 20 V, including any transients that may be present. If this
voltage is exceeded, device damage is likely. This is in contrast to the predecessor bipolar devices, which could
survive up to 30 V on the input bias pin. Also, because no internal clamp is included in the device, the VDD
pin must be protected from external sources which could exceed the 20 V level. If containing the start-up and
bootstrap supply voltage from the auxiliary winding NA below 20 V under all line and load conditions can not be
achieved, use a zener protection diode from VDD to GND. Depending on the impedance and arrangement of the
bootstrap supply, this may require adding a resistor, RVDD, in series with the auxiliary winding to limit the current
into the zener as shown in Figure 8-1. Ensure that over all tolerances and temperatures, the minimum zener
voltage is higher than the highest UVLO upper turnon threshold. To prevent noise related problems, filter VDD
with a ceramic bypass capacitor to GND. The VDD pin must be decoupled as close to the GND pin as possible.
NP
NS
RSTART
To
Input
RVDD
NA
DBIAS
VDD
OUT
CVCC
DZCLAMP
CVDDbp
0.1 PF
GND
RCS
Figure 8-1. VDD Protection
Although nominal VDD operating current is only 2.3 mA, the total supply current is higher, depending on the
OUT current. Total VDD current is the sum of quiescent VDD current and the average OUT current. Knowing the
operating frequency and the MOSFET gate charge (Qg), average OUT current can be calculated from Equation
1.
IOUT = Q g × fSW
(1)
8.3.1.8 VREF
VREF is the voltage reference for the error amplifier and also for many other internal circuits in the IC. The
5-V reference tolerance is ±1% for the UCCx8C4x family. The high-speed switching logic uses VREF as the
logic power supply. The reference voltage is divided down internally to 2.5 V ±1% and connected to the error
amplifier's noninverting input for accurate output voltage regulation. The reference voltage sets the internal
bias currents and thresholds for functions such as the oscillator upper and lower thresholds along with the
overcurrent limiting threshold. The output short-circuit current is 55 mA (maximum). To avoid device over-heating
and damage, do not pull VREF to ground as a means to terminate switching. For reference stability and to
prevent noise problems with high-speed switching transients, bypass VREF to GND with a ceramic capacitor
close to the IC package. A ceramic capacitor with a minimum value of 0.1 µF is required. Additional VREF
bypassing is required for external loads on the reference. An electrolytic capacitor may also be used in addition
to the ceramic capacitor.
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8.3.2 Undervoltage Lockout
Six sets of UVLO thresholds are available with turn-on and turnoff thresholds of: (14.5 V and 9 V), (8.4 V and
7.6 V), (7 V and 6.6 V) respectively. The first set is primarily intended for off-line and 48-V distributed power
applications, where the wider hysteresis allows for lower frequency operation and longer soft-starting time of the
converter. The second group of UVLO options is ideal for high frequency DC-DC converters typically running
from a 12-VDC input. The third, and newest, set has been added to address battery powered and portable
applications. Table 8-2 shows the maximum duty cycle and UVLO thresholds by device.
Table 8-2. UVLO Options
MAXIMUM
DUTY CYCLE (%)
UVLO ON
(V)
UVLO OFF
(V)
DEVICE
NUMBER
100
100
14.5
9
UCCx8C42
8.4
7.6
UCCx8C43
100
7
6.6
UCCx8C40
50
14.5
9
UCCx8C44
50
8.4
7.6
UCCx8C45
50
7
6.6
UCCx8C41
During UVLO the IC draws less than 100 µA of supply current. After crossing the turnon threshold, the device
supply current increases to a maximum of 3 mA, typically 2.3 mA. This low start-up current allows the power
supply designer to optimize the selection of the startup resistor value to provide a more efficient design. In
applications where low component cost overrides maximum efficiency, the low run current of 2.3 mA (typical)
allows the control device to run directly through the single resistor to (+) rail, rather than requiring a bootstrap
winding on the power transformer, along with a rectifier. The start and run resistor for this case must also pass
enough current to allow driving the primary switching MOSFET, which may be a few milliamps in small devices.
< 3 mA
IVDD
< 100 µA
VOFF
VON
VVDD
Figure 8-2. UVLO ON and OFF Profile
8.3.3 ±1% Internal Reference Voltage
The BiCMOS internal reference of 2.5 V has an enhanced design, and uses production trim to allow initial
accuracy of ±1% at room temperature and ±2% over the full temperature range. This reference voltage can be
used to eliminate an external reference in applications that do not require the extreme accuracy afforded by the
additional device. This reference voltage is useful for non-isolated DC-DC applications, where the control device
is referenced to the same common as the output. It is also applicable in off-line designs that regulate on the
primary side of the isolation boundary by looking at a primary bias winding, or from a winding on the output
inductor of a buck-derived circuit.
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8.3.4 Current Sense and Overcurrent Limit
An external series resistor (RCS) senses the current and converts this current into a voltage that becomes the
input to the CS pin. The CS pin is the noninverting input to the PWM comparator. The device compares the CS
input with a signal proportional to the error amplifier output voltage. The gain of the current sense amplifier is
typically 3 V/V. The peak ISENSE current is determined using Equation 2
ISENSE =
VCS
R CS
(2)
The typical value for VCS is 1 V. A small RC filter (RCSF and CCSF) may be required to suppress switch transients
caused by the reverse recovery of a secondary side diode or equivalent capacitive loading in addition to parasitic
circuit impedances. The time constant of this filter should be considerably less than the switching period of the
converter.
Error
Amplifier
COMP
2R
R
1V
PWM
Comparator
ISENSE
RCSF
RCS
CS
CCSF
GND
Figure 8-3. Current-Sense Circuit Schematic
Cycle-by-cycle pulse width modulation performed at the PWM comparator essentially compares the error
amplifier output to the current sense input. This is not a direct volt-to-volt comparison, as the error amplifier
output network incorporates two diodes in series with a resistive divider network before connecting to the PWM
comparator. The two-diode drop adds an offset voltage that enables zero duty cycle to be achieved with a low
amplifier output. The 2R/R resistive divider facilitates the use of a wider error amplifier output swing that can be
more symmetrically centered on the 2.5-V noninverting input voltage.
The 1-V Zener diode associated with the PWM comparator input from the error amplifier is not an actual diode
in the device design, but an indication that the maximum current sense input amplitude is 1 V (typical). When
this threshold is reached, regardless of the error amplifier output voltage, cycle-by-cycle current limiting occurs,
and the output pulse width is terminated within 35 ns (typical). The minimum value for this current limit threshold
is 0.9 V with a 1.1-V maximum. In addition to the tolerance of this parameter, the accuracy of the current sense
resistor, or current sense circuitry, must be taken into account. It is advised to factor in the worst case of primary
and secondary currents when sizing the ratings and worst-case conditions in all power semiconductors and
magnetic components.
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8.3.5 Reduced-Discharge Current Variation
The oscillator design for the UCCx8C4x controllers incorporates a trimmed discharge current to accurately
program maximum duty cycle and operating frequency. In its basic operation, a timing capacitor (CCT) is charged
by a current source, formed by the timing resistor (RRT) connected to the device reference voltage (VREF).
The oscillator design incorporates comparators to monitor the amplitude of the timing capacitor voltage. The
exponentially shaped waveform charges up to a specific amplitude representing the oscillator upper threshold of
3 V. After the controller reaches this level, an internal current sink to ground turns on and the capacitor begins
to discharge. This discharge continues until the oscillator lower threshold has reached 0.7 V at which point the
current sink is turned off. Next, the timing capacitor starts charging again and a new switching cycle begins.
VREF
VDDON
VDDOFF
RRT
CCT
CCT
RT/CT
tON
tOFF
GND
tPERIOD
8.4 mA
Figure 8-4. Oscillator Circuit
While the device discharges the timing capacitor, resistor RRT continues attempting to charge CCT. It is the exact
ratio of these two currents, the discharging versus the charging current, which specifies the maximum duty cycle.
During the discharge time of CCT, the device output is always off. This represents an ensured minimum off
time of the switch, commonly referred to as dead-time. To program an accurate maximum duty cycle, use the
information provided in Maximum Duty Cycle vs Oscillator Frequency for maximum duty cycle versus oscillator
frequency. Any number of maximum duty cycles can be programmed for a given frequency by adjusting the
values of RRT and CCT. After selecting the value of RRT, find the oscillator timing capacitance using the curves in
Oscillator Frequency vs Timing Resistance and Capacitance. However, because resistors are available in more
precise increments, typically 1%, and capacitors are only available in 5% accuracy, it might be more practical to
select the closest capacitor value first and then calculate the timing resistor value.
8.3.6 Oscillator Synchronization
Synchronization is best achieved by forcing the timing capacitor voltage above the oscillator internal upper
threshold. A small resistor is placed in series with CCT to GND. This resistor serves as the input for the sync
pulse which raises the CCT voltage above the oscillator internal upper threshold. The PWM is allowed to run
at the frequency set by RRT and CCT until the sync pulse appears. This scheme offers several advantages
including having the local ramp available for slope compensation. The UCCx8C4x oscillator must be set to a
lower frequency than the sync pulse stream, typically 20 percent with a 0.5-V pulse applied across the resistor.
VREF
RRT
CCT + SYNC
CCT
RT/CT
SYNC
SYNC
50
GND
CCT
Copyright © 2016, Texas Instruments Incorporated
Figure 8-5. Oscillator Synchronization Circuit
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8.3.7 Soft-Start Timing
The soft-start timing is the technique to gradually power up the converter in a well-controlled fashion by slowly
increasing the effective duty cycle starting at zero and gradually rising. Following start-up of the PWM, the error
amplifier inverting input is low, commanding the error amplifier’s output to go high. The output stage of the
amplifier can source 1 mA typically, which is enough to drive most high impedance compensation networks, but
not enough for driving large loads quickly. Soft-start timing is achieved by charging a fairly large value, >1-µF,
capacitor (CSS) connected to the error amplifier output through a PNP transistor as shown in Figure 8-6
VREF
RSS
COMP
ZF
+
2N2907
CSS
FB
ZI
To VOUT
Figure 8-6. Soft-Start Implementation
The limited charging current of the amplifier into the capacitor translates into a dv/dt limitation on the error
amplifier output. This directly corresponds to some maximum rate of change of primary current in a current mode
controlled system as one of the PWM comparator inputs gradually rises. The values of RSS and CSS must be
selected to bring the COMP pin up at a controlled rate, limiting the peak current supplied by the power stage.
After the soft-start interval is complete, the capacitor continues to charge to VREF, effectively removing the PNP
transistor from the circuit consideration. Soft-start timing offers a different, frequently preferred function in current
mode controlled systems than it does in voltage mode control. In current mode, soft start controls the rising of
the peak switch current. In voltage mode control, soft start gradually widens the duty cycle, regardless of the
primary current or rate of ramp-up.
The purpose of resistor RSS and the diode is to remove the soft-start capacitor from the error amplifier path
during normal operation, after the soft-start period completes and the capacitor charges fully. The optional
diode in parallel with the resistor forces a soft-start period each time the PWM goes through UVLO condition
that forces VREF to go low. Without the diode, the capacitor remains charged during a brief loss of supply or
brown-out, and the device does not emable a soft-start function upon re-application of VDD.
8.3.8 Enable and Disable
There are several ways to enable or disable the UCCx8C4x devices, depending on which type of restart is
required. The two basic techniques use external transistors to either pull the error amplifier output low (< 2 VBE)
or pull the current sense input high (> 1.1 V). Application of the disable signal causes the output of the PWM
comparator to be high. The PWM latch is reset dominant so that the output remains low until the next clock
cycle after the shutdown condition at the COMP or CS pin is removed. Another choice for restart without a
soft-start period is to pull the current sense input above the cycle-by-cycle current limiting threshold. A logic level
P-channel FET from the reference voltage to the current sense input can be used.
COMP
DISABLE
Figure 8-7. Disable Circuit
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19
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
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8.3.9 Slope Compensation
With current mode control, slope compensation is required to stabilize the overall loop with duty cycles
exceeding 50%. Although not required, slope compensation also improves stability in applications using below
a 50% maximum duty cycle. Slope compensation is introduced by injecting a portion of the oscillator waveform
to the actual sensed primary current. The two signals are summed together at the current sense input (CS)
connection at the filter capacitor. To minimize loading on the oscillator, it is best to buffer the timing capacitor
waveform with a small transistor whose collector is connected to the reference voltage.
VREF
0.1 µF
RRT
RT/CT
CCT
RRAMP
RCSF
ISENSE
CS
RCS
CCSF
Figure 8-8. Slope Compensation Circuit
8.3.10 Voltage Mode
In certain applications, voltage mode control may be a preferred control strategy for a variety of reasons. Voltage
mode control is easily executable with any current mode controller, especially the UCCx8C4x family members.
Implementation requires generating a 0-V to 0.9-V sawtooth shaped signal to input to the current sense pin
(CS) which is also one input to the PWM comparator. This is compared to the divided down error amplifier
output voltage at the other input of the PWM comparator. As the error amplifier output is varied, it intersects the
sawtooth waveform at different points in time, thereby generating different pulse widths. This is a straightforward
method of linearly generating a pulse whose width is proportional to the error voltage.
Implementation of voltage mode control is possible by using a fraction of the oscillator timing capacitor
(CCT) waveform. This value can be divided down and fed to the current sense pin as shown in Figure 8-9.
The oscillator timing components must be selected to approximate as close to a linear sawtooth waveform
as possible. Although exponentially charged, large values of timing resistance and small values of timing
capacitance help approximate a more linear shaped waveform. A small transistor is used to buffer the oscillator
timing components from the loading of the resistive divider network. Due to the offset of the oscillator’s lower
timing threshold, a DC blocking capacitor is added.
VREF
RRT
2N2222
RT/CT
CS
CCT
Figure 8-9. Current Mode PWM Used as a Voltage Mode PWM
20
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UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
8.4 Device Functional Modes
8.4.1 Normal Operation
During normal operating mode, the controller can be used in peak current mode or voltage mode control.
When the converter is operating in peak current mode, the controller regulates the converter's peak current and
duty cycle. When used in voltage mode control, the controller regulates the power converter's duty cycle. The
regulation of the system's peak current and duty cycle can be achieved with the use of the integrated error
amplifier and external feedback circuitry.
8.4.2 UVLO Mode
During the system start-up, VDD voltage starts to rise from 0 V. Before the VDD voltage reaches its
corresponding turn-on threshold, the IC is operating in UVLO mode. During UVLO mode operation, the VREF
pin voltage is not generated. When VDD is above 1 V and below the turn-on threshold, the VREF pin is actively
pulled low. This behavior allows VREF to be used as a logic signal to indicate UVLO mode. If the bias voltage to
VDD drops below the UVLO-OFF threshold, the PWM switching stops and VREF returns to 0 V. The device can
be restarted by applying a voltage greater than the UVLO-ON threshold to the VDD pin.
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21
UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
9 Application and Implementation
Note
Information in the following applications sections is not part of the TI component specification,
and TI does not warrant its accuracy or completeness. TI’s customers are responsible for
determining suitability of components for their purposes, as well as validating and testing their design
implementation to confirm system functionality.
9.1 Application Information
The UCCx8C4x controllers are peak current mode pulse width modulators. These controllers have an onboard
amplifier and can be used in isolated and nonisolated power supply designs. The onboard totem pole gate
driver is capable of delivering 1 A of peak current. This high-speed PWM is capable of operating at switching
frequencies up to 1 MHz. . Figure 9-1 shows a typical off-line application using UCC38C44.
D50
F1
12 V
OUT
T1
R10
AC Input
C52
C3
C12
+
C55
R56
BR1
100 VAC – 240 VAC
EMI Filter
Required
D2
R11
C1A
L50
D51
C18
5V
OUT
R12
RT1
C53
C54
D6
R55
C5
SEC
COMMON
R6
IC2
R50
UCC38C44
1 COMP
REF 8
2 FB
VCC 7
3 CS
OUT 6
4 RT/CT
GND 5
R16
Q1
R57
IC2
R53
R52
C50
C13
C51
R50
K
IC3
A
R
R54
Copyright © 2016, Texas Instruments Incorporated
Figure 9-1. Typical Off-Line Application
22
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UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
Figure 9-2 shows a forward converter with synchronous rectification. This application provides 48 V to 3.3 V at
10 A with over 85% efficiency, and uses the UCC38C42as the secondary-side controller and UCC3961 as the
primary-side startup control device.
L1
4.7uH
3r3V
C2
C18
1nF
T1
4700pF
+
C21
Q4
C19
470uF
R21
10k
10
+
R7
VinP
C17
4700pF
R20
10
D2
0.1uF
C20
470uF
Q3
PWRGND
R1
32.4k
C25
R27
0.047uF
R26
+
C1
4.7
4.7
D1
R2
470uF
1.2k
Q1
R4
R5
1.5k
R19
20
D5
76.8k
BAR74
R3
2.4k
VinN
R28
D3
R8
5.1k
1
100
BAR74
R9
0.33
2
U1
1
C3
10nF
2
3
OVS
UVS
UCC3961
SD
ST
SS
VDD
FB
OUT
14
3
R6
4.7
Q2
13
C22
4.7nF
4
5
RT
PGND
6
C5
7
REF
CS
AGND
VS
HIDR
DT
BTLO
4 V
CC
R23
LODR
7
C26
6
2uF
5
402
11
C9
C8
1uF
10
R10
1k
R16
0.1uF
21.5k
C23
C24
0.1uF
680pF
U2
1
R11
46.4k
PGND
12
C4
0.22uF
U4
TPS2832
8
IN
BOOT
COMP
REF
FB
VCC
5.6nF
20k
2
8
R22
UCC38C4x
CS
OUT
RT/CT
GND
7
4
100pF
R24
100
6
50k
470pF
0.22uF
+
C7
3
R15
C6
0.1uF
8
C16
R17
9
C13
5
C14
1uF
C15
1uF
BZX84C15LT1
C12
20k
D6
3300pF
R14
R18
20k 40%
1
T2
R12
200
R13
2
7.5k
C11
3
4
C10
2.7nF
R25
1500pF
20k
300
Copyright © 2016, Texas Instruments Incorporated
Figure 9-2. Forward Converter with Synchronous Rectification Using the UCC38C42 as the SecondarySide Controller
Copyright © 2022 Texas Instruments Incorporated
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23
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
9.2 Typical Application
A typical application for the UCC28C42 controller in an off-line flyback converter is shown in Figure 9-3. The
controller uses an inner current control loop that contains a small current sense resistor which senses the
primary inductor current ramp. This current sense resistor transforms the inductor current waveform to a voltage
signal that is input directly into the primary side PWM comparator. This inner loop determines the response to
input voltage changes. An outer voltage control loop involves comparing a portion of the output voltage to a
reference voltage at the input of an error amplifier. When used in an off-line isolated application, the voltage
feedback of the isolated output is accomplished using a secondary-side error amplifier and adjustable voltage
reference, such as the TL431. The error signal crosses the primary to secondary isolation boundary using an
opto-isolator whose collector is connected to the VREF pin and the emitter is connected to FB. The outer voltage
control loop determines the response to load changes.
D CLAMP
~
VIN = 85 VAC
to 265 VAC
±
D BRIDGE
CSS
CSNUB
10 nF
RSNUB
50 k
DOUT
+
CIN
180µF
~
RSTART
420k
NP
DBIAS
NS
COUT
2200µF
RVDD
22
VOUT
12 V,
4A
NA
CVDD
120µF
RSS
R COMPp
10 k
C COMPp
10 nF
RRT
15. 4 k
CRAMP
10 nF
1
UCC28C42
COMP
VREF
LP =1. 5 mH
NP:NS = 10
NP:NA = 10
8
2 FB
VDD
7
3 CS
OUT
6
4 RT/CT
GND
5
RG
10
QSW
DZ CVDDbp
18 V 0. 1 µF
CCT
1000pF
CVREF
1 µF
R BLEEDER
10 k
RRAMP
24.9 k
CCSF
100 pF
RLED
1.3 k
RCSF
3. 8 k
RP
Not Populated
RCS
0. 75
RTLbias
1k
OPTOCOUPLER
10 V
RFBG
4. 99 k
RFBU
9. 53 k
R COMPz C COMPz
88. 7 k
0. 01 µF
ROPTO
1k
TL431
RFBB
2. 49 k
Copyright © 2016, Texas Instruments Incorporated
Figure 9-3. Typical Application Design Schematic
9.2.1 Design Requirements
Table 9-1 shows a typical set of performance requirements for an off-line flyback converter capable of providing
48 W at 12-V output voltage from a universal AC input. The design uses peak primary current control in a
continuous current mode PWM converter.
Table 9-1. Design Parameters
PARAMETER
TEST CONDITIONS
VIN
Input Voltage
fLINE
Line Frequency
VOUT
Output Voltage
IVOUT(min) ≤ IVOUT ≤ IVOUT(max)
VRIPPLE
Output Ripple Voltage
IVOUT(min) ≤ IVOUT ≤ IVOUT(max)
IVOUT
Output Current
fSW
Switching Frequency
η
Efficiency
24
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MIN
NOM
MAX
UNIT
85
115/230
265
VRMS
47
50/60
63
Hz
11.75
12
12.25
V
0
4
100
110
mVpp
A
kHz
85%
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UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
9.2.2 Detailed Design Procedure
This procedure outlines the steps to design an off-line universal input continuous current mode (CCM) flyback
converter. See Figure 9-3 for component names referred to in the design procedure.
9.2.2.1 Input Bulk Capacitor and Minimum Bulk Voltage
Bulk capacitance may consist of one or more capacitors connected in parallel, often with some inductance
between them to suppress differential-mode conducted noise. The value of the input capacitor sets the minimum
bulk voltage;. Setting the bulk voltage lower by using minimal input capacitance results in higher peak primary
currents leading to more stress on the MOSFET switch, the transformer, and the output capacitors. Setting
the bulk voltage higher by using a larger input capacitor results in higher peak current from the input source
and the capacitor itself is physically larger. Compromising between size and component stresses determines
the acceptable minimum input voltage. The total required value for the primary-side bulk capacitance (CIN) is
selected based upon the power level of the converter (POUT), the efficiency target (η), the minimum input voltage
(VIN(min)), and is chosen to maintain an acceptable minimum bulk voltage level (VBULK(min)), using Equation 3.
CIN =
2 × PIN × F0.25 +
VBULK (min )
1
× arcsin F
GG
N
¾2 × VIN (min )
2
2
k2 × VIN
(min ) F VBULK (min ) o × fLINE (min )
(3)
where
•
VIN(min) is the RMS value of the minimum AC input voltage (85 VRMS) whose minimum line frequency is
denoted as fLINE(min), equal to 47 Hz
Based on Equation 3, to achieve a minimum bulk voltage of 75 V, assuming 85% converter efficiency, the bulk
capacitor must be larger than 126 µF. this design uses a value of 180 µF, with consideration for component
tolerances and efficiency estimation.
9.2.2.2 Transformer Turns Ratio and Maximum Duty Cycle
The transformer design begins with selecting a suitable switching frequency for the given application. The
UCC28C42 is capable of switching up to 1 MHz but considerations such as overall converter size, switching
losses, core loss, system compatibility, and interference with communication frequency bands generally
determine an optimum frequency that should be used. For this off-line converter, the switching frequency (fSW) is
selected to be 110 kHz as a compromise to minimize the transformer size and the EMI filter size, and still have
acceptable losses.
The transformer primary to secondary turns ratio (NPS) can be selected based on the desired MOSFET voltage
rating and the secondary diode voltage rating. Because the maximum input voltage is 265 VRMS, the peak bulk
input voltage can be calculated as shown in Equation 4.
VBULK (max ) = ¾2 × VIN (max ) N 375 V
(4)
To minimize the cost of the system, a readily available 650-V MOSFET is selected. Derating the maximum
voltage stress on the drain to 80% of its rated value and allowing for a leakage inductance voltage spike of up
to 30% of the maximum bulk input voltage, the reflected output voltage must be less than 130 V as shown in
Equation 5.
VREFLECTED
0.8 u VDS(rated) 1.3 u VBULK(max)
130.2 V
(5)
The maximum primary to secondary transformer turns ratio (NPS) for a 12 V output can be selected as
NPS =
VREFLECTED
= 10.85
VOUT
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25
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A turns ratio of NPS = 10 is used in the design example.
The auxiliary winding is used to supply bias voltage to the controller. Maintaining the bias voltage above the VDD
minimum operating voltage after turnon is required for stable operation. The minimum VDD operating voltage for
the controller selected for this design is 10 V. The auxiliary winding is selected to support a 12 V bias voltage so
that it is above the minimum operating level but maintains a low level of losses in the IC. The primary to auxiliary
turns ratio (NPA) can be calculated from Equation 7:
NPA = NPS ×
VOUT
= 10
VBIAS
(7)
The output diode experiences a voltage stress that is equal to the output voltage plus the reflected input voltage:
VDIODE =
VBULK :max ;
+ VOUT = 49.5 V
NPS
(8)
TI recommends a Schottky diode with a rated blocking voltage greater than 60 V to allow for voltage spikes due
to ringing. The forward voltage drop (VF) of this diode is estimated to be equal to 0.6 V
To avoid high peak currents, the flyback converter in this design operates in continuous conduction mode. Once
NPS is determined, the maximum duty cycle (DMAX) can be calculated using the transfer function for a CCM
flyback converter:
DMAX
1
VOUT + VF
p×l
p
=l
NPS
1 F DMAX
VBULK :min ;
DMAX
NPS u VOUT
VF
VBULK(min) NPS u VOUT
VF
(9)
0.627
(10)
Because the maximum duty cycle exceeds 50%, and the design is an off-line (AC-input) application, the
UCC28C42 is best suited for this application.
9.2.2.3 Transformer Inductance and Peak Currents
For this design example, the transformer magnetizing inductance is selected based upon the CCM condition. An
inductance value that allows the converter to stay in CCM over a wider operating range before transitioning into
discontinuous current mode is used to minimize losses due to otherwise high currents and also to decrease the
output ripple. The design of the transformer in this example sizes the inductance so the converter enters CCM
operation at approximately 10% load and minimum bulk voltage to minimize output ripple.
The inductor (LP) for a CCM flyback can be calculated using Equation 11.
2
LP =
1
×
2
2
NPS × VOUT
p
VBULK :min ; + NPS × VOUT
0.1 × PIN × fSW
kVBULK :min ; o × l
(11)
where
•
•
PIN is estimated by dividing the maximum output power (POUT) by the target efficiency (η)
fSW is the switching frequency of the converter
For the UCC28C42 the switching frequency is equal to the oscillator frequency and is set to 110 kHz. Selecting
fSW to be 110 kHz provides a good compromise between size of magnetics, switching losses, and places
the first harmonic below the 150-kHz lower limit of EN55022. Therefore, the transformer inductance must be
approximately 1.8 mH. A 1.5 mH inductance is chosen as the magnetizing inductance, LP, value for this design.
26
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UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
Based on calculated inductor value and the switching frequency, the current stress of the MOSFET and output
diode can be calculated.
The peak current in the primary-side MOSFET of a CCM flyback can be calculated as shown in Equation 12.
NPS × VOUT
VBULK (min ) VBULK :min ; + :NPS × VOUT ;
PIN
=
+n
×
r
NPS × VOUT
2 × Lm
fSW
VBULK :min ; ×
VBULK :min ; + :NPS × VOUT ;
IPK MOSFET
(12)
The MOSFET peak current is 1.36 A. The RMS current of the MOSFET is calculated to be 0.97 A as shown in
Equation 13. Therefore, IRFB9N65A is selected to be used as the primary-side switch.
IRM S MOSFET
DMAX 2 × IPK MOSFET × VBULK (min )
VBULK (min ) 2
DMAX 3
p FF
×l
G + kDMAX × IPK MOSFET 2 o
=¨
LP × fSW
LP × fSW
3
(13)
The output diode peak current is equal to the MOSFET peak current reflected to the secondary side.
IPK DIODE = NPS × IPK MOSFET = 13.634 A
(14)
The diode average current is equal to the total output current (4 A) combined with a required 60-V rating and
13.6-A peak current requirement, a 48CTQ060-1 is selected for the output diode.
9.2.2.4 Output Capacitor
The total output capacitance is selected based upon the output voltage ripple requirement. In this design, 0.1%
voltage ripple is assumed. Based on the 0.1% ripple requirement, the capacitor value can be selected using
Equation 15.
NPS × VOUT
VBULK :min ; + NPS × VOUT
= 1865 JF
0.001 × VOUT × fSW
IOUT ×
COUT R
(15)
To design for device tolerances, a 2200-µF capacitor was selected.
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27
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9.2.2.5 Current Sensing Network
The current sensing network consists of the primary-side current sensing resistor (RCS), filtering components
RCSF and CCSF, and optional RP. Typically, the direct current sense signal contains a large amplitude leading
edge spike associated with the turnon of the main power MOSFET, reverse recovery of the output rectifier, and
other factors including charging and discharging of parasitic capacitances. Therefore, CCSF and RCSF form a
low-pass filter that provides immunity to suppress the leading edge spike. For this converter, CCSF is chosen to
be 100 pF.
Without RP, RCS sets the maximum peak current in the transformer primary based on the maximum amplitude of
the CS pin, which is specified to be 1 V. To achieve 1.36-A primary side peak current, a 0.75-Ω resistor is chosen
for RCS.
The high current sense threshold of CS helps to provide better noise immunity to the system but also results
in higher losses in the current sense resistor. These current sense losses can be minimized by injecting an
offset voltage into the current sense signal using RP. RP and RCSF form a resistor divider network from the
current sense signal to the reference voltage of the controller (VVREF) which adds an offset to the current
sense voltage. This technique still achieves current mode control with cycle-by-cycle over-current protection. To
calculate required offset value (VOFFSET), use Equation 16.
VOFFSET =
R CSF
× VREF
R CSF + R P
(16)
After adding the RP resistance, adjust the RCS value accordingly.
9.2.2.6 Gate Drive Resistor
RG is the gate driver resistor for the power switch (QSW). The selection of this resistor value must be done in
conjunction with EMI compliance testing and efficiency testing. Using a larger resistor value for RG slows down
the turnon and turnoff of the MOSFET. A slower switching speed reduces EMI but also increases the switching
loss. A tradeoff between switching loss and EMI performance must be carefully performed. For this design, a
10‑Ω resistor was chosen for the gate drive resistor.
9.2.2.7 VREF Capacitor
A precision 5-V reference voltage performs several important functions. The reference voltage is divided down
internally to 2.5 V and connected to the error amplifier’s noninverting input for accurate output voltage regulation.
Other duties of the reference voltage are to set internal bias currents and thresholds for functions such as
the oscillator upper and lower thresholds. Therefore, the reference voltage must be bypassed with a ceramic
capacitor. A 1-µF, 16-V ceramic capacitor was selected for this converter. Placement of this capacitor on the
physical printed-circuit board layout must be as close as possible to the respective VREF and GND pins.
9.2.2.8 RT/CT
The internal oscillator uses a timing capacitor (CCT) and a timing resistor (RRT) to program the oscillator
frequency and maximum duty cycle. The operating frequency can be programmed based the curves in Figure
7-1, where the timing resistor can be found once the timing capacitor is selected. It is best for the timing
capacitor to have a flat temperature coefficient, typical of most COG or NPO type capacitors. For this converter,
15.4 kΩ and 1000 pF were selected for RRT and CCT to operate at 110-kHz switching.
28
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
9.2.2.9 Start-Up Circuit
At start-up, the IC gets its power directly from the high-voltage bulk, through a high-voltage resistor (RSTART).
The selection of the start-up resistor is the tradeoff between power loss and start-up time. The current flowing
through R START at the minimum input voltage must be higher than the VDD current under UVLO conditions (100
µA at its maximum value). A resistance of 420-kΩ was chosen for RSTART, providing 250 µA of start-up current
at low-line conditions. The start-up resistor is physically comprised of two 210-kΩ resistors in series to meet the
high voltage requirements and power rating at high-line.
After VDD is charged up above the UVLO-ON threshold, the UCC28C42 starts to consume full operating
current. The VDD capacitor is required to provide enough energy to prevent its voltage from dropping below
the UVLO-OFF threshold during start-up, before the output is able to reach its regulated level. A large bulk
capacitance would hold more energy but would result in slower start-up time. In this design, a 120-µF capacitor
is chosen to provide enough energy and maintain a start-up time of approximately 7 seconds. For faster start-up,
the bulk capacitor value may be decreased or the RSTART resistor modified to a lower value.
9.2.2.10 Voltage Feedback Compensation
Feedback compensation, also called closed-loop control, can reduce or eliminate steady state error, reduce
the sensitivity of the system to parametric changes, change the gain or phase of a system over some desired
frequency range, reduce the effects of small signal load disturbances and noise on system performance, and
create a stable system from an unstable system. A system is stable if its response to a perturbation is that the
perturbation eventually dies out. A peak current mode flyback uses an outer voltage feedback loop to stabilize
the converter. To adequately compensate the voltage loop, the open-loop parameters of the power stage must
be determined.
9.2.2.10.1 Power Stage Poles and Zeroes
The first step in compensating a fixed frequency flyback is to verify if the converter is continuous conduction
mode (CCM) or discontinuous conduction mode (DCM). If the primary inductance (LP) is greater than the
inductance for DCM or CCM boundary mode operation, called the critical inductance (LPcrit), then the converter
operates in CCM:
LP > LPcrit , then CCM
LPcrit =
(17)
2
R OUT × :NPS ;2
VIN
p
×l
2 × fSW
VIN + VOUT × NPS
(18)
For the entire input voltage range, the selected inductor has a value larger than the critical inductor. Therefore,
the converter operates in CCM and the compensation loop requires design based on CCM flyback equations.
The current-to-voltage conversion is done externally with the ground-referenced RCS and the internal 2R/R
resistor divider which sets up the internal current sense gain, ACS = 3. The exact value of these internal resistors
is not critical but the IC provides tight control of the resistor divider ratio, so regardless of the actual resistor
value variations their relative value to each other is maintained.
The DC open-loop gain (GO) of the fixed-frequency voltage control loop of a peak current mode control CCM
flyback converter shown in Equation 19 is approximated by first using the output load (ROUT), the primary to
secondary turns ratio (NPS), and the maximum duty cycle (D) as calculated in Equation 20.
GO =
R OUT × NPS
1
×
2
:1 F D;
R CS × ACS
+ :2 × M; + 1
RL
(19)
In Equation 19, D is calculated with Equation 20, τL is calculated with Equation 21, and M is calculated with
Equation 22.
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
D=
NPS × VOUT
VBULKmin + :NPS × VOUT ;
RL =
M=
www.ti.com
(20)
2 × LP × fSW
R OUT × :NPS ;2
(21)
VOUT × NPS
VBULKmin
(22)
For this design, a converter with an output voltage (VOUT) of 12 V, and 48 W relates to an output load (ROUT)
equal to 3 Ω at full load. With a maximum duty cycle of 0.627, a current sense resistance of 0.75 Ω, and a
primary to secondary turns-ratio of 10, the open-loop gain calculates to 3.082 or 9.776 dB.
A CCM flyback has two zeroes that are of interest. The ESR and the output capacitance contribute a left-half
plane zero (ωESRz) to the power stage, and the frequency of this zero (fESRz), are calculated with Equation 23
and Equation 24.
1
× COUT
(23)
1
2 × N × R ESR × COUT
(24)
XESRz =
fESRz =
R ESR
The fESRz zero for an output capacitance of 2200 µF and a total ESR of 43 mΩ is located at 1.682 kHz.
CCM flyback converters have a zero in the right-half plane (RHP) in their transfer function. A RHP zero has the
same 20 dB per decade rising gain magnitude with increasing frequency just like a left-half plane zero, but it
adds a 90° phase lag instead of lead. This phase lag tends to limit the overall loop bandwidth. The frequency
location (fRHPz) of the RHP zero (ωRHPz) is a function of the output load, the duty cycle, the primary inductance
(LP), and the primary to secondary side turns ratio (NPS).
XRHPz =
fRHPz =
R OUT × :1 F D;2 × :NPS ;2
LP × D
(25)
R OUT × :1 F D;2 × :NPS ;2
2 × N × LP × D
(26)
The right-half plane zero frequency increases with higher input voltage and lighter load. Generally, the design
requires consideration of the worst case of the lowest right-half plane zero frequency and the converter must be
compensated at the minimum input and maximum load condition. With a primary inductance of 1.5 mH, at 75-V
DC input, the RHP zero frequency (fRHPz) is equal to 7.07 kHz at maximum duty cycle, full load.
The power stage has one dominate pole (ωP1) which is in the region of interest, located at a lower frequency
(fP1); which is related to the duty cycle, the output load, and the output capacitance, and calculated with Equation
28. There is also a double pole placed at half the switching frequency of the converter (fP2) calculated with
Equation 30. For this example, pole fP1 is located at 40.37 Hz and fP2 is at 55 kHz.
XP1
30
:1 F D;3
+1+D
RL
=
R OUT × COUT
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(27)
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fP1
SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
:1 F D;3
+1+D
RL
=
2 × N × R OUT × COUT
(28)
XP2 = N × fSW
(29)
fSW
2
(30)
fP2 =
9.2.2.10.2 Slope Compensation
Slope compensation is the large signal subharmonic instability that can occur with duty cycles that may extend
beyond 50% where the rising primary side inductor current slope may not match the falling secondary side
current slope. The subharmonic oscillation would result in an increase in the output voltage ripple and may even
limit the power handling capability of the converter.
The target of slope compensation is to achieve an ideal quality coefficient (QP), equal to 1 at half of the switching
frequency. The QP is calculated with Equation 31.
1
N × >MC × :1 F D; F 0.5?
QP =
(31)
where
•
•
D is the primary side switch duty cycle
MC is the slope compensation factor, which is defined with Equation 32
MC =
Se
+1
Sn
(32)
where
•
•
Se is the compensation ramp slope
Sn is the inductor rising slope
The optimal goal of the slope compensation is to achieve QP = 1; upon rearranging Equation 32 the ideal value
of slope compensation factor is determined:
Mideal
1
+ 0.5
N
=
1FD
(33)
For this design to have adequate slope compensation, MC must be 2.193 when D reaches it maximum value of
0.627.
The inductor rising slope (Sn) at the CS pin is calculated with Equation 34.
Sn =
VINmin × R CS
V
= 0.038
Js
LP
(34)
The compensation slope (Se) is calculated with Equation 35.
Se = :MC F 1; × Sn = 44.74
mV
Js
(35)
The compensation slope is added into the system through RRAMP and RCSF. The CRAMP is an AC-coupling
capacitor that allows the voltage ramp of the oscillator to be used without adding an offset to the current sense;
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
select a value to approximate a high-frequency short circuit, such as 10 nF, as a starting point and make
adjustments if required. The RRAMP and RCSF resistors form a voltage divider from the oscillator charge slope
and this proportional ramp is injected into the CS pin to add slope compensation. Choose the value of RRAMP to
be much larger than the RRT resistor so that it does not load down the internal oscillator and result in a frequency
shift. The oscillator charge slope is calculated using the peak-to-peak voltage of the RT/CT sawtooth waveform
(VOSCpp) equal to 1.9 V, and the minimum ON time, as shown in Equation 37.
D
t ONmin =
SOSC =
fSW
(36)
VOSCpp
1.9 V
mV
=
= 333
5.7 Js
Js
t ONmin
(37)
To achieve a 44.74-mV/µs compensation slope, RCSF is calculated with Equation 38. In this design, RRAMP is
selected as 24.9 kΩ, a 3.8-kΩ resistor was selected for RCSF.
R CSF =
R RAMP
SOSC
F1
Se
(38)
9.2.2.10.3 Open-Loop Gain
Once the power stage poles and zeros are calculated and the slope compensation is determined, the power
stage open-loop gain and phase of the CCM flyback converter can be plotted as a function of frequency. The
power stage transfer function can be characterized with Equation 39.
HOPEN :s; = G0 ×
l1 +
s:f;
s:f;
p × l1 F
p
1
XESRz
XRHPz
×
s:f;
s:f;
s:f;2
1+
1
+
+
XP1
XP2 × Q P :XP2 ;2
(39)
The bode for the open-loop gain and phase can be plotted by using Equation 40.
GainOPEN :s; = 20 × log: HOPEN :s; ;
(40)
See Figure 9-4 and Figure 9-5.
10
0
5
-45
Phase (q)
Gain (dB)
0
-5
-10
-15
-90
-135
-20
-180
-25
1
10
100
1000
frequency (Hz)
10000
100000
D001
Figure 9-4. Converter Open-Loop Bode Plot - Gain
32
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1
10
100
1000
frequency (Hz)
10000
100000
D002
Figure 9-5. Converter Open-Loop Bode Plot Phase
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9.2.2.10.4 Compensation Loop
The design of the compensation loop involves selecting the appropriate components so that the required gain,
poles, and zeros can be designed to result in a stable system over the entire operating range. There are three
distinct portions of the loop: the TL431, the opto-coupler, and the error amplifier. Each of these stages combines
with the power stage to result in a stable robust system.
For good transient response, the bandwidth of the finalized design must be as large as possible. The bandwidth
of a CCM flyback, fBW, is limited to ¼ of the RHP zero frequency, or approximately 1.77 kHz using Equation 41.
fBW =
fRHPz
4
(41)
The gain of the open-loop power stage at fBW can be calculated using Equation 40 or can be observed on the
Bode plot (Figure 9-4) and is equal to –19.55 dB and the phase at fBW is equal to –58°.
The secondary side portion of the compensation loop begins with establishing the regulated steady state output
voltage. To set the regulated output voltage, a TL431 adjustable precision shunt regulator is ideally suited for
use on the secondary side of isolated converters due to its accurate voltage reference and internal op-amp.
The resistors used in the divider from the output terminals of the converter to the TL431 REF pin are selected
based upon the desired power consumption. Because the REF input current for the TL431 is only 2 µA, selecting
the resistors for a divider current (IFB_REF) of 1 mA results in minimal error. The top divider resistor (RFBU) is
calculated:
R FBU =
VOUT F REFTL431
IFB _REF
(42)
The TL431 reference voltage (REFTL431) has a typical value of 2.495 V. A 9.53-kΩ resistor is chosen for RFBU. To
set the output voltage to 12 V, 2.49 kΩ is used for RFBB.
R FBB =
REFTL431
× R FBU
VOUT F REFTL431
(43)
For good phase margin, a compensator zero (fCOMPz) is required and should be placed at 1/10th the desired
bandwidth:
fCOMPz =
fBW
10
(44)
XCOMPz = 2 × N × fCOMPz
(45)
With this converter, fCOMPz should be set at approximately 177 Hz. A series resistor (RCOMPz) and capacitor
(CCOMPz) placed across the TL431 cathode to REF sets the compensator zero location. Setting CCOMPz to
0.01 µF, RCOMPz is calculated:
R COMPz =
1
XCOMPz × CCOMPz
(46)
Using a standard value of 88.7 kΩ for RZ and a 0.01 µF for CZ results in a zero placed at 179 Hz.
In Figure 9-3, RTLbias provides cathode current to the TL431 from the regulated voltage provided from the Zener
diode (DREG). For robust performance, 10 mA is provided to bias the TL431 by way of the 10-V Zener and a
1-kΩ resistor is used for RTLbias.
The gain of the TL431 portion of the compensation loop is calculated with Equation 47.
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GTL431 :s; = lR COMPz +
1
1
p×
R FBU
s(f) × CZCOMPz
www.ti.com
(47)
A compensation pole is required at the frequency of right half plane zero or the ESR zero, whichever is lowest.
Based previous the analysis, the right half plane zero (fRHPz) is located at 7.07 kHz and the ESR zero (fESRz)
is at 1.68 kHz; therefore, for this design, the compensation pole must be put at 1.68 kHz. The opto-coupler
contains a parasitic pole that is difficult to characterize over frequency so the opto-coupler is set up with a
pull-down resistor (ROPTO) equal to 1 kΩ, which moves the parasitic opto-coupler pole further out and beyond the
range of interest for this design.
The required compensation pole can be added to the primary side error amplifier using RCOMPp and CCOMPp.
Choosing RCOMPp as 10 kΩ, the required value of CCOMPp is determined using Equation 48.
CCOMPp =
1
2 × N × fESRz × R COMPp
= 9.46 nF
(48)
A 10-nF capacitor is used for CCOMPp setting the compensation pole at 1.59 kHz.
Adding a DC gain to the primary-side error amplifier may be required to obtain the required bandwidth and
helps to adjust the loop gain as needed. Using 4.99 kΩ for RFBG sets the DC gain on the error amplifier to 2.
At this point the gain transfer function of the error amplifier stage (GEA(s)) of the compensation loop can be
characterized using Equation 49.
GEA :s; = l
R COMPp
1
p×F
G
1 + s:f; × CCOMPp × R COMPp
R FBG
(49)
Using an opto-coupler whose current transfer ratio (CTR) is typically at 100% in the frequency range of interest
so that CTR = 1, the transfer function of the opto-coupler stage (GOPTO(s)) is found using Equation 50.
GOPTO (s) =
CTR × R OPTO
R LED
(50)
The bias resistor (RLED) to the internal diode of the opto-coupler and the pull-down resistor on the opto emitter
(ROPTO) sets the gain across the isolation boundary. ROPTO has already been set to 1 kΩ but the value of RLED
has not yet been determined.
The total closed loop gain (GTOTAL(s)) is the combination of the open-loop power stage (Ho(s)), the opto gain
(GOPTO(s)), the error amplifier gain (GEA(s)), and the gain of the TL431 stage (GTL431(s)), as shown in Equation
51.
GTOTAL :s; = HOPEN :s; × GOPTO :s; × GEA :s; × GTL431 :s;
(51)
The required value for RLED can be selected to achieve the desired crossover frequency (fBW). By setting the
total loop gain equal to 1 at the desired crossover frequency and rearranging Equation 51, the optimal value for
RLED can be determined, as shown in Equation 52.
R LED Q HOPEN :s; × CTR × COPTO × GEA :s; × GTL431 :s;
(52)
A 1.3-kΩ resistor suits the requirement for RLED.
Based on the compensation loop structure, the entire compensation loop transfer function is written as Equation
53.
34
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GCLOSED :s; = HOPEN :s; × l
×n
R COMPz
R COMPp
CTR × R OPTO
1
p×l
p×F
G
R LED
R FBG
1 + ks × CCOMPp × R COMPp o
1
A
+@
s × CCOMPz
r
R FBU
(53)
The final closed-loop bode plots are show in Figure 9-6 and Figure 9-7. The converter achieves a crossover
frequency of approximately 1.8 kHz and has a phase margin of approximately 67°.
TI recommends checking the loop stability across all the corner cases including component tolerances to ensure
system stability.
80
0
60
-45
Degrees (q)
Gain (dB)
40
20
-90
0
-135
-20
-180
-40
1
10
100
1000
frequency (Hz)
10000
100000
D003
Figure 9-6. Converter Closed-Loop Bode Plot –
Gain
1
10
100
1000
frequency (Hz)
10000
100000
D001
D004
Figure 9-7. Converter Closed-Loop Bode Plot –
Phase
9.2.3 Application Curves
Figure 9-8. Primary Side MOSFET Drain to Source
Voltage at 240-V AC Input (100 V/div)
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Figure 9-9. Primary Side MOSFET Drain to Source
Voltage at 120-V AC Input (100 V/div)
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CH1: Output Voltage AC Coupled, 200 mV/div
CH4: Output Current, 1 A/div
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Figure 9-11. Output Voltage Ripple at Full Load
(100 mV/div)
Figure 9-10. Output Voltage During 0.9-A to 2.7-A
Load Transient
Figure 9-12. Output Voltage Behavior at Full Load Start-up (5 V/div)
9.3 Power Supply Recommendations
The absolute maximum supply voltage is 20 V of UCC28C42, including any transients that may be present. If
this voltage is exceeded, device damage is likely. This damage risk is in contrast to the predecessor bipolar
devices, which could survive up to 30 V. Thus, the supply pin must be decoupled as close to the GND pin as
possible.
Because no clamp is included in the device, the supply pin must be protected from external sources which could
exceed the 20-V level.
To prevent false triggering due to leading edge noises, an RC current sense filter may be required on CS. Keep
the time constant of the RC filter well below the minimum on-time pulse width.
To prevent noise problems with high-speed switching transients, bypass VREF to ground with a ceramic
capacitor close to the IC package. A minimum of 0.1-µF ceramic capacitor is required. Additional VREF
bypassing is required for external loads on the reference. An electrolytic capacitor may also be used in addition
to the ceramic capacitor.
36
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9.4 Layout
9.4.1 Layout Guidelines
9.4.1.1 Precautions
Careful layout of the printed board is a necessity for high-frequency power supplies. As the device-switching
speeds and operating frequencies increase, the layout of the converter becomes increasingly important.
This 8-pin device has only a single ground for the logic and power connections. This forces the gate-drive
current pulses to flow through the same ground that the control circuit uses for reference. Thus, the interconnect
inductance must be minimized as much as possible. One implication is to place the device (gate driver) circuitry
close to the MOSFET it is driving. This can conflict with the need for the error amplifier and the feedback path to
be away from the noise generating components.
The single most critical item in a PWM controlled printed-circuit board layout is the placement of the timing
capacitor. While both the supply and reference bypass capacitor locations are important, the timing capacitor
placement is far more critical. Any noise spikes on the CCT waveform due to lengthy printed circuit trace
inductance or pick-up noise from being in proximity to high power switching noise causes a variety of operational
problems. Dilemmas vary from incorrect operating frequency caused by pre-triggering the oscillator due to noise
spikes to frequency jumping with varying duty cycles, also caused by noise spikes. The placement of the timing
capacitor must be treated as the most important layout consideration. Keep PC traces as short as possible to
minimize added series inductance.
9.4.1.2 Feedback Traces
Try to run the feedback trace as far from the inductor and noisy power traces as possible. You would also like
the feedback trace to be as direct as possible and somewhat thick. These two sometimes involve a trade-off, but
keeping it away from EMI and other noise sources is the more critical of the two. If possible, run the feedback
trace on the side of the PCB opposite of the inductor with a ground plane separating the two.
9.4.1.3 Bypass Capacitors
When using a low value ceramic bypass capacitor, it must be placed as close to the VDD pin of the device
as possible. This eliminates as much trace inductance effects as possible and give the internal device rail a
cleaner voltage supply. Using surface mount capacitors also reduces lead length and lessens the chance of
noise coupling into the effective antenna created by through-hole components.
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SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
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9.4.1.4 Compensation Components
For best stability, external compensation components must be placed close to the IC. Keep FB lead length as
short as possible and FB stray capacitance as small as possible. TI recommends surface mount components
here as well for the same reasons discussed for the filter capacitors. These must not be placed very close to
traces with high switching noise.
9.4.1.5 Traces and Ground Planes
Make all of the power (high current) traces as short, direct, and thick as possible. It is good practice on a
standard PCB board to make the traces an absolute minimum of 15 mils (0.381 mm) per ampere. The inductor,
output capacitors, and output diode must be as close to each other possible. This helps reduce the EMI radiated
by the power traces due to the high switching currents through them. This also reduces lead inductance and
resistance as well, which in turn reduces noise spikes, ringing, and resistive losses that produce voltage errors.
The grounds of the IC, input capacitors, output capacitors, and output diode, if applicable, must be connected
close together directly to a ground plane. It would also be a good idea to have a ground plane on both sides
of the PCB. This reduces noise as well by reducing ground loop errors as well as by absorbing more of the
EMI radiated by the inductor. For multi-layer boards with more than two layers, a ground plane can be used
to separate the power plane, where the power traces and components are, and the signal plane, where the
feedback and compensation and components are, for improved performance. On multi-layer boards the use of
vias is required to connect traces and different planes. It is good practice to use one standard via per 200 mA of
current if the trace conducts a significant amount of current from one plane to the other.
Arrange the components so that the switching current loops curl in the same direction. Due to the way switching
regulators operate, there are two power states. One state when the switch is ON and one when the switch is
OFF. During each state there is a current loop made by the power components that are currently conducting.
Place the power components so that during each of the two states the current loop is conducting in the same
direction. This prevents magnetic field reversal caused by the traces between the two half-cycles and reduces
radiated EMI.
38
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Copyright © 2022 Texas Instruments Incorporated
Product Folder Links: UCC28C40 UCC28C41 UCC28C42 UCC28C43 UCC28C44 UCC28C45 UCC38C40
UCC38C41 UCC38C42 UCC38C43 UCC38C44 UCC38C45
UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
www.ti.com
SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
9.4.2 Layout Example
MOSFET Heatsink
Track To
D
½ PRI Winding
Track To
GND
TRANSFORMER
22AWG
Jumper
Wire
Aux Cap
1
22AWG Jumper Wires
E
K
C
A
PCB Bo om-side View
Figure 9-13. UCCx8C4x Layout Example
Copyright © 2022 Texas Instruments Incorporated
Submit Document Feedback
Product Folder Links: UCC28C40 UCC28C41 UCC28C42 UCC28C43 UCC28C44 UCC28C45 UCC38C40
UCC38C41 UCC38C42 UCC38C43 UCC38C44 UCC38C45
39
UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
www.ti.com
10 Device and Documentation Support
10.1 Device Support
10.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
10.2 Documentation Support
10.2.1 Related Documentation
For related documentation see the following:
UC384x Provides Low-Cost Current-Mode Control (SLUA143)
10.3 Receiving Notification of Documentation Updates
To receive notification of documentation updates, navigate to the device product folder on ti.com. Click on
Subscribe to updates to register and receive a weekly digest of any product information that has changed. For
change details, review the revision history included in any revised document.
10.4 Support Resources
TI E2E™ support forums are an engineer's go-to source for fast, verified answers and design help — straight
from the experts. Search existing answers or ask your own question to get the quick design help you need.
Linked content is provided "AS IS" by the respective contributors. They do not constitute TI specifications and do
not necessarily reflect TI's views; see TI's Terms of Use.
40
Submit Document Feedback
Copyright © 2022 Texas Instruments Incorporated
Product Folder Links: UCC28C40 UCC28C41 UCC28C42 UCC28C43 UCC28C44 UCC28C45 UCC38C40
UCC38C41 UCC38C42 UCC38C43 UCC38C44 UCC38C45
UCC28C40, UCC28C41, UCC28C42, UCC28C43, UCC28C44, UCC28C45, UCC38C40, UCC38C41,
UCC38C42, UCC38C43, UCC38C44, UCC38C45
www.ti.com
SLUS458H – JULY 2000 – REVISED NOVEMBER 2022
10.5 Trademarks
TI E2E™ is a trademark of Texas Instruments.
All trademarks are the property of their respective owners.
10.6 Electrostatic Discharge Caution
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric changes could cause the device not to meet its published
specifications.
10.7 Glossary
TI Glossary
This glossary lists and explains terms, acronyms, and definitions.
11 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Copyright © 2022 Texas Instruments Incorporated
Submit Document Feedback
Product Folder Links: UCC28C40 UCC28C41 UCC28C42 UCC28C43 UCC28C44 UCC28C45 UCC38C40
UCC38C41 UCC38C42 UCC38C43 UCC38C44 UCC38C45
41
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UCC28C40D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C40
Samples
UCC28C40DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C40
Samples
UCC28C40DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C40
Samples
UCC28C40DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C40
Samples
UCC28C41D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C41
Samples
UCC28C41DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C41
Samples
UCC28C41DGK
ACTIVE
VSSOP
DGK
8
100
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C41
Samples
UCC28C41DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C41
Samples
UCC28C41DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C41
Samples
UCC28C42D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C42
Samples
UCC28C42DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C42
Samples
UCC28C42DGK
ACTIVE
VSSOP
DGK
8
100
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C42
Samples
UCC28C42DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C42
Samples
UCC28C42DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C42
Samples
UCC28C42DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C42
Samples
UCC28C43D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C43
Samples
UCC28C43DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C43
Samples
UCC28C43DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C43
Samples
UCC28C43DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C43
Samples
UCC28C43DGKRG4
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
-40 to 105
28C43
Samples
Addendum-Page 1
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UCC28C43DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C43
Samples
UCC28C43DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C43
Samples
UCC28C44D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C44
Samples
UCC28C44DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C44
Samples
UCC28C44DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C44
Samples
UCC28C44DGKG4
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 85
28C44
Samples
UCC28C44DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C44
Samples
UCC28C44DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C44
Samples
UCC28C44DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C44
Samples
UCC28C45D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C45
Samples
UCC28C45DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C45
Samples
UCC28C45DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
-40 to 105
28C45
Samples
UCC28C45DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C45
Samples
UCC28C45DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
-40 to 105
28C45
Samples
UCC38C40D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C40
Samples
UCC38C40DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C40
Samples
UCC38C40DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C40
Samples
UCC38C40DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C40
Samples
UCC38C41D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C41
Samples
UCC38C41DGK
ACTIVE
VSSOP
DGK
8
100
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C41
Samples
UCC38C41DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C41
Samples
Addendum-Page 2
PACKAGE OPTION ADDENDUM
www.ti.com
14-Oct-2022
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UCC38C41DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C41
Samples
UCC38C42D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C42
Samples
UCC38C42DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C42
Samples
UCC38C42DGK
ACTIVE
VSSOP
DGK
8
100
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C42
Samples
UCC38C42DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C42
Samples
UCC38C42DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C42
Samples
UCC38C42DRG4
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C42
Samples
UCC38C43D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C43
Samples
UCC38C43DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C43
Samples
UCC38C43DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C43
Samples
UCC38C43DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C43
Samples
UCC38C43DGKRG4
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI
Level-2-260C-1 YEAR
0 to 70
38C43
Samples
UCC38C43DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C43
Samples
UCC38C44D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C44
Samples
UCC38C44DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C44
Samples
UCC38C44DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C44
Samples
UCC38C44DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C44
Samples
UCC38C44DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C44
Samples
UCC38C45D
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C45
Samples
UCC38C45DG4
ACTIVE
SOIC
D
8
75
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C45
Samples
UCC38C45DGK
ACTIVE
VSSOP
DGK
8
80
RoHS & Green
NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C45
Samples
Addendum-Page 3
PACKAGE OPTION ADDENDUM
www.ti.com
Orderable Device
14-Oct-2022
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
Samples
(4/5)
(6)
UCC38C45DGKR
ACTIVE
VSSOP
DGK
8
2500
RoHS & Green
Call TI | NIPDAUAG
Level-2-260C-1 YEAR
0 to 70
38C45
Samples
UCC38C45DR
ACTIVE
SOIC
D
8
2500
RoHS & Green
NIPDAU
Level-1-260C-UNLIM
0 to 70
38C45
Samples
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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