UCD74106
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SLUSAJ5A – MAY 2011 – REVISED DECEMBER 2012
Synchronous-Buck Power Stage
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FEATURES
1
•
•
•
•
•
•
•
•
•
DESCRIPTION
Fully Integrated Power Switches With Drivers
for Single and Multiphase Synchronous Buck
Converters
Full Compatibility With TI Fusion Digital Power
Supply Controllers, (UCD91xx and UCD92xx
Families)
Compatible With Analog Domain Controllers
Wide Input Voltage Range:
– 4.5 V to 18 V
– Operational Down to 2.2-V Input With an
External Bias Supply
Up to 6-A Output Current
Operational to 2-MHz Switching Frequency
Current Limit With Current Limit Flag
Onboard Regulated 6-V Driver Supply From
VIN
Thermal Protection and Monitoring
The UCD74106 is a complete power system ready to
drive a buck power supply (Figure 1). High-side
MOSFETs, low-side MOSFETs, drivers, current
sensing circuitry and necessary protection functions
are all integrated into one monolithic solution to
facilitate minimum size and maximum efficiency.
Driver circuits provide high charge and discharge
current for the high-side NMOS switch and the lowside NMOS synchronous rectifier in a synchronous
buck circuit. The MOSFET gates are driven to 6.25 V
by an internally regulated VGG supply. The internal
VGG regulator can be disabled to permit the user to
supply an independent gate drive voltage. This
flexibility allows a wide power conversion input
voltage range of 2.2 V to 18 V. Internal Under
Voltage Lockout (UVLO) logic insures VGG is good
before allowing chip operation.
A drive logic block allows operation in one of two
modes. In synchronous mode, the logic block uses
the PWM signal to control both the high-side and lowside gate drive signals. Dead time is optimized to
prevent cross conduction. The synchronous rectifier
enable (SRE) pin controls whether or not the low-side
FET is turned on when the PWM signal is low.
APPLICATIONS
•
•
•
Digitally-Controlled Synchronous-Buck Power
Stage for Single and Multi-Phase Applications
High Efficiency Small Size Regulators for
Desktop, Server, Telecom and Notebook
Applications
Synchronous-Buck Power Stages
Simplified Application Diagram
UCD74106
VIN 13
4
PWM
3
SRE
C1
TMON 10
2
FLT
12 FLTRST
BST
1
SW
6
PGND
7
R2
C3
11 IMON
R1
L1
C3
5
C2
VGG
BP3
AGND
9
8
C4
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2011–2012, Texas Instruments Incorporated
UCD74106
SLUSAJ5A – MAY 2011 – REVISED DECEMBER 2012
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DESCRIPTION (CONT.)
On-board current sense amplifiers monitor the current to safeguard the power stage from sudden high current
loads. In the event of an over-current fault, the output power stage is turned off and the Fault Flag (FLT) is
asserted to alert the controller.
Output current is measured and monitored by a precision integrated current sense element. This method
provides an accuracy of ±5%. The amplified signal is available for use by the controller on the IMON pin. The
IMON pin has a positive offset so that both positive (sourcing) and negative (sinking) current can be sensed.
If the die temperature exceeds 150°C, the temperature sensor initiates a thermal shutdown that halts output
switching and sets the FLT flag. Normal operation resumes when the die temperature falls below the thermal
hysteresis band and the Fault Flag is re-set by the controller.
This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling and installation procedures can cause damage.
ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes could cause the device not to meet its published specifications.
Table 1. ORDERING INFORMATION
OPERATING
TEMPERATURE
RANGE, TA
PIN COUNT
–40°C to 125°C
13-pin
ORDERABLE PART
NUMBER
SUPPLY
UCD74106RGMR
Reel of 2500
UCD74106RGMT
Reel of 250
PACKAGE
TOP SIDE MARKING
QFN
UCD74106
ABSOLUTE MAXIMUM RATINGS (1)
over operating free-air temperature range (unless otherwise noted)
Supply voltage, VIN
Boot voltage, BST
DC
AC
MIN
MAX
-0.3
20
-0.3
SW + 7
(2)
Gate supply voltage, VGG
34
-0.3
7
-2
VIN + 1
Analog outputs, TMON, IMON
-0.3
3.6
Digital I/O’s, PWM, SRE, FLT, FLTRST
-0.3
5.5
Junction temperature, TJ
-55
150
Storage temperature, Tstg
-65
150
Switch voltage, SW
DC
AC
(2)
ESD rating, Human Body Model (HBM)
Lead temperature
(1)
(2)
2
500
Reflow soldering, 10 sec
V
34
2000
ESD rating, Charged Device Model (CDM)
UNIT
300
V
°C
V
°C
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only and functional operation of the device at these or any other condition beyond those indicated is not implied. Exposure to absolutemaximum-rated conditions for extended periods may affect device reliability. All voltages are with respect to GND. Currents are positive
into, negative out of the specified terminal. Consult company packaging information for thermal limitations and considerations of
packages.
AC levels are limited to within 5 ns.
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THERMAL INFORMATION
UCD74106
THERMAL METRIC (1)
RGM
UNITS
13 PINS
Junction-to-ambient thermal resistance (2)
θJA
70.2
(3)
θJCtop
Junction-to-case (top) thermal resistance
θJB
Junction-to-board thermal resistance (4)
11.0
ψJT
Junction-to-top characterization parameter (5)
0.9
ψJB
Junction-to-board characterization parameter (6)
11.0
θJCbot
Junction-to-case (bottom) thermal resistance (7)
0.9
(1)
(2)
(3)
(4)
(5)
(6)
(7)
47.3
°C/W
For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
The junction-to-ambient thermal resistance under natural convection is obtained in a simulation on a JEDEC-standard, high-K board, as
specified in JESD51-7, in an environment described in JESD51-2a.
The junction-to-case (top) thermal resistance is obtained by simulating a cold plate test on the package top. No specific JEDECstandard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
The junction-to-board thermal resistance is obtained by simulating in an environment with a ring cold plate fixture to control the PCB
temperature, as described in JESD51-8.
The junction-to-top characterization parameter, ψJT, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA, using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-board characterization parameter, ψJB, estimates the junction temperature of a device in a real system and is extracted
from the simulation data for obtaining θJA , using a procedure described in JESD51-2a (sections 6 and 7).
The junction-to-case (bottom) thermal resistance is obtained by simulating a cold plate test on the exposed (power) pad. No specific
JEDEC standard test exists, but a close description can be found in the ANSI SEMI standard G30-88.
Spacer
RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
MIN
TYP
MAX
Internally generated VGG
4.5
12
18
Externally supplied VGG
2.2
12
18
4.5
6.2
VIN
Power input voltage
VG
Externally supplied gate drive voltage
UNIT
V
G
TJ
Operating junction temperature range
-40
125
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ELECTRICAL CHARACTERISTICS
VIN = 12 V; 1 μF from BP3 to GND, 0.22 μF from BST to SW, 4.7 µF from VGG to PGND, TA = TJ = -40°C to 125°C (unless
otherwise noted).
PARAMETER
TEST CONDITION
MIN
TYP
MAX UNITS
Supply
Supply current
Outputs not switching, VIN = 2.2 V, VGG = 5 V
4
Supply current
Outputs not switching, VIN = 12 V,
4
mA
Gate Drive Under Voltage Lockout
VGG UVLO ON
BP3 rising
4.0
VGG UVLO OFF
BP3 falling
3.8
VGG UVLO hysteresis
V
200
mV
VGG Supply Generator
VGG
VIN = 7 to 14 V
VGG drop out
VIN = 4.5 to 7 V, IVGG < 20 mA
5.2
6.25
6.8
V
200
mV
3.45
V
BP3 Supply Voltage
BP3
IDD = 0 to 10 mA
3.15
3.3
Input Signal (PWM, SRE)
VIH
Positive-going input threshold
voltage
VIL
Negative-going input threshold
voltage
2.3
Tristate condition
tHLD_R
1.4
3-state hold-off time
VPWM = 1.65 V
IPWM input current
VPWM = 5.0 V
VPWM = 3.3 V
VPWM = 0 V
ISRE input current
4
V
1
1.9
200
165
-165
VSRE = 5.0 V
1
VSRE = 3.3 V
1
VSRE = 0 V
1
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ns
250
μA
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ELECTRICAL CHARACTERISTICS (continued)
VIN = 12 V; 1 μF from BP3 to GND, 0.22 μF from BST to SW, 4.7 µF from VGG to PGND, TA = TJ = -40°C to 125°C (unless
otherwise noted).
PARAMETER
TEST CONDITION
MIN
TYP
MAX UNITS
FAULT Flag (FLT)
FLT
Output high level
IOH = 500 µA
Output low level
IOL = 500 µA
2.7
V
0.6
Current Limit
Over current threshold
PWM frequency = 1 MHz, VIN = 12 V, VOUT = 1.2
V
6.7
7.5
8.2
4.106
4.322
4.538
A
Current Sense Amplifier
Gain (1)
IMON/ISW, 0.3 ≤ V(IMON) ≤ 1.3 V
Zero amp load offset
0 ≤ V(IMON) ≤ 3.1 V, RIMON = 22.6 kΩ
μA/A
μA
22.1
Thermal Sense
Thermal shutdown (1)
Thermal shutdown hysteresis
155
(1)
°C
30
Temperature sense T (1)
Gain
Temperature sense T offset
TJ = 25 °C, -100 μA ≤ ITMON ≤ 100 µA
10
mV/°C
750
mV
POWER Drive Train
Propagation delay from PWM to
switch node going high (1)
20
High-side MOSFET turn on –
dead Time (1)
3
5
15
Low-side MOSFET turn on –
dead time (1)
3
7
15
Min PWM pulse width (1)
(1)
(2)
ns
20 (2)
As designed and characterized, not fully tested in production.
There is no inherent limit on the minimum pulse width. Depending on the board layout, partial enhancement of the high-side FET may
be observed for shorter pulse widths.
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DEVICE INFORMATION
RGM Package
(Top View)
BST
1
FLT
2
SRE
3
PWM
4
VGG
5
13 VIN
12
FLTRST
11 IMON
SW 6
7 PGND
10
TMON
9
Bp3
8
AGND
BLOCK DIAGRAM
PWM
UCD74106
TMON
Thermal
Sense
4
FLTRST
12
12
SRE
R2
FLT
3
Drive Logic
2
VIN
VIN
13
C1
IMON
Currennt
Sense
Processor
VGG
Generator
11
R1
BST
1
VIN
C3
Driver
VGG
SW
5
C2
6
VOUT
C4
3.3-V
Generator
Driver
BP3
9
L1
PGND
AGND
7
8
C4
Figure 1. Typical Block Diagram
6
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TERMINAL FUNCTIONS
TERMINAL
I/O
FUNCTION
NAME
NO.
PWM
4
I
20-kΩ input capable of accepting 3.3-V or 5-V logic level signals up to 2 MHz. A
Schmitt trigger input comparator desensitizes this pin from external noise. This pin
controls the state of the high-side MOSFET and the low-side MOSFET when SRE is
high. When PWM is in HiZ state the output power stage is turned off within 200 ns.
SRE
3
I
Synchronous rectifier enable input. High impedance digital input capable of accepting
3.3-V or 5-V logic level signals used to control the synchronous rectifier switch. An
appropriate anti-cross-conduction delay is used during synchronous mode.
BST
1
I
Charge pump capacitor connection. It provides a floating supply for the high-side
driver. Connect a 0.22-µF ceramic capacitor from this pin to SW.
VGG
5
I/O
Gate drive voltage for the power MOSFETs. For VIN > 4.5 V, the internal VGG
generator can be used. For VIN < 4.5 V, this pin should be driven from an external
bias supply. In all cases, bypass this pin with a 4.7-µF (min), 10-V (min) ceramic
capacitor to PGND.
BP3
9
O
Output of internal 3.3-V LDO regulator for powering internal logic circuits. Bypass this
pin with 1 µF (min) to AGND . This LDO is supplied by the VGG pin.
O
Current sense monitor output. Provides a current source output that is proportional to
the current flowing in the low-side MOSFET. The gain on this pin is equal to 4.32
µA/A. The IMON pin should be connected to a 22.6-kΩ resistor to AGND to produce
a voltage proportional to the power-stage load current. The IMON pin sources 22.1
µA at no load. This provides a pedestal that permits the reporting of negative
(sinking) current.
O
Temperature sense pin. The voltage on this pin is proportional to the die temperature.
The gain is 10 mV/°C . At TJ = 25°C, the output voltage has an offset of 0.75 V.
When the die temperature reaches the thermal shutdown threshold, this pin is pulled
to BP3 and power FETs are switched off. Normal operation resumes when the die
temperature falls below the thermal hysteresis band.
IMON
TMON
11
10
FLT
2
O
Fault flag. This signal is a 3.3-V digital output which is latched high when the load
current exceeds the current limit trip point. When tripped both high side and low side
are latched off. See FLT clear protocol as defined by FLTRST. Additionally, if the die
temperature exceeds 150°C, VIN and/or VGG is outside of UVLO limits, the output
switching will be halted and FLT flag is set. Normal operation resumes after fault
clear sequence is complete.
FLTRST
12
I
Fault reset mode.
PGND
7
-
Shared power ground return for the buck power stage
SW
6
-
Switching node of the buck power stage and square wave input to the buck inductor.
Electrically this is the connection of the high-side MOSFET source to the low-side
MOSFET drain.
VIN
13
-
Input voltage to the buck power stage and driver circuit
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TYPICAL CHARACTERISTICS
Typical Efficiency
DRIVER EFFICIENCY
vs
LOAD CURRENT
DRIVER DISSIPATION
vs
LOAD CURRENT
100
1.8
90
1.6
80
1.4
Driver Dissipation (W)
70
Driver Efficiency (%)
12 VIN ->3.3 VOUT at 1 MHz
12 VIN ->2.5 VOUT at 1 MHz
8 VIN ->1.2 VOUT at 1 MHz
12 VIN ->1.2 VOUT at 1 MHz
12 VIN ->800 mVOUT at 1 MHz
60
50
40
30
12 VIN ->3.3 VOUT at 1 MHz
12 VIN ->2.5 VOUT at 1 MHz
8 VIN ->1.2 VOUT at 1 MHz
12 VIN ->1.2 VOUT at 1 MHz
12 VIN ->800 mVOUT at 1 MHz
20
10
0
0
1
3
2
4
5
1.2
1.0
0.8
0.6
0.4
0.2
0
6
0
1
Load Current (A)
2
3
4
5
6
Load Current (A)
Figure 2.
Figure 3.
PWM and SRE Behavior
The PWM and SRE (Synchronous Rectifier Enable) pins control the high-side and low-side drivers, as described
in Table 2.
Table 2. PWM and SRE Behavior
SRE = High
SRE = Low
8
PWM = High
HS = ON, LS = OFF
HS = ON, LS = OFF
PWM = Low
HS = OFF, LS = ON
HS = OFF, LS = OFF
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PWM = HiZ
HS = OFF, LS = OFF
HS = OFF, LS = OFF
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Fault Modes
Fault Reset Mode
The Fault Reset Mode can be programmed with the FLTRST pin, as described in Table 3
Table 3.
Mode
1
2
3
FLTRST
GND
BP3
Open
FLT Clear
PWM = HiZ
PWM = 0, SRE = 0
PWM = 1 pulse
MODE 1
UCD74106
ILOAD
SW
FLT
UCD92k
PWM
INT
UCD74106
Tri Stated
UCD PWM EN
UCD PWM EN
UDG-11067
PWM DIS HiZ
UCD74106
INT
FLT
UCD9X
2
Figure 4. Fault Handshake Protocol, (PWM HiZ)
NOTE
Handshake Sequence:
1. UCD74106 detects fault condition – FLT flag is set
2. FLT flag generates UCD interrupt
3. UCD releases PWM -HiZ state
4. UCD74106 detects (HiZ) as FLT clear; if fault condition is no longer present then flag is cleared
5. If FLT clears UCD responds to the Start command, Else PWM stays in HiZ state
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MODE 2
SRE
PWM
INT_UCD
FLT
UCD PWM Enabled
PWM
DIS
UCD74106
Tri Stated
PWM EN
Figure 5. Fault Handshake Protocol, (PWM SRE)
NOTE
Handshake Sequence:
1. UCD74106 detects fault condition – FLT is set
2. FLT flag generates interrupt
3. UCD sets SRE low AND stops PWM
4. UCD74106 reads (SRE and PWM) = low as FLT clear; if fault condition is no longer present
then flag is cleared.
5. If FLT clears UCD responds to the Start command, Else (PWM and SRE) = low
10
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MODE 3
PWM
FLT
FLT_FLAG
UCD PWM Enabled
UCD74106 Tri Stated
UDG-11083
Area of Uncertainty
Figure 6. Fault Handshake Protocol, (PWM pulses)
NOTE
No Handshake Re-Set:
1. UCD74106 detects fault condition – FL_FLAG is set
2. No action on UCD side.
3. UCD74106 reads one complete PWM pulse without FLT being present and re-sets FLT_FLAG
signal on the falling edge of PWM.
4. Within the area of uncertainty: FLT rising edge to FLT_FLAG rising edge delay is zero (gate
delay only).
5. PWM falling edge to FLT_FLAG falling edge delay is zero (gate delay only).
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The high-side current limit fault behavior shown in Figure 7.
PWM
ILIMIT
IL
HS
LS
FLT
UDG-11069
Figure 7.
In general, FLT is always cleared by the first complete PWM pulse (a rising and a falling edge) without a fault
present. This is true for all faults including UV and OT. The only exception to this occurs during start up where
FLT will self clear once UVLO is disabled, as shown in Figure 8. However, if a subsequent under voltage
condition occurs the fault must be cleared by one complete PWM pulse without a fault, as shown in Figure 7.
PWM
SRE
HS
LS
UVL
O
FLT
UDG-11070
? t4
Figure 8.
12
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IMON Behavior
The plot in Figure 9 shows how the voltage on the IMON pin will behave with a 22.6-kΩ resistor. The solid dark
line represents the typical behavior and the shaded region represents the tolerance band due too gain.
VIMON
vs
LOAD CURRENT
VIMON (V)
1.0
0.5
0
-2
0
2
4
6
Load Current (A)
Figure 9.
MTTF
vs
DUTY CYCLE
100000
100
110
10000
120
130
140
MTTF (years)
1000
150
100
10
1
0
1
10
100
Duty Cycle (%)
Figure 10.
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TMON Behavior
The voltage on this pin is proportional to the die temperature:
TMON = TOFFSET + TGAIN ´ TJ
(1)
Table 4. Temperature Sense Definitions
NAME
TMON
TOFFSET
TGAIN
TJ
DESCRIPTION
Voltage from TMON pin to GND
Thermal sense T offset
Thermal sense T gain
Device internal junction temperature
TMON
vs
JUNCTION TEMPERATURE
2
1.8
1.6
TMON Voltage (V)
1.4
1.2
1
0.8
0.6
0.4
0.2
0
-40
-20
0
20
40
60
80
100
120
Junction Temperature (°C)
Figure 11.
If the junction temperature exceeds approximately 155°C, the device will enter thermal shutdown. This will assert
the FLT pin, both MOSFETs will be turned off and the switch node will become high impedance. When the
junction temperature cools by approximately 30°C, the device will exit thermal shutdown and resume switching
as directed by the PWM and SRE pins.
14
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APPLICATION INFORMATION
Operating Frequency
Switching frequency is a key place to start the design of any DC/DC converter. This will set performance limits on
things such as: maximum efficiency, minimum size, and achievable closed loop bandwidth. A higher switching
frequency is, generally, going to yield a smaller design at the expense of a lower efficiency. The size benefit is
principally a result of the smaller inductor and capacitor energy storage elements needed to maintain ripple and
transient response requirements. The additional losses result from a variety of factors, however, one of the
largest contributors is the loss incurred by switching the MOSFETs on and off. The integrated nature of the
UCD74106 makes these losses drastically smaller and subsequently enables excellent efficiency from a few
hundred kHz up to the low MHz. For a reasonable trade off of size versus efficiency, 750 kHz is a good place to
start.
VGG
If 4.5 V < VIN ≤ 6 V then a simple efficiency enhancement can be achieved by connecting VGG directly to VIN.
This allows the solution to bypass the drop-out voltage of the internal VGG linear regulator, subsequently
improving the enhancement of the MOSFETs. When doing this it is critical to make sure that VGG never exceeds
the absolute maximum rating of 7 V.
Inductor selection
There are three main considerations in the selection of an inductor once the switching frequency has been
determined. Any real world design is an iterative trade off of each of these factors.
1. The electrical value which in turn is driven by:
(a) RMS current
(b) The maximum desired output ripple voltage
(c) The desired transient response of the converter
2. Losses
(a) Copper (PCu)
(b) Core (Pfe)
3. Saturation characteristics of the core
Inductance Value
The principle equation used to determine the inductance is:
di (t)
vL (t) = L L
dt
(2)
During the on time of the converter the inductance can be solved to be:
V - VOUT D
L = IN
fS
DI
(3)
Table 5. Definitions
VIN
VOUT
Input voltage
Output voltage
fS
Switching frequency
D
Duty cycle (VOUT/VIN for a buck converter)
ΔI
The target peak-to-peak inductor current
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In general, it is desirable to make ΔI large to improve transient response and small to reduce output ripple
voltage and RMS current. A number of considerations go into this however, ΔI = 0.4 IOUT results in a small ILRMS
without an unnecessary penalty on transient response. It also creates a reasonable ripple current that most
practical capacitor banks can handle. Here IOUT is defined as the maximum expected steady state current.
Plugging these assumptions into the above inductance equation results in:
V - VOUT D
L = 5 IN
2 ´ IOUT fS
(4)
For example, plotting this result as a function of VIN and VOUT results in:
•
•
•
•
•
•
IOUT = 6.0 A
ƒS = 1.0 MHz
NCRIT = 5
ΔI/IOUT = 40%
ΔI = 2.4 A
VOUT (V)
Figure 12. VIN and VOUT
NOTE
The maximum inductance occurs at the maximum VIN and VOUT shown in Figure 12. In
general, this inductance value should be used in order to keep the inductor ripple current
from becoming too large over the range of supported VIN and VOUT.
16
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Inductor Losses and Saturation
The current rating of an inductor is based on two things, the current necessary to raise the component
temperature by 40°C and the current level necessary to reduce the inductance to 80% of its initial value
(saturation current ). The current rating is the lower of these two numbers. Both of these factors are influenced by
the choice of core material. Popular materials currently in use are: ferrite, powdered alloy and powdered iron.
Ferrite is regarded as the highest performance material and as such is the lowest loss and the highest cost. Solid
ferrite all by itself will saturate with a relatively small amount of current. This can be addressed by inserting a gap
into the core. This, in effect, makes the inductor behave in a linear manner over a wide DC current range.
However, once the inductance begins to roll off, these gapped materials exhibit a “sharp” saturation
characteristic. In other words the inductance value reduces rapidly with increases in current above the saturation
level. This can be dangerous if not carefully considered, in that the current can rise to dangerous levels.
Powdered iron has the advantage of lower cost and a soft saturation characteristic; however, its losses can be
very large as switching frequencies increase. This can make it undesirable for a UCD74106 based application
where higher switching frequency may be desired. It’s also worth noting that many powdered iron cores exhibit
an aging characteristic where the core losses increase over time. This is a wear out mechanism that needs to be
considered when using these materials.
The powdered alloy cores bring the soft saturation characteristics of powered iron with considerable
improvements in loss without the wear-out mechanism observed in powered iron. These benefits come at a cost
premium.
In general the following relative figure of merits can be made:
Table 6. Core Material Choices
FERRITE
POWDERED ALLOY
POWDERED IRON
COST
High
Medium
Low
LOSS
Low
Medium
High
SATURATION
Rapid
Soft
Soft
When selecting an inductor with an appropriate core it’s important to have in mind the following:
• ILRMS, maximum RMS current
• ΔI, maximum peak-to-peak current
• IMAX, maximum peak current
The RMS current can be determined by the following equation:
ILRMS = IOUT 2 +
DI2
12
(5)
When the 40% ripple constraint is used at maximum load current, this equation simplifies to: ILRMS ≈ IOUT.
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It is widely recognized that the Steinmetz equation (Pfe) is a good representation of core losses for sinusoidal
stimulation. It is important to recognize that this approximation applies to sinusoidal excitation only. This is a
reasonable assumption when working with converters whose duty cycles are near 50%, however, when the duty
cycle becomes narrow this estimate may no longer be valid and considerably more loss may result.
Pfe = k ´ f a ´ B ACb
(6)
The principle drivers in this equation are the material and its respective geometry (k, α, β), the peak AC flux
density (BAC) and the excitation frequency (f). The frequency is simply the switching frequency of the converter
while the constant k, can be computed based on the effective core volume (Ve) and a specific material constant
(kfe).
k = k fe ´ Ve
(7)
The AC flux density (BAC) is related to the conventional inductance specifications by the following relationship:
L DI
B AC =
Ae ´ N 2
(8)
Where L is the inductance, Ae, is the effective cross sectional area that the flux takes through the core and N is
the number of turns.
Some inductor manufactures use the inductor ΔI as a figure of merit for this loss, since all of the other terms are
a constant for a given component. They may provide a plot of core loss versus ΔI for various frequencies where
ΔI can be calculated as:
V - VOUT D
DI = IN
L
fS
(9)
IMAX has a direct impact on the saturation level. A good rule of thumb is to add 15% of head room to the
maximum steady state peak value to provide some room for transients.
DI ö
æ
IMAX = 1.15 ´ ç IOUT + ÷
2ø
è
(10)
For example for a 6-A design has the following:
Table 7. 6-A Design: Inductor Current Requirements
IOUT
6A
ILRMS
6A
ΔI
2.4 A
IMAX
9.6 A
Armed with this data one can now approach the inductor datasheet to select a part with a saturation limit above 9
A and current heating limit above 6 A. Furthermore total losses can be estimated based on the datasheet DCR
value (ILRMS2 x DCR) and the core loss curves for a given frequency and ΔI.
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Input Capacitance
Due to the non-zero impedance of the power planes of the input voltage rail, it is necessary to add some local
capacitance near the UCD74106 to ensure that the voltage at this node is quiet and stable. The primary things to
consider are:
• The radiated fields generated by the di/dt and dv/dt from this node
• RMS currents capability needed in the capacitors
• The AC voltage present and respective susceptibility of any device connected to this node
ICINRMS = IOUT 2 ´ D ´ (1 - D) +
DI2
´D
12
(11)
As a point of reference if ΔI = 0.4 IOUT this places the worst case ICINRMS at approximately 3 A. This corresponds
to a duty cycle of approximately 50%. Other duty cycles can result in a significantly lower RMS current.
A good input capacitor would be a 22-µF X5R ceramic capacitor. Equally important as selecting the proper
capacitor is placing and routing that capacitor. It is crucial that the decoupling be placed as close as possible to
both the power pin (VIN) and ground (PGND). It is important to recognize that each power stage should have its
own local decoupling. One 22-µF capacitor should be placed across each VIN and PGND pair. The proximity of
the capacitance to these pins will reduce the radiated fields mentioned above.
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Output Capacitance
The goal of the output capacitor bank is to keep the output voltage within regulation limits during steady state
and transient conditions.
The total AC RMS current flowing through the capacitor bank can be calculated as:
DI
ICOUTRMS =
12
(12)
For a single type of output capacitor the output ripple voltage wave form can be approximated by the following
equation:
1 t
VOUT (t) = IC (t) ´ esr +
IC ( t) ´ dt
C 0
(13)
ò
Where:
DI ´ fs
ì
D
DI
t<
´tï
D
2
fs
ï
IC (t) = í
ï DI ´ fS ´ æ t - D ö + DI otherwise
÷
ï 1- D ç
fS ø 2
è
î
(14)
After substitution and simplification yields:
ì
æ DI ´ fS
DI ö 1 æ t ´ DI ´ (fS ´ t - D ) DI ´ (1 - 2 ´ D ) ö
D
ï
´t - ÷ + ´ç
esr ´ ç
t<
÷
ç
÷
´
´
D
2
C
2
D
12
f
f
ïï
è
ø
S
S
è
ø
VOUT (t) = í
æ DI ´ fS æ
ï
D ö DI ö 1 æ DI ´ (fS ´ t - 1)´ (D - fS ´ t ) DI ´ (1 - 2 ´ D ) ö
÷ otherwise
ïesr ´ çç 1 - D ´ ç t - f ÷ + 2 ÷÷ + C ´ çç
÷
2 ´ (1 - D )´ fS
12 ´ fS
S ø
è
ïî
è
ø
è
ø
(15)
The term in this equation multiplied by the esr gives the ripple voltage component due to esr and the term
multiplied by 1/C gives the ripple voltage component due to the change in charge on the capacitor plates. In the
case were the esr component dominates the peak-to-peak output voltage can be approximated as:
VPPesr » DI ´ esr
(16)
When the charge term dominates the peak-to-peak voltage ripple becomes:
DI
VPPQ »
8 ´ C ´ fS
(17)
It’s tempting to simply add these two results together for the case where the voltage ripple is significantly
influenced by both the capacitance and the esr. However, this will yield an overly pessimistic result, in that it
does not account for the phase difference between these terms.
Using the ripple voltage equations and the RMS current equation should give a design that safely meets the
steady state output requirements. However, additional capacitance is often needed to meet transient
requirements and the specific local decoupling requirements of any device that is being powered off of this
voltage. This is not just a function of the capacitor bank but also the dynamics of the control loop. See the
UCD9240 Compensation Cookbook for additional details (TI Literature Number SLUA497).
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Decoupling
It is necessary that VGG and BP3 have their own local capacitance as physically close as possible to these pins.
The VGG capacitor should be connected as close as possible to the VGG pin and PGND with a 4.7-µF ceramic
capacitor. The BP3 capacitor should be connected as close as possible to BP3 pin and AGND with a 1-µF
ceramic capacitor.
The UCD74106 also supports the ability to operate from input voltages down to 2.2 V. In these cases an
additional supply rail must be connected to VGG. Potential external bias supply generators for low VIN operation:
TPS63000, TPS61220.
Current Sense
An appropriate resistor must be connected to the current sense output pins to convert the IMON current to a
voltage. In the case of the UCD92xx digital controllers, these parts have a full scale current monitor range of 0 V
to 2 V. This range can be maximized to make full use of the current monitoring resolution inside the controller.
VMON(min)
VMON(max)
£ RMON £
IOFFSET + IMIN ´ IGAIN
IOFFSET + IMAX ´ IGAIN
(18)
Table 8. Current Sense Definitions
NAME
RMON
DESCRIPTION
Resistor from IMON pin to GND
VMON(min)
Minimum voltage for IMON (typically, 0.2 V)
VMON(max)
Maximum voltage for IMON (typically, 1.8 V)
IMIN
Minimum load current to sense
IMAX
Maximum load current to sense
IOFFSET
IGAIN
Current sense amplifier zero amp load offset
Current sense amplifier gain
The recommended 22.6-kΩ resistor can be used to keep IMON within range for sensing load currents below -2 A
to above 6 A.
In some applications it may be necessary to filter the IMON signal. The UCD74106 IMON pin is a current source
output, so a capacitor to ground in parallel with the current-to-voltage conversion resistor is all that is required.
As a rule of thumb, placing the corner frequency of the filter at 20% of the switching frequency should be
sufficient.
For example, if the switching frequency is 500 kHz or higher, the ripple frequency will be easily rejected with a
corner frequency of approximately 100 kHz. With a 100-kHz pole point, the filter time constant is 1.6 µs. A fast
current transient should be detected within 4.8 µs.
1
CMON =
2 ´ p ´ RMON ´ 20% ´ fS
(19)
Layout Recommendations
The primary thermal cooling path is from the VIN, GND, and the SW stripes on the bottom of the package. Wide
copper traces should connect to these nodes. 1-ounce copper should be the minimum thickness of the top layer;
however, 2 ounce is better. Multiple thermal vias should be placed near the GND stripes which connect to a PCB
ground plane. There is room to place multiple 10 mil (0.25 mm) diameter vias next to the VIN and GND stripes
under the package.
For input bypassing, the 22-µF input ceramic caps should be connected as close as possible to the VIN and
GND stripes. If possible, the input caps should be placed directly under the UCD74106 using multiple 10-mil vias
to bring the VIN and GND connections to the back side of the board. Minimizing trace inductance in the bypass
path is extremely important to reduce the amplitude of ringing on the switching node.
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REVISION HISTORY
Changes from Original (May, 2011) to Revision A
Page
•
Changed Wide Input Voltage Range from 14 V to 18 V. ...................................................................................................... 1
•
Added updated ABSOLUTE MAXIMUM RATINGS information. .......................................................................................... 2
•
Changed Power input voltage max rating from 16 V to 18 V. .............................................................................................. 3
•
Changed Output high level max rating from 4 mA to 500 µA. .............................................................................................. 5
•
Changed Output low level max rating from -4 mA to -500 µA. ............................................................................................. 5
•
Changed GND to AGND. ...................................................................................................................................................... 7
•
Changed 4.4 to 4.32. ............................................................................................................................................................ 7
•
Changed GND to AGND. ...................................................................................................................................................... 7
•
Changed 12 mV/°C to 10 mV/°C. ......................................................................................................................................... 7
•
Changed LS = ON to LS = OFF. .......................................................................................................................................... 8
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PACKAGE OPTION ADDENDUM
www.ti.com
10-Dec-2020
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
(2)
Lead finish/
Ball material
MSL Peak Temp
Op Temp (°C)
Device Marking
(3)
(4/5)
(6)
UCD74106RGMR
ACTIVE
VQFN-HR
RGM
13
3000
RoHS-Exempt
& Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
74106
UCD74106RGMT
ACTIVE
VQFN-HR
RGM
13
250
RoHS-Exempt
& Green
NIPDAU
Level-1-260C-UNLIM
-40 to 125
74106
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of
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