VCA2616

VCA 261 6 VCA2616 VCA2611 SBOS234E – MARCH 2002 – REVISED NOVEMBER 2004 Dual, Variable-Gain Amplifier with Low-Noise Preamp FEATURES q LOW-NOISE PREAMP: – Low Input Noise: 0.95nV/√Hz – Active Termination Noise Reduction – Switchable Termination Value – 80MHz Bandwidth – 5dB to 25dB Gain – Differential In and Out q LOW-NOISE VARIABLE GAIN AMPLIFIER: – Low-Noise VCA – Up to 40dB Gain Range – 40MHz Bandwidth – Differential In and Out q LOW CROSSTALK: 66dB at Max Gain, 5MHz q HIGH-SPEED VARIABLE GAIN ADJUST q SWITCHABLE EXTERNAL PROCESSING DESCRIPTION The VCA2616 and VCA2611 are dual, Low-Noise Preamplifiers (LNP), plus low-noise Variable Gain Amplifiers (VGA). The VCA2611 is an upgraded version of the VCA2616. The only difference between the VCA2616 and the VCA2611 is the input structure to the LNP. The VCA2616 is limited to –0.3V negativegoing input spikes; the VCA2611 is limited to –2.0V negativegoing input spikes. This change allows the user to use slower and less expensive input clamping diodes prior to the LNP input. In some designs, input clamping may not be required. The combination of Active Termination (AT) and Maximum Gain Select (MGS) allow for the best noise performance. The VCA2616 and VCA2611 also feature low crosstalk and outstanding distortion performance. The LNP has differential input and output capability and is strappable for gains of 5dB, 17dB, 22dB, or 25dB. Low input impedance is achieved by AT, resulting in as much as a 4.6dB improvement in noise figure over conventional shunt termination. The termination value can also be switched to accommodate different sources. The output of the LNP is available for external signal processing. The variable gain is controlled by an analog voltage whose gain varies from 0dB to the gain set by the MGS. The ability to program the variable gain also allows the user to optimize dynamic range. The VCA input can be switched from the LNP to external circuits for different applications. The output can be used in either a single-ended or differential mode to drive highperformance Analog-to-Digital (A/D) converters, and is cleanly limited for optimum overdrive recovery. Maximum Gain Select VCACNTL MGS1 MGS2 MGS3 APPLICATIONS q ULTRASOUND SYSTEMS q WIRELESS RECEIVERS q TEST EQUIPMENT FBCNTL RF2 RF1 LNPOUTN VCAINN FBSW FB VCA2616 (1 of 2 Channels) Analog Control Maximum Gain Select The combination of low noise, gain, and gain range programmability makes the VCA2616 and VCA2611 versatile building blocks in a number of applications where noise performance is critical. The VCA2616 and VCA2611 are available in a TQFP-48 package. Input LNPINP VCAOUTN LNPGS1 LNPGS2 LNPGS3 LNPINN VCAOUTP Programmable Gain Amplifier 24 to 45dB LNP Gain Set Low Noise Preamp 5dB to 25dB Voltage Controlled Attenuator LNPOUTP VCAINP SEL Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2002-2004, Texas Instruments Incorporated www.ti.com ABSOLUTE MAXIMUM RATINGS(1) Power Supply (+VS) ............................................................................. +6V VCA2616 Analog Input ............................................ –0.3V to (+VS + 0.3V) VCA2611 Analog Input ............................................ –2.0V to (+VS + 0.3V) Logic Input ............................................................... –0.3V to (+VS + 0.3V) Case Temperature ......................................................................... +100°C Junction Temperature .................................................................... +150°C Storage Temperature ...................................................... –40°C to +150°C NOTE: (1) Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. Exposure to absolute maximum conditions for extended periods may affect device reliability. ELECTROSTATIC DISCHARGE SENSITIVITY This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. PACKAGE/ORDERING INFORMATION(1) PACKAGE DESIGNATOR PFB SPECIFIED TEMPERATURE RANGE –40°C to +85°C PRODUCT VCA2616 PACKAGE-LEAD TQFP-48 PACKAGE MARKING VCA2616 ORDERING NUMBER VCA2616YT VCA2616YR VCA2611Y/250 VCA2611Y/2K TRANSPORT MEDIA, QUANTITY Tape and Reel, 250 Tape and Reel, 2000 Tape and Reel, 250 Tape and Reel, 2000 " VCA2611 " TQFP-48 " PFB " –40°C to +85°C " VCA2611 " " " " " NOTE: (1) For the most current package and ordering information, see the Package Option Addendum located at the end of this data sheet. ELECTRICAL CHARACTERISTICS At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted. The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. VCA2616Y, VCA2611Y PARAMETER PREAMPLIFIER Input Resistance Input Capacitance Input Bias Current CMRR Maximum Input Voltage Input Voltage Noise(1) Input Current Noise Noise Figure, RS = 75Ω, RIN = 75Ω(1) Bandwidth CONDITIONS MIN TYP 600 15 1 50 1 112 4.2 0.95 0.35 6.2 80 2 40 300 2 1 ±40 –71 –63 –75 –75 –66 ±2 2.5 10.9 ±50 21 50 0.2 to 3.0 1 0.2 –40 4.75 Operating, Both Channels +85 5.25 495 ±1(2) 24 53 MAX UNITS kΩ pF nA dB VPP mVPP nV/ √Hz nV/ √Hz pA/ √Hz dB MHz VPP MHz V/µs VPP Ω mA dBc dBc dBc dBc dB ns V dB/V dB mV dB dB V MΩ µs °C V mW f = 1MHz, VCACNTL = 0.2V Preamp Gain = +5dB Preamp Gain = +25dB Preamp Gain = +5dB Preamp Gain = +25dB Independent of Gain RF = 550Ω, Preamp Gain = 22dB, PGA Gain = 39dB Gain = 22dB PROGRAMMABLE VARIABLE GAIN AMPLIFIER Peak Input Voltage Differential –3dB Bandwidth Slew Rate Output Signal Range RL ≥ 500Ω Each Side to Ground Output Impedance f = 5MHz Output Short-Circuit Current 3rd-Harmonic Distortion f = 5MHz, VOUT = 1VPP, VCACNTL = 3.0V 2nd-Harmonic Distortion f = 5MHz, VOUT = 1VPP, VCACNTL = 3.0V IMD, 2-Tone VOUT = 2VPP, f = 1MHz VOUT = 2VPP, f = 10MHz Crosstalk VCACNTL = 0.2V Group Delay Variation 1MHz < f < 10MHz, Full Gain Range DC Output Level, VIN = 0 ACCURACY Gain Slope Gain Error Output Offset Voltage Total Gain GAIN CONTROL INTERFACE Input Voltage (VCACNTL) Range Input Resistance Response Time POWER SUPPLY Operating Temperature Range Specified Operating Range Power Dissipation NOTES: (1) For preamp driving VGA. (2) Referenced to best fit dB-linear curve. –45 –45 VCACNTL = 0.2V VCACNTL = 3.0V 18 47 40dB Gain Change, MGS = 111 5.0 410 2 VCA2616, VCA2611 www.ti.com SBOS234E PIN CONFIGURATION Top View VCAOUTNA VCAOUTPB VCAINSEL VCAOUTNB VCAOUTPA FBSWCNTL TQFP VCACNTL GNDA MGS1 MGS2 48 VDDA NC NC VCAINNA VCAINPA LNPOUTNA LNPOUTPA SWFBA FBA 1 2 3 4 5 6 7 8 9 47 46 45 44 43 42 41 40 MGS3 39 38 37 36 VDDB 35 NC 34 NC 33 VCAINNB 32 VCAINPB GNDB 31 LNPOUTNB 30 LNPOUTPB 29 SWFBB 28 FBB 27 COMP1B 26 COMP2B 25 LNPINNB 24 VCA2616 VCA2611 COMP1A 10 COMP2A 11 LNPINNA 12 13 14 15 16 17 18 19 20 21 22 23 LNPGS1B LNPGS2B LNPGS2A LNPGS1A PIN DESCRIPTIONS PIN 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 21 22 23 24 DESIGNATOR VDDA NC NC VCAINNA VCAINPA LNPOUTNA LNPOUTPA SWFBA FBA COMP1A COMP2A LNPINNA LNPGS3A LNPGS2A LNPGS1A LNPINPA VDDR VBIAS VCM GNDR LNPINPB LNPGS1B LNPGS2B LNPGS3B DESCRIPTION Channel A +Supply Do Not Connect Do Not Connect Channel A VCA Negative Input Channel A VCA Positive Input Channel A LNP Negative Output Channel A LNP Positive Output Channel A Switched Feedback Output Channel A Feedback Output Channel A Frequency Compensation 1 Channel A Frequency Compensation 2 Channel A LNP Inverting Input Channel A LNP Gain Strap 3 Channel A LNP Gain Strap 2 Channel A LNP Gain Strap 1 Channel A LNP Noninverting Input +Supply for Internal Reference 0.01µF Bypass to Ground 0.01µF Bypass to Ground Ground for Internal Reference Channel B LNP Noninverting Input Channel B LNP Gain Strap 1 Channel B LNP Gain Strap 2 Channel B LNP Gain Strap 3 PIN 25 26 27 28 29 30 31 32 33 34 35 36 37 38 39 40 41 42 43 44 45 46 47 48 DESIGNATOR LNPINNB COMP2B COMP1B FBB SWFBB LNPOUTPB LNPOUTNB VCAINPB VCAINNB NC NC VDDB GNDB VCAOUTNB VCAOUTPB MGS3 MGS2 MGS1 VCACNTL VCAINSEL FBSWCNTL VCAOUTPA VCAOUTNA GNDA DESCRIPTION Channel B LNP Inverting Input Channel B Frequency Compensation 2 Channel B Frequency Compensation 1 Channel B Feedback Output Channel B Switched Feedback Output Channel B LNP Positive Output Channel B LNP Negative Output Channel B VCA Positive Input Channel B VCA Negative Input Do Not Connect Do Not Connect Channel B +Analog Supply Channel B Analog Ground Channel B VCA Negative Output Channel B VCA Positive Output Maximum Gain Select 3 (LSB) Maximum Gain Select 2 Maximum Gain Select 1 (MSB) VCA Control Voltage VCA Input Select, HI = External Feedback Switch Control: HI = ON Channel A VCA Positive Output Channel A VCA Negative Output Channel A Analog Ground VCA2616, VCA2611 SBOS234E LNPGS3B LNPGS3A LNPINPB LNPINPA VDDR VBIAS VCM GNDR www.ti.com 3 TYPICAL CHARACTERISTICS At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted. The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal amplitude compared to differential operation. 65 60 55 50 Gain (dB) GAIN vs VCACNTL MGS = 101 MGS = 100 GAIN ERROR vs TEMPERATURE 2.0 1.5 MGS = 110 MGS = 111 Gain Error (dB) 1.0 0.5 0.0 –0.5 –1.0 –40°C +85°C +25°C 45 40 35 30 25 20 15 MGS = 010 MGS = 011 MGS = 000 MGS = 001 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) –1.5 –2.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) GAIN ERROR vs VCACNTL 2.0 1.5 1.0 1MHz 5MHz 10MHz 2.0 1.5 1.0 GAIN ERROR vs VCACNTL Gain Error (dB) 0.5 0.0 –0.5 –1.0 –1.5 –2.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) Gain Error (dB) MGS = 000 0.5 0 –0.5 –1.0 MGS = 111 –1.5 –2.0 MGS = 011 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) 50 45 40 35 30 Units GAIN MATCH 0.2V CHA to CHB 60 50 40 GAIN MATCH 3.0V CHA to CHB Units 0.23 –0.29 –0.42 0.29 0.36 25 20 15 10 5 0 0.10 –0.22 –0.03 0.16 –0.16 –0.09 –0.48 0.04 –0.55 –0.35 30 20 10 0 –0.14 0.12 –0.20 –0.10 0.01 0.05 0.07 0.04 –0.23 Delta Gain (dB) –0.26 –0.17 Delta Gain (dB) 4 VCA2616, VCA2611 www.ti.com SBOS234E 0.02 0.09 0.15 0.18 TYPICAL CHARACTERISTICS (Cont.) At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted. The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal amplitude compared to differential operation. GAIN vs FREQUENCY GAIN vs FREQUENCY VCA (Pre-Amp) (VCACNTL = 0.2V) 30 5.0 LNA = 25dB LNA = 22dB 4.0 25 3.0 20 2.0 Gain (dB) 1.0 0.0 –1.0 –2.0 –3.0 LNA = 5dB 0 100k 1M Frequency (Hz) 10M 100M –4.0 –5.0 100k MGS = 000 MGS = 011 MGS = 111 MGS = 100 Gain (dB) 15 LNA = 17dB 10 5 1M Frequency (Hz) 10M 100M 45 40 35 30 GAIN vs FREQUENCY VCA (VCACNTL = 3.0V) MGS = 111 GAIN vs FREQUENCY LNA and VCA (VCACNTL = 3.0V) 60 50 LNP = 25dB LNP = 22dB MGS = 100 40 MGS = 011 Gain (dB) 25 20 15 10 5 0 100k 1M Frequency (Hz) 10M 100M MGS = 000 Gain (dB) LNP = 17dB 30 LNP = 5dB 20 10 0 100k 1M Frequency (Hz) 10M 100M GAIN vs FREQUENCY LNA and VCA (LNP = 22dB) 60 VCNTL = 3.0V 50 2000 1800 1600 OUTPUT-REFERRED NOISE vs VCACNTL (LNP = 25dB) RS= 50Ω Noise (nV/√Hz) 40 Gain (dB) 30 20 10 VCNTL = 1.6V 1400 1200 1000 800 600 400 MGS = 111 VCNTL = 0.2V 0 100k 1M Frequency (Hz) 10M 100M 200 0 MGS = 011 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) VCA2616, VCA2611 SBOS234E www.ti.com 5 TYPICAL CHARACTERISTICS (Cont.) At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted. The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended unless otherwise noted. This results in a 6dB reduction in signal amplitude compared to differential operation. INPUT-REFERRED NOISE vs VCACNTL (LNP = 25dB) 24 22 20 18 16 14 12 10 8 6 4 2 0 RS = 50Ω 10.0 INPUT-REFERRED NOISE vs RS (LNP = 25dB) Noise (nV/√Hz) Noise (nV√Hz) MGS = 111 1.0 MGS = 111 MGS = 011 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) 0.1 1 10 RS ( Ω ) 100 1k NOISE FIGURE vs RS (LNP = 25dB) 9 8 7 Noise Figure (dB) 6 5 4 3 2 1 0 10 100 RS (Ω) 1k MGS = 111 Noise Figure (dB) 30 28 26 24 22 20 18 16 14 12 10 8 6 4 2 0 NOISE FIGURE vs VCACNTL (LNP = 25dB) MGS = 111 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) –30 –35 –40 –45 DISTORTION vs FREQUENCY MGS = 000 2VPP DIFFERENTIAL Harmonic (dBc) Distortion (dBc) –50 –55 –60 –65 –70 –75 –80 –85 100k VC = 0.2, H2 VC = 0.2, H3 VC = 3.0, H2 VC = 3.0, H3 –30 –35 –40 –45 –50 –55 –60 –65 –70 –75 –80 –85 –90 DISTORTION vs FREQUENCY MGS = 011 2VPP DIFFERENTIAL VC = 0.2, H2 VC = 0.2, H3 VC = 3.0, H2 VC = 3.0, H3 1M Frequency (Hz) 10M 1M Frequency (Hz) 10M 100k 6 VCA2616, VCA2611 www.ti.com SBOS234E TYPICAL CHARACTERISTICS (Cont.) At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted. The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal amplitude compared to differential operation. –30 –35 –40 Distortion (dBc) DISTORTION vs FREQUENCY MGS = 111 2VPP DIFFERENTIAL VC = 0.2, H2 VC = 0.2, H3 Distortion (dBc) –45 –50 –55 –60 –65 –70 –75 –80 100k VC = 3.0, H3 1M VC = 3.0, H2 –30 –35 –40 –45 –50 –55 –60 –65 –70 –75 –80 –85 –90 DISTORTION vs FREQUENCY MGS = 000 1VPP SINGLE-ENDED VC = 0.2, H3 VC = 0.2, H2 VC = 3.0, H2 VC = 3.0, H3 10M 100k Frequency (Hz) 1M Frequency (Hz) 10M –30 –35 –40 –45 –50 –55 –60 –65 –70 –75 –80 –85 –90 DISTORTION vs FREQUENCY MGS = 011 1VPP SINGLE-ENDED VC = 0.2, H3 VC = 0.2, H2 Distortion (dBc) VC = 3.0, H2 VC = 3.0, H3 –30 –35 –40 –45 –50 –55 –60 –65 –70 –75 –80 –85 –90 100k DISTORTION vs FREQUENCY MGS = 111 1VPP SINGLE-ENDED VC = 0.2, H2 VC = 0.2, H3 Distortion (dBc) VC = 3.0, H2 VC = 3.0, H3 100k 1M Frequency (Hz) 10M 1M Frequency (Hz) 10M DISTORTION vs VCACNTL 2VPP DIFFERENTIAL –45 MGS = 111, H2 –50 DISTORTION vs VCACNTL 1VPP SINGLE-ENDED –45 MGS = 111, H2 –50 Distortion (dBc) Distortion (dBc) –55 MGS = 011, H2 –60 –65 MGS = 000, H3 –70 MGS = 111, H3 –75 –80 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) MGS = 011, H3 MGS = 000, H2 –55 MGS = 011, H2 MGS = 000, H2 –60 –65 –70 MGS = 111, H3 –75 –80 MGS = 000, H3 MGS = 011, H3 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 VCACNTL (V) VCA2616, VCA2611 SBOS234E www.ti.com 7 TYPICAL CHARACTERISTICS (Cont.) At TA = +25°C, VDDA = VDDB = VDDR = +5V, load resistance = 500Ω on each output to ground, MGS = 011, LNP = 22dB and fIN = 5MHz, unless otherwise noted. The input to the preamp (LNP) is single-ended, and the output from the VCA is single-ended, unless otherwise noted. This results in a 6dB reduction in signal amplitude compared to differential operation. CROSSTALK vs FREQUENCY 1VPP SINGLE-ENDED MGS = 011 –5 –15 80.50 80.00 ICC vs TEMPERATURE VCACNTRL 0V –25 Crosstalk (dB) VCACNTRL 1.5V 79.50 79.00 ICC (dBFS) –35 –45 –55 –65 –75 –85 1M 10M Frequency (Hz) 100M VCACNTRL 3.0V 78.50 78.00 77.50 77.00 76.50 76.00 –40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90 Temperature (°C) 16 14 GROUP DELAY vs FREQUENCY Group Delay (nS) 12 10 8 VC = 3.0 VC = 0.2 6 4 2 0 1M 10M Frequency (Hz) 100M 8 VCA2616, VCA2611 www.ti.com SBOS234E THEORY OF OPERATION The VCA2616 and VCA2611 are dual-channel systems consisting of three primary blocks: an LNP, a VCA, and a Programmable Gain Amplifier (PGA). For greater system flexibility, an onboard multiplexer is provided for the VCA inputs, selecting either the LNP outputs or external signal inputs. Figure 1 shows a simplified block diagram of the dualchannel system. op amp. The VCM node shown in Figure 2 is the VCM output (pin 19). Typical R and C values are shown, yielding a highpass time constant similar to that of the LNP. If a different common-mode referencing method is used, it is important that the common-mode level be within 10mV of the VCM output for proper operation. 1kΩ External InA Input Signal Channel A Input LNP VCA PGA Channel A Output 47nF To VCAIN 1kΩ VCM (+2.5V) VCA Control Analog Control Maximum Gain Select MGS FIGURE 2. Recommended Circuit for Coupling an External Signal into the VCA Inputs. Channel B Output Channel B Input LNP VCA PGA VCA—OVERVIEW The magnitude of the differential VCA input signal (from the LNP or an external source) is reduced by a programmable attenuation factor, set by the analog VCA Control Voltage (VCACNTL) at pin 43. The maximum attenuation factor is further programmable by using the three MGS bits (pins 40-42). Figure 3 illustrates this dual-adjustable characteristic. Internally, the signal is attenuated by having the analog VCACNTL vary the channel resistance of a set of shunt-connected FET transistors. The MGS bits effectively adjust the overall size of the shunt FET by switching parallel components in or out under logic control. At any given maximum gain setting, the analog variable gain characteristic is linear in dB as a function of the control voltage, and is created as a piecewise approximation of an ideal dB-linear transfer function. The VCA gain control circuitry is common to both channels of the VCA2616 and VCA2611. External InB FIGURE 1. Simplified Block Diagram of the VCA2616. LNP—OVERVIEW The LNP input may be connected to provide active-feedback signal termination, achieving lower system noise performance than conventional passive shunt termination. Further lower noise performance is obtained if signal termination is not required. The unterminated LNP input impedance is 600kΩ. The LNP can process fully differential or singleended signals in each channel. Differential signal processing results in significantly reduced 2nd-harmonic distortion and improved rejection of common-mode and power-supply noise. The first gain stage of the LNP is AC-coupled into its output buffer with a 4.8µs time constant (33kHz high-pass characteristic). The buffered LNP outputs are designed to drive the succeeding VCA directly or, if desired, external loads as low as 135Ω with minimal impact on signal distortion. The LNP employs very low impedance local feedback to achieve stable gain with the lowest possible noise and distortion. Four pin-programmable gain settings are available: 5dB, 17dB, 22dB, and 25dB. Additional intermediate gains can be programmed by adding trim resistors between the Gain Strap programming pins. The common-mode DC level at the LNP output is nominally 2.5V, matching the input common-mode requirement of the VCA for simple direct coupling. When external signals are fed to the VCA, they should also be set up with a 2.5VDC common-mode level. Figure 2 shows a circuit that demonstrates the recommended coupling method using an external 0 VCA Attenuation (dB) Minimum Attenuation –24 –45 0 Maximum Attenuation 3.0 Control Voltage (V) FIGURE 3. Swept Attenuator Characteristic. VCA2616, VCA2611 SBOS234E www.ti.com 9 PGA OVERVIEW AND OVERALL DEVICE CHARACTERISTICS The differential output of the VCA attenuator is then amplified by the PGA circuit block. This post-amplifier is programmed by the same MGS bits that control the VCA attenuator, yielding an overall swept-gain amplifier characteristic in which the VCA × PGA gain varies from 0dB (unity) to a programmable peak gain of 24-, 27-, 30-, 33-, 36-, 39-, 42-, or 45dB. The Gain vs VCACNTL curve in the Typical Characteristics shows the composite gain control characteristic of the entire VCA2616. Setting VCACNTL to 3.0V causes the digital MGS gain control to step in 3dB increments. Setting VCACNTL to 0V causes all the MGS-controlled gain curves to converge at one point. The gain at the convergence point is the LNP gain less 6dB, because the measurement setup looks at only one side of the differential PGA output, resulting in 6dB lower signal amplitude. the circuit. This reduces the susceptibility to power-supply variation, ripple, and noise. In addition, separate power supply and ground connections are provided for each channel and for the reference circuitry, further reducing interchannel crosstalk. Further details regarding the design, operation, and use of each circuit block are provided in the following sections. LOW-NOISE PREAMPLIFIER (LNP)—DETAIL The LNP is designed to achieve a low-noise figure, especially when employing active termination. Figure 4 is a simplified schematic of the LNP, illustrating the differential input and output capability. The input stage employs low resistance local feedback to achieve stable low-noise, lowdistortion performance with very high input impedance. Normally, low noise circuits exhibit high power consumption as a result of the large bias currents required in both input and output stages. The LNP uses a patented technique that combines the input and output stages such that they share the same bias current. Transistors Q4 and Q5 amplify the signal at the gate-source input of Q4, the +IN side of the LNP. The signal is further amplified by the Q1 and Q2 stage, and then by the final Q3 and RL gain stage, which uses the same bias current as the input devices Q4 and Q5. Devices Q6 through Q10 play the same role for signals on the –IN side. The differential gain of the LNP is given in Equation 1: ADDITIONAL FEATURES—OVERVIEW Overload protection stages are placed between the attenuator and the PGA, providing a symmetrically clipped output whenever the input becomes large enough to overload the PGA. A comparator senses the overload signal amplitude and substitutes a fixed DC level to prevent undesirable overload recovery effects. As with the previous stages, the VCA is AC-coupled into the PGA. In this case, the coupling time constant varies from 5µs at the highest gain (45dB) to 59µs at the lowest gain (25dB). The VCA2616 includes a built-in reference, common to both channels, to supply a regulated voltage for critical areas of R  Gain = 2 ×  L   RS  (1) COMP2A VDD COMP1A Q2 CCOMP 4.7pF (External Capacitor) RL 93Ω LNPOUTN Buffer Q3 RS1 105Ω RS2 34Ω LNPGS2 RS3 17Ω LNPOUTP Buffer RL 93Ω Q9 To Bias Circuitry Q8 RW LNPINP Q4 LNPGS1 Q7 RW LNPINN Q1 LNPGS3 Q10 To Bias Circuitry Q5 Q6 FIGURE 4. Schematic of the Low-Noise Preamplifier (LNP). 10 VCA2616, VCA2611 www.ti.com SBOS234E where RL is the load resistor in the drains of Q3 and Q8, and RS is the resistor connected between the sources of the input transistors Q4 and Q7. The connections for various RS combinations are brought out to device pins LNPGS1, LNPGS2, and LNPGS3 (pins 13-15 for channel A, 22-24 for channel B). These Gain Strap pins allow the user to establish one of four fixed LNP gain options as shown in Table I. LNP PIN STRAPPING LNPGS1, LNPGS2, LNPGS3 Connected Together LNPGS1 Connected to LNPGS3 LNPGS1 Connected to LNPGS2 All Pins Open LNP GAIN (dB) 25 22 17 5 NOISE (nV/√ Hz) LNP GAIN (dB) 25 22 17 5 Input-Referred 1.35 1.41 1.63 4.28 Output-Referred 2260 1650 1060 597 TABLE II. Equivalent Noise Performance for MGS = 111 and VCACNTL = 3.0V with 50Ω source impedance. The LNP is capable of generating a 2VPP differential signal. The maximum signal at the LNP input is therefore 2VPP divided by the LNP gain. An input signal greater than this would exceed the linear range of the LNP, an especially important consideration at low LNP gain settings. The VCA2611 is an upgraded version of the VCA2616. The only difference between the VCA2616 and the VCA2611 is the input structure to the LNP. The VCA2616 is limited to –0.3V negative-going input spikes; the VCA2611 is limited to –2.0V negative-going input spikes. This change allows the user to use slower and less expensive input clamping diodes prior to the LNA input. In some designs, input clamping may not be required. TABLE I. Pin Strappings of the LNP for Various Gains. It is also possible to create other gain settings by connecting an external resistor between LNPGS1 on one side, and LNPGS2 and/or LNPGS3 on the other. In that case, the internal resistor values (see Figure 4) should be combined with the external resistor to calculate the effective value of RS for use in Equation 1. The resulting expression for external resistor value is given in Equation 2: REXT = 2RS1RL + 2RFIXRL – Gain × RS1RFIX Gain × RS1 – 2RL (2) ACTIVE FEEDBACK WITH THE LNP One of the key features of the LNP architecture is the ability to employ active-feedback termination to achieve superior noise performance. Active-feedback termination achieves a lower noise figure than conventional shunt termination, essentially because no signal current is wasted in the termination resistor itself. Another way to understand this is to consider first that the input source, at the far end of the signal cable, has a cable-matching source resistance of RS. Using conventional shunt termination at the LNP input, a second terminating resistor of value RS is connected to ground. Therefore, the signal loss is 6dB due to the voltage divider action of the series and shunt RS resistors. The effective source resistance has been reduced by the same factor of 2, but the noise contribution has been reduced by only the √2 , only a 3dB reduction. Therefore, the net theoretical SNR degradation is 3dB, assuming a noise-free amplifier input. (In practice, the amplifier noise contribution will degrade both the unterminated and the terminated noise figures, somewhat reducing the distinction between them.) See Figure 5 for an amplifier using active feedback. This diagram appears very similar to a traditional inverting amplifier. However, the analysis is somewhat different because the gain A in this case is not a very large open-loop op amp gain; rather, it is the relatively low and controlled gain of the LNP itself. Thus, the impedance at the inverting amplifier terminal will be reduced by a finite amount, as given in the familiar relationship of Equation 3: RIN = RF (1 + A) where REXT is the externally selected resistor value needed to achieve the desired gain setting, RS1 is the fixed parallel resistor in Figure 4, and RFIX is the effective fixed value of the remaining internal resistors: RS2, RS3, or (RS2 || RS3), depending on the pin connections. Note that the best process and temperature stability will be achieved by using the pre-programmed fixed-gain options of Table I, since the gain is then set entirely by internal resistor ratios, which are typically accurate to ±0.5%, and track quite well over process and temperature. When combining external resistors with the internal values to create an effective RS value, note that the internal resistors have a typical temperature coefficient of +700ppm/°C and an absolute value tolerance of approximately ±5%, yielding somewhat less predictable and stable gain settings. With or without external resistors, the board layout should use short Gain Strap connections to minimize parasitic resistance and inductance effects. The overall noise performance of the VCA2616 and VCA2611 will vary as a function of gain. Table II shows the typical inputand-output-referred noise densities of the entire VCA2616 and VCA2611 for maximum VCA and PGA gain; that is, VCACNTL set to 3.0V and all MGS bits set to 1. Note that the inputreferred noise values include the contribution of a 50Ω fixed source impedance, and are therefore somewhat larger than the intrinsic input noise. As the LNP gain is reduced, the noise contribution from the VCA/PGA portion becomes more significant, resulting in higher input-referred noise. However, the output-referred noise, which is indicative of the overall SNR at that gain setting, is reduced. To preserve the low-noise performance of the LNP, the user should take care to minimize resistance in the input lead. A parasitic resistance of only 10Ω will contribute 0.4nV/ √Hz . (3) where RF is the feedback resistor (supplied externally between the LNPINP and FB terminals for each channel), A is VCA2616, VCA2611 SBOS234E www.ti.com 11 RF VCA NOISE = 3.8nV√Hz, LNP GAIN = 20dB 14 LNP Noise nV/√Hz 6.0E-10 8.0E-10 1.0E-09 1.2E-09 1.4E-09 1.6E-09 1.8E-09 2.0E-09 RS LNPIN Noise Figure (dB) 12 A 10 8 6 4 2 RIN Active Feedback RIN = RF 1+A = RS RS A RS 0 0 100 200 300 400 500 600 700 800 900 1000 Source Impedance (Ω) FIGURE 7. Noise Figure for Conventional Termination. Conventional Cable Termination FIGURE 5. Configurations for Active Feedback and Conventional Cable Termination. the user-selected gain of the LNP, and RIN is the resulting amplifier input impedance with active feedback. In this case, unlike the conventional termination above, both the signal voltage and the RS noise are attenuated by the same factor of 2 (6dB) before being re-amplified by the A gain setting. This avoids the extra 3dB degradation due to the square-root effect described earlier, the key advantage of the active termination technique. This previous explanation ignored the input noise contribution of the LNP itself. Also, the noise contribution of the feedback resistor must be included for a completely correct analysis. The curves given in Figures 6 and 7 allow the VCA2616 and VCA2611 user to compare the achievable VCA NOISE = 3.8nV√Hz, LNP GAIN = 20dB 9 8 7 Noise Figure (dB) 6 5 4 3 2 1 0 0 100 200 300 400 500 600 700 800 Source Impedance (Ω) 900 1000 LNP Noise nV/√Hz 6.0E-10 8.0E-10 1.0E-09 1.2E-09 1.4E-09 1.6E-09 1.8E-09 2.0E-09 noise figure for active and conventional termination methods. The left-most set of data points in each graph give the results for typical 50Ω cable termination, showing the worst noise figure but also the greatest advantage of the active feedback method. A switch, controlled by the FBSWCNTL signal on pin 45, enables the user to reduce the feedback resistance by adding an additional parallel component, connected between the LNPINP and SWFB terminals. The two different values of feedback resistance will result in two different values of active-feedback input resistance. Thus, the active-feedback impedance can be optimized at two different LNP gain settings. The switch is connected at the buffered output of the LNP and has an ON resistance of approximately 1Ω. When employing active feedback, the user should be careful to avoid low-frequency instability or overload problems. Figure 8 illustrates the various low-frequency time constants. RF VCM CF 0.001µF 44pF CC 200kΩ Buffer LNPOUTN RS 44pF LNPOUTP Gain Stage VCM (VCA) LNP 200kΩ Buffer FIGURE 6. Noise Figure for Active Termination. FIGURE 8. Low-Frequency LNP Time Constants. 12 VCA2616, VCA2611 www.ti.com SBOS234E Referring again to the input resistance calculation of Equation (3), and considering that the gain term “A” falls off below 21kHz, it is evident that the effective LNP input impedance will rise below 3.6kHz, with a DC limit of approximately RF. To avoid interaction with the feedback pole/zero at low frequencies, and to avoid the higher signal levels resulting from the rising impedance characteristic, it is recommended that the external RFCC time constant be set to about 5µs. Achieving the best active-feedback architecture is difficult with conventional op amp circuit structures. The overall gain A must be negative in order to close the feedback loop, the input impedance must be high to maintain low current noise and good gain accuracy, but the gain ratio must be set with very low value resistors to maintain good voltage noise. Using a two-amplifier configuration (noninverting for high impedance plus inverting for negative feedback reasons) results in excessive phase lag and stability problems when the loop is closed. The VCA2616 and VCA2611 use a patented architecture that achieves these requirements, with the additional benefits of low power dissipation and differential signal handling at both input and output. For greatest flexibility and lowest noise, the user may wish to shape the frequency response of the LNP. The COMP1 and COMP2 pins for each channel (pins 10 and 11 for channel A, pins 26 and 27 for channel B) correspond to the drains of Q3 and Q8, see Figure 4. A capacitor placed between these pins will create a single-pole low-pass response, in which the effective R of the RC time constant is approximately 186Ω. RF RI Input C A Output FIGURE 10. LNP with Compensation Capacitor. AVOIDING UNSTABLE PERFORMANCE The VCA2612 and the VCA2616 are very similar in performance in all respects, except in the area of noise performance. See Figure 4 for a schematic of the LNP. This brings the input noise of the VCA2616 and VCA2611 down to 1.0nV/√Hz compared to the input on the VCA2612 1.25nV/√Hz impedance at the gate of either Q4 or Q7, as can be approximated by the network shown in Figure 11. The resistive component shown in Figure 11 is negative, which gives rise to unstable behavior when the signal source resistance has both inductive and capacitive elements. It should be noted that this negative resistance is not a physical resistor, but an equivalent resistance that is a function of the devices shown in Figure 4. Normally, when an inductor and capacitor are placed in series or parallel, there is a positive resistance in the loop that prevents unstable behavior. COMPENSATION WHEN USING ACTIVE FEEDBACK The typical open-loop gain versus frequency characteristic for the LNP is shown in Figure 9. The –3dB bandwidth is approximately 180MHz and the phase response is such that when feedback is applied, the LNP will exhibit a peaked response or might even oscillate. One method of compensating for this undesirable behavior is to place a compensation capacitor at the input to the LNP, as shown in Figure 10. This method is effective when the desired –3dB bandwidth is much less than the open-loop bandwidth of the LNP. This compensation technique also allows the total compensation capacitor to include any stray or cable capacitance that is associated with the input connection. Equation 4 relates the bandwidth to the various impedances that are connected to the LNP. 24pF –93Ω 57pF FIGURE 11. VCA2616 and VCA2611 Input Impedance. For the VCA2616 and VCA2611, the situation can be remedied by placing an external resistor with a value of approximately 15Ω or higher in series with the input lead. The net series resistance will be positive, and there will be no observed instability. Although this technique will prevent oscillations, it is not recommended, as it will also increase the input noise. A 4.7pF external capacitor must be placed between pins COMP2A (pin 11) and LNPINPA (pin 16), and between pins COMP2B (pin 26) and LNPINPB (pin 21). This has the result of making the input impedance always capacitive due to the feedback effect of the compensation capacitor and the gain of the LNP. Using capacitive feedback, the LNP becomes unconditionally stable, as there is no longer a negative component to the input impedance. The compensation capacitor mentioned above will be reflected to the input by the formula: CIN = (A + 1)CCOMP (5) BW = (A + 1) RI + RF 2πC(RI )(RF ) (4) –3dB Bandwidth 25dB Gain 180MHz FIGURE 9. Open-Loop Gain Characteristic of LNP. VCA2616, VCA2611 SBOS234E www.ti.com 13 The capacitance that is determined in Equation 5 should be added to the capacitance of Equation 4 to determine the overall bandwidth of the LNP. The LNPINNA (pin 12) and the LNPINNB (pin 25) should be bypassed to ground by the shortest means possible to avoid any inductance in the lead. The attenuator is comprised of two sections, with five parallel clipping amplifier/FET combinations in each. Special reference circuitry is provided so that the (VCM – VT) limit voltage will track temperature and IC process variations, minimizing the effects on the attenuator control characteristic. In addition to the analog VCACNTL gain setting input, the attenuator architecture provides digitally programmable adjustment in eight steps, via the three MGS bits. These adjust the maximum achievable gain (corresponding to minimum attenuation in the VCA, with VCACNTL = 3.0V) in 3dB increments. This function is accomplished by providing multiple FET sub-elements for each of the Q1 to Q10 FET shunt elements (see Figure 12). In the simplified diagram of Figure 13, each shunt FET is shown as two sub-elements, QNA and QNB. Selector switches, driven by the MGS bits, activate either or both of the sub-element FETs to adjust the maximum RON and thus achieve the stepped attenuation options. The VCA can be used to process either differential or singleended signals. Fully differential operation will reduce 2ndharmonic distortion by about 10dB for full-scale signals. Input impedance of the VCA will vary with gain setting, due to the changing resistances of the programmable voltage divider structure. At large attenuation factors (that is, low gain settings), the impedance will approach the series resistor value of approximately 135Ω. As with the LNP stage, the VCA output is AC-coupled into the PGA. This means that the attenuation-dependent DC common-mode voltage will not propagate into the PGA, and so the PGA’s DC output level will remain constant. Finally, note that the VCACNTL input consists of FET gate inputs. This provides very high impedance and ensures that multiple VCA2616 and VCA2611 devices may be connected in parallel with no significant loading effects. The nominal voltage range for the VCACNTL input spans from 0V to 3V. Overdriving this input (≤ 5V) does not affect the performance. LNP OUTPUT BUFFER The differential LNP output is buffered by wideband class AB voltage followers which are designed to drive low impedance loads. This is necessary to maintain LNP gain accuracy, since the VCA input exhibits gain-dependent input impedance. The buffers are also useful when the LNP output is brought out to drive external filters or other signal processing circuitry. Good distortion performance is maintained with buffer loads as low as 135Ω. As mentioned previously, the buffer inputs are AC-coupled to the LNP outputs with a 3.6kHz high-pass characteristic, and the DC common-mode level is maintained at the correct VCM for compatibility with the VCA input. VOLTAGE-CONTROLLED ATTENUATOR (VCA)—DETAIL The VCA is designed to have a dB-linear attenuation characteristic; that is, the gain loss in dB is constant for each equal increment of the VCACNTL control voltage. See Figure 1 for a block diagram of the VCA. The attenuator is essentially a variable voltage divider consisting of one series input resistor, RS, and ten identical shunt FETs, placed in parallel and controlled by sequentially activated clipping amplifiers. Each clipping amplifier can be thought of as a specialized voltage comparator with a soft transfer characteristic and well-controlled output limit voltages. The reference voltages V1 through V10 are equally spaced over the 0V to 3.0V control voltage range. As the control voltage rises through the input range of each clipping amplifier, the amplifier output will rise from 0V (FET completely ON) to VCM – VT (FET nearly OFF), where VCM is the common source voltage and VT is the threshold voltage of the FET. As each FET approaches its OFF state and the control voltage continues to rise, the next clipping amplifier/FET combination takes over for the next portion of the piecewise-linear attenuation characteristic. Thus, low control voltages have most of the FETs turned ON, while high control voltages have most turned OFF. Each FET acts to decrease the shunt resistance of the voltage divider formed by RS and the parallel FET network. INPUT OVERLOAD RECOVERY One of the most important applications for the VCA2616 and VCA2611 is processing signals in an ultrasound system. The ultrasound signal flow begins when a large signal is applied to a transducer, which converts electrical energy to acoustic energy. It is not uncommon for the amplitude of the electrical signal that is applied to the transducer to be ±50V or greater. RS INPUT Q1A VCM Q1B Q2A Q2B Q3A Q3B Q4A Q4B Q5A OUTPUT Q5B A1 B1 B2 A2 A3 A4 A5 Programmable Attenuator Section FIGURE 13. Programmable Attenuator Section. 14 VCA2616, VCA2611 www.ti.com SBOS234E Attenuator Input RS Q1 A1 C1 V1 A2 C2 V2 Q2 A3 C3 V3 Q3 A4 C4 V4 Q4 A5 A1-A10 Attenuator Stages QS Q5 A6 C5 V5 C6 V6 Q6 A7 C7 V7 Q7 A8 C8 V8 Q8 A9 C9 V9 Q9 Attenuator Output VCM Q10 A10 C10 V10 Control Input C1-C10 Clipping Amplifiers 0dB –4.5dB Attenuation Characteristic of Individual FETs VCM-VT 0 V1 V2 V3 V4 V5 V6 V7 V8 V9 Characteristic of Attenuator Control Stage Output V10 Overall Control Characteristics of Attenuator 0dB –45dB 0.3V Control Signal 3V FIGURE 12. Piecewise Approximation to Logarithmic Control Characteristics. VCA2616, VCA2611 SBOS234E www.ti.com 15 VDD (= maximum attenuation). For VCACNTL = 3V (no attenuation), the VCA + PGA gain will be controlled by the programmed PGA gain (24 to 45 dB in 3dB steps). For clarity, the gain and attenuation factors are detailed in Table III. RF CF MGS SETTING Protection Network LNPINP LNP LNPOUTN ATTENUATOR GAIN VCACNTL = 0V to 3V –24dB to 0dB –27dB to 0dB –30dB to 0dB –33dB to 0dB –36dB to 0dB –39dB to 0dB –42dB to 0dB –45dB to 0dB DIFFERENTIAL PGA GAIN 24dB 27dB 30dB 33dB 36dB 39dB 42dB 45dB ATTENUATOR + DIFF. PGA GAIN 0dB to 24dB 0dB to 27dB 0dB to 30dB 0dB to 33dB 0dB to 36dB 0dB to 39dB 0dB to 42dB 0dB to 45dB 000 001 010 011 100 101 110 111 ESD Diode FIGURE 14. VCA2616 and VCA2611 Diode Bridge Protection Circuit. To prevent damage, it is necessary to place a protection circuit between the transducer and the VCA2616 and VCA2611 (see Figure 14). Care must be taken to prevent any signal from turning the ESD diodes on. Turning on the ESD diodes inside the VCA2616 and VCA2611 could cause the input coupling capacitor (CC) to charge to the wrong value. TABLE III. MGS Settings. The PGA architecture consists of a differential, programmable-gain voltage to current converter stage followed by transimpedance amplifiers to create and buffer each side of the differential output. The circuitry associated with the voltage to current converter is similar to that previously described for the LNP, with the addition of eight selectable PGA gain-setting resistor combinations (controlled by the MGS bits) in place of the fixed resistor network used in the LNP. Low input noise is also a requirement of the PGA design due to the large amount of signal attenuation which can be inserted between the LNP and the PGA. At minimum VCA attenuation (used for small input signals) the LNP noise dominates; at maximum VCA attenuation (large input signals) the PGA noise dominates. Note that if the PGA output is used single-ended, the apparent gain will be 6dB lower. PGA POST-AMPLIFIER—DETAIL Figure 15 shows a simplified circuit diagram of the PGA block. As described previously, the PGA gain is programmed with the same MGS bits which control the VCA maximum attenuation factor. Specifically, the PGA gain at each MGS setting is the inverse (reciprocal) of the maximum VCA attenuation at that setting. Therefore, the VCA + PGA overall gain will always be 0dB (unity) when the analog VCACNTL input is set to 0V VDD To Bias Circuitry RL Q1 Q11 Q12 Q9 RL VCAOUTP VCM Q3 RS1 RS2 +In Q2 Q5 Q6 Q4 Q14 Q13 Q8 VCM VCAOUTN Q7 –In Q10 To Bias Circuitry FIGURE 15. Simplified Block Diagram of the PGA Section Within the VCA2616 and VCA2611. 16 VCA2616, VCA2611 www.ti.com SBOS234E PACKAGE OPTION ADDENDUM www.ti.com 18-Mar-2005 PACKAGING INFORMATION Orderable Device VCA2611Y/250 VCA2611Y/2K VCA2616YR VCA2616YT (1) Status (1) ACTIVE ACTIVE ACTIVE ACTIVE Package Type TQFP TQFP TQFP TQFP Package Drawing PFB PFB PFB PFB Pins Package Eco Plan (2) Qty 48 48 48 48 250 2000 2000 250 TBD TBD TBD TBD Lead/Ball Finish CU SNPB CU SNPB CU SNPB CU SNPB MSL Peak Temp (3) Level-2-220C-1 YEAR Level-2-220C-1 YEAR Level-3-235C-168 HR Level-3-235C-168 HR The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS) or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release. In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis. Addendum-Page 1 MECHANICAL DATA MTQF019A – JANUARY 1995 – REVISED JANUARY 1998 PFB (S-PQFP-G48) PLASTIC QUAD FLATPACK 0,50 36 25 0,27 0,17 0,08 M 37 24 48 13 0,13 NOM 1 5,50 TYP 7,20 SQ 6,80 9,20 SQ 8,80 0,05 MIN 1,05 0,95 Seating Plane 0,75 0,45 Gage Plane 0,25 0°– 7° 12 1,20 MAX 0,08 4073176 / B 10/96 NOTES: A. All linear dimensions are in millimeters. B. This drawing is subject to change without notice. C. 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