MP400FC_09

P r oMP400ov a t i o n FFr room d u c t IIn n o v a nn m MP400FC MP400 General DescriPtion Power Operational Amplifiers FeatUres ♦ Low Cost ♦ Wide Common Mode Range ♦ Standard Supply Voltage ♦ Single Supply: 10 V to 50 V ♦ Output Current - 150 mA Continuous ♦ Output Voltage 50-350 V ♦ 350 V/µS Slew Rate ♦ 200 kHz Power Bandwidth The MP400FC combines a high voltage, high speed precision power op amp with a supply voltage boost function in an integrated thermally conductive module. The voltage boost function uses a switch mode power supply (SMPS) to boost the input power supply voltage. This allows the user the benefits of using his standard 12 V or 24 V buss without the need to design a high voltage supply to power the op amp. The SMPS voltage is adjustable from 50-350 V, allowing for op amp output voltages up to 340 V. External phase compensation provides the user with the flexibility to tailor gain, slew rate and bandwidth for a specific application. The unique design of this amplifier provides extremely high slew rates in pulse applications while maintaining low quiescent current. The output stage is well protected with a user defined current limit. Safe Operating Area (SOA) must be observed for reliable operation. applications ♦ ♦ ♦ ♦ Piezoelectric positioning and Actuation Electrostatic Deflection Deformable Mirror Actuators Chemical and Biological Stimulators eqUivalent circUit DiaGraM 21 Vin (10V to 50V) 22 23 25 Vbias 24 L1 D1 SMPS CONTROLLER Q2 R7 R8 L2 R17 8 Vboost 6 LFin 4 +Vs 37 Cr+ 36 Cc+ 1 Out R15 C6 AMP C8 34 Rset 26 27 28 29 30 31 32 33 35 40 39 41 Analog -IN +IN -Vs Power GND GND R19 2 Ilim 42 Cc38 Cr15 14 13 12 R14 MP400U http://www.cirrus.com Copyright © Cirrus Logic, Inc. 2009 (All Rights Reserved) aUG 2009 1 APEX − MP400UREVE MP400 P r o d u c t I n n o v a t i o nF r o m 1. characteristics anD sPeciFications aBsolUte MaxiMUM ratinGs Parameter SUPPLY VOLTAGE, +VCC to GND OUTPUT CURRENT, peak within SOA POWER DISSIPATION, internal, DC, Amplifier OUTPUT POWER, SMPS INPUT VOLTAGE, Differential INPUT VOLTAGE, Common Mode TEMPERATURE, pin solder, 10s TEMPERATURE, junction TEMPERATURE RANGE, storage OPERATING TEMPERATURE, case (Note 2) −40 −40 -16 -16 symbol Min Max 50 200 14.2 67 16 16 225 150 105 85 Units V mA W W V V °C °C °C °C sPeciFications Parameter aMPliFier inPUt OFFSET VOLTAGE OFFSET VOLTAGE vs. temperature OFFSET VOLTAGE vs. supply BIAS CURRENT, initial OFFSET CURRENT, initial INPUT RESISTANCE, DC COMMON MODE VOLTAGE RANGE, pos. COMMON MODE VOLTAGE RANGE, neg. COMMON MODE REJECTION, DC NOISE aMPliFier Gain OPEN LOOP @ 15 Hz GAIN BANDWIDTH PRODUCT @ 1 MHz PHASE MARGIN aMPliFier oUtPUt VOLTAGE SWING VOLTAGE SWING VOLTAGE SWING CURRENT, continuous, DC SLEW RATE SETTLING TIME, to 0.1% RESISTANCE, No load POWER BANDWIDTH, 300 VP-P CURRENT, quiescent, amplifier only 2 V Step RLIM = 6.2 Ω +VS = 160 V, −VS = -160 V 0.2 IO = 10 mA IO = 100 mA IO = 150 mA 150 100 350 1 44 200 0.7 2.5 |VS| - 2 |VS| - 8.6 |VS| - 10 |VS| - 12 V V V mA V/µS µS Ω kHz mA Full temperature range 89 120 1 50 dB MHz ° 700KHz bandwidth 90 (Note 3) 8.5 12 106 +VS - 2 -VS + 5.5 118 418 0 to 85°C (Case) 8 -63 32 200 400 40 mV µV/°C µV/V pA pA Ω V V dB µV RMS test conditions Min typ Max Units 2 MP400U P r o d u c t I n n o v a t i o nF r o m MP400 sPeciFications, (cont). Parameter sMPs INPUT VOLTAGE, VIN SMPS OUTPUT VOLTAGE, VB SMPS OUTPUT CURRENT, IS OUTPUT VOLTAGE TOLERANCE VOLTAGE BOOST therMal RESISTANCE, DC, junction to case RESISTANCE, junction to air TEMPERATURE RANGE, case Full temperature range, f300 V. 100µF Vboost SMPS CIRCUIT PGND 470µF R4 0-5V DAC 5.62K R5 5.62K Rset 0.0Ω INK DROPLETS CC DEFLECTION PLATE 4 MP400U P r o d u c t I n n o v a t i o nF r o m MP400 tyPical PerForMance GraPhs GAIN = -50 POWER SUPPLY REJECTION (dB) 350 POWER RESPONSE 160 CURRENT LIMIT CURRENT LIMIT, ILIM (mA) +VS 100 80 60 40 20 POWER SUPPLY REJECTION -VS +VS OUTPUT VOLTAGE, (V) 300 250 200 150 100 50 0 1 NO COMPENSATION 100 10 FREQUENCY, (KHz) 1000 GAIN = -100 120 80 -VS 40 0 0 2 4 6 8 10 RESISTOR ( ) 12 14 0 100 1K FREQUENCY, (Hz) 10K INTERNAL POWER DISSIPATION, PD (W) VOLTAGE DROP FROM SUPPLY (V) AMPLIFIER INTERNAL POWER DERATING 16 14 12 10 8 6 4 2 0 0 25 50 75 100 CASE TEMPERATURE, TC (°C) OUTPUT VOLTAGE SWING COMMON MODE REJECTION (dB) 12 10 8 6 4 2 0 50 100 150 200 PEAK TO PEAK LOAD CURRENT (mA) 0 -VS SIDE DROP 140 120 100 COMMON MODE REJECTION 80 60 40 20 0 1 10 100 1K 10K 100K FREQUENCY (Hz) +VS SIDE DROP EFFICIENCY Vs. SMPS CURRENT 70 EFFICIENCY Vs. SMPS CURRENT 70 EFFICIENCY Vs. SMPS CURRENT 70 VIN = 48V VB = 350V EFFICIENCY (%) EFFICIENCY (%) 65 EFFICIENCY (%) VIN = 24V VB = 240V 0.1 IO 0.15 0.2 65 65 60 60 VIN = 12V VB = 120V 55 0.05 0.1 0.15 0.2 60 55 IO 50 0.05 55 0.05 0.1 IO 0.15 0.2 SMPS CURRENT VS. SMPS VOLTAGE 0.45 SMPS OUTPUT POWER, PO (W) 0.5 0.4 CBOOST = 470uF 70 60 50 40 30 20 10 0 0 SMPS POWER DERATING SMPS CURRENT, IS (A) 8V =4 V IN 4V =2 V IN 0.35 0.3 0.25 0.2 0.15 0.1 V IN 0.05 Limit of LFIN filter inductor, and MP400 amplifier. 0 50 100 150 200 250 300 350 BOOST SUPPLY VOLTAGE, VB (V) =1 2V 50 75 100 25 CASE TEMPERATURE, TC (°C) MP400U 5 MP400 P r o d u c t I n n o v a t i o nF r o m 100 80 GAIN, Db SMALL SIGNAL OPEN LOOP GAIN RC = OPEN, CC = 0pF RC = 3.3K, CC = 1pF RC = 3.3K, CC = 2.2pF RC = 3.3K, CC = 5pF CS = 68pF PIN = -40dBm RBIAS = OPEN RS = 48.7Ω VS = ±50V 1 60 40 20 0 -20 RC = 3.3K, CC = 10pF RC = 3.3K, CC = 22pF 10 100 FREQUENCY, KHz 1000 180 150 120 PHASE, ° 180 150 120 PHASE, ° SMALL SIGNAL OPEN LOOP PHASE, VO = 250mVP-P RC = 3.3K, CC = 22pF RC = 3.3K, CC = 10pF SMALL SIGNAL OPEN LOOP PHASE RC = 3.3K, CC = 22pF RC = 3.3K, CC = 10pF 90 60 30 0 CS = 68pF -30 PIN = -40dBm R = OPEN -60 RBIAS 48.7Ω = S -90 1 RC = 3.3K, CC = 5pF RC = 3.3K, CC = 2.2pF RC = 3.3K, CC = 1pF RC = OPEN, CC = 0pF 10 100 FREQUENCY, KHz 1000 90 60 CS = 68pF 0 PIN = -40dBm -30 RBIAS = 100K R = 48.7Ω -60 V S = ±50V S -90 1 30 RC = 3.3K, CC = 5pF RC = 3.3K, CC = 2.2pF RC = 3.3K, CC = 1pF RC = OPEN, CC = 0pF 10 100 FREQUENCY, KHz 1000 45 35 25 GAIN, dB 15 5 -5 -15 -25 GAIN vs. INPUT/OUTPUT SIGNAL LEVEL 35 25 SMALL SIGNAL GAIN vs. COMPENSATION, VO = 5VP-P CC = 0pF 500 mVP-P GAIN,dB A V = +51 RBIAS = 100K RC = OPEN RF = 75K RG = 1.5K RL = 50K VS = ±50V 10 5 VP-P 15 5 -5 -15 -25 10K -35 10 CC = 1pF A V = +26 RBIAS = 100K RF = 35.7K RG = 1.5K RL = 50K VS = ±50V CC = 2.2pF CC = 5pF CC = 10pF CC = 22pF 100 1K FREQUENCY, KHz 10K 50 VP-P 100 1K FREQUENCY, KHz CC = 0pF 45 35 25 GAIN,dB 15 5 -5 -15 -25 SMALL SIGNAL GAIN vs. COMPENSATION, VO = 500mVP-P 35 25 15 GAIN,dB LARGE SIGNAL GAIN vs. COMPENSATION, VO = 50VP-P CC = 0pF -35 10 A V = +26 RBIAS = 100K RC = 3.3K RF = 35.7K RG = 1.5K RL = 50K VS = ±50V CC = 1pF CC = 2.2pF CC = 5pF CC = 10pF CC = 22pF 100 1K FREQUENCY, KHz 10K 5 -5 -15 -25 -35 10 A V = +26 RBIAS = 100K RF = 35.7K RG = 1.5K RL = 50K VS = ±50V CC = 1pF CC = 2.2pF CC = 5pF CC = 10pF CC = 22pF 10K 100 1K FREQUENCY, KHz 6 MP400U P r o d u c t I n n o v a t i o nF r o m MP400 SR+/SR- (25% - 75%) 1000 800 SR, V/µs 600 400 200 0 0 2 SR+/SR- (25% - 75%) SRSR+ A V = +101 CL = 8pF RF = 25K RG = 250Ω RL = 50K VS = ±150V 4 6 8 10 12 PEAK-TO-PEAK INPUT VOLTAGE 14 16 1000 SR, V/µs A V = +26 C = 8pF 800 RL = 35.6K F RG = 1.5K 600 RL = 50K VS = ±150V 400 200 0 0 2 SR+ SR- 4 6 8 10 12 PEAK-TO-PEAK INPUT VOLTAGE 14 16 1000 800 SR, V/µs 600 400 200 0 0 2 SR+/SR- (25% -75%) SR+ SRA V = +51 CL = 8pF RF = 75K RG = 1.5K RL = 50K VS = ±150V 4 6 8 10 12 PEAK-TO-PEAK INPUT VOLTAGE 14 16 Time, µs 1 0.8 0.6 0.4 0.2 0 0 2 RISE AND FALL TIME (10% - 90%) A V = +51 CL = 8pF RF = 75K RG = 1.5K RL = 50K VS = ±150V TF TR 4 6 8 10 12 PEAK-TO-PEAK INPUT VOLTAGE 14 16 15 OUTPUT VOLTAGE, V TRANSIENT RESPONSE 1VP-P input1 INPUT VOLTAGE, V OUTPUT VOLTAGE, V 5 0 -5 -10 -15 -4 30 -2 0 2 50 0 -50 input10 4 2 0 -2 -4 -6 -0.4 -0.8 4 TIME, µs 2VP-P input2 6 8 10 -1.2 12 1.5 -100 -150 -4 -2 0 2 4 6 TIME, µs 8 10 -8 12 3.0 2.4 1.8 1.2 0.6 0 -0.6 -1.2 -1.8 -2.4 -3.0 TRANSIENT RESPONSE OUTPUT VOLTAGE, V OUTPUT VOLTAGE, V 10 0 -10 -20 -30 -4 -2 0 2 -0.5 -1 4 TIME, µs 6 8 10 -1.5 12 INPUT VOLTAGE, V 5 6 7 8 MP400U INPUT VOLTAGE, V 20 A V = +26 CC = 2.2pF 1 CL = 8pF RC = 3.3K 0.5 RF = 35.7K RG = 1.5K 0 RL = 50K 150 Out - 0pF 120 input 90 60 Out - 1pF & 3.3K 30 0 -30 -60 Out - 5pF & 3.3K -90 -120 -150 -2 -1 0 1 2 3 4 TIME, µs PULSE RESPONSE vs. CC AND RC A V = +51 CC = 68pF CL = 330pF RC = 48Ω RF = 75K RG = 1.5K RL = OPEN VS = ±150V INPUT VOLTAGE, V 10 1.2 A V = +26 CC = 2.2pF 0.8 CL = 8pF RC = 3.3K 0.4 RF = 35.7K RG = 1.5K 0 RL = 50K 150 100 TRANSIENT RESPONSE 10VP-P A V = +26 CC = 2.2pF CL = 8pF RC = 3.3K RF = 35.7K RG = 1.5K RL = 50K 8 6 7 MP400 PULSE RESPONSE vs. CAP LOAD 300pf, 3VP-P 200pf, 3VP-P 100pf, 3VP-P P r o d u c t I n n o v a t i o nF r o m 140 120 100 80 60 40 20 0 -20 -40 -60 -80 -6 -4 -2 0 A V = -50 RF = 75K RG = 1.5K RL = 50K VS = ±150V 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 TIME, µs 300pF, 2VP-P 200pF, 2VP-P 100pF, 2VP-P 140 120 100 80 60 40 20 0 -20 -40 -60 -80 -6 -4 -2 0 PULSE RESPONSE vs. CAP LOAD 300pF, 1VP-P 200pF, 1VP-P 100pF, 1VP-P OUTPUT, V OUTPUT, V A V = -50 RF = 75K RG = 1.5K RL = 50K VS = ±150V 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 TIME, µs 140 120 100 80 60 40 20 0 -20 -40 -60 -80 -6 -4 -2 0 PULSE RESPONSE vs. CAP LOAD OUTPUT, V A V = -50 RF = 75K RG = 1.5K RL = 50K VS = ±150V CL = 8pF 2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 TIME, µs 300 OUTPUT VOLTAGE, V 0.2 0.15 IS, A PULSE RESPONSE A V = +51 CL = 8pF RF = 75K RG = 1.5K RL = 50K VS = ±150V OVERDRIVE RECOVERY A V = +51 CC = OPEN CL = 8pF RC = OPEN RF = 75K RG = 1.5K RL = 50K VS = ±150V INPUT 6 4 200 100 0 0.1 OUTPUT 2 0 -2 -4 0.05 0 -0.05 -1 0 1 2 3 TIME,µs 4 5 6 -100 -200 -300 -6 -4 -2 0 2 4 TIME, µs 6 8 10 -6 12 18 16 14 12 IS, mA IS vs. VIN A V = +51 CL = 8pF CS = 68pF RF = 75K RG = 1.5K RL = 50K RS = 48.7Ω VS = ±150V 30 25 20 SUPPLY CURRENT vs. FREQUENCY A V = +51 CL = 8pF CS = 68pF RF = 75K RG = 1.5K RL = 50K RS = 48.7Ω VS = ±150V IS, mA 10 8 6 4 2 0 0 15 10 5 VIN = 6VP VIN = 3VP 1 2 3 4 5 6 VIN, VP-P (100KHz sine wave) 7 8 9 0 10 100 Frequency, (KHz sine wave) 1000 1200 SR+/SR- (25%-75%) SR+(A V = -25) SR-(A V = -25) SR+(A V = +26) SR-(A V = +26) SR+/SR- V/µs V/µs RF = 75K 1000 RG = 1.5K RL = 50K 800 VS = ±150V CL = 8pF 600 400 200 1600 SR+/SR- (25%-75%) R = 75K 1400 RF = 1.5K G 1200 RL = 50K VS = ±150V 1000 C = 8pF L 800 600 400 200 8 SR+(A V = -50) SR-(A V = -50) SR+(A V = +51) MP400U INPUT VOLTAGE, V 14 CS = 68pF RF = 75K 12 RG = 1.5K 10 RL = 50K 8 RS = 48.7Ω VS = ±150V 6 4 2 0 0 1 2 20 15 10 CS = 68pF RF = 75K RG = 1.5K RL = 50K RS = 48.7Ω VS = ±150V IS, mA IS, mA VIN = 6VP VIN = 3VP P r o d u c t I n n o v a t i o nF r o m 3 4 5 6 VIN, VP-P (100KHz sine wave) 7 8 9 5 0 10 MP400 1000 100 Frequency, (KHz sine wave) 1200 SR+/SR- (25%-75%) SR+(A V = -25) SR-(A V = -25) SR+(A V = +26) SR-(A V = +26) SR+/SR- V/µs V/µs RF = 75K 1000 RG = 1.5K RL = 50K 800 VS = ±150V CL = 8pF 600 400 200 0 0 1 2 3 1600 SR+/SR- (25%-75%) R = 75K 1400 RF = 1.5K G 1200 RL = 50K VS = ±150V 1000 C = 8pF L 800 600 400 200 0 SR+(A V = -50) SR-(A V = -50) SR+(A V = +51) SR-(A V = +51) 0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 INPUT VOLTAGE, VOLTS PEAK-TO-PEAK 4 5 6 7 8 9 10 11 12 13 14 15 PEAK TO PEAK INPUT VOLTAGE General Please read Application note 1 “General operating considerations” which covers stability, power supplies, heat sinking, mounting, current limit, SOA interpretation, and specification interpretation. Visit www.cirrus.com for design tools that help automate tasks such as calculations for stability, internal power dissipation, and current limit. There you will also find a complete application notes library, technical seminar workbook, and evaluation kits. theory oF oPeration The PA78 is designed specifically as a high speed pulse amplifier. In order to achieve high slew rates with low idle current, the internal design is quite different from traditional voltage feedback amplifiers. Basic op amp behaviors like high input impedance and high open loop gain still apply. But there are some notable differences, such as signal dependent supply current, bandwidth and output impedance, among others. The impact of these differences varies depending on application performance requirements and circumstances. These different behaviors are ideal for some applications but can make designs more challenging in other circumstances. sUPPly cUrrent anD ByPass caPacitance A traditional voltage feedback amplifier relies on fixed current sources in each stage to drive the parasitic capacitances of the next stage. These currents combine to define the idle or quiescent current of the amplifier. By design, these fixed currents are often the limiting parameter for slew rate and bandwidth of the amplifier. Amplifiers which are high voltage and have fast slew rates typically have high idle currents and dissipate notable power with no signal applied to the load. At the heart of the PA78 design is a signal dependent current source which strikes a new balance between supply current and dynamic performance. With small input signals, the supply current of the PA78 is very low, idling at less than 1 mA. With large transient input signals, the supply currents increase dramatically to allow the amplifier stages to respond quickly. The Pulse Response plot in the typical performance section of this datasheet describes the dynamic nature of the supply current with various input transients. Choosing proper bypass capacitance requires careful consideration of the dynamic supply currents. High frequency ceramic capacitors of 0.1 µF or more should be placed as close as possible to the amplifier supply pins. The inductance of the routing from the supply pins to these ceramic capacitors will limit the supply of peak current during transients, thus reducing the slew rate of the PA78. The high frequency capacitance should be supplemented by additional bypass capacitance not more than a few centimeters from the amplifier. This additional bypass can be a slower capacitor technology, such as electrolytic, and is necessary to keep the supplies stable during sustained output currents. Generally, a few microfarad is sufficient. sMall siGnal PerForMance The small signal performance plots in the typical performance section of this datasheet describe the behavior when the dynamic current sources described previously are near the idle state. The selection of compensation capacitor directly affects the open loop gain and phase performance. Depending on the configuration of the amplifier, these plots show that the phase margin can diminish to very low levels when left uncompensated. This is due to the amount of bias current in the input stage when the part is in standby. An increase in the idle current in the output stage of the amplifier will improve phase margin for small signals although will increase the overall supply current. MP400U 9 MP400 P r o d u c t I n n o v a t i o nF r o m Current can be injected into the output stage by adding a resistor, RBIAS, between CC- and VS+. The size of RBIAS will depend upon the application but 500 µA (50 V V+ supply/100K) of added bias current shows significant improvement in the small signal phase plots. Adding this resistor has little to no impact on small signal gain or large signal performance as under these conditions the current in the input stage is elevated over its idle value. It should also be noted that connecting a resistor to the upper supply only injects a fixed current and if the upper supply is fixed and well bypassed. If the application includes variable or adjustable supplies, a current source diode could also be used. These two terminal components combine a JFET and resistor connected within the package to behave like a current source. As a second stability measure, the PA78 is externally compensated and performance can be optimized to the application. Unlike the RBIAS technique, external phase compensation maintains the low idle current but does affect the large signal response of the amplifier. Refer to the small and large signal response plots as a guide in making the tradeoffs between bandwidth and stability. Due to the unique design of the PA78, two symmetric compensation networks are required. The compensation capacitor CC must be rated for a working voltage of the full operating supply voltage (+VS to –VS). NPO capacitors are recommended to maintain the desired level of compensation over temperature. The PA78 requires an external 33 pF capacitor between CC- and -VS to prevent oscillations in the falling edge of the output. This capacitor should be rated for the full supply voltage (+VS to -VS). As the amplitude of the input signal increases, the internal dynamic current sources increase the operation bandwidth of the amplifier. This unique performance is apparent in its slew rate, pulse response, and large signal performance plots. Recall the previous discussion about the relationships between signal amplitude, supply current, and slew rate. As the amplitude of the input amplitude increases from 1 VP-P to 15 VP-P, the slew rate increases from 50 V/µs to well over 350 V/µs. Notice the knee in the Rise and Fall times plot, at approximately 6 VP-P input voltage. Beyond this point the output becomes clipped by the supply rails and the amplifier is no longer operating in a closed loop fashion. The rise and fall times become faster as the dynamic current sources are providing maximum current for slewing. The result of this amplifier architecture is that it slews fast, but allows good control of overshoot for large input signals. This can be seen clearly in the large signal Transient Response plots. larGe siGnal PerForMance heatsinkinG anD saFe oPeratinG area SOA The MOSFET output stage of the PA78 is not limited by second break160 down considerations as in bipolar output stages. Only thermal considerations of the package and current handling capabilities limit the 140 Safe Operating Area. The SOA plots include power dissipation limita120 25°C tions which are dependent upon case temperature. Keep in mind that 100 the dynamic current sources which drive high slew rates can increase 75°C the operating temperature of the amplifier during periods of repeated 80 slewing. The plot of supply current vs. input signal amplitude for a 100 60 125°C kHz signal provides an indication of the supply current with repeated 40 slewing conditions. This application dependent condition must be considered carefully. 20 The output stage is self-protected against transient flyback by the para0 sitic body diodes of the output stage. However, for protection against 10 100 1000 sustained high energy flyback, external, fast recovery diodes must be SUPPLY TO OUTPUT DIFFERENTIAL, VS-VO (V) used. For proper operation, the current limit resistor, RLIM, must be connected as shown in the external connections diagram. For maximum reliability and protection, the largest resistor value should be used. The maximum practical value for RLIM is about 12 Ω. However, refer to the SOA curves for each package type to assist in selecting the optimum value for Rlim in the intended application. Current limit may not protect against short circuit conditions with supply voltages over 200 V. 10 MP400U cUrrent liMit OUTPUT CURRENT (mA) P r o d u c t I n n o v a t i o nF r o m MP400 layoUt consiDerations The PA78 is built on a dielectrically isolated process and the package tab is therefore not electrically connected to the amplifier. For high speed operation, the package tab should be connected to a stable reference to reduce capacitive coupling between amplifier nodes and the floating tab. It is often convenient to directly connect the tab to GND or one of the supply rails, but an AC connection through a 1µF capacitor to GND is also sufficient if a DC connection is undesirable Care should be taken to position the RC / CC compensation networks close SMPS Output vs. RSET to the amplifier compensation pins. Long loops in these paths pick up noise 100000 and increase the likelihood of LC interactions and oscillations. The MP400FC is designed to operate off of a standard voltage rail. Typical values include 12 V, 24 V, or 48 V. The addition of the on-board SMPS eliminates the need to design or purchase a high voltage power supply. The only inputs required by the SMPS are the VIN source. Input and output filter capacitor, and boost voltage set resistor (RSET). The SMPS output can be adjusted between a minimum of 50 V to a maximum of 350 V. The voltage boost adjustment is independent of VIN. Adjustment to the boost level is made through a resistor from the RSET pin to ground. The resistor value is: RSET = 1.85 • 105 - 615 VBOOST - 49.95 sMPs oPeration 10000 1000 100 10 RSET (Ω) 1 50 100 150 200 250 300 350 VBoost (V) Where VBOOST = desired SMPS voltage. Example: 1) Desired VBOOST = 160 V 2) RSET = 1K (1066 by equation) If RSET is open, VBOOST will be 50 V. If RSET is shorted to ground VBOOST will be limited to 350 V. Note that while the MP400 SMPS generates a positive voltage from 50 V to 350 V, the amplifier may operate from a variety of supply voltages. Symmetric, asymmetrical and single supply configurations can be used so long as the total supply voltage from +VS to -VS does not exceed 350 V. The amplifier performance graphs in this datasheet include some plots taken with symmetrical supplies, but those plots generally apply to all supply configurations. sMPs oUtPUt caPacitor An external SMPS output filter capacitor is required for proper operation. ESR considerations prevail in the choice of the output filter capacitor. Select the highest value capacitor that meets the following ESR requirement. The minimum value for CBOOST is 100 µF. ESR = dVo/ILPK Where, dVo ILPK L VIN ton VBOOST IO FSW = The maximum acceptable output ripple voltage = Peak inductor current = (1/L) • VIN • ton = 10-6 if the internal inductor is used. = Input voltage of the application. = √(2 • Io • L • ((Vo + 0.6 - VIN)/(FSW • VIN2))) = The boost supply voltage of the application. = The maximum continuous output current for the application. = 100 KHz switching frequency of the MP400FC boost supply. MP400U 11 MP400 sMPs inPUt caPacitor P r o d u c t I n n o v a t i o nF r o m An external input capacitor is required. This capacitor should be at least 100 µF. therMal consiDerations For reliable operation the MP400FC will require a heatsink for most applications. When choosing the heatsink the power dissipation in the op amp and the SMPS MOSFET switch (Q2) are both considered. The power dissipation of the op amp is determined in the same manner as any power op amp. The power dissipation of the MOSFET switch (Q2) is the sum of the power dissipation due to conduction and the switching power. PD(Q2) = (IIN(pk)2 • RDS(ON) • D) + (IIN(pk) • VIN • tr • FSW) Where: VIN VB IO FSW RDS(ON) tr D t1 = = SMPS input voltage = SMPS output voltage = Total SMPS output current = 100 KHz = 0.621 Ω = 82 x 10-9s = t1 • FSW 2 • IO • 10 x 10-6 • VB • td 10 x 10-6 ( VB - VIN FSW • VIN2 ) IIN(pk) = td = t1 • ( VB VB - VIN ) - t1 contactinG cirrUs loGic sUPPort For all Apex Precision Power product questions and inquiries, call toll free 800-546-2739 in North America. For inquiries via email, please contact apex.support@cirrus.com. International customers can also request support by contacting their local Cirrus Logic Sales Representative. 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Cirrus Logic, Cirrus, and the Cirrus Logic logo designs, Apex Precision Power, Apex and the Apex Precision Power logo designs are trademarks of Cirrus Logic, Inc. All other brand and product names in this document may be trademarks or service marks of their respective owners. 12 MP400U