19-3652; Rev 1; 9/06
KIT
ATION
EVALU
E
L
B
A
AVAIL
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
The MAX1214 is a monolithic, 12-bit, 210Msps analogto-digital converter (ADC) optimized for outstanding
dynamic performance at high-IF frequencies up to
300MHz. The product operates with conversion rates
up to 210Msps while consuming only 820mW.
At 210Msps and an input frequency up to 250MHz, the
MAX1214 achieves a spurious-free dynamic range
(SFDR) of 77.2dBc. Its excellent signal-to-noise ratio
(SNR) of 66dB at 10MHz remains flat (within 2dB) for
input tones up to 300MHz. This ADC yields an excellent
low noise floor of -67.6dBFS, which makes it ideal for
wideband applications such as cable-head end
receivers and power-amplifier predistortion in cellular
base-station transceivers.
The MAX1214 requires a single 1.8V supply. The analog
input is designed for either differential or single-ended
operation and can be AC- or DC-coupled. The ADC also
features a selectable on-chip divide-by-2 clock circuit,
which allows the user to apply clock frequencies as high
as 340MHz. This helps to reduce the phase noise of the
input clock source. A low-voltage differential signal
(LVDS) sampling clock is recommended for best performance. The converter’s digital outputs are LVDS compatible and the data format can be selected to be either
two’s complement or offset binary.
The MAX1214 is available in a 68-pin QFN package
with exposed paddle (EP) and is specified over the
industrial (-40°C to +85°C) temperature range.
See the Pin-Compatible Versions table for a complete
selection of 8-bit, 10-bit, and 12-bit high-speed DACs in
this family (with and without input buffers).
Features
♦ 210Msps Conversion Rate
♦ Low Noise Floor of -67.6dBFS
♦ Excellent Low-Noise Characteristics
SNR = 65.6dB at fIN = 100MHz
SNR = 65dB at fIN = 250MHz
♦ Excellent Dynamic Range
SFDR = 74.2dBc at fIN = 100MHz
SFDR = 77.2dBc at fIN = 250MHz
♦ 59.5dB NPR for fNOTCH = 28.8MHz and a Noise
Bandwidth of 50MHz
♦ Single 1.8V Supply
♦ 820mW Power Dissipation at fSAMPLE = 210MHz
and fIN = 100MHz
♦ On-Chip Track-and-Hold Amplifier
♦ Internal 1.23V-Bandgap Reference
♦ On-Chip Selectable Divide-by-2 Clock Input
♦ LVDS Digital Outputs with Data Clock Output
♦ MAX1214 EV Kit Available
Ordering Information
PART
TEMP RANGE
PIN-PACKAGE
MAX1214EGK-D
-40°C to +85°C
68 QFN-EP*
MAX1214EGK+D
-40°C to +85°C
68 QFN-EP*
*EP = Exposed paddle.
+Denotes lead-free package.
D = Dry pack.
Applications
Pin-Compatible Versions
Base-Station Power-Amplifier Linearization
PART
RESOLUTION
(BITS)
SPEED GRADE
(Msps)
ON-CHIP
BUFFER
Wireless and Wired Broadband Communication
MAX1121
8
250
Yes
Communications Test Equipment
MAX1122
10
170
Yes
Radar and Satellite Subsystems
MAX1123
10
210
Yes
MAX1124
10
250
Yes
MAX1213
12
170
Yes
MAX1215
12
250
Yes
MAX1213N
12
170
No
MAX1214N
12
210
No
MAX1215N
12
250
No
Cable-Head End Receivers
Pin Configuration appears at end of data sheet.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1214
General Description
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
ABSOLUTE MAXIMUM RATINGS
AVCC to AGND ......................................................-0.3V to +2.1V
OVCC to OGND .....................................................-0.3V to +2.1V
AVCC to OVCC .......................................................-0.3V to +2.1V
AGND to OGND ....................................................-0.3V to +0.3V
INP, INN to AGND ....................................-0.3V to (AVCC + 0.3V)
All Digital Inputs to AGND........................-0.3V to (AVCC + 0.3V)
REFIO, REFADJ to AGND ........................-0.3V to (AVCC + 0.3V)
All Digital Outputs to OGND ....................-0.3V to (OVCC + 0.3V)
Continuous Power Dissipation (TA = +70°C, multi-layer board)
68-Pin QFN-EP (derate 41.7mW/°C
above +70°C)........................................................3333mW/°C
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature .....................................................+150°C
Storage Temperature Range ............................-60°C to +150°C
Maximum Current into Any Pin .........................................±50mA
Lead Temperature (soldering,10s) ..................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(AVCC = OVCC = 1.8V, AGND = OGND = 0, fSAMPLE = 210MHz, differential sine-wave clock input drive, 0.1µF capacitor on REFIO,
internal reference, digital output pins differential RL = 100Ω ±1%, TA = TMIN to TMAX, unless otherwise noted. Typical values are at
TA = +25°C.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
-1.75
±0.75
+1.75
LSB
-1
±0.4
+1
LSB
DC ACCURACY
Resolution
12
Integral Nonlinearity (Note 2)
INL
fIN = 10MHz, TA = +25°C
Differential Nonlinearity (Note 2)
DNL
TA = +25°C, no missing codes
Transfer Curve Offset
VOS
TA = +25°C (Note 2)
Bits
-3.5
+3.5
Offset Temperature Drift
40
mV
µV/°C
ANALOG INPUTS (INP, INN)
Full-Scale Input Voltage Range
VFS
TA = +25°C (Note 2)
1320
1454
Full-Scale Range Temperature
Drift
130
Common-Mode Input Range
VCM
Input Capacitance
CIN
Differential Input Resistance
RIN
Full-Power Analog Bandwidth
1590
Internally self-biased
mVP-P
ppm/°C
1.365 ±0.15
V
2.5
pF
3.0
4.2
FPBW
6.3
700
kΩ
MHz
REFERENCE (REFIO, REFADJ)
Reference Output Voltage
VREFIO
TA = +25°C, REFADJ = AGND
1.18
1.23
Reference Temperature Drift
REFADJ Input High Voltage
90
VREFADJ
Used to disable the internal reference
AVCC - 0.3
1.30
V
ppm/°C
V
SAMPLING CHARACTERISTICS
Maximum Sampling Rate
fSAMPLE
Minimum Sampling Rate
fSAMPLE
Clock Duty Cycle
210
Set by clock-management circuit
MHz
20
MHz
40 to 60
%
Aperture Delay
tAD
Figures 4, 11
620
ps
Aperture Jitter
tAJ
Figure 11
0.2
psRMS
2
_______________________________________________________________________________________
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
(AVCC = OVCC = 1.8V, AGND = OGND = 0, fSAMPLE = 210MHz, differential sine-wave clock input drive, 0.1µF capacitor on REFIO,
internal reference, digital output pins differential RL = 100Ω ±1%, TA = TMIN to TMAX, unless otherwise noted. Typical values are at
TA = +25°C.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
200
500
mVP-P
1.15 ±0.25
V
CLOCK INPUTS (CLKP, CLKN)
Differential Clock Input Amplitude
(Note 3)
Clock Input Common-Mode
Voltage Range
Internally self-biased
Clock Differential Input
Resistance
RCLK
11
±25%
kΩ
Clock Differential Input
Capacitance
CCLK
5
pF
DYNAMIC CHARACTERISTICS (at -1dBFS)
Signal-to-Noise
Ratio
SNR
fIN = 10MHz, TA ≥ +25°C
64
66
fIN = 100MHz, TA ≥ +25°C
64
65.6
fIN = 200MHz
fIN = 250MHz
Signal-to-Noise
and Distortion
SINAD
65
fIN = 10MHz, TA ≥ +25°C
64
65.8
fIN = 100MHz, TA ≥ +25°C
63.3
65
fIN = 200MHz
SFDR
Worst Harmonics
(HD2 or HD3)
Two-Tone Intermodulation
Distortion
Noise-Power Ratio
TTIMD
NPR
dB
63.7
fIN = 250MHz
Spurious-Free
Dynamic Range
dB
65.3
64.4
fIN = 10MHz, TA ≥ +25°C
73
fIN = 100MHz, TA ≥ +25°C
68
80.7
74.2
fIN = 200MHz
69.1
fIN = 250MHz
77.2
dBc
fIN = 10MHz, TA ≥ +25°C
-82
-73
fIN = 100MHz, TA ≥ +25°C
-74.2
-68
fIN = 200MHz
-73.5
fIN = 250MHz
-72.5
fIN1 = 99MHz at -7dBFS,
fIN2 = 101MHz at -7dBFS
fNOTCH = 28.8MHz ±1MHz,
noise BW = 50MHz, AIN = -9.1dBFS
dBc
-74
dBc
59.5
dB
LVDS DIGITAL OUTPUTS (D0P/N–D11P/N, ORP/N)
Differential Output Voltage
|VOD|
RL = 100Ω ±1%
250
400
mV
Output Offset Voltage
OVOS
RL = 100Ω ±1%
1.125
1.310
V
_______________________________________________________________________________________
3
MAX1214
ELECTRICAL CHARACTERISTICS (continued)
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
ELECTRICAL CHARACTERISTICS (continued)
(AVCC = OVCC = 1.8V, AGND = OGND = 0, fSAMPLE = 210MHz, differential sine-wave clock input drive, 0.1µF capacitor on REFIO,
internal reference, digital output pins differential RL = 100Ω ±1%, TA = TMIN to TMAX, unless otherwise noted. Typical values are at
TA = +25°C.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
LVCMOS DIGITAL INPUTS (CLKDIV, T/B)
Digital Input-Voltage Low
VIL
Digital Input-Voltage High
VIH
0.2 x AVCC
0.8 x AVCC
V
V
TIMING CHARACTERISTICS
CLK-to-Data Propagation Delay
tPDL
Figure 4
1.77
ns
CLK-to-DCLK Propagation Delay
tCPDL
Figure 4
4.31
ns
DCLK-to-Data Propagation Delay
tPDL - tCPDL Figure 4 (Note 3)
2.09
2.54
2.91
ns
LVDS Output Rise Time
tRISE
20% to 80%, CL = 5pF
460
LVDS Output Fall Time
tFALL
20% to 80%, CL = 5pF
460
ps
11
Clock
cycles
Output Data Pipeline Delay
tLATENCY
Figure 4
ps
POWER REQUIREMENTS
Analog Supply Voltage Range
AVCC
1.70
1.80
1.90
V
Digital Supply Voltage Range
OVCC
1.70
1.80
1.90
V
mA
Analog Supply Current
IAVCC
fIN = 100MHz
390
460
Digital Supply Current
IOVCC
fIN = 100MHz
64
75
mA
Analog Power Dissipation
PDISS
fIN = 100MHz
820
963
mW
Power-Supply Rejection Ratio
(Note 4)
PSRR
Offset
1.8
mV/V
Gain
1.5
%FS/V
Note 1: ≥+25°C guaranteed by production test, 20MHz to work appropriately and according to data sheet specifications.
Data Clock Outputs (DCLKP, DCLKN)
The MAX1214 features a differential clock output, which
can be used to latch the digital output data with an
external latch or receiver. Additionally, the clock output
can be used to synchronize external devices (e.g.,
FPGAs) to the ADC. DCLKP and DCLKN are differential
outputs with LVDS-compatible voltage levels. There is a
4.31ns delay time between the rising (falling) edge of
CLKP (CLKN) and the rising edge of DCLKP (DCLKN).
See Figure 4 for timing details.
Clock Inputs (CLKP, CLKN)
Designed for a differential LVDS clock input drive, it is
recommended to drive the clock inputs of the MAX1214
with an LVDS- or PECL-compatible clock to achieve the
best dynamic performance. The clock signal source
must be a high-quality, low phase noise with fast edge
rates to avoid any degradation in the noise performance
of the ADC. The clock inputs (CLKP, CLKN) are internally
biased to 1.15V, accept a typical 0.5VP-P differential signal swing, and are usually driven in AC-coupled configuration. See the Differential, AC-Coupled PECLCompatible Clock Input section for more circuit details
on how to drive CLKP and CLKN appropriately. Although
not recommended, the clock inputs also accept a singleended input signal.
Divide-by-2 Clock Control (CLKDIV)
The MAX1214 offers a clock control line (CLKDIV),
which supports the reduction of clock jitter in a system.
Connect CLKDIV to OGND to enable the ADC’s internal
divide-by-2 clock divider. Data is now updated at onehalf the ADC’s input clock rate. CLKDIV has an internal
pulldown resistor and can be left open for applications
that require this divide-by-2 mode. Connecting CLKDIV
to OVCC disables the divide-by-2 mode.
REFT
ADC FULL SCALE = REFT - REFB
1V
G
REFERENCE
SCALING AMPLIFIER
REFB
REFERENCE
BUFFER
REFIO
0.1μF
REFADJ
CONTROL LINE TO
DISABLE REFERENCE BUFFER
100Ω*
MAX1214
AVCC
*REFADJ MAY
BE SHORTED TO
AGND DIRECTLY
AVCC/2
REFT: TOP OF REFERENCE LADDER.
REFB: BOTTOM OF REFERENCE LADDER.
Figure 3. Simplified Reference Architecture
______________________________________________________________________________________
11
MAX1214
On-Chip Reference Circuit
The MAX1214 features an internal 1.23V bandgap reference circuit (Figure 3), which in combination with an internal reference-scaling amplifier determines the FSR of the
MAX1214. Bypass REFIO with a 0.1µF capacitor to
AGND. To compensate for gain errors or increase the
ADC’s FSR, the voltage of this bandgap reference can be
indirectly adjusted by adding an external resistor (e.g.,
100kΩ trim potentiometer) between REFADJ and AGND
or REFADJ and REFIO. See the Applications Information
section for a detailed description of this process.
To disable the internal reference, connect REFADJ to
AVCC. In this configuration, an external, stable reference must be applied to REFIO to set the converter’s
full scale. To enable the internal reference, connect
REFADJ to AGND.
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
System Timing Requirements
Figure 4 depicts the relationship between the clock
input and output, analog input, sampling event, and
data output. The MAX1214 samples on the rising
(falling) edge of CLKP (CLKN). Output data is valid on
the next rising (falling) edge of the DCLKP (DCLKN)
clock, but has an internal latency of 11 clock cycles.
Digital Outputs (D0P/N–D11P/N, DCLKP/N,
ORP/N) and Control Input T/B
Digital outputs D0P/N–D11P/N, DCLKP/N, and ORP/N
are LVDS compatible, and data on D0P/N–D11P/N is
presented in either binary or two’s-complement format
(Table 1). The T/B control line is an LVCMOS-compatible input, which allows the user to select the desired
output format. Pulling T/B low outputs data in two’s
complement and pulling it high presents data in offset
binary format on the 12-bit parallel bus. T/B has an
internal pulldown resistor and may be left unconnected
in applications using only two’s-complement output
SAMPLING EVENT
SAMPLING EVENT
format. All LVDS outputs provide a typical voltage
swing of 0.325V around a common-mode voltage of
1.15V, and must be terminated at the far end of each
transmission line pair (true and complementary) with
100Ω. The LVDS outputs are powered from a separate
power supply, which can be operated between 1.7V
and 1.9V.
The MAX1214 offers an additional differential output
pair (ORP, ORN) to flag out-of-range conditions, where
out-of-range is above positive or below negative full
scale. An out-of-range condition is identified with ORP
(ORN) transitioning high (low).
Note: Although a differential LVDS output architecture
reduces single-ended transients to the supply and
ground planes, capacitive loading on the digital outputs should still be kept as low as possible. Using
LVDS buffers on the digital outputs of the ADC when
driving larger loads may improve overall performance
and reduce system-timing constraints.
SAMPLING EVENT
SAMPLING EVENT
INN
INP
tCH
tAD
tCL
CLKN
N
N+9
N+8
N+1
CLKP
tCPDL
tLATENCY
DCLKP
N-8
N+1
N
N-7
DCLKN
tCPDL - tPDL
tPDL
D0P/N–
D11P/N
ORP/N
N-8
N-7
N-1
N
tCPDL - tPDL~ 0.4 x tSAMPLE WITH tSAMPLE = 1/fSAMPLE
NOTE: THE ADC SAMPLES ON THE RISING EDGE OF CLKP. THE RISING EDGE OF DCLKP CAN BE USED TO EXTERNALLY LATCH THE OUTPUT DATA.
Figure 4. System and Output Timing Diagram
12
______________________________________________________________________________________
N+1
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
INP ANALOG
INPUT VOLTAGE
LEVEL
INN ANALOG
INPUT VOLTAGE
LEVEL
OUT-OF-RANGE
ORP (ORN)
> VCM + VFS / 4
< VCM - VFS / 4
1 (0)
1111 1111 1111
(exceeds +FS, OR set)
0111 1111 1111
(exceeds +FS, OR set)
VCM + VFS / 4
VCM - VFS / 4
0 (1)
1111 1111 1111 (+FS)
0111 1111 1111 (+FS)
VCM
VCM
0 (1)
1000 0000 0000 or
0111 1111 1111 (FS/2)
0000 0000 0000 or
1111 1111 1111 (FS/2)
VCM - VFS / 4
VCM + VFS / 4
0 (1)
0000 0000 0000 (-FS)
1000 0000 0000 (-FS)
1 (0)
00 0000 0000
(exceeds -FS, OR set)
10 0000 0000
(exceeds -FS, OR set)
< VCM + VFS / 4
> VCM - VFS / 4
BINARY DIGITAL OUTPUT
CODE (D11P/N–D0P/N)
TWO’S COMPLEMENT DIGITAL
OUTPUT CODE (D11P/N–D0P/N)
OVCC
REFT
ADC FULL SCALE = REFT - REFB
REFERENCE
BUFFER
G
REFERENCESCALING
AMPLIFIER
REFB
1V
VOP
VON
REFIO
0.1μF
2.2kΩ
13kΩ TO
1MΩ
REFADJ
2.2kΩ
CONTROL LINE
TO DISABLE
REFERENCE BUFFER
Applications Information
FSR Adjustments Using the Internal
Bandgap Reference
The MAX1214 supports a full-scale adjustment range of
10% (±5%). To decrease the full-scale signal range, an
external resistor value ranging from 13kΩ to 1MΩ may
be added between REFADJ and AGND. A similar
approach can be taken to increase the ADC’s full-scale
range FSR. Adding a variable resistor, potentiometer,
or predetermined resistor value between REFADJ and
REFIO increases the FSR of the data converter. Figure
6a shows the two possible configurations and their
impact on the overall full-scale range adjustment of the
MAX1214. Do not use resistor values of less than 13kΩ
to avoid instability of the internal gain regulation loop
for the bandgap reference. See Figure 6b for the
results of the adjustment range for a selection of resistors used to trim the full-scale range of the MAX1214.
AVCC
AVCC/2
REFT: TOP OF REFERENCE LADDER.
REFB: BOTTOM OF REFERENCE LADDER.
Figure 6a. Circuit Suggestions to Adjust the ADC’s Full-Scale
Range
FS VOLTAGE vs. FS ADJUST RESISTOR
1.57
MAX1213 fig06b
Figure 5. Simplified LVDS Output Architecture
MAX1214
1.55
1.53
RESISTOR VALUE APPLIED BETWEEN
REFADJ AND REFIO INCREASES VFS
1.51
VFS (V)
OGND
13kΩ TO
1MΩ
1.49
1.47
1.45
1.43
RESISTOR VALUE APPLIED BETWEEN
REFADJ AND AGND DECREASES VFS
1.41
1.39
1.37
0
125 250 375 500 625 750 875 1000
FS ADJUST RESISTOR (kΩ)
Figure 6b. FS Adjustment Range vs. FS Adjustment Resistor
______________________________________________________________________________________
13
MAX1214
Table 1. MAX1214 Digital Output Coding
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
Differential, AC-Coupled,
LVPECL-Compatible Clock Input
The MAX1214 dynamic performance depends on the
use of a very clean clock source. The phase noise floor
of the clock source has a negative impact on the SNR
performance. Spurious signals on the clock signal
source also affect the ADC’s dynamic range. The preferred method of clocking the MAX1214 is differentially
with LVDS- or LVPECL-compatible input levels. The fast
data transition rates of these logic families minimize the
clock-input circuitry’s transition uncertainty, thereby
improving the SNR performance. To accomplish this, a
50Ω reverse-terminated clock signal source with low
phase noise is AC-coupled into a fast differential
receiver such as the MC100LVEL16D (Figure 7). The
receiver produces the necessary LVPECL output levels
to drive the clock inputs of the data converter.
Transformer-Coupled,
Differential Analog Input Drive
In general, the MAX1214 provides the best SFDR and
THD with fully differential input signals and it is not
recommended to drive the ADC inputs in single-ended
configuration. In differential input mode, even-order
harmonics are usually lower since INP and INN are balanced, and each of the ADC inputs only requires half
the signal swing compared to a single-ended configuration. Wideband RF transformers provide an excellent
solution to convert a single-ended signal to a fully differential signal, required by the MAX1214 to reach its
optimum dynamic performance.
A secondary-side termination of a 1:1 transformer (e.g.,
Mini-Circuit’s ADT1-1WT) into two separate 24.9Ω ±1%
resistors (use tight resistor tolerances to minimize
effects of imbalance; 0.5% would be an ideal choice)
placed between top/bottom and center tap of the transformer is recommended to maximize the ADC’s dynamic range. This configuration optimizes THD and SFDR
performance of the ADC by reducing the effects of
transformer parasitics. However, the source impedance combined with the shunt capacitance provided
by a PCB and the ADC’s parasitic capacitance limit the
ADC’s full-power input bandwidth to approximately
600MHz.
VCLK
0.1μF
SINGLE-ENDED
INPUT TERMINAL
8
0.1μF
0.1μF
2
7
150Ω
50Ω
MC100LVEL16D
0.1μF
6
3
510Ω
150Ω
510Ω
AVCC OVCC
4
5
0.01μF
INP
CLKN CLKP
D0P/N–D11P/N
MAX1214
INN
12
AGND OGND
Figure 7. Differential, AC-Coupled, LVPECL-Compatible Clock Input Configuration
14
______________________________________________________________________________________
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
Single-Ended, AC-Coupled Analog Inputs
Although not recommended, the MAX1214 can be used
in single-ended mode (Figure 9). Analog signals can be
AC-coupled to the positive input INP through a 0.1µF
capacitor and terminated with a 49.9Ω resistor to AGND.
The negative input should be reverse terminated with
49.9Ω resistors and AC-grounded with a 0.1µF capacitor.
Grounding, Bypassing, and
Board Layout Considerations
The MAX1214 requires board layout design techniques
suitable for high-speed data converters. This ADC provides separate analog and digital power supplies. The
analog and digital supply voltage pins accept 1.7V to
1.9V input voltage ranges. Although both supply types
can be combined and supplied from one source, it is
recommended to use separate sources to cut down on
performance degradation caused by digital switching
currents, which can couple into the analog supply network. Isolate analog and digital supplies (AVCC and
OVCC) where they enter the PCB with separate networks of ferrite beads and capacitors to their corresponding grounds (AGND, OGND).
AVCC
SINGLE-ENDED
INPUT TERMINAL
10Ω
0.1μF
ADT1-1WT
OVCC
INP
ADT1-1WT
D0P/N–D11P/N
25Ω
MAX1214
25Ω
12
INN
10Ω
0.1μF
AGND
OGND
Figure 8. Analog Input Configuration with Back-to-Back Transformers and Secondary-Side Termination
AVCC
SINGLE-ENDED
INPUT TERMINAL
0.1μF
OVCC
INP
D0P/N–D11P/N
49.9Ω
1%
0.1μF
MAX1214
INN
12
49.9Ω
1%
AGND
OGND
Figure 9. Single-Ended AC-Coupled Analog Input Configuration
______________________________________________________________________________________
15
MAX1214
To further enhance THD and SFDR performance at high
input frequencies (>100MHz), a second transformer
(Figure 8) should be placed in series with the singleended-to-differential conversion transformer. This transformer reduces the increase of even-order harmonics
at high frequencies.
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
To achieve optimum performance, provide each supply
with a separate network of a 47µF tantalum capacitor
and parallel combinations of 10µF and 1µF ceramic
capacitors. Additionally, the ADC requires each supply
pin to be bypassed with separate 0.1µF ceramic
capacitors (Figure 10). Locate these capacitors directly
at the ADC supply pins or as close as possible to the
MAX1214. Choose surface-mount capacitors, whose
preferred location should be on the same side as the
converter to save space and minimize the inductance.
If close placement on the same side is not possible,
these bypassing capacitors may be routed through
vias to the bottom side of the PCB.
Multilayer boards with separated ground and power
planes produce the highest level of signal integrity.
Consider the use of a split ground plane arranged to
match the physical location of analog and digital
ground on the ADC’s package. The two ground planes
should be joined at a single point so the noisy digital
ground currents do not interfere with the analog ground
plane. The dynamic currents that may need to travel
long distances before they are recombined at a common-source ground, resulting in large and undesirable
ground loops, are a major concern with this approach.
Ground loops can degrade the input noise by coupling
back to the analog front-end of the converter, resulting
in increased spurious activity, leading to decreased
noise performance.
Alternatively, all ground pins could share the same
ground plane, if the ground plane is sufficiently isolated
from any noisy, digital systems ground. To minimize the
coupling of the digital output signals from the analog
input, segregate the digital output bus carefully from the
analog input circuitry. To further minimize the effects of
digital noise coupling, ground return vias can be positioned throughout the layout to divert digital switching
currents away from the sensitive analog sections of the
ADC. This approach does not require split ground
planes, but can be accomplished by placing substantial
ground connections between the analog front-end and
the digital outputs.
The MAX1214 is packaged in a 68-pin QFN-EP package (package code: G6800-4), providing greater
design flexibility, increased thermal dissipation, and
optimized AC performance of the ADC. The exposed
paddle (EP) must be soldered down to AGND.
In this package, the data converter die is attached to
an EP lead frame with the back of this frame exposed
at the package bottom surface, facing the PCB side of
the package. This allows a solid attachment of the
package to the board with standard infrared (IR) flow
soldering techniques.
Thermal efficiency is one of the factors for selecting a
package with an exposed pad for the MAX1214. The
exposed pad improves thermal and ensures a solid
ground connection between the DAC and the PCB’s
analog ground layer.
Considerable care must be taken when routing the digital output traces for a high-speed, high-resolution data
converter. It is recommended running the LVDS output
traces as differential lines with 100Ω matched impedance from the ADC to the LVDS load device.
BYPASSING—ADC LEVEL
BYPASSING—BOARD LEVEL
OVCC
AVCC
0.1μF
AVCC
0.1μF
1μF
AGND
10μF
47μF
ANALOG POWERSUPPLY SOURCE
10μF
47μF
DIGITAL/OUTPUT
DRIVER POWERSUPPLY SOURCE
OGND
D0P/N–D11P/N
OVCC
MAX1214
12
1μF
AGND
OGND
NOTE: EACH POWER-SUPPLY PIN (ANALOG
AND DIGITAL) SHOULD BE DECOUPLED WITH
AN INDIVIDUAL 0.1μF CAPACITOR AS CLOSE
AS POSSIBLE TO THE ADC.
Figure 10. Grounding, Bypassing, and Decoupling Recommendations for the MAX1214
16
______________________________________________________________________________________
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
Integral Nonlinearity (INL)
Integral nonlinearity is the deviation of the values on an
actual transfer function from a straight line. This straight
line can be either a best straight-line fit or a line drawn
between the end points of the transfer function, once
offset and gain errors have been nullified. However, the
static linearity parameters for the MAX1214 are measured using the histogram method with a 10MHz input
frequency.
Differential Nonlinearity (DNL)
Differential nonlinearity is the difference between an
actual step width and the ideal value of 1LSB. A DNL
error specification of less than 1LSB guarantees no
missing codes and a monotonic transfer function. The
MAX1214’s DNL specification is measured with the histogram method based on a 10MHz input tone.
Dynamic Parameter Definitions
Aperture Jitter
Figure 11 depicts the aperture jitter (tAJ), which is the
sample-to-sample variation in the aperture delay.
Aperture Delay
Aperture delay (tAD) is the time defined between the
rising edge of the sampling clock and the instant when
an actual sample is taken (Figure 11).
CLKP
tion error only and results directly from the ADC’s resolution (N bits):
SNR[max] = 6.02 x N + 1.76
In reality, other noise sources such as thermal noise,
clock jitter, signal phase noise, and transfer function
nonlinearities are also contributing to the SNR calculation and should be considered when determining the
signal-to-noise ratio in ADC.
Signal-to-Noise Plus Distortion (SINAD)
SINAD is computed by taking the ratio of the RMS signal to all spectral components excluding the fundamental and the DC offset. In the case of the MAX1214,
SINAD is computed from a curve fit.
Spurious-Free Dynamic Range (SFDR)
SFDR is the ratio of RMS amplitude of the carrier frequency (maximum signal component) to the RMS value
of the next-largest noise or harmonic distortion component. SFDR is usually measured in dBc with respect to
the carrier frequency amplitude or in dBFS with respect
to the ADC’s full-scale range.
Intermodulation Distortion (IMD)
IMD is the ratio of the RMS sum of the intermodulation
products to the RMS sum of the two fundamental input
tones. This is expressed as:
⎛
VIM12 + VIM22 + ...... + VIM32 + VIMn2
IMD = 20 × log⎜
⎜
V12 + V22
⎝
⎞
⎟
⎟
⎠
CLKN
The fundamental input tone amplitudes (V1 and V2) are at
-7dBFS. The intermodulation products are the amplitudes
of the output spectrum at the following frequencies:
• Second-order intermodulation products: fIN1 + fIN2,
fIN2 - fIN1
ANALOG
INPUT
tAD
tAJ
SAMPLED
DATA (T/H)
T/H
TRACK
HOLD
TRACK
Figure 11. Aperture Jitter/Delay Specifications
Signal-to-Noise Ratio (SNR)
For a waveform perfectly reconstructed from digital
samples, the theoretical maximum SNR is the ratio of
the full-scale analog input (RMS value) to the RMS
quantization error (residual error). The ideal, theoretical
minimum analog-to-digital noise is caused by quantiza-
• Third-order intermodulation products: 2 x fIN1 - fIN2,
2 x fIN2 - fIN1, 2 x fIN1 + fIN2, 2 x fIN2 + fIN1
• Fourth-order intermodulation products: 3 x fIN1 - fIN2,
3 x fIN2 - fIN1, 3 x fIN1 + fIN2, 3 x fIN2 + fIN1
• Fifth-order intermodulation products: 3 x fIN1 - 2 x fIN2,
3 x fIN2-2 x fIN1, 3 x fIN1+2 x fIN2, 3 x fIN2 + 2 x fIN1
Full-Power Bandwidth
A large -1dBFS analog input signal is applied to an
ADC and the input frequency is swept up to the point
where the amplitude of the digitized conversion result
has decreased by 3dB. The -3dB point is defined as
the full-power input bandwidth frequency of the ADC.
______________________________________________________________________________________
17
MAX1214
Static Parameter Definitions
Noise-Power Ratio (NPR)
requires sufficiently long data records to guarantee a
suitable number of samples inside the notch. NPR for the
MAX1214 was determined for 50MHz noise bandwidth
signals, simulating a typical cable signal environment
(see the Typical Operating Characteristics for test details
and results), and with a notch frequency of 28.8MHz.
NPR is commonly used to characterize the return path of
cable systems where the signals are typically individual
quadrature amplitude-modulated (QAM) carriers with a
frequency spectrum similar to noise. Numerous such
carriers are operated in a continuous spectrum, generating a noise-like signal, which covers a relatively broad
bandwidth. To test the MAX1214 for NPR, a “noise-like”
signal is passed through a high-order bandpass filter to
produce an approximately square spectral pedestal of
noise with about the same bandwidth as the signals
being simulated. Following the bandpass filter, the signal
is passed through a narrow band-reject filter to produce
a deep notch at the center of the noise pedestal. Finally,
this signal is applied to the MAX1214 and its digitized
results analyzed. The RMS noise power of the signal
inside the notch is compared with the RMS noise level
outside the notch using an FFT. Note that the NPR test
Pin-Compatible, LowerSpeed/Resolution Versions
Applications that require lower resolution, a choice of
buffered or non-buffered outputs, and/or higher speed
can refer to other family members of the MAX1214.
Adjusting an application to a lower resolution has been
simplified by maintaining an identical pinout for all members of this high-speed family. See the Pin-Compatible
Versions table on the first page of this data sheet for a
selection of different resolution and speed grades.
Pin Configuration
63 62 61 60 59 58
D9P
D9N
D10N
D10P
D11N
D11P
ORN
ORP
OVCC
OGND
AVCC
67 66 65 64
AVCC
AGND
68
AVCC
AGND
AGND
T/B
TOP VIEW
57 56 55 54 53 52
AVCC
1
51
D8P
AGND
2
50
D8N
REFIO
3
49
D7P
REFADJ
4
48
D7N
AGND
5
47
D6P
AVCC
6
46
D6N
AGND
7
45
OGND
INP
8
44
OVCC
INN
9
43
DCLKP
AGND 10
42
DCLKN
AVCC
11
41
OVCC
AVCC 12
40
D5P
AVCC 13
39
D5N
D4P
EP
MAX1214
AVCC 14
38
AGND 15
37
D4N
AGND 16
36
D3P
35
D3N
CLKDIV 17
D2P
D2N
D1P
D1N
D0P
D0N
OVCC
OVCC
AVCC
OGND
AGND
CLKN
CLKP
AGND
AVCC
AGND
18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 33 34
AGND
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
QFN
18
______________________________________________________________________________________
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
68L QFN.EPS
For the MAX1214, the package code is G6800-4.
PACKAGE OUTLINE, 68L QFN, 10x10x0.9 MM
1
C
21-0122
2
______________________________________________________________________________________
19
MAX1214
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information
go to www.maxim-ic.com/packages.)
MAX1214
1.8V, 12-Bit, 210Msps ADC for
Broadband Applications
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information
go to www.maxim-ic.com/packages.)
PACKAGE OUTLINE, 68L QFN, 10x10x0.9 MM
1
C
21-0122
2
Revision History
Pages changed at Rev 1: 1, 2, 12–16, 18, 20
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2006 Maxim Integrated Products
is a registered trademark of Maxim Integrated Products, Inc.
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