19-3244; Rev 0; 4/04
KIT
ATION
EVALU
E
L
B
AVAILA
TFT-LCD DC-DC Converters with
Operational Amplifiers
The MAX1516/MAX1517/MAX1518 include a high-performance step-up regulator, two linear-regulator controllers,
and high-current operational amplifiers for active-matrix
thin-film transistor (TFT) liquid-crystal displays (LCDs).
Also included is a logic-controlled, high-voltage switch
with adjustable delay.
The step-up DC-DC converter provides the regulated
supply voltage for the panel source driver ICs. The converter is a high-frequency (1.2MHz) current-mode regulator with an integrated 14V n-channel MOSFET that
allows the use of ultra-small inductors and ceramic
capacitors. It provides fast transient response to pulsed
loads while achieving efficiencies over 85%.
The gate-on and gate-off linear-regulator controllers
provide regulated TFT gate-on and gate-off supplies
using external charge pumps attached to the switching
node. The MAX1518 includes five high-performance
operational amplifiers, the MAX1517 includes three,
and the MAX1516 includes one operational amplifier.
These amplifiers are designed to drive the LCD backplane (VCOM) and/or the gamma-correction divider
string. The devices feature high output current
(±150mA), fast slew rate (13V/µs), wide bandwidth
(12MHz), and rail-to-rail inputs and outputs.
The MAX1516/MAX1517/MAX1518 are available in 32pin thin QFN packages with a maximum thickness of
0.8mm for ultra-thin LCD panels.
Applications
Notebook Computer Displays
LCD Monitor Panels
Features
♦ 2.6V to 5.5V Input Supply Range
♦ 1.2MHz Current-Mode Step-Up Regulator
Fast Transient Response to Pulsed Load
High-Accuracy Output Voltage (1.5%)
Built-In 14V, 2.4A, 0.16Ω N-Channel MOSFET
High Efficiency (90%)
♦ Linear-Regulator Controllers for VGON and VGOFF
♦ High-Performance Operational Amplifiers
±150mA Output Short-Circuit Current
13V/µs Slew Rate
12MHz, -3dB Bandwidth
Rail-to-Rail Inputs/Outputs
♦ Logic-Controlled, High-Voltage Switch with
Adjustable Delay
♦ Timer-Delay Fault Latch for All Regulator Outputs
♦ Thermal-Overload Protection
♦ 0.6mA Quiescent Current
Minimal Operating Circuit
VCN
VCP
VIN
VMAIN
LX
IN
FB
STEP-UP
CONTROLLER
PGND
COMP
AGND
VCP
MAX1518
Automotive Displays
DRVP
GATE-ON
CONTROLLER
Ordering Information
FBP
VGON
SRC
DEL
PART
TEMP RANGE
PIN-PACKAGE
MAX1516ETJ
-40°C to +100°C
32 Thin QFN 5mm x 5mm
MAX1517ETJ
-40°C to +100°C
32 Thin QFN 5mm x 5mm
MAX1518ETJ
-40°C to +100°C
32 Thin QFN 5mm x 5mm
COM
SWITCH
CONTROL
CTL
VCN
DRN
DRVN
GATE-OFF
CONTROLLER
VGOFF
SUP
NEG1
OUT1
FBN
OP1
REF
POS1
NEG2
OUT2
REF
NEG4
OP2
OP4
POS2
OUT4
POS4
NEG5
OUT3
OP3
OP5
POS3
OUT5
POS5
BGND
Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1516/MAX1517/MAX1518
General Description
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
ABSOLUTE MAXIMUM RATINGS
IN, CTL to AGND ......................................................-0.3V to +6V
COMP, FB, FBP, FBN, DEL, REF to AGND ....-0.3V to (VIN + 0.3V)
PGND, BGND to AGND ......................................................±0.3V
LX to PGND ............................................................-0.3V to +14V
SUP to AGND .........................................................-0.3V to +14V
DRVP, SRC to AGND..............................................-0.3V to +30V
POS_, NEG_, OUT_ to AGND ...................-0.3V to (VSUP + 0.3V)
POS1 to NEG1, POS2 to NEG2, POS3 to NEG3,
POS4 to NEG4, POS5 to NEG5 ...............................-6V to +6V
DRVN to AGND ...................................(VIN - 30V) to (VIN + 0.3V)
COM, DRN to AGND ................................-0.3V to (VSRC + 0.3V)
DRN to COM............................................................-30V to +30V
OUT_ Maximum Continuous Output Current....................±75mA
LX Switch Maximum Continuous RMS Output Current .........1.6A
Continuous Power Dissipation (TA = +70°C)
32-Pin Thin QFN (derate 21.2mW/°C above +70°C) ..1702mW
Operating Temperature Range .........................-40°C to +100°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25µA, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
IN Supply Range
IN Undervoltage-Lockout
Threshold
SYMBOL
CONDITIONS
VIN
VUVLO
MIN
VIN rising, typical hysteresis = 150mV
2.3
VFB = VFBP = 1.4V, VFBN = 0,
LX not switching
IN Quiescent Current
IIN
TYP
MAX
UNITS
5.5
V
2.5
2.7
V
0.6
0.8
6
11
2.6
mA
VFB = 1.1V, VFBP = 1.4V, VFBN = 0,
LX switching
Duration to Trigger Fault
Condition
55
REF Output Voltage
-2µA < IREF < 50µA, VIN = 2.6V to 5.5V
1.231
Temperature rising
Thermal Shutdown
1.250
ms
1.269
+160
Hysteresis
V
°C
15
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
VIN
13
V
Operating Frequency
fOSC
1020
1200
1380
kHz
84
87
90
%
Oscillator Maximum Duty Cycle
FB Regulation Voltage
VFB
No load
TA = +25°C to +85°C
1.221
1.233
1.245
TA = 0°C to +85°C
1.218
1.233
1.247
0.96
1.00
1.04
V
±0.15
%/ V
+40
nA
FB Fault Trip Level
VFB falling
FB Load Regulation
0 < IMAIN < full load, transient only
FB Line Regulation
VIN = 2.6V to 5.5V
FB Input Bias Current
VFB = 1.4V
-40
FB Transconductance
∆ICOMP = 5µA
75
FB Voltage Gain
FB to COMP
2
-1.6
+0.04
150
600
_______________________________________________________________________________________
V
%
280
µS
V/ V
TFT-LCD DC-DC Converters with
Operational Amplifiers
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25µA, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
LX On-Resistance
SYMBOL
CONDITIONS
MIN
RLX(ON)
LX Leakage Current
ILX
VLX = 13V
LX Current Limit
ILIM
VFB = 1V, duty cycle = 65%
Current-Sense
Transconductance
Soft-Start Period
TYP
MAX
UNITS
160
250
mΩ
0.02
40
µA
2.5
3.0
3.5
A
3.0
3.8
5
S
tSS
Soft-Start Step Size
14
ms
ILIM / 8
A
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
4.5
MAX1518
SUP Supply Current
ISUP
Buffer configuration,
VPOS_ = 4V, no load
3
0.7
1.1
0
12
mV
±50
nA
VSUP
V
(VNEG_, VPOS_, VOUT_) ≅ VSUP / 2,
TA = +25°C
Input Bias Current
IBIAS
(VNEG_ , VPOS_, VOUT_) ≅ VSUP / 2
Input Common-Mode Range
VCM
+1
0
0 ≤ (VNEG_, VPOS_) ≤ VSUP
45
Open-Loop Gain
dB
125
Output Voltage Swing, High
Output Voltage Swing, Low
IOUT_ = 100µA
VSUP 15
VSUP 3
IOUT_ = 5mA
VSUP 150
VSUP 80
dB
mV
VOH
VOL
mA
2
MAX1516
VOS
CMRR
V
4.8
MAX1517
Input Offset Voltage
Common-Mode Rejection Ratio
13.0
3.2
IOUT_ = -100µA
2
15
IOUT_ = -5mA
70
150
Short-Circuit Current
To VSUP / 2, source or sink
50
Output Source and Sink Current
(VNEG_ , VPOS_, VOUT_) ≅ VSUP / 2,
|∆VOS| < 10mV
40
mA
DC, 6V ≤ VSUP ≤ 13V,
(VNEG_, VPOS_) ≅ VSUP/2
60
dB
Power-Supply Rejection Ratio
PSRR
Slew Rate
-3dB Bandwidth
Gain-Bandwidth Product
GBW
150
mV
mA
13
V/µs
RL = 10kΩ, CL = 10pF, buffer configuration
12
MHz
Buffer configuration
8
MHz
GATE-ON LINEAR-REGULATOR CONTROLLER
FBP Regulation Voltage
VFBP
FBP Fault Trip Level
FBP Input Bias Current
FBP Effective Load-Regulation
Error (Transconductance)
IFBP
IDRVP = 100µA
1.231
1.250
1.269
V
VFBP falling
0.96
1.00
1.04
V
VFBP = 1.4V
-50
+50
nA
-1.5
%
VDRVP = 10V, IDRVP = 50µA to 1mA
-0.7
_______________________________________________________________________________________
3
MAX1516/MAX1517/MAX1518
ELECTRICAL CHARACTERISTICS (continued)
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25µA, TA = 0°C to +85°C. Typical values are at TA = +25°C, unless otherwise noted.)
PARAMETER
SYMBOL
FBP Line (IN) Regulation Error
DRVP Sink Current
MIN
IDRVP = 100µA, 2.6V < VIN < 5.5V
IDRVP
DRVP Off-Leakage Current
Soft-Start Period
CONDITIONS
VFBP = 1.1V, VDRVP = 10V
1
VFBP = 1.4V, VDRVP = 28V
TYP
MAX
UNITS
±1.5
±5
mV
5
0.01
tSS
Soft-Start Step Size
mA
10
µA
14
ms
VREF /
128
V
GATE-OFF LINEAR-REGUALTOR CONTROLLER
FBN Regulation Voltage
VFBN
FBN Fault Trip Level
FBN Input Bias Current
IFBN
IDRVN = 100µA
235
250
265
mV
VFBN rising
370
420
470
mV
VFBN = 0
-50
+50
nA
11
25
mV
+0.7
±5
mV
FBN Effective Load-Regulation
Error (Transconductance)
VDRVN = -10V, IDRVN = 50µA to 1mA
FBN Line (IN) Regulation Error
IDRVN = 0.1mA, 2.6V < VIN < 5.5V
DRVN Source Current
IDRVN
DRVN Off-Leakage Current
Soft-Start Period
VFBN = 500mV, VDRVN = -10V
1
VFBN = 0V, VDRVN = -25V
4
-0.01
tSS
Soft-Start Step Size
mA
-10
µA
14
ms
VREF /
128
V
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1V
VTH(DEL)
4
5
6
µA
1.19
1.25
1.31
V
DEL Discharge Switch OnResistance
During UVLO, VIN = 2.2V
CTL Input Low Voltage
VIN = 2.6V to 5.5V
CTL Input High Voltage
VIN = 2.6V to 5.5V
2
CTL Input Leakage Current
CTL = AGND or IN
-1
0.6
CTL-to-SRC Propagation Delay
+1
100
µA
ns
28
ISRC
V
V
SRC Input Voltage Range
SRC Input Current
Ω
20
VDEL = 1.5V, CTL = IN
50
100
VDEL = 1.5V, CTL = AGND
15
30
V
µA
SRC to COM Switch OnResistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
6
12
Ω
DRN to COM Switch OnResistance
RDRN(ON)
VDEL = 1.5V, CTL = AGND
35
70
Ω
COM to PGND Switch OnResistance
RCOM(ON)
VDEL = 1.1V
1000
1800
Ω
4
350
_______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25µA, TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
IN Supply Range
SYMBOL
CONDITIONS
VIN
IN Undervoltage-Lockout
Threshold
VUVLO
VIN rising, typical hysteresis = 150mV
MIN
MAX
UNITS
2.6
5.5
V
2.265
2.715
V
VFB = VFBP = 1.4V, VFBN = 0,
LX not switching
IN Quiescent Current
IIN
REF Output Voltage
0.8
mA
VFB = 1.1V, VFBP = 1.4V, VFBN = 0,
LX switching
-2µA < IREF < 50µA, VIN = 2.6V to 5.5V
11
1.222
1.269
V
VIN
13
V
1020
1380
kHz
1.212
1.250
V
±0.15
%/ V
nA
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
Operating Frequency
fOSC
FB Regulation Voltage
VFB
No load
FB Line Regulation
VIN = 2.6V to 5.5V
FB Input Bias Current
VFB = 1.4V
-40
+40
FB Transconductance
∆ICOMP = 5µA
75
300
µS
250
mΩ
2.5
3.5
A
4.5
13.0
V
LX On-Resistance
RLX(ON)
LX Current Limit
ILIM
VFB = 1V, duty cycle = 65%
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
SUP Supply Current
ISUP
Input Offset Voltage
VOS
Input Common-Mode Range
VCM
Output Voltage Swing, High
Output Voltage Swing, Low
Buffer configuration,
VPOS_ = 4V, no load
MAX1518
4.8
MAX1517
3.0
MAX1516
1.1
(VNEG_, VPOS_, VOUT_) ≅ VSUP / 2
0
IOUT_ = 100µA
VSUP 15
IOUT_ = 5mA
VSUP 150
VOH
VOL
mA
12
mV
VSUP
V
mV
IOUT_ = -100µA
15
IOUT_ = -5mA
150
Source
50
Sink
50
Short-Circuit Current
To VSUP / 2
Output Source and Sink Current
(VNEG_ , VPOS_, VOUT_) ≅ VSUP / 2,
|∆VOS| < 10mV
mV
mA
40
mA
GATE-ON LINEAR-REGULATOR CONTROLLER
FBP Regulation Voltage
VFBP
IDRVP = 100µA
1.218
1.269
V
_______________________________________________________________________________________
5
MAX1516/MAX1517/MAX1518
ELECTRICAL CHARACTERISTICS
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VSUP = 8V, PGND = AGND = BGND = 0, IREF = 25µA, TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
SYMBOL
FBP Effective Load-Regulation
Error (Transconductance)
MIN
VDRVP = 10V, IDRVP = 50µA to 1mA
FBP Line (IN) Regulation Error
DRVP Sink Current
CONDITIONS
IDRVP = 100µA, 2.6V < VIN < 5.5V
IDRVP
VFBP = 1.1V, VDRVP = 10V
MAX
UNITS
-2
%
5
mV
1
mA
GATE-OFF LINEAR-REGULATOR CONTROLLER
FBN Regulation Voltage
VFBN
FBN Effective Load-Regulation
Error (Transconductance)
235
VDRVN = -10V, IDRVN = 50µA to 1mA
FBN Line (IN) Regulation Error
DRVN Source Current
IDRVN = 100µA
IDRVN = 0.1mA, 2.6V < VIN < 5.5V
IDRVN
VFBN = 500mV, VDRVN = -10V
265
mV
25
mV
5
mV
1
mA
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1V
VTH(DEL)
CTL Input Low Voltage
VIN = 2.6V to 5.5V
CTL Input High Voltage
VIN = 2.6V to 5.5V
4
6
µA
1.19
1.31
V
0.6
V
2
SRC Input Voltage Range
SRC Input Current
V
28
ISRC
VDEL = 1.5V, CTL = IN
100
VDEL = 1.5V, CTL = AGND
30
V
µA
SRC to COM Switch OnResistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
12
Ω
DRN to COM Switch OnResistance
RDRN(ON)
VDEL = 1.5V, CTL = AGND
70
Ω
COM to PGND Switch OnResistance
RCOM(ON)
VDEL = 1.1V
1800
Ω
350
Note 1: Specifications to -40°C are guaranteed by design, not production tested.
6
_______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
70
60
50
40
1.3
NO LOAD, SUP DISCONNECTED,
R1 = 95.3kΩ, R2 = 10.2kΩ
8
1.2
1.1
MAX1516 toc03
MAX1516 toc02
10
SUPPLY CURRENT (mA)
VIN = 3.3V
1.4
SWITCHING FREQUENCY (MHz)
VIN = 5.0V
80
EFFICIENCY (%)
MAX1516 toc01
100
90
STEP-UP SUPPLY CURRENT
vs. SUPPLY VOLTAGE
SWITCHING FREQUENCY
vs. INPUT VOLTAGE
STEP-UP EFFICIENCY
vs. LOAD CURRENT
CURRENT INTO INDUCTOR
6
4
CURRENT INTO IN PIN
2
VOUT = 13V
0
1.0
30
1
10
100
1000
3.0
2.5
3.5
4.0
4.5
5.0
5.5
2.5
STEP-UP REGULATOR SOFT-START
(HEAVY LOAD)
3.0
3.5
4.0
4.5
5.0
5.5
SUPPLY VOLTAGE (V)
INPUT VOLTAGE (V)
LOAD CURRENT (mA)
STEP-UP REGULATOR PULSED
LOAD-TRANSIENT RESPONSE
MAX1516 toc05
MAX1516 toc04
A
0V
A
200mA
B
0V
13V
B
C
0A
C
0A
2ms/div
A: VIN, 5V/div
B: VMAIN, 5V/div
C: INDUCTOR CURRENT, 1A/div
10µs/div
A: LOAD CURRENT, 1A/div
B: VMAIN, 200mV/div, AC-COUPLED
C: INDUCTOR CURRENT, 1A/div
_______________________________________________________________________________________
7
MAX1516/MAX1517/MAX1518
Typical Operating Characteristics
(Circuit of Figure 1. VIN = 5V, VMAIN = 13V, VGON = 24V, VGOFF = -8V, VOUT1 = VOUT2 = VOUT3 = VOUT4 = VOUT5 = 6.5V, TA = +25°C
unless otherwise noted.)
Typical Operating Characteristics (continued)
(Circuit of Figure 1. VIN = 5V, VMAIN = 13V, VGON = 24V, VGOFF = -8V, VOUT1 = VOUT2 = VOUT3 = VOUT4 = VOUT5 = 6.5V, TA = +25°C
unless otherwise noted.)
REF VOLTAGE LOAD REGULATION
A
GATE-ON REGULATOR LINE REGULATION
MAX1516 toc07
1.253
1.252
REF VOLTAGE (V)
0V
B
55ms
1.251
1.250
1.249
1.248
0A
10
20
30
40
50
VGON = 23.5V
IGON = 20mA
23
24
25
-0.15
-0.20
-0.25
-0.30
0.50
0.25
0
20
29
30
-0.2
-0.4
-0.6
-0.8
-0.25
15
28
GATE-OFF REGULATOR LOAD REGULATION
MAX1516 toc10
0.75
27
0
VOLTAGE ERROR (%)
-0.10
VGOFF = -8V
IGOFF = 50mA
26
INPUT VOLTAGE (V)
GATE-OFF REGULATOR LINE REGULATION
1.00
OUTPUT VOLTAGE ERROR (%)
MAX1516 toc09
-0.05
8
-0.6
MAX1516 toc11
GATE-ON REGULATOR LOAD REGULATION
LOAD CURRENT (mA)
-0.4
LOAD CURRENT (µA)
0
10
-0.2
-1.0
0
A: VMAIN, 5V/div
B: INDUCTOR CURRENT, 1A/div
5
0
-0.8
1.247
10ms/div
0
0.2
OUTPUT VOLTAGE ERROR (%)
MAX1516 toc06
MAX1516 toc08
TIMER DELAY LATCH
RESPONSE TO OVERLOAD
VOLTAGE ERROR (%)
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
-1.0
-16
-14
-12
INPUT VOLTAGE (V)
-10
-8
0
10
20
30
LOAD CURRENT (mA)
_______________________________________________________________________________________
40
50
TFT-LCD DC-DC Converters with
Operational Amplifiers
OPERATIONAL-AMPLIFIER
RAIL-TO-RAIL INPUT/OUTPUT
MAX1518 OPERATIONAL-AMPLIFIER
SUPPLY CURRENT vs. SUPPLY VOLTAGE
POWER-UP SEQUENCE
MAX1516 toc12
A
B
0V
0V
C
5
SUPPLY CURRENT (mA)
0V
A
0V
3
2
B
NO-LOAD
BUFFER CONFIGURATION
VPOS1 TO VPOS5 = VSUP / 2
1
0V
0V
0
4.5
A: VMAIN, 10V/div
B: VSRC, 20V/div
C: VGOFF, 10V/div
D: VGON, 20V/div
VSUP = 6V
4
D
4ms/div
MAX1516 toc14
MAX1516 toc13
6
6.0
6.5
7.0
7.5
8.0
40µs/div
8.5
A: INPUT SIGNAL, 2V/div
B: OUTPUT SIGNAL, 2V/div
SUPPLY VOLTAGE (V)
OPERATIONAL-AMPLIFIER
SMALL-SIGNAL STEP RESPONSE
OPERATIONAL-AMPLIFIER
LARGE-SIGNAL STEP RESPONSE
OPERATIONAL-AMPLIFIER
LOAD-TRANSIENT RESPONSE
MAX1516 toc17
MAX1516 toc16
MAX1516 toc15
VSUP = 6V
0V A
A
A
0V
0V
+50mA
B
0
B
B
-50mA
0V
0V
400ns/div
A: OUTPUT VOLTAGE, 1V/div, AC-COUPLED
B: OUTPUT CURRENT, 50mA/div
1µs/div
A: INPUT SIGNAL, 2V/div
B: OUTPUT SIGNAL, 2V/div
400ns/div
A: INPUT SIGNAL, 100mV/div
B: OUTPUT SIGNAL, 100mV/div
_______________________________________________________________________________________
9
MAX1516/MAX1517/MAX1518
Typical Operating Characteristics (continued)
(Circuit of Figure 1. VIN = 5V, VMAIN = 13V, VGON = 24V, VGOFF = -8V, VOUT1 = VOUT2 = VOUT3 = VOUT4 = VOUT5 = 6.5V, TA = +25°C
unless otherwise noted.)
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
Pin Description
PIN
10
NAME
FUNCTION
MAX1516
MAX1517
MAX1518
1
SRC
SRC
SRC
Switch Input. Source of the internal high-voltage p-channel MOSFET. Bypass SRC to
PGND with a minimum 0.1µF capacitor close to the pins.
2
REF
REF
REF
Reference Bypass Terminal. Bypass REF to AGND with a minimum of 0.22µF close to
the pins.
3
AGND
AGND
AGND
Analog Ground for Step-Up Regulator and Linear Regulators. Connect to power
ground (PGND) underneath the IC.
4
PGND
PGND
PGND
Power Ground. PGND is the source of the main step-up n-channel power MOSFET.
Connect PGND to the input-capacitor ground terminals through a short, wide PC board
trace. Connect to analog ground (AGND) underneath the IC.
5
OUT1
OUT1
OUT1
Operational-Amplifier 1 Output
6
NEG1
NEG1
NEG1
Operational-Amplifier 1 Inverting Input
7
POS1
POS1
POS1
Operational-Amplifier 1 Noninverting Input
8
N.C.
OUT2
OUT2
Operational-Amplifier 2 Output for the MAX1518 and MAX1517. Not Internally
Connected for the MAX1516.
9
N.C.
NEG2
NEG2
Operational-Amplifier 2 Inverting Input for the MAX1518 and MAX1517. Not Internally
Connected for the MAX1516.
10
I. C.
POS2
POS2
Operational-Amplifier 2 Noninverting Input for the MAX1518 and MAX1517. Internally
Connected for the MAX1516. Connect this pin to GND for the MAX1516.
11
BGND
BGND
BGND
Analog Ground for Operational Amplifiers. Connect to power ground (PGND)
underneath the IC.
12
N.C.
N.C.
POS3
Operational-Amplifier 3 Noninverting Input for the MAX1518. Not Internally Connected
for the MAX1517 and MAX1516.
13
N.C.
N.C.
OUT3
Operational-Amplifier 3 Output. Not Internally Connected for the MAX1517 and
MAX1516.
14
SUP
SUP
SUP
Operational-Amplifier Power Input. Positive supply rail for the operational amplifiers.
Typically connected to VMAIN. Bypass SUP to BGND with a 0.1µF capacitor.
15
N.C.
POS3
POS4
Operational-Amplifier 4 Noninverting Input for the MAX1518. Operational-Amplifier 3
Noninverting Input for the MAX1517. Not Internally Connected for the MAX1516.
16
N.C.
NEG3
NEG4
Operational-Amplifier 4 Inverting Input for the MAX1518. Operational-Amplifier 3
Inverting Input for the MAX1517. Not Internally Connected for the MAX1516.
17
N.C.
OUT3
OUT4
Operational-Amplifier 4 Output for the MAX1518. Operational-Amplifier 3 Output for the
MAX1517. Not Internally Connected for the MAX1516.
18
I. C.
I. C.
POS5
Operational-Amplifier 5 Noninverting Input for the MAX1518. Internally Connected for
the MAX1517 and MAX1516. Connect this pin to GND for the MAX1517 and MAX1516.
19
N.C.
N.C.
NEG5
Operational-Amplifier 5 Inverting Input. Not Internally Connected for the MAX1517 and
MAX1516.
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
PIN
NAME
FUNCTION
MAX1516
MAX1517
MAX1518
20
N.C.
N.C.
OUT5
21
LX
LX
LX
N-Channel Power MOSFET Drain and Switching Node. Connect the inductor and
Schottky diode to LX and minimize the trace area for lowest EMI.
22
IN
IN
IN
Supply Voltage Input. IN can range from 2.6V to 5.5V.
23
FB
FB
FB
Step-Up Regulator Feedback Input. Regulates to 1.236V (nominal). Connect a resistive
voltage-divider from the output (VMAIN) to FB to analog ground (AGND). Place the
divider within 5mm of FB.
24
COMP
COMP
COMP
Operational-Amplifier 5 Output. Not Internally Connected for the MAX1517 and
MAX1516.
Step-Up Regulator Error-Amplifier Compensation Point. Connect a series RC from
COMP to AGND. See the Loop Compensation section for component selection
guidelines.
Gate-On Linear-Regulator Feedback Input. FBP regulates to 1.25V (nominal). Connect
FBP to the center of a resistive voltage-divider between the regulator output and AGND
to set the gate-on linear-regulator output voltage. Place the resistive voltage-divider
close to the pin.
25
FBP
FBP
FBP
26
DRVP
DRVP
DRVP
Gate-On Linear-Regulator Base Drive. Open drain of an internal n-channel MOSFET.
Connect DRVP to the base of an external pnp pass transistor. See the Pass-Transistor
Selection section.
27
FBN
FBN
FBN
Gate-Off Linear-Regulator Feedback Input. FBN regulates to 250mV (nominal).
Connect FBN to the center of a resistive voltage-divider between the regulator output
and REF to set the gate-off linear-regulator output voltage. Place the resistive voltagedivider close to the pin.
28
DRVN
DRVN
DRVN
Gate-Off Linear-Regulator Base Drive. Open drain of an internal p-channel MOSFET.
Connect DRVN to the base of an external npn pass transistor. See the Pass-Transistor
Selection section.
29
DEL
DEL
DEL
High-Voltage Switch Delay Input. Connect a capacitor from DEL to AGND to set the
high-voltage switch startup delay.
30
CTL
CTL
CTL
High-Voltage Switch Control Input. When CTL is high, the high-voltage switch between
COM and SRC is on and the high-voltage switch between COM and DRN is off. When
CTL is low, the high-voltage switch between COM and SRC is off and the high-voltage
switch between COM and DRN is on. CTL is inhibited by the undervoltage lockout and
when the voltage on DEL is less than 1.25V.
31
DRN
DRN
DRN
Switch Input. Drain of the internal high-voltage back-to-back p-channel MOSFETs
connected to COM.
32
COM
COM
COM
Internal High-Voltage MOSFET Switch Common Terminal. Do not allow the voltage on
COM to exceed VSRC.
______________________________________________________________________________________
11
MAX1516/MAX1517/MAX1518
Pin Description (continued)
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
Typical Operating Circuit
The MAX1518 Typical Operating Circuit (Figure 1) is a
complete power-supply system for TFT LCDs. The circuit generates a +13V source-driver supply and +24V
and -8V gate-driver supplies. The input voltage range
for the IC is from +2.6V to +5.5V. The listed load currents in Figure 1 are available from a +4.5V to +5.5V
supply. Table 1 lists some recommended components,
and Table 2 lists the contact information of component
suppliers.
Table 1. Component List
DESIGNATION
Detailed Description
Main Step-Up Regulator
The main step-up regulator employs a current-mode,
fixed-frequency PWM architecture to maximize loop
bandwidth and provide fast transient response to
pulsed loads typical of TFT-LCD panel source drivers.
The 1.2MHz switching frequency allows the use of lowprofile inductors and ceramic capacitors to minimize
the thickness of LCD panel designs. The integrated
high-efficiency MOSFET and the IC’s built-in digital
soft-start functions reduce the number of external components required while controlling inrush currents. The
output voltage can be set from VIN to 13V with an external resistive voltage-divider. To generate an output voltage greater than 13V, an external cascoded MOSFET
is needed. See the Generating Output Voltages > 13V
section in the Design Procedures.
The regulator controls the output voltage and the power
delivered to the output by modulating the duty cycle (D)
of the internal power MOSFET in each switching cycle.
The duty cycle of the MOSFET is approximated by:
V
−V
D ≈ MAIN IN
VMAIN
C1
22µF, 6.3V X5R ceramic capacitor (1210)
TDK C3225X5R0J227M
C2
22µF, 16V X5R ceramic capacitor (1812)
TDK C4532X5X1C226M
D1
3A, 30V Schottky diode (M-flat)
Toshiba CMS02
D2, D3
The MAX1516/MAX1517/MAX1518 contain a highperformance step-up switching regulator, two low-cost
linear-regulator controllers, multiple high-current operational amplifiers, and startup timing and level-shifting
functionality useful for active-matrix TFT LCDs. Figure 2
shows the MAX1518 Functional Diagram.
DESCRIPTION
200mA, 100V, dual ultra-fast diodes (SOT23)
Fairchild MMBD4148SE
L1
3.0µH, 3A inductor
Sumida CDRH6D28-3R0
Q1
200mA, 40V pnp bipolar transistor (SOT23)
Fairchild MMBT3906
Q2
200mA, 40V npn bipolar transistor (SOT23)
Fairchild MMBT3904
Figure 3 shows the Functional Diagram of the step-up
regulator. An error amplifier compares the signal at FB
to 1.236V and changes the COMP output. The voltage
at COMP sets the peak inductor current. As the load
varies, the error amplifier sources or sinks current to the
COMP output accordingly to produce the inductor peak
current necessary to service the load. To maintain stability at high duty cycles, a slope-compensation signal
is summed with the current-sense signal.
On the rising edge of the internal clock, the controller
sets a flip-flop, turning on the n-channel MOSFET and
applying the input voltage across the inductor. The current through the inductor ramps up linearly, storing
energy in its magnetic field. Once the sum of the current-feedback signal and the slope compensation
exceeds the COMP voltage, the controller resets the
flip-flop and turns off the MOSFET. Since the inductor
current is continuous, a transverse potential develops
across the inductor that turns on the diode (D1). The
voltage across the inductor then becomes the difference between the output voltage and the input voltage.
Table 2. Component Suppliers
SUPPLIER
PHONE
FAX
WEBSITE
Fairchild
408-822-2000
408-822-2102
www.fairchildsemi.com
Sumida
847-545-6700
847-545-6720
www.sumida.com
TDK
847-803-6100
847-390-4405
www.component.tdk.com
Toshiba
949-455-2000
949-859-3963
www.toshiba.com/taec
12
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
D1
C2
22µF
C1
22µF
R10
10Ω
VMAIN
13V/500mA
R1
95.3kΩ
1%
LX
IN
MAX1516/MAX1517/MAX1518
LX
L1
3.0µH
VIN
4.5V TO 5.5V
FB
C18
0.1µF
LX
R1
10.2kΩ
1%
180kΩ
0.1µF
AGND
COMP
PGND
220µF
0.1µF
D2
LX
0.1µF
MAX1518
6.8kΩ
D3
6.8kΩ
DRVP
0.1µF
Q2
R7
332kΩ
1%
VGOFF
-8V/50mA
DRVN
R4
192kΩ
1%
Q1
FBP
R5
10.0kΩ
1%
FBN
0.22µF
R8
40.2kΩ
1%
0.47µF
SRC
REF
VGON
24V/20mA
COM
0.22µF
DRN
DEL
CTL
0.033µF
SUP
0.1µF
BGND
NEG1
OUT1
NEG2
OUT2
POS1
POS2
OUT3
TO VCOM
BACKPLANE
POS3
NEG4
POS4
OUT4
POS5
NEG5
OUT5
Figure 1. Typical Operating Circuit
______________________________________________________________________________________
13
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
VCN
VCP
VIN
VMAIN
LX
IN
FB
STEP-UP
CONTROLLER
PGND
COMP
AGND
VCP
MAX1518
DRVP
GATE-ON
CONTROLLER
FBP
VGON
SRC
DEL
COM
SWITCH
CONTROL
CTL
VCN
DRN
DRVN
GATE-OFF
CONTROLLER
VGOFF
SUP
NEG1
OUT1
FBN
OP1
REF
POS1
NEG2
OUT2
REF
NEG4
OP2
OP4
OUT4
POS4
POS2
NEG5
OUT3
OP3
OP5
OUT5
POS5
POS3
BGND
Figure 2. MAX1518 Functional Diagram
14
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
MAX1516/MAX1517/MAX1518
LX
RESET DOMINANT
CLOCK
S
R
PGND
Q
ILIM
COMPARATOR
SOFTSTART
VLIMIT
SLOPE COMP
PWM
COMPARATOR
Σ
CURRENT
SENSE
OSCILLATOR
FAULT
COMPARATOR
TO FAULT LATCH
1.0V
ERROR AMP
FB
1.236V
COMP
Figure 3. Step-Up Regulator Functional Diagram
This discharge condition forces the current through the
inductor to ramp back down, transferring the energy
stored in the magnetic field to the output capacitor and
the load. The MOSFET remains off for the rest of the
clock cycle.
Gate-On Linear-Regulator Controller, REG P
The gate-on linear-regulator controller (REG P) is an
analog gain block with an open-drain n-channel output.
It drives an external pnp pass transistor with a 6.8kΩ
base-to-emitter resistor (Figure 1). Its guaranteed basedrive sink current is at least 1mA. The regulator including
Q1 in Figure 1 uses a 0.47µF ceramic output capacitor
and is designed to deliver 20mA at 24V. Other output
voltages and currents are possible with the proper pass
transistor and output capacitor. See the Pass-Transistor
Selection and Stability Requirements sections.
REG P is typically used to provide the TFT-LCD gate
drivers’ gate-on voltage. Use a charge pump with as
many stages as necessary to obtain a voltage exceeding the required gate-on voltage (see the Selecting the
Number of Charge-Pump Stages section). Note the
voltage rating of the DRVP is 28V. If the charge-pump
output voltage can exceed 28V, an external cascode
npn transistor should be added as shown in Figure 4.
Alternately, the linear regulator can control an intermediate charge-pump stage while regulating the final
charge-pump output (Figure 5).
REG P is enabled after the REF voltage exceeds 1.0V.
Each time it is enabled, the controller goes through a
soft-start routine that ramps up its internal reference
DAC in 128 steps.
______________________________________________________________________________________
15
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
LX
FROM CHARGE-PUMP
OUTPUT
VMAIN
0.1µF
VMAIN
13V
0.1µF
DRVP
NPN CASCODE
TRANSISTOR
PNP PASS
TRANSISTOR
6.8kΩ
VGON
MAX1516
MAX1517
MAX1518
MAX1516
MAX1517
MAX1518
FBP
Q1
DRVP
VGON
35V
0.47µF
0.22µF
267kΩ
1%
FBP
10.0kΩ
1%
Figure 4. Using Cascoded npn for Charge-Pump Output
Voltages >28V
Figure 5. The linear regulator controls the intermediate chargepump stage.
Gate-Off Linear-Regulator Controller, REG N
Short-Circuit Current Limit
The operational amplifiers limit short-circuit current to
approximately ±150mA if the output is directly shorted to
SUP or to BGND. If the short-circuit condition persists, the
junction temperature of the IC rises until it reaches the
thermal-shutdown threshold (+160°C typ). Once the junction temperature reaches the thermal-shutdown threshold,
an internal thermal sensor immediately sets the thermal
fault latch, shutting off all the IC’s outputs. The device
remains inactive until the input voltage is cycled.
The gate-off linear-regulator controller (REG N) is an
analog gain block with an open-drain p-channel output.
It drives an external npn pass transistor with a 6.8kΩ
base-to-emitter resistor (Figure 1). Its guaranteed basedrive source current is at least 1mA. The regulator
including Q2 in Figure 1 uses a 0.47µF ceramic output
capacitor and is designed to deliver 50mA at -8V. Other
output voltages and currents are possible with the proper
pass transistor and output capacitor (see the PassTransistor Selection and Stability Requirements sections).
REG N is typically used to provide the TFT-LCD gate
drivers’ gate-off voltage. A negative voltage can be
produced using a charge-pump circuit as shown in
Figure 1. REG N is enabled after the voltage on REF
exceeds 1.0V. Each time it is enabled, the control goes
through a soft-start routine that ramps down its internal
reference DAC from VREF to 250mV in 128 steps.
Operational Amplifiers
The MAX1518 has five operational amplifiers, the
MAX1517 has three operational amplifiers, and the
MAX1516 has one operational amplifier. The operational
amplifiers are typically used to drive the LCD backplane
(VCOM) or the gamma-correction divider string. They
feature ±150mA output short-circuit current, 13V/µs slew
rate, and 12MHz bandwidth. The rail-to-rail input and
output capability maximizes system flexibility.
16
Driving Pure Capacitive Load
The operational amplifiers are typically used to drive
the LCD backplane (VCOM) or the gamma-correction
divider string. The LCD backplane consists of a distributed series capacitance and resistance, a load that can
be easily driven by the operational amplifier. However,
if the operational amplifier is used in an application with
a pure capacitive load, steps must be taken to ensure
stable operation.
As the operational amplifier’s capacitive load increases,
the amplifier’s bandwidth decreases and gain peaking
increases. A 5Ω to 50Ω small resistor placed between
OUT_ and the capacitive load reduces peaking but also
reduces the gain. An alternative method of reducing
peaking is to place a series RC network (snubber) in parallel with the capacitive load. The RC network does not
continuously load the output or reduce the gain. Typical
values of the resistor are between 100Ω and 200Ω, and
the typical value of the capacitor is 10nF.
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
VIN
2.5V
VREF
1.05V
VMAIN
VGON
Reference Voltage (REF)
The reference output is nominally 1.25V and can
source at least 50µA (see the Typical Operating
Characteristics). Bypass REF with a 0.22µF ceramic
capacitor connected between REF and AGND.
Power-Up Sequence and Soft-Start
Once the voltage on IN exceeds approximately 1.7V,
the reference turns on. With a 0.22µF REF bypass
capacitor, the reference reaches its regulation voltage
of 1.25V in approximately 1ms. When the reference
voltage exceeds 1.0V, the ICs enable the main step-up
regulator, the gate-on linear-regulator controller, and
the gate-off linear-regulator controller simultaneously.
The IC employs soft-start for each regulator to minimize
inrush current and voltage overshoot and to ensure a
well-defined startup behavior. During the soft-start, the
main step-up regulator directly limits the peak inductor
current. The current-limit level is increased through the
soft-start period from 0 up to the full current-limit value
in eight equal current steps (ILIM / 8). The maximum
load current is available after the output voltage reaches regulation (which terminates soft-start), or after the
soft-start timer expires. Both linear-regulator controllers
use a 7-bit soft-start DAC. For the gate-on linear regulator, the DAC output is stepped in 128 steps from zero
up to the reference voltage. For the gate-off linear regulator, the DAC output steps from the reference down to
250mV in 128 steps. The soft-start duration is 14ms
(typ) for all three regulators.
A capacitor (CDEL) from DEL to AGND determines the
switch-control-block startup delay. After the input voltage exceeds the UVLO threshold (2.5V typ) and the
soft-start routine for each regulator is complete and
there is no fault detected, a 5µA current source starts
charging CDEL. Once the capacitor voltage exceeds
12ms
INPUT SOFT- SOFTVOLTAGE START START
OK BEGINS ENDS
1.25V
VGOFF
VDEL
SWITCH
CONTROL
ENABLED
Figure 6. Power-Up Sequence
1.25V (typ), the switch-control block is enabled as
shown in Figure 6. After the switch-control block is
enabled, COM can be connected to SRC or DRN
through the internal p-channel switches, depending
upon the state of CTL. Before startup and when IN is
less than VUVLO, DEL is internally connected to AGND
to discharge CDEL. Select CDEL to set the delay time
using the following equation:
CDEL = DELAY _ TIME ×
5µA
1.25V
Switch-Control Block
The switch-control input (CTL) is not activated until all
four of the following conditions are satisfied: the input
voltage exceeds VUVLO, the soft-start routine of all the
regulators is complete, there is no fault condition
detected, and VDEL exceeds its turn-on threshold. As
shown in Figure 7, COM is pulled down to PGND
through a 1kΩ resistor when the switch control is not
activated. Once activated and if CTL is high, the 5Ω
internal p-channel switch (Q1) between COM and SRC
turns on and the 30Ω p-channel switch (Q2) between
DRN and COM turns off. If CTL is low, Q1 turns off and
Q2 turns on.
______________________________________________________________________________________
17
MAX1516/MAX1517/MAX1518
Undervoltage Lockout (UVLO)
The undervoltage-lockout (UVLO) circuit compares the
input voltage at IN with the UVLO threshold (2.5V rising,
2.35V falling, typ) to ensure the input voltage is high
enough for reliable operation. The 150mV (typ) hysteresis
prevents supply transients from causing a restart. Once
the input voltage exceeds the UVLO rising threshold,
startup begins. When the input voltage falls below the
UVLO falling threshold, the controller turns off the main
step-up regulator, turns off the linear-regulator outputs,
and disables the switch control block; the operationalamplifier outputs are high impedance.
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
IN
MAX1516
MAX1517
MAX1518
5µA
2.5V
FB OK
FBP OK
FBN OK
Q1
SRC
DEL
REF
COM
1kΩ
CTL
Q2
Q3
DRN
Figure 7. Switch-Control Block
Fault Protection
Thermal-Overload Protection
During steady-state operation, if the output of the main
regulator or any of the linear-regulator outputs does not
exceed its respective fault-detection threshold, the
MAX1516/MAX1517/MAX1518 activate an internal fault
timer. If any condition or combination of conditions indicates a continuous fault for the fault-timer duration
(55ms typ), the MAX1516/MAX1517/MAX1518 set the
fault latch to shut down all the outputs except the reference. Once the fault condition is removed, cycle the
input voltage (below the UVLO falling threshold) to
clear the fault latch and reactivate the device. The faultdetection circuit is disabled during the soft-start time.
Thermal-overload protection prevents excessive power
dissipation from overheating the MAX1516/MAX1517/
MAX1518. When the junction temperature exceeds TJ =
+160°C, a thermal sensor immediately activates the
fault protection, which shuts down all outputs except
the reference, allowing the device to cool down. Once
the device cools down by approximately 15°C, cycle
the input voltage (below the UVLO falling threshold) to
clear the fault latch and reactivate the device.
The thermal-overload protection protects the controller
in the event of fault conditions. For continuous operation, do not exceed the absolute maximum junction
temperature rating of TJ = +150°C.
18
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
Main Step-Up Regulator
Inductor Selection
The minimum inductance value, peak current rating,
and series resistance are factors to consider when
selecting the inductor. These factors influence the converter’s efficiency, maximum output load capability,
transient-response time, and output voltage ripple. Size
and cost are also important factors to consider.
The maximum output current, input voltage, output voltage, and switching frequency determine the inductor
value. Very high inductance values minimize the current ripple and therefore reduce the peak current,
which decreases core losses in the inductor and I2R®
losses in the entire power path. However, large inductor values also require more energy storage and more
turns of wire, which increases size and can increase
I 2 R losses in the inductor. Low inductance values
decrease the size but increase the current ripple and
peak current. Finding the best inductor involves choosing the best compromise between circuit efficiency,
inductor size, and cost.
The equations used here include a constant LIR, which
is the ratio of the inductor peak-to-peak ripple current
to the average DC inductor current at the full load current. The best trade-off between inductor size and circuit efficiency for step-up regulators generally has an
LIR between 0.3 and 0.5. However, depending on the
AC characteristics of the inductor core material and
ratio of inductor resistance to other power-path resistances, the best LIR can shift up or down. If the inductor resistance is relatively high, more ripple can be
accepted to reduce the number of turns required and
increase the wire diameter. If the inductor resistance is
relatively low, increasing inductance to lower the peak
current can decrease losses throughout the power
path. If extremely thin high-resistance inductors are
used, as is common for LCD-panel applications, the
best LIR can increase to between 0.5 and 1.0.
Once a physical inductor is chosen, higher and lower
values of the inductor should be evaluated for efficiency improvements in typical operating regions.
Calculate the approximate inductor value using the typical input voltage (VIN), the maximum output current
(IMAIN(MAX)), the expected efficiency (ηTYP) taken from
an appropriate curve in the Typical Operating
Characteristics section, and an estimate of LIR based
on the above discussion:
η
V
2
VMAIN − VIN
TYP
L = IN
×
V
I
f
LIR
MAIN MAIN(MAX) OSC
Choose an available inductor value from an appropriate
inductor family. Calculate the maximum DC input current at the minimum input voltage (VIN(MIN)) using conservation of energy and the expected efficiency at that
operating point (ηMIN) taken from the appropriate curve
in the Typical Operating Characteristics:
IIN(DCMAX
,
)=
IMAIN(MAX) × VMAIN
VIN(MIN) × ηMIN
Calculate the ripple current at that operating point and
the peak current required for the inductor:
IRIPPLE =
VIN(MIN) × (VMAIN − VIN(MIN) )
L × VMAIN × fOSC
IRIPPLE
IPEAK = IIN(DCMAX
,
)+
2
The inductor’s saturation current rating and the
MAX1516/MAX1517/MAX1518s’ LX current limit (ILIM)
should exceed IPEAK, and the inductor’s DC current
rating should exceed IIN(DC,MAX). For good efficiency,
choose an inductor with less than 0.1Ω series resistance.
Considering the Typical Operating Circuit, the maximum
load current (IMAIN(MAX)) is 500mA with a 13V output and
a typical input voltage of 5V. Choosing an LIR of 0.5 and
estimating efficiency of 85% at this operating point:
5V 2 13V − 5V 0.85
L=
≈ 3.3µH
13V 0.5A × 1.2MHz 0.5
Using the circuit’s minimum input voltage (4.5V) and
estimating efficiency of 80% at that operating point:
IIN(DCMAX
,
)=
0.5A × 13V
≈ 1.8A
4.5V × 0.8
The ripple current and the peak current are:
4.5V × (13V − 4.5V)
≈ 0.74A
3.3µH × 13V × 1.2MHz
0.74A
IPEAK = 1.8A +
≈ 2.2A
2
IRIPPLE =
I2R is a registered trademark of Instruments for Research and
Industry, Inc.
______________________________________________________________________________________
19
MAX1516/MAX1517/MAX1518
Design Procedure
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
Output-Capacitor Selection
The total output voltage ripple has two components: the
capacitive ripple caused by the charging and discharging of the output capacitance, and the ohmic ripple due
to the capacitor’s equivalent series resistance (ESR).
VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR)
V
I
−V
VRIPPLE(C) ≈ MAIN MAIN IN , and
COUT VMAINfOSC
VRIPPLE(ESR) ≈ IPEAKRESR(COUT)
where I PEAK is the peak inductor current (see the
Inductor Selection section). For ceramic capacitors, the
output voltage ripple is typically dominated by
VRIPPLE(C). The voltage rating and temperature characteristics of the output capacitor must also be considered.
Input-Capacitor Selection
The input capacitor (CIN) reduces the current peaks
drawn from the input supply and reduces noise injection into the IC. A 22µF ceramic capacitor is used in the
Typical Applications Circuit (Figure 1) because of the
high source impedance seen in typical lab setups.
Actual applications usually have much lower source
impedance since the step-up regulator often runs
directly from the output of another regulated supply.
Typically, CIN can be reduced below the values used in
the Typical Applications Circuit. Ensure a low-noise
supply at IN by using adequate C IN . Alternately,
greater voltage variation can be tolerated on CIN if IN is
decoupled from CIN using an RC lowpass filter (see
R10 and C18 in Figure 1).
Rectifier Diode
The MAX1516/MAX1517/MAX1518s’ high switching frequency demands a high-speed rectifier. Schottky
diodes are recommended for most applications
because of their fast recovery time and low forward
voltage. In general, a 2A Schottky diode complements
the internal MOSFET well.
Output-Voltage Selection
The output voltage of the main step-up regulator can be
adjusted by connecting a resistive voltage-divider from
the output (VMAIN) to AGND with the center tap connected to FB (see Figure 1). Select R2 in the 10kΩ to 50kΩ
range. Calculate R1 with the following equation:
V
R1 = R2 × MAIN − 1
VFB
20
where VFB, the step-up regulator’s feedback set point,
is 1.236V. Place R1 and R2 close to the IC.
Generating Output Voltages >13V
The maximum output voltage of the step-up regulator is
13V, which is limited by the absolute maximum rating of
the internal power MOSFET. To achieve higher output
voltages, an external n-channel MOSFET can be cascoded with the internal FET (Figure 8). Since the gate of the
external FET is biased from the input supply, use a logiclevel FET to ensure that the FET is fully enhanced at the
minimum input voltage. The current rating of the FET
needs to be higher than the IC’s internal current limit.
Loop Compensation
Choose RCOMP to set the high-frequency integrator
gain for fast transient response. Choose CCOMP to set
the integrator zero to maintain loop stability.
For low-ESR output capacitors, use the following equations to obtain stable performance and good transient
response:
RCOMP ≈
315 × VIN × VOUT × COUT
L × IMAIN(MAX)
CCOMP ≈
VOUT × COUT
10 × IMAIN(MAX) × RCOMP
To further optimize transient response, vary RCOMP in
20% steps and CCOMP in 50% steps while observing
transient-response waveforms.
Charge Pumps
Selecting the Number of Charge-Pump Stages
For highest efficiency, always choose the lowest number of charge-pump stages that meet the output
requirement. Figures 9 and 10 show the positive and
negative charge-pump output voltages for a given
VMAIN for one-, two-, and three-stage charge pumps.
The number of positive charge-pump stages is given by:
V
+V
−V
nPOS = GON DROPOUT MAIN
VMAIN − 2 × VD
where nPOS is the number of positive charge-pump
stages, VGON is the gate-on linear-regulator REG P output, VMAIN is the main step-up regulator output, VD is
the forward-voltage drop of the charge-pump diode,
and VDROPOUT is the dropout margin for the linear regulator. Use VDROPOUT = 0.3V.
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
VIN
LX
nitely has a negligible effect on output-current capability because the internal switch resistance and the diode
impedance place a lower limit on the source impedance. A 0.1µF ceramic capacitor works well in most
low-current applications. The flying capacitor’s voltage
rating must exceed the following:
FB
VCX > n × VMAIN
STEP-UP
CONTROLLER
PGND
MAX1516
MAX1517
MAX1518
Figure 8. Operation with Output Voltages >13V Using
Cascoded MOSFET
where n is the stage number in which the flying capacitor appears, and VMAIN is the output voltage of the
main step-up regulator.
Charge-Pump Output Capacitor
Increasing the output capacitance or decreasing the
ESR reduces the output ripple voltage and the peak-topeak transient voltage. With ceramic capacitors, the
output voltage ripple is dominated by the capacitance
value. Use the following equation to approximate the
required capacitor value:
The number of negative charge-pump stages is given by:
−V
+ VDROPOUT
nNEG = GOFF
VMAIN − 2 × VD
where nNEG is the number of negative charge-pump
stages, VGOFF is the gate-off linear-regulator REG N
output, VMAIN is the main step-up regulator output, VD
is the forward-voltage drop of the charge-pump diode,
and VDROPOUT is the dropout margin for the linear regulator. Use VDROPOUT = 0.3V.
The above equations are derived based on the
assumption that the first stage of the positive charge
pump is connected to VMAIN and the first stage of the
negative charge pump is connected to ground.
Sometimes fractional stages are more desirable for better efficiency. This can be done by connecting the first
stage to VIN or another available supply. If the first
charge-pump stage is powered from V IN , then the
above equations become:
V
+V
+V
nPOS = GON DROPOUT IN
VMAIN − 2 × VD
−V
+ VDROPOUT + VIN
nNEG = GOFF
VMAIN − 2 × VD
Flying Capacitors
Increasing the flying-capacitor (CX) value lowers the
effective source impedance and increases the outputcurrent capability. Increasing the capacitance indefi-
COUT _ CP ≥
ILOAD _ CP
2fOSC VRIPPLE _ CP
where COUT_CP is the output capacitor of the charge
pump, I LOAD_CP is the load current of the charge
pump, and VRIPPLE_CP is the peak-to-peak value of the
output ripple.
Charge-Pump Rectifier Diodes
Use low-cost silicon switching diodes with a current rating equal to or greater than two times the average
charge-pump input current. If it helps avoid an extra
stage, some or all of the diodes can be replaced with
Schottky diodes with an equivalent current rating.
Linear-Regulator Controllers
Output-Voltage Selection
Adjust the gate-on linear-regulator (REG P) output voltage by connecting a resistive voltage-divider from the
REG P output to AGND with the center tap connected
to FBP (Figure 1). Select the lower resistor of the divider
R5 in the range of 10kΩ to 30kΩ. Calculate the upper
resistor R4 with the following equation:
V
R4 = R5 × GON − 1
VFBP
where VFBP = 1.25V (typ).
______________________________________________________________________________________
21
MAX1516/MAX1517/MAX1518
VMAIN
>13V
POSITIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. VMAIN
NEGATIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. VMAIN
60
-0
VD = 0.3V TO 1V
3-STAGE CHARGE PUMP
1-STAGE
CHARGE PUMP
-5
50
-10
40
-15
G_OFF (V)
G_ON (V)
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
2-STAGE CHARGE PUMP
30
20
-20
2-STAGE
CHARGE PUMP
-25
-30
3-STAGE
CHARGE PUMP
-35
10
-40
1-STAGE CHARGE PUMP
0
VD = 0.3V TO 1V
-45
2
4
6
8
10
12
14
VMAIN (V)
2
4
6
8
10
12
14
VMAIN (V)
Figure 9. Positive Charge-Pump Output Voltage vs. VMAIN
Figure 10. Negative Charge-Pump Output Voltage vs. VMAIN
Adjust the gate-off linear-regulator REG N output voltage by connecting a resistive voltage-divider from
VGOFF to REF with the center tap connected to FBN
(Figure 1). Select R8 in the range of 20kΩ to 50kΩ.
Calculate R7 with the following equation:
Therefore, transistors with current gain over 100 at the
maximum output current can be difficult to stabilize and
are not recommended unless the high gain is needed to
meet the load-current requirements.
The transistor’s saturation voltage at the maximum output current determines the minimum input-to-output
voltage differential that the linear regulator can support.
Also, the package’s power dissipation limits the usable
maximum input-to-output voltage differential. The maximum power-dissipation capability of the transistor’s
package and mounting must exceed the actual power
dissipated in the device. The power dissipated equals
the maximum load current (ILOAD(MAX)_LR) multiplied
by the maximum input-to-output voltage differential:
V
−V
R7 = R8 × FBN GOFF
VREF − VFBN
where VFBN = 250mV, VREF = 1.25V. Note that REF can
only source up to 50µA; using a resistor less than 20kΩ
for R8 results in higher bias current than REF can supply.
Pass-Transistor Selection
The pass transistor must meet specifications for current
gain (hFE), input capacitance, collector-emitter saturation
voltage and power dissipation. The transistor’s current
gain limits the guaranteed maximum output current to:
V
ILOAD(MAX) = IDRV − BE × hFE(MIN)
RBE
where IDRV is the minimum guaranteed base-drive current, VBE is the transistor’s base-to-emitter forward voltage drop, and RBE is the pullup resistor connected
between the transistor’s base and emitter. Furthermore,
the transistor’s current gain increases the linear regulator’s DC loop gain (see the Stability Requirements section), so excessive gain destabilizes the output.
22
P = ILOAD(MAX)_ LR × (VIN(MAX)_ LR − VOUT _ LR )
where VIN(MAX)_LR is the maximum input voltage of the
linear regulator, and VOUT_LR is the output voltage of
the linear regulator.
Stability Requirements
The MAX1516/MAX1517/MAX1518 linear-regulator controllers use an internal transconductance amplifier to
drive an external pass transistor. The transconductance
amplifier, the pass transistor, the base-emitter resistor,
and the output capacitor determine the loop stability.
The following applies to both linear-regulator controllers
in the MAX1516/MAX1517/MAX1518.
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
10 I
×h
A V _ LR ≅ × 1 + BIAS FE × VREF
VT ILOAD _ LR
where VT is 26mV at room temperature, and IBIAS is the
current through the base-to-emitter resistor (RBE). For
the MAX1516/MAX1517/MAX1518, the bias currents for
both the gate-on and gate-off linear-regulator controllers
are 0.1mA. Therefore, the base-to-emitter resistor for
both linear regulators should be chosen to set 0.1mA
bias current:
RBE =
0.7V
VBE
=
≈ 6.8kΩ
0.1mA 0.1mA
The output capacitor and the load resistance create the
dominant pole in the system. However, the internal
amplifier delay, pass transistor’s input capacitance,
and the stray capacitance at the feedback node create
additional poles in the system, and the output capacitor’s ESR generates a zero. For proper operation, use
the following equations to verify the linear regulator is
properly compensated:
1) First, determine the dominant pole set by the linear
regulator’s output capacitor and the load resistor:
fPOLE _ LR =
ILOAD(MAX)_ LR
2π × COUT _ LR × VOUT _ LR
gm is the transconductance of the pass transistor,
and fT is the transition frequency. Both parameters
can be found in the transistor’s data sheet. Because
RBE is much greater than RIN, the above equation
can be simplified:
fPOLE _ IN =
1
2π × CIN × RIN
Substituting for CIN and RIN yields:
f
fPOLE _ IN = T
hFE
4) Next, calculate the pole set by the linear regulator’s feedback resistance and the capacitance
between FB_ and AGND (including stray capacitance):
fPOLE _ FB =
1
2π × CFB × (RUPPER || RLOWER )
where C FB is the capacitance between FB_ and
AGND, RUPPER is the upper resistor of the linear
regulator’s feedback divider, and RLOWER is the
lower resistor of the divider.
5) Next, calculate the zero caused by the output
capacitor’s ESR:
fPOLE _ ESR =
1
2π × COUT _ LR × RESR
The unity-gain crossover of the linear regulator is:
fCROSSOVER = AV_LR ✕ fPOLE_LR
2) The pole created by the internal amplifier delay is
approximately 1MHz:
fPOLE_AMP = 1MHz
3) Next, calculate the pole set by the transistor’s
input capacitance, the transistor’s input resistance,
and the base-to-emitter pullup resistor:
fPOLE _ IN =
1
2π × CIN × (RBE || RIN )
where CIN =
where RESR is the equivalent series resistance of
COUT_LR.
To ensure stability, choose COUT_LR large enough so
the crossover occurs well before the poles and zero
calculated in steps 2 to 5. The poles in steps 3 and 4
generally occur at several megahertz, and using
ceramic capacitors ensures the ESR zero occurs at
several megahertz as well. Placing the crossover below
500kHz is sufficient to avoid the amplifier-delay pole
and generally works well, unless unusual component
choices or extra capacitances move one of the other
poles or the zero below 1MHz.
gm
h
, RIN = FE ,
2πfT
gm
______________________________________________________________________________________
23
MAX1516/MAX1517/MAX1518
The transconductance amplifier regulates the output
voltage by controlling the pass transistor’s base current. The total DC loop gain is approximately:
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
Applications Information
Power Dissipation
An IC’s maximum power dissipation depends on the
thermal resistance from the die to the ambient environment and the ambient temperature. The thermal resistance depends on the IC package, PC board copper
area, other thermal mass, and airflow.
The MAX1516/MAX1517/MAX1518, with their exposed
backside pad soldered to 1in2 of PC board copper,
can dissipate about 1.7W into +70°C still air. More PC
board copper, cooler ambient air, and more airflow
increase the possible dissipation, while less copper or
warmer air decreases the IC’s dissipation capability.
The major components of power dissipation are the
power dissipated in the step-up regulator and the
power dissipated by the operational amplifiers.
Step-Up Regulator
The largest portions of power dissipation in the step-up
regulator are the internal MOSFET, the inductor, and the
output diode. If the step-up regulator has 90% efficiency,
about 3% to 5% of the power is lost in the internal
MOSFET, about 3% to 4% in the inductor, and about 1%
in the output diode. The remaining 1% to 3% is distributed among the input and output capacitors and the PC
board traces. If the input power is about 5W, the power
lost in the internal MOSFET is about 150mW to 250mW.
Operational Amplifier
The power dissipated in the operational amplifiers
depends on their output current, the output voltage,
and the supply voltage:
PDSOURCE = IOUT _(SOURCE) × (VSUP − VOUT _ )
PDSINK = IOUT _(SINK) × VOUT _
where IOUT_(SOURCE) is the output current sourced by
the operational amplifier, and IOUT_(SINK) is the output
current that the operational amplifier sinks.
In a typical case where the supply voltage is 13V and
the output voltage is 6V with an output source current
of 30mA, the power dissipated is 180mV.
PC Board Layout and Grounding
Careful PC board layout is important for proper operation. Use the following guidelines for good PC board
layout:
• Minimize the area of high-current loops by placing
the inductor, the output diode, and the output
capacitors near the input capacitors and near the
LX and PGND pins. The high-current input loop
goes from the positive terminal of the input capacitor
24
to the inductor, to the IC’s LX pin, out of PGND, and
to the input capacitor’s negative terminal. The highcurrent output loop is from the positive terminal of
the input capacitor to the inductor, to the output
diode (D1), and to the positive terminal of the output
capacitors, reconnecting between the output capacitor and input capacitor ground terminals. Connect
these loop components with short, wide connections. Avoid using vias in the high-current paths. If
vias are unavoidable, use many vias in parallel to
reduce resistance and inductance.
• Create a power-ground island (PGND) consisting of
the input and output capacitor grounds, PGND pin,
and any charge-pump components. Connect all of
these together with short, wide traces or a small
ground plane. Maximizing the width of the powerground traces improves efficiency and reduces output voltage ripple and noise spikes. Create an
analog ground plane (AGND) consisting of the
AGND pin, all the feedback-divider ground connections, the operational-amplifier divider ground connections, the COMP and DEL capacitor ground
connections, and the device’s exposed backside
pad. Connect the AGND and PGND islands by connecting the PGND pin directly to the exposed backside pad. Make no other connections between these
separate ground planes.
• Place all feedback voltage-divider resistors as close
to their respective feedback pins as possible. The
divider’s center trace should be kept short. Placing
the resistors far away causes their FB traces to
become antennas that can pick up switching noise.
Take care to avoid running any feedback trace near
LX or the switching nodes in the charge pumps.
• Place the IN pin and REF pin bypass capacitors as
close to the device as possible. The ground connection of the IN bypass capacitor should be connected
directly to the AGND pin with a wide trace.
• Minimize the length and maximize the width of the
traces between the output capacitors and the load
for best transient responses.
• Minimize the size of the LX node while keeping it
wide and short. Keep the LX node away from feedback nodes (FB, FBP, and FBN) and analog ground.
Use DC traces to shield if necessary.
Refer to the MAX1518 evaluation kit for an example of
proper PC board layout.
Chip Information
TRANSISTOR COUNT: 4608
PROCESS: BiCMOS
______________________________________________________________________________________
TFT-LCD DC-DC Converters with
Operational Amplifiers
COM
DRN
CTL
DEL
DRVN
FBN
DRVP
FBP
TOP VIEW
32
31
30
29
28
27
26
25
SRC
1
24
COMP
REF
2
23
FB
AGND
3
22
IN
PGND
4
21
LX
MAX1516
I.C.
17
N.C.
9
10
11
12
13
14
15
16
N.C.
18
8
N.C.
7
N.C.
SUP
POS1
N.C.
N.C.
N.C.
N.C.
19
I.C.
20
6
BGND
5
NEG1
N.C.
OUT1
THIN QFN
5mm x 5mm
25
FBP
26
DRVP
FBP
27
FBN
DRVP
28
DRVN
FBN
29
DEL
DRVN
30
CTL
DEL
31
DRN
CTL
32
TOP VIEW
COM
DRN
TOP VIEW
COM
N.C. = NOT INTERNALLY CONNECTED
I.C. = INTERNALLY CONNECTED
32
31
30
29
28
27
26
25
SRC
1
24
COMP
SRC
1
24
COMP
REF
2
23
FB
REF
2
23
FB
AGND
3
22
IN
AGND
3
22
IN
PGND
4
21
LX
PGND
4
21
LX
MAX1517
MAX1518
POS5
17
OUT4
9
10
11
12
13
14
15
16
9
10
11
12
13
THIN QFN
5mm x 5mm
14
15
16
NEG4
18
8
SUP
7
OUT2
POS4
POS1
OUT3
OUT3
I.C.
17
POS3
18
8
POS2
7
OUT2
BGND
POS1
NEG2
NEG5
NEG3
OUT5
19
POS3
20
6
SUP
5
NEG1
N.C.
OUT1
N.C.
N.C.
N.C.
19
POS2
20
6
BGND
5
NEG1
NEG2
OUT1
THIN QFN
5mm x 5mm
N.C. = NOT INTERNALLY CONNECTED
I.C. = INTERNALLY CONNECTED
______________________________________________________________________________________
25
MAX1516/MAX1517/MAX1518
Pin Configurations
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
D2
0.15 C A
D
b
CL
0.10 M C A B
D2/2
D/2
PIN # 1
I.D.
QFN THIN.EPS
MAX1516/MAX1517/MAX1518
TFT-LCD DC-DC Converters with
Operational Amplifiers
k
0.15 C B
PIN # 1 I.D.
0.35x45∞
E/2
E2/2
CL
(NE-1) X e
E
E2
k
L
DETAIL A
e
(ND-1) X e
DETAIL B
e
L1
L
CL
CL
L
L
e
e
0.10 C
A
C
A1
0.08 C
A3
PACKAGE OUTLINE
16, 20, 28, 32, 40L, THIN QFN, 5x5x0.8mm
E
21-0140
COMMON DIMENSIONS
A
A3
b
D
E
k
L
0
0.20 REF.
0.02 0.05
0
0.20 REF.
0.02 0.05
0.02 0.05
0
0.20 REF.
0.20 REF.
0
-
0.05
0.20 REF.
0.25 0.30 0.35 0.25 0.30 0.35 0.20 0.25 0.30 0.20 0.25 0.30 0.15 0.20 0.25
4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10
4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10 4.90 5.00 5.10
0.65 BSC.
0.80 BSC.
e
L1
0.02 0.05
0.50 BSC.
0.50 BSC.
0.40 BSC.
- 0.25 - 0.25
- 0.25 0.35 0.45
0.25 - 0.25 0.30 0.40 0.50 0.45 0.55 0.65 0.45 0.55 0.65 0.30 0.40 0.50 0.40 0.50 0.60
-
-
-
-
-
-
N
ND
NE
16
4
4
20
5
5
JEDEC
WHHB
WHHC
-
-
-
-
-
WHHD-1
-
0.30 0.40 0.50
32
8
8
40
10
10
WHHD-2
-
28
7
7
D2
E2
DOWN
BONDS
PKG.
CODES
MIN.
NOM. MAX.
T1655-1
T1655-2
3.00
3.00
3.10 3.20 3.00
3.10 3.20 3.00
3.10 3.20
3.10 3.20
T2055-2
T2055-3
3.00
3.00
3.10 3.20 3.00
3.10 3.20 3.00
3.10 3.20
3.10 3.20
T2055-4
T2855-1
T2855-2
T2855-3
T2855-4
T2855-5
T2855-6
T2855-7
T3255-2
T3255-3
T3255-4
3.00
3.15
2.60
3.15
2.60
2.60
3.15
2.60
3.00
3.00
3.00
3.10
3.25
2.70
3.25
2.70
2.70
3.25
2.70
3.10
3.10
3.10
3.10
3.25
2.70
3.25
2.70
2.70
3.25
2.70
3.10
3.10
3.10
T4055-1
3.20
3.30 3.40 3.20
0.70 0.75 0.80 0.70 0.75 0.80 0.70 0.75 0.80 0.70 0.75 0.80 0.70 0.75 0.80
0
2
EXPOSED PAD VARIATIONS
PKG.
20L 5x5
28L 5x5
32L 5x5
40L 5x5
16L 5x5
SYMBOL MIN. NOM. MAX. MIN. NOM. MAX. MIN. NOM. MAX. MIN. NOM. MAX. MIN. NOM. MAX.
A1
1
3.20
3.35
2.80
3.35
2.80
2.80
3.35
2.80
3.20
3.20
3.20
MIN.
3.00
3.15
2.60
3.15
2.60
2.60
3.15
2.60
3.00
3.00
3.00
NOM. MAX. ALLOWED
3.20
3.35
2.80
3.35
2.80
2.80
3.35
2.80
3.20
3.20
3.20
3.30 3.40
NO
YES
NO
YES
NO
NO
NO
YES
YES
NO
NO
YES
NO
YES
NO
YES
NOTES:
1. DIMENSIONING & TOLERANCING CONFORM TO ASME Y14.5M-1994.
2. ALL DIMENSIONS ARE IN MILLIMETERS. ANGLES ARE IN DEGREES.
3. N IS THE TOTAL NUMBER OF TERMINALS.
4. THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION SHALL CONFORM TO JESD 95-1
SPP-012. DETAILS OF TERMINAL #1 IDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE
ZONE INDICATED. THE TERMINAL #1 IDENTIFIER MAY BE EITHER A MOLD OR MARKED FEATURE.
5. DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.25 mm AND 0.30 mm
FROM TERMINAL TIP.
6. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY.
7. DEPOPULATION IS POSSIBLE IN A SYMMETRICAL FASHION.
8. COPLANARITY APPLIES TO THE EXPOSED HEAT SINK SLUG AS WELL AS THE TERMINALS.
9. DRAWING CONFORMS TO JEDEC MO220, EXCEPT EXPOSED PAD DIMENSION FOR T2855-1,
T2855-3 AND T2855-6.
10. WARPAGE SHALL NOT EXCEED 0.10 mm.
PACKAGE OUTLINE
16, 20, 28, 32, 40L, THIN QFN, 5x5x0.8mm
21-0140
E
2
2
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2004 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
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