19-2741; Rev 0; 4/03
KIT
ATION
EVALU
LE
B
A
IL
A
AV
TFT LCD DC-to-DC Converter with
Operational Amplifiers
The two easy-to-use, high-performance operational
amplifiers can drive the LCD backplane (VCOM) and/or
the gamma correction divider string. The devices feature high short-circuit current (150mA), fast slew rate
(7.5V/µs), wide bandwidth (12MHz), and Rail-to-Rail®
inputs and outputs.
The MAX1542/MAX1543 are available in 20-pin thin
QFN packages with a maximum thickness of 0.8mm for
ultra-thin LCD panel design.
Applications
Notebook Computer Displays
Features
♦ Ultra-High-Performance Step-Up Regulator
Fast Transient Response to Pulsed Load Using
Current-Mode Control Architecture
High-Accuracy Output Voltage (1.3%)
Built-In 14V, 1.2A, 0.2Ω N-Channel Power
MOSFET with Lossless Current-Sensing
High Efficiency (85%)
8-Step Current-Controlled Digital Soft-Start
♦ Two High-Performance Operational Amplifiers
150mA Output Short-Circuit Current
7.5V/µs Slew Rate
12MHz -3dB Bandwidth
Rail-to-Rail Inputs/Outputs
Unity Gain Stable
♦ Logic-Controlled High-Voltage Switch with
Adjustable Delay
♦ Timer Delay Latch FB Fault Protection
♦ Thermal Protection
♦ 2.6V to 5.5V Input Operating Voltage Range
♦ 3.6mA (Switching), 0.45mA (Not Switching)
Quiescent Current
♦ Ultra-Thin 20-Pin Thin QFN Package
(5mm x 5mm x 0.8mm)
LCD Monitor Panels
PDAs
Car Navigation Displays
Ordering Information
PART
TEMP RANGE
PIN-PACKAGE
FB
MAX1542ETP
-40°C to +85°C
20 Thin QFN (5mm x 5mm)
16
DEL
COMP
17
18
19
20
DRN
TOP VIEW
CTL
Pin Configurations
MAX1543ETP
-40°C to +85°C
20 Thin QFN (5mm x 5mm)
COM
1
15
FREQ
SRC
2
14
IN
I.C.
3
13
LX
PGND
4
12
SUP
AGND
5
11
POS2
7
8
9
NEG1
OUT1
OUT2
NEG2 10
6
POS1
MAX1543
THIN QFN (5mm x 5mm)
Pin Configurations continued at end of data sheet.
Rail-to-Rail is a registered trademark of Nippon Motorola, Ltd.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1542/MAX1543
General Description
The MAX1542/MAX1543 include a high-performance
boost regulator and two high-current operational amplifiers for active matrix, thin-film transistor (TFT), liquidcrystal displays (LCDs). Also included is a logiccontrolled, high-voltage switch with adjustable delay.
The MAX1543 includes an additional high-voltage load
switch and features pin-selectable boost regulator
switching frequency.
The step-up DC-to-DC converter is a high-frequency
640kHz (MAX1543)/1.2MHz (MAX1542/MAX1543) current-mode regulator with a built-in power MOSFET that
allows the use of ultra-small inductors and ceramic
capacitors. It provides fast transient response to pulsed
loads while producing efficiencies over 85%.
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
ABSOLUTE MAXIMUM RATINGS
IN, CTL, COMP, FB, DEL, FREQ (MAX1543)
to AGND ...............................................................-0.3V to +6V
COMP, FB, DEL to AGND .............................-0.3V to (IN + 0.3V)
PGND to AGND ..................................................................±0.3V
LX to PGND ............................................................-0.3V to +14V
SUP, POS1, NEG1, OUT1, POS2,
NEG2, OUT2 to AGND .......................................-0.3V to +14V
POS1, NEG1, OUT1, POS2, NEG2,
OUT2 to AGND ......................................-0.3V to (SUP + 0.3V)
SRC, COM to AGND...............................................-0.3V to +30V
SRC to COM ...........................................................-0.3V to +30V
SRC to DRN (MAX1543).........................................-0.3V to +30V
COM to AGND ...........................................-0.3V to (SRC + 0.3V)
DRN (MAX1543) to AGND .........................-0.3V to (SRC + 0.3V)
DRN (MAX1543) to COM.........................................-30V to +30V
MAX1542 COM RMS Output Current ...............................+75mA
MAX1543 COM RMS Output Current ...............................±50mA
OUT1, OUT2 Continuous Output Current.........................±75mA
Continuous Power Dissipation (TA = +70°C)
20-Pin Thin QFN 5mm x 5mm
(derate 20.8mW/°C above +70°C) .............................1667mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature .....................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VSUP = 8V, VSRC = 28V, FREQ = IN (MAX1543), PGND = AGND = 0, TA = 0°C to +85°C, typical values at TA = +25°C,
unless otherwise noted.)
PARAMETER
IN Supply Range
IN Undervoltage Lockout
Threshold
IN Quiescent Current
SYMBOL
CONDITIONS
VIN
VUVLO
IIN
Duration to Trigger Fault
Condition
TYP
2.6
MAX
UNITS
5.5
V
VIN rising
2.3
2.5
2.7
VIN falling
2.2
2.35
2.5
VFB = 1.3V, LX not switching
0.45
0.65
VFB = 1.1V, LX switching
3.6
6.5
MAX1542
55
MAX1543
Thermal Shutdown
MIN
FREQ = AGND
51
FREQ = IN
55
Rising edge
160
Hysteresis
15
V
mA
ms
°C
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
Operating Frequency
fOSC
VIN
MAX1542
MAX1543
1020
1200
FREQ = AGND
512
600
768
FREQ = IN
1020
1200
1380
82
87
Oscillator Maximum Duty Cycle
FREQ Input Low Voltage
MAX1543, VIN = 2.6V to 5.5V
FREQ Input High Voltage
MAX1543, VIN = 2.6V to 5.5V
FREQ Pulldown Current
MAX1543, VFREQ = 1.0V
FB Regulation Voltage
VFB
No load
FB Fault Trip Level
VFB falling
FB Load Regulation
0 ≤ IMAIN ≤ full load
FB Line Regulation
VIN = 2.6V to 5.5V
2
13
1380
kHz
92
%
0.3 x VIN
V
0.7 x
VIN
V
3.5
5
6.5
TA = +85°C
1.224
1.240
1.256
TA = 0°C to +85°C
1.222
1.240
1.258
1
1.04
0.96
V
-1
-0.08
_______________________________________________________________________________________
µA
V
V
%
±0.15
%/V
TFT LCD DC-to-DC Converter with
Operational Amplifiers
(VIN = 3V, VSUP = 8V, VSRC = 28V, FREQ = IN (MAX1543), PGND = AGND = 0, TA = 0°C to +85°C, typical values at TA = +25°C,
unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
+40
nA
FB Input Bias Current
VFB = 1.5V
-40
FB Transconductance
∆ICOMP = 5µA
75
FB Voltage Gain
FB to COMP
700
210
400
mΩ
0.01
20
µA
1.2
1.5
1.8
A
0.30
0.50
0.65
Ω
LX On-Resistance
RLX(ON)
LX Leakage Current
ILX
VLX = 13V
LX Current Limit
ILIM
VFB = 1V, duty cycle = 65%
Current-Sense Transresistance
MAX1542
Soft-Start Period
tSS
MAX1543
160
280
µS
V/V
14
FREQ = AGND
13
FREQ = IN
ms
14
Soft-Start Step Size
ILIM / 8
A
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
SUP Supply Current
ISUP
Buffer configuration, VPOS_ = 4V, no load
1.3
1.9
mA
Input Offset Voltage
VOS
VCM = VSUP/2, TA = +25°C
0
12
mV
Input Bias Current
IBIAS
NEG1, NEG2, POS1, POS2
+1
±50
nA
Input Common-Mode Voltage
Range
VCM
VSUP
V
Common-Mode Rejection Ratio
CMRR
4.5
0
0 ≤ VNEG_, VPOS_ ≤ VSUP
Output Voltage Swing Low
90
dB
125
dB
IOUT_ = 100µA
VSUP 15
VSUP 2
IOUT_ = 5mA
VSUP 150
VSUP 80
mV
VOH
VOL
IOUT_ = -100µA
2
15
IOUT_ = -5mA
80
150
Source
50
150
Sink
50
140
Short-Circuit Current
To VSUP/2
Output Source-and-Sink Current
Buffer configuration, VPOS_ = 4V,
|∆VOS| < 10mV
40
DC, 6V ≤ VSUP ≤ 13V, VNEG_, VPOS_ =
VSUP/2
60
Power-Supply Rejection Ratio
PSRR
Gain-Bandwidth Product
GBW
mV
mA
mA
100
dB
7.5
V/µs
RL = 10kΩ, CL =10pF, buffer configuration
12
MHz
Buffer configuration
8
MHz
Slew Rate
-3dB Bandwidth
V
50
Open-Loop Gain
Output Voltage Swing High
13.0
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
During startup, VDEL = 1V
4
5
6
µA
_______________________________________________________________________________________
3
MAX1542/MAX1543
ELECTRICAL CHARACTERISTICS (continued)
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VSUP = 8V, VSRC = 28V, FREQ = IN (MAX1543), PGND = AGND = 0, TA = 0°C to +85°C, typical values at TA = +25°C,
unless otherwise noted.)
PARAMETER
SYMBOL
DEL Turn-On Threshold
VTH(DEL)
CONDITIONS
MIN
TYP
MAX
UNITS
1.178
1.240
1.302
V
DEL Discharge Switch OnResistance
During UVLO, VIN = 2.2V
CTL Input Low Voltage
VIN = 2.6V to 5.5V
CTL Input High Voltage
VIN = 2.6V to 5.5V
2
CTL Input Leakage Current
CTL = AGND or IN
-1
0.6
CTL-to-SRC Propagation Delay
DRN Input Current
V
V
+1
µA
100
SRC Input Voltage Range
SRC Input Current
Ω
20
ns
28
ISRC
IDRC
V
MAX1542
70
130
MAX1543
100
180
VDRN = 8V, CTL = AGND, VDEL = 1.5V
15
30
VDRN = 8V, CTL = AGND, VDEL = 1.5V,
MAX1543
90
150
MAX1542
5
10
MAX1543
15
30
30
60
Ω
1000
1800
Ω
VDRN = 8V, CTL = IN,
VDEL = 1.5V
SRC to COM Switch OnResistance
RSRC(ON)
VDEL = 1.5V,
CTL = IN
DRN to COM Switch OnResistance (MAX1543)
RDRN(ON)
VDEL = 1.5V, CTL = AGND
COM to PGND Switch OnResistance (MAX1543)
RCOM(ON)
VDEL = 1.1V
350
µA
µA
Ω
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VSUP = 8V, VSRC = 28V, FREQ = IN (MAX1543), PGND = AGND = 0, TA = -40°C to +85°C, unless otherwise noted.)
PARAMETER
IN Supply Range
IN Undervoltage Lockout
Threshold
IN Quiescent Current
SYMBOL
CONDITIONS
MAX
UNITS
2.6
5.5
V
VIN rising
2.3
2.7
VIN falling
2.2
2.5
VIN
VUVLO
IIN
MIN
TYP
VFB = 1.3V, LX not switching
0.65
VFB = 1.1V, LX switching
6.5
V
mA
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
Operating Frequency
fOSC
MAX1542
FB Regulation Voltage
13
1000
1400
FREQ = AGND
512
768
FREQ = IN
1000
1400
V
kHz
No load
1.215
1.260
FB Fault Trip Level
VFB falling
0.96
1.04
V
FB Line Regulation
VIN = 2.6V to 5.5V
0.15
%/V
FB Transconductance
∆ICOMP = 5µA
300
µS
400
mΩ
LX On-Resistance
4
VFB
MAX1543
VIN
75
RLX(ON)
_______________________________________________________________________________________
V
TFT LCD DC-to-DC Converter with
Operational Amplifiers
(VIN = 3V, VSUP = 8V, VSRC = 28V, FREQ = IN (MAX1543), PGND = AGND = 0, TA = -40°C to +85°C, unless otherwise noted.)
PARAMETER
LX Current Limit
SYMBOL
ILIM
CONDITIONS
VFB = 1V, duty cycle = 65%
Current-Sense Transresistance
MIN
MAX
UNITS
1.2
TYP
1.8
A
0.30
0.65
Ω
4.5
13.0
V
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
SUP Supply Current
ISUP
Buffer configuration, VPOS_ = 4V, no load
2.1
mA
Input Offset Voltage
VOS
VCM = VSUP/2, TA = +25ºC
12
mV
Input Bias Current
IBIAS
NEG1, NEG2, POS1, POS2
±50
nA
Input Common-Mode Voltage
Range
VCM
VSUP
V
Output Voltage Swing High
Output Voltage Swing Low
0
IOUT_ = 100µA
VSUP 15
IOUT_ = 5mA
VSUP 150
mV
VOH
VOL
IOUT_ = -100µA
15
IOUT_ = -5mA
150
Source
50
Sink
50
Short-Circuit Current
To VSUP/2
Output Source-and-Sink Current
Buffer configuration, VPOS_ = 4V,
| ∆VOS | < 10mV
mV
mA
40
mA
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1.0V
VTH (DEL)
CTL Input Low Voltage
VIN = 2.6V to 5.5V
CTL Input High Voltage
VIN = 2.6V to 5.5V
4
6
µA
1.178
1.302
V
0.6
V
2
SRC Input Voltage Range
SRC Input Current
ISRC
DRN Input Current
V
28
IDRN
VDRN = 8V, CTL = IN,
VDEL = 1.5V
MAX1542
130
MAX1543
180
VDRN = 8V, CTL = AGND, VDEL = 1.5V
30
VDRN = 8V, CTL = AGND, VDEL = 1.5V,
MAX1543
150
MAX1542
10
MAX1543
30
SRC to COM Switch OnResistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
DRN to COM Switch OnResistance (MAX1543)
RDRN(ON)
VDEL = 1.5V, CTL = AGND
COM to PGND Switch OnResistance (MAX1543)
RCOM(ON)
VDEL = 1.1V
350
V
µA
µA
Ω
60
Ω
1800
Ω
_______________________________________________________________________________________
5
MAX1542/MAX1543
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics
(VIN = 3.3V, VMAIN = 8V, fOSC = 1.2MHz, TA = +25°C, unless otherwise noted.)
70
VIN = 2.7V
80
VIN = 5V
70
VIN = 2.7V
65
60
60
55
55
7.9
7.8
7.7
7.6
1000
1
10
0.6
SUPPLY CURRENT (mA)
CURRENT INTO IN PIN
0.4
0.3
NO LOAD
fOSC = 1.2MHz
R1 = 75kΩ
R2 = 13.7kΩ
SUP DISCONNECTED
0.2
0.1
0
2.5
3.0
3.5
SWITCHING FREQUENCY
vs. INPUT VOLTAGE
NO LOAD
VIN = 3.3V
fOSC = 1.2MHz
R1 = 75kΩ
R2 = 13.7kΩ
SUP DISCONNECTED
1.2
CURRENT INTO INDUCTOR
0.8
CURRENT INTO IN PIN
0.4
4.5
5.0
5.5
MAX1543
IMAIN = 200mA
1200
FREQ = IN
1000
FREQ = AGND
800
600
400
-40
VIN (V)
-15
10
35
85
60
2.5
3.0
3.5
TEMPERATURE (°C)
4.5
SUP SUPPLY CURRENT
vs. TEMPERATURE
2.0
MAX1542 toc07
NO LOAD
BUFFER CONFIGURATION
POS_ = VSUP/2
4.0
VIN (V)
SUP SUPPLY CURRENT
vs. SUP VOLTAGE
1.75
1400
0
4.0
VSUP = 13V
1.50
ISUP (mA)
ISUP (mA)
1.6
1.25
VSUP = 8V
1.2
1.00
NO LOAD
BUFFER CONFIGURATION
VPOS = VSUP/2
VSUP = 5V
0.8
0.75
4.5
6.0
7.5
9.0
VSUP (V)
6
10.5
12.0
1000
STEP-UP REGULATOR SUPPLY CURRENT
vs. TEMPERATURE
1.6
CURRENT INTO INDUCTOR
0.5
100
LOAD CURRENT (mA)
2.0
MAX1542 toc04
0.7
10
1
1000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
STEP-UP REGULATOR SUPPLY CURRENT
vs. SUPPLY VOLTAGE
100
MAX1542 toc05
100
VIN = 3.3V
fOSC = 1.2MHz
7.5
50
10
1
SWITCHING FREQUENCY (kHz)
50
VIN = 3.3V
75
MAX1542 toc08
65
8.0
MAX1542 toc06
VIN = 3.3V
75
8.1
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
EFFICIENCY (%)
85
VIN = 5V
80
VIN = 5V
MAX1543
fOSC = 640kHz
L = 10µH
90
MAX1542 toc02
MAX1543
fOSC = 1.2MHz
L = 4.7µH
85
95
MAX1542 toc01
95
90
STEP-UP REGULATOR OUTPUT VOLTAGE
vs. LOAD CURRENT (VMAIN = 8V)
STEP-UP REGULATOR EFFICIENCY
vs. LOAD CURRENT (VMAIN = 8V)
MAX1542 toc03
STEP-UP REGULATOR EFFICIENCY
vs. LOAD CURRENT (VMAIN = 8V)
SUPPLY CURRENT (mA)
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
13.5
-40
-15
10
35
60
TEMPERATURE (°C)
_______________________________________________________________________________________
85
5.0
5.5
TFT LCD DC-to-DC Converter with
Operational Amplifiers
OPERATIONAL AMPLIFIER FREQUENCY
RESPONSE FOR VARIOUS CLOAD
OVERSHOOT (%)
56pF
15pF
-10
-20
100
FALLING EDGE
40
0
1k
10k
1
100k
100
LOAD CAPACITANCE (pF)
OPERATIONAL AMPLIFIER
OUTPUT HIGH VOLTAGE vs. LOAD
OPERATIONAL AMPLIFIER
OUTPUT LOW VOLTAGE vs. LOAD
160
MAX1542 toc11
VSUP = 8V
AV = 1
VSUP = 8V
AV = 1
120
VOL (mV)
120
80
40
80
40
0
0
0
2
4
6
10
8
0
2
IOUT_ (mA)
PSRR (dB)
80
60
40
400
350
RISING EDGE
MAX1542 toc14
450
VSUP = 8V
AV = +1
RL = 10kΩ
CL = 10pF
VCM = 4V
300
250
FALLING EDGE
1
200mA
IMAIN
200mA/div
20mA
IL
500mA/div
VMAIN
AC-COUPLED
100mV/div
150
L = 4.7µH
RCOMP = 120kΩ
CCOMP = 470pF
50
VSUP = 8V
0.1
10
200
100
20
8
MAX1542 toc15
500
SETTLING TIME (ns)
100
6
STEP-UP REGULATOR
LOAD-TRANSIENT RESPONSE
OPERATIONAL AMPLIFIER
SETTLING TIME vs. STEP SIZE
MAX1542 toc13
120
4
IOUT_ (mA)
POWER-SUPPLY REJECTION RATIO
vs. FREQUENCY
0
1000
FREQUENCY (Hz)
160
VSUP - VOUT (mV)
60
20
VSUP = 8V
AV = 1
RL = 10kΩ
-30
RISING EDGE
MAX1542 toc12
MAGNITUDE (dB)
80
1000pF
0
VSUP = 8V
RL = 10kΩ
AV = 1
POS_ = 4V ±50mV
MAX1542 toc10
100pF
10
100
MAX1542 toc09
20
OPERATIONAL AMPLIFIER OVERSHOOT
vs. LOAD CAPACITANCE
MAX1542/MAX1543
Typical Operating Characteristics (continued)
(VIN = 3.3V, VMAIN = 8V, fOSC = 1.2MHz, TA = +25°C, unless otherwise noted.)
0
10
100
FREQUENCY (Hz)
1k
10k
0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
40µs/div
STEP SIZE (V)
_______________________________________________________________________________________
7
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
Typical Operating Characteristics (continued)
(VIN = 3.3V, VMAIN = 8V, fOSC = 1.2MHz, TA = +25°C, unless otherwise noted.)
STEP-UP REGULATOR
PULSED LOAD-TRANSIENT RESPONSE
STARTUP SEQUENCE
MAX1542 toc16
MAX1542 toc17
VIN
2V/div
IMAIN
1A/div
VMAIN
8V/div
IL
500mA/div
VMAIN
100mV/div
AC-COUPLED
VGOFF
5V/div
L = 4.7µH
RCOMP = 120kΩ
CCOMP = 470pF
VCOM
10V/div
10µs/div
1ms/div
TIMER DELAY LATCH
RESPONSE TO OVERLOAD
HEAVY-LOAD
SOFT-START WAVEFORMS
OPERATIONAL AMPLIFIER
RAIL-TO-RAIL I/O PERFORMANCE
MAX1542 toc19
MAX1542 toc18
MAX1542 toc20
VMAIN
5V/div
VIN
5V/div
VMAIN
5V/div
IL
2A/div
IL
500mA/div
VOUT1
5V/div
VCOM
20V/div
RLOAD = 10Ω
VPOS1
5V/div
VOUT1
5V/div
VSUP = 8V
BUFFER CONFIGURATION
10ms/div
2ms/div
100µs/div
OPERATIONAL AMPLIFIER
LOAD-TRANSIENT RESPONSE
OPERATIONAL AMPLIFIER
LARGE-SIGNAL STEP RESPONSE
MAX1542 toc21
MAX1542 toc22
VOUT1
1V/div
AC-COUPLED
4V
VPOS1
500mV/div
AC-COUPLED
4V
+50
IOUT1
50mA/div
0
-50
VSUP = 8V
AV = 10
BUFFER CONFIGURATION
1µs/div
8
1µs/div
_______________________________________________________________________________________
VOUT1
2V/div
TFT LCD DC-to-DC Converter with
Operational Amplifiers
OPERATIONAL AMPLIFIER
LARGE-SIGNAL STEP RESPONSE
OPERATIONAL AMPLIFIER
SMALL-SIGNAL STEP RESPONSE
MAX1542 toc23
MAX1542 toc24
POS_
50mV/div
AC-COUPLED
CH2 + OVER
6.234%
CHI AMPL
4.86V
VOUT_
1V/div
CH2 - OVER
2.352%
CHI + OVER
4.970%
OUT_
50mV/div
AC-COUPLED
AV = 1
VSUP = 8V, AV = 1
200ns/div
1µs/div
Pin Description
PIN
NAME
FUNCTION
MAX1542
MAX1543
1
1
COM
Internal High-Voltage MOSFET Switch Common Terminal. Do not allow the voltage on
COM to exceed VSRC.
2
2
SRC
Switch Input. Source of the internal high-voltage P-channel MOSFET. Bypass SRC to
PGND with a minimum of 0.1µF close to the pins.
3, 15, 20
—
N.C.
No Connection. Not internally connected.
—
3
I.C.
Internal Connection. Make no connection to this pin.
4
4
PGND
Power Ground. PGND is the source of the main boost N-channel power MOSFET. Connect
PGND to the output capacitor ground terminals through a short, wide PC board trace.
Connect to analog ground (AGND) underneath the IC.
5
5
AGND
Analog Ground. Connect to power ground (PGND) underneath the IC.
6
6
POS1
Operational Amplifier 1 Noninverting Input
7
7
NEG1
Operational Amplifier 1 Inverting Input
8
8
OUT1
Operational Amplifier 1 Output
9
9
OUT2
Operational Amplifier 2 Output
10
10
NEG2
Operational Amplifier 2 Inverting Input
11
11
POS2
Operational Amplifier 2 Noninverting Input
12
12
SUP
13
13
LX
Operational Amplifier Power Input. Positive supply rail for the OUT1 and OUT2 amplifiers.
Typically connected to VMAIN. Bypass SUP to AGND with a 0.1µF capacitor.
Power MOSFET N-Channel Drain and Switching Node. Connect the inductor and catch
diode to LX and minimize the trace area for lowest EMI.
_______________________________________________________________________________________
9
MAX1542/MAX1543
Typical Operating Characteristics (continued)
(VIN = 3.3V, VMAIN = 8V, fOSC = 1.2MHz, TA = +25°C, unless otherwise noted.)
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
Pin Description (continued)
PIN
NAME
FUNCTION
MAX1542
MAX1543
14
14
IN
—
15
FREQ
16
16
FB
Step-Up Converter Feedback Input. Regulates to 1.24V (nominal). Connect a resistordivider from the output (VMAIN) to FB to analog ground (AGND). Place the resistor-divider
within 5mm of FB.
17
17
COMP
Step-Up Regulator Error Amplifier Compensation Point. Connect a series RC from COMP
to AGND. See the Loop Compensation section for component selection guidelines.
Supply Voltage. IN can range from 2.6V to 5.5V.
Oscillator Frequency Select Input. Pull FREQ low or leave it unconnected for 640kHz
operation. Connect FREQ high for 1.2MHz operation. This input has a 5µA pulldown
current.
High-Voltage Switch Delay Input. Connect a capacitor from DEL to AGND to set the highvoltage switch startup delay. A 5µA current source charges CDEL.
18
18
DEL
For the MAX1542, the high-voltage switch between SRC and COM is disabled until VDEL
exceeds 1.24V. Following the delay period, CTL controls the state of the high-voltage
switch.
For the MAX1543, the switches between SRC, COM, and DRN are disabled and a 1kΩ
pulldown between COM and PGND is enabled until VDEL exceeds 1.24V. Following the
delay period, the 1kΩ pulldown is released and CTL controls the state of the high-voltage
switches (see the Delay Control Circuit section).
19
19
CTL
High-Voltage Switch Control Input. When CTL is high, the high-voltage switch between
COM and SRC is on and the high-voltage switches between COM and DRN (MAX1543)
are off. When CTL is low, the high-voltage switch between COM and SRC is off and the
high-voltage switches between COM and DRN (MAX1543) are on. CTL is inhibited by the
undervoltage lockout and when VDEL is less than 1.24V.
—
20
DRN
Switch Input. Drain of the internal high-voltage back-to-back P-channel MOSFETs
connected to COM.
Typical Application Circuits
The MAX1542 typical application circuit (Figure 1) and
the MAX1543 typical application circuit (Figure 2) generate an +8V source driver supply and approximately
+22V and -7V gate driver supplies for TFT displays. The
input voltage is from +2.6V to +5.5V. Table 1 lists recommended components and Table 2 lists contact information for component suppliers.
Detailed Description
The MAX1542/MAX1543 include a high-performance
step-up regulator, two high-current operational amplifiers, and startup timing and level-shifting functionality
useful for active matrix TFT LCDs. Figure 3 shows the
MAX1542/MAX1543 functional diagram.
10
Main Step-Up Converter
The MAX1542/MAX1543 main step-up converter
switches at 1.2MHz or 640kHz (MAX1543 only) (see the
Oscillator Frequency (FREQ) section). The devices
employ a current-mode, fixed-frequency, pulse-width
modulation (PWM) architecture to maximize loop bandwidth providing fast transient response to pulsed loads
found in source drivers for TFT LCD panels. The highswitching frequency also allows the use of low-profile
inductors and capacitors to minimize the thickness of
LCD panel designs. The integrated high-efficiency
MOSFET and the IC’s built-in digital soft-start function
reduce the number of external components required
while controlling inrush current. The output voltage of
the main step-up converter (VMAIN) can be set from VIN
to 13V with an external resistive voltage-divider at FB.
______________________________________________________________________________________
TFT LCD DC-to-DC Converter with
Operational Amplifiers
MAX1542/MAX1543
Table 1. Component List
DESIGNATION
DESCRIPTION
PART
C1
10µF ±10%, 6.3V X5R ceramic capacitor
TDK C3216X5R0J106K
C8, C9
4.7µF ±10%, 10V X5R ceramic capacitors
TDK C3225X5R1A475K
1A, 30V Schottky diode
Toshiba CRS02
D1
D2, D3, D4
L1
200mA, 100V dual ultra-fast diodes
Fairchild MMBD4148SE
4.7µH, 1.3A inductor
Sumida CLS5D11HP-4R7
Table 2. Component Suppliers
SUPPLIER
PHONE
FAX
WEBSITE
847-956-0666
847-956-0702
www.sumida.com
847-803-6100
847-803-6296
www.component.tdk.com
Fairchild
888-522-5372
408-822-2104
www.fairchildsemi.com
Toshiba
949-455-2000
949-859-3963
www.toshiba.com/taec/
Inductors
Sumida USA
Capacitors
TDK
Diodes
The regulator controls the output voltage and the power
delivered to the outputs by modulating the duty cycle
(D) of the power MOSFET in each switching cycle. The
duty cycle of the MOSFET is approximated by:
V
−V
D ≈ MAIN IN
VMAIN
The device regulates the output voltage through a combination of an error amplifier, two comparators, and
several signal generators (Figure 3). The error amplifier
compares the signal at FB to 1.24V and varies the
COMP output. The voltage at COMP determines the
current trip point each time the internal MOSFET turns
on. As the load varies, the error amplifier sources or
sinks current to the COMP output accordingly to produce the inductor peak current necessary to service
the load. To maintain stability at high duty cycles, a
slope compensation signal is summed with the currentsense signal.
Operational Amplifiers
The MAX1542/MAX1543 include two operational amplifiers that are typically used to drive the LCD backplane
VCOM and/or the gamma correction divider string. The
operational amplifiers feature ±150mA output short-circuit current, 7.5V/µs slew rate, and 12MHz bandwidth.
The rail-to-rail inputs and outputs maximize flexibility.
Short-Circuit Current Limit
The MAX1542/MAX1543 operational amplifiers limit
short-circuit current to ±150mA if the output is directly
shorted to SUP or AGND. In such a condition, the junction temperature of the IC rises until it reaches the thermal shutdown threshold, typically +160°C. Once it
reaches this threshold, the IC shuts down and remains
inactive until IN falls below VUVLO.
Driving Pure Capacitive Loads
The operational amplifiers are typically used to drive
the LCD backplane (VCOM) or the gamma correction
divider string. The LCD backplane consists of a distributed series capacitance and resistance, a load easily
driven by the operational amplifiers. However, if the
operational amplifiers are used in an application with a
pure capacitive load, steps must be taken to ensure
stable operation.
As the operational amplifier’s capacitive load increases,
the amplifier bandwidth decreases and gain peaking
increases. A small 5Ω to 50Ω resistance placed between
OUT_ and the capacitive load reduces peaking but
reduces the amplifier gain. An alternative method of
reducing peaking is the use of a snubber circuit. A 150Ω
and 10nF (typ) shunt load, or snubber, does not continuously load the output or reduce amplifier gain.
______________________________________________________________________________________
11
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
C4
0.1µF
D4
G_ON
C3
0.1µF
G_OFF
-7V AT 20mA
D2
C2
0.1µF
+22V AT 20mA
C5
0.1µF
C6
0.1µF
D3
C7
0.1µF
VIN
2.6V TO 5.5V
VMAIN
L1
4.7µF
C1
10µF
IN
D1
+8V AT 250mA
C8
4.7µF
R1
75kΩ
C9
4.7µF
LX
FB
R2
13.7kΩ
COMP
SUP
R8
100kΩ
R5
40kΩ
MAX1542
C11
220pF
R3
40kΩ
POS1
POS2
R6
40kΩ
SRC
CTL
C10
33nF
COM
DEL
PGND
NEG1
OUT1
NEG2
OUT2
AGND
R4
40kΩ
TO VCOM
BACKPLANE
Figure 1. MAX1542 Typical Application Circuit
Delay Control Circuit
A capacitor from DEL to AGND selects the switch control
block supply startup delay. After the input voltage
exceeds VUVLO, a 5µA current source charges CDEL.
Once the capacitor voltage exceeds the turn-on threshold (1.24V) COM can be connected to SRC, depending
on the state of CTL. Before startup and when IN is less
than VUVLO, DEL is internally connected to AGND to discharge CDEL. Select CDEL using the following equation:
CDEL = (DELAY TIME) ×
5µA
1.24 V
MAX1542 Control Block Switch
The switch control input (CTL) is not activated until
VDEL exceeds the turn-on voltage (1.24V) and the input
voltage (VIN) exceeds VUVLO (2.5V). Once activated,
12
CTL controls the P-channel MOSFET, between COM
and SRC. A high at CTL turns on Q1 between SRC and
COM, and a low at CTL turns Q1 off (Figure 4).
MAX1543 Control Block Switch
The switch control input (CTL) is not activated until the
input voltage (VIN) exceeds VUVLO (2.5V) and VDEL
exceeds the turn-on voltage (1.24V). During UVLO or
when DEL is below the turn-on threshold, COM is
pulled low to PGND through Q3 and a 1kΩ resistance.
Once activated, CTL controls the COM MOSFETs,
switching COM between SRC and DRN. A high at CTL
turns on Q1 and disables Q2. A low at CTL turns on Q2
and turns off Q1 (Figure 4).
Undervoltage Lockout (UVLO)
The UVLO comparator of the MAX1542/MAX1543 compares the input voltage at IN with the UVLO threshold
______________________________________________________________________________________
TFT LCD DC-to-DC Converter with
Operational Amplifiers
D4
C3
0.1µF
G_OFF
-7V AT 20mA
D2
C2
0.1µF
C5
0.1µF
VIN
2.6V TO 5.5V
L1
4.7µF
C1
10µF
C6
0.1µF
D3
C7
0.1µF
C8
4.7µF
C9
4.7µF
LX
FB
FREQ
COMP
R2
13.7kΩ
SUP
R8
100kΩ
C11
220pF
G_ON
+22V AT 20mA
VMAIN
+8V AT 250mA
D1
R1
75kΩ
IN
MAX1542/MAX1543
C4
0.1µF
R5
40kΩ
R3
40kΩ
R6
40kΩ
R4
40kΩ
MAX1543
POS1
POS2
CTL
SRC
C10
33nF
COM
DRN
DEL
PGND
NEG1
OUT1
NEG2
OUT2
AGND
TO VCOM
BACKPLANE
Figure 2. MAX1543 Typical Application Circuit
(2.5V rising, 2.35V falling, typ) to ensure that the input
voltage is high enough for reliable operation. The
150mV (typ) hysteresis prevents supply transients from
causing a restart. Once the input voltage exceeds the
UVLO threshold, startup begins. When the input voltage falls below the UVLO threshold, the controller turns
off the N-channel MOSFET, the switch control block
turns off Q1, and the operational amplifier outputs float.
For the MAX1543, the switch control block also turns off
Q2 and turns on Q3 when the input voltage falls below
the UVLO threshold (Figure 4).
Oscillator Frequency (FREQ)
The MAX1542 internal oscillator is preset to 1.2MHz. The
internal oscillator frequency is pin programmable for the
MAX1543. Connect FREQ to ground or leave it unconnected for 640kHz operation and connect it to VIN for
1.2MHz operation. FREQ has a 5µA (typ) pulldown current.
Fault Protection
Once the soft-start routine is complete, if the output of
the main regulator is below the fault detection threshold,
the MAX1542/MAX1543 activate the fault timer. If the
fault condition continuously exists throughout the fault
timer duration, the MAX1542/MAX1543 set the fault
latch, which shuts down the device. After removing the
fault condition, cycle the input voltage (IN) below VUVLO
to clear the fault latch and reactivate the device.
Digital Soft-Start
The MAX1542/MAX1543 digital soft-start period duration is 14ms (typ). During this time, the MAX1542/
MAX1543 directly limit the peak inductor current, allowing from zero up to the full current-limit value in eight
equal current steps (ILIM/8). The maximum load current
is available after output voltage reaches the full regulation threshold (which terminates soft-start), or after the
soft-start timer expires.
______________________________________________________________________________________
13
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
SOFTSTART
COMP
ERROR
AMPLIFIER
IN
ERROR
COMPARATOR
FB
LX
CONTROL
AND DRIVER
LOGIC
1.24V
N
CLOCK
PGND
OSCILLATOR
FREQ*
SLOPE
COMPENSATION
CURRENT
SENSE
∑
AGND
5µA
SUP
EXT*
NEG1
5µA
OUT1
NEG2
SWITCH
CONTROL
(SEE FIGURE 4)
OUT2
POS1
POS2
MAX1542
MAX1543
DEL
SRC
COM
DRN*
CTL
*MAX1543 ONLY.
Figure 3. Functional Diagram
Thermal-Overload Protection
Thermal-overload protection prevents excessive power
dissipation from overheating the MAX1542/MAX1543.
When the junction temperature exceeds TJ = +160°C, a
thermal sensor immediately activates the fault protection, which shuts down the device, allowing the IC to
cool. The input voltage must fall (below VUVLO) to clear
the fault latch and reactivate the controller.
Thermal-overload protection protects the controller in
the event of fault conditions. For continuous operation,
do not exceed the absolute maximum junction-temperature rating of TJ = +150°C.
14
Applications Information
Inductor Selection
The primary considerations in inductor selection are
inductor physical shape, circuit efficiency, and cost.
The factors that determine the inductance value are
input voltage, output voltage, switching frequency, and
maximum output current. Final inductor selection
includes ensuring the chosen inductor meets the application’s peak current and RMS current requirements.
Very high inductance values minimize the current ripple
and therefore reduce the peak current, which decreases core losses in the inductor and I2R losses in the circuit’s entire power path. However, large inductance
______________________________________________________________________________________
TFT LCD DC-to-DC Converter with
Operational Amplifiers
MAX1542/MAX1543
IN
MAX1542
MAX1543
5µA
2.5V
N
SRC
Q1
P
DEL
COM
REF
P
CTL
Q2
P
1kΩ
DRN
Q3
N
MAX1543 ONLY
Figure 4. Switch Control
values also require more energy storage and more
turns of wire, which increase physical size and can
increase I2R losses in the inductor. Low inductance values decrease the physical size but increase the current
ripple and peak current. Finding the best inductor
involves choosing the best compromise between circuit
efficiency, inductor size, and cost.
The equations used here include a constant, LIR, which
is the ratio of the inductor peak-to-peak ripple current to
the average DC inductor current at the full output current. The best trade-off between inductor size and circuit efficiency for step-up converters generally has an
LIR between 0.3 and 0.5. However, depending on the
AC characteristics of the inductor core material and
ratio of inductor resistance to other power path resistances, the best LIR can shift up or down. If the inductor
resistance is relatively high, more ripple can be accepted to reduce the number of turns required and increase
the wire diameter. If the inductor resistance is relatively
low, increasing inductance to lower the peak current
can decrease losses throughout the power path. If
extremely thin, high-resistance inductors are used, as is
common for LCD panel applications, the best LIR can
increase to between 0.5 and 1.0.
Once a physical inductor is chosen, higher and lower
values of that inductor should be evaluated for efficiency improvements in typical operating regions.
Calculate the approximate inductor value using the typical input voltage (VIN), the maximum output current
(IMAIN(MAX)), the expected efficiency (ηTYP) taken from
an appropriate curve in the Typical Operating
Characteristics, and an estimate for LIR based on the
above paragraphs:
L ≅ VIN2 x ηTYP x (VMAIN − VIN ) /
(VMAIN2 x LIR x IMAIN(MAX) x fOSC )
Choose an available inductor value from an appropriate
inductor family. Calculate the maximum DC input current at the minimum input voltage VIN(MIN) using conservation of energy and the expected efficiency at that
______________________________________________________________________________________
15
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
operating point (ηMIN) taken from an appropriate curve
in the Typical Operating Characteristics:
IIN(DC,MAX) = IMAIN(MAX) ✕ VMAIN / (VIN(MIN) ✕ ηMIN)
Calculate the ripple current at that operating point and
the peak current required for the inductor:
IRIPPLE = VIN(MIN) ✕ (VMAIN -VIN(MIN)) / (L ✕ fOSC ✕
VMAIN)
IPEAK = IIN(DC,MAX) + (IRIPPLE) / 2
The inductor’s saturation current rating and the
MAX1542/MAX1543s’ LX current limit (I LIM ) should
exceed I PEAK and the inductor’s DC current rating
should exceed IIN(DC,MAX). For reasonable efficiency,
choose an inductor with less than 0.5Ω series resistance.
Considering the Typical Application Circuits, the maximum load current (IMAIN(MAX)) is 200mA with an 8V
output and a typical input voltage of 3.3V.
Input Capacitor Selection
The input capacitor (CIN) reduces the current peaks
drawn from the input supply and reduces noise injection into the device. A 10µF ceramic capacitor is used
in the Typical Application Circuits (Figures 1 and 2)
because of the high source impedance seen in typical
lab setups. Actual applications usually have much
lower source impedance since the step-up regulator
often runs directly from the output of another regulated
supply. Typically, CIN can be reduced below the values
used in the Typical Application Circuits. Ensure a lownoise supply at IN by using adequate CIN.
Output Voltage
The MAX1542/MAX1543 operate with an adjustable output from VIN to 13V. Connect a resistive voltage-divider
to FB (Typical Application Circuits) from the output
(VMAIN) to AGND. Select the resistor values as follows:
Choosing an LIR of 0.6 and estimating efficiency of
85% at this operating point:
V
R1 = R2 MAIN − 1
VFB
L = (3.3V)2 ✕ 0.85 ✕ (8V - 3.3V) / ((8V)2 ✕ 0.6 ✕ 0.2A ✕
1.2MHz) = 4.7µH
Using the circuit’s minimum input voltage (2.7V) and
estimating efficiency of 80% at that operating point,
IIN(DC,MAX) = (0.2A ✕ 8V / (2.7V ✕ 0.8)) = 741mA
where VFB, the step-up converter feedback set point, is
1.24V. Since the input bias current into FB is typically
zero, R2 can have a value up to 100kΩ without sacrificing accuracy, although lower values provide better
noise immunity. Connect the resistor-divider as close to
the IC as possible.
The ripple current and the peak current are:
IRIPPLE = 2.7V ✕ (8V - 2.7V) / (4.7µH ✕ 1.2MHz ✕ 8V)
= 317mA
IPEAK = 741mA + (317mA / 2) = 900mA
Output Capacitor Selection
The total output voltage ripple has two components: the
capacitive ripple caused by the charging and discharging of the output capacitance, and the ohmic ripple due to the capacitor’s equivalent series resistance
(ESR):
VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C)
VRIPPLE(ESR) ≅ IPEAK x RESR(COUT) , and
V
I
−V
VRIPPLE(C) ≅ MAIN MAIN IN
COUT VMAIN × ƒ OSC
where I PEAK is the peak inductor current (see the
Inductor Selection section). For ceramic capacitors, the
output voltage ripple is typically dominated by VRIPPLE(C). The voltage rating and temperature characteristics of the output capacitor must also be considered.
16
Loop Compensation
Choose RCOMP to set the high-frequency integrator
gain for fast transient response. Choose CCOMP to set
the integrator zero to maintain loop stability.
For low-ESR output capacitors, use the following equations to obtain stable performance and good transient
response:
RCOMP ≅
500 x VIN x VOUT x COUT
L x IMAIN(MAX)
CCOMP ≅
VOUT x COUT
10 x IMAIN(MAX) x RCOMP
To further optimize transient response, vary RCOMP in
20% steps and CCOMP in 50% steps while observing
transient response waveforms.
Charge Pumps
Selecting the Number of Charge-Pump Stages
For highest efficiency, always choose the lowest number of charge-pump stages that meet the output
requirements. Figures 5 and 6 show the positive and
______________________________________________________________________________________
TFT LCD DC-to-DC Converter with
Operational Amplifiers
MAX1542/MAX1543
POSITIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. VMAIN
NEGATIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. VMAIN
60
-0
VD = 0.3V TO 1V
3-STAGE CHARGE-PUMP
1-STAGE
CHARGE-PUMP
-5
50
-10
-15
G_OFF (V)
G_ON (V)
40
2-STAGE CHARGE-PUMP
30
20
-20
2-STAGE
CHARGE-PUMP
-25
-30
3-STAGE
CHARGE-PUMP
-35
10
-40
1-STAGE CHARGE-PUMP
0
VD = 0.3V TO 1V
-45
2
4
6
8
10
12
14
2
4
VMAIN (V)
6
8
10
12
14
VMAIN (V)
Figure 5. Positive Charge-Pump Output Voltage vs. VMAIN
Figure 6. Negative Charge-Pump Output Voltage vs. VMAIN
negative charge-pump output voltages for a given
VMAIN for one-, two-, and three-stage charge pumps,
based on the following equations:
ing greater than 8V. The flying capacitor in the second
stage (C4) requires a voltage rating greater than 16V.
G _ ON = VMAIN + n(VMAIN − VD )
G _ OFF = − n(VMAIN − VD )
where G_ON is the positive charge-pump output voltage, G_OFF is the negative charge-pump output voltage, n is the number of charge-pump stages, and VD is
the voltage drop across each diode.
VD is the forward voltage drop of the charge-pump
diodes.
Flying Capacitors
Increasing the flying capacitor (C3, C4, and C5) value
increases the output current capability. Increasing the
capacitance indefinitely has a negligible effect on output current capability because the internal switch resistance and the diode impedance limit the source
impedance. A 0.1µF ceramic capacitor works well in
most low-current applications. The flying capacitor’s
voltage rating must exceed the following:
VCX > n ✕ VMAIN
Where n is the stage number in which the flying capacitor appears, and VMAIN is the main output voltage. For
example, the two-stage positive charge pump in the
Typical Application Circuits (Figures 1 and 2) where
VMAIN = 8V contains two flying capacitors. The flying
capacitor in the first stage (C5) requires a voltage rat-
Charge-Pump Output Capacitor
Increasing the output capacitance or decreasing the
ESR reduces the output ripple voltage and the peak-topeak transient voltage. With ceramic capacitors, the
output voltage ripple is dominated by the capacitance
value. Use the following equation to approximate the
required capacitor value:
COUT ≥
ILOAD
2 × FOSC × VRIPPLE
where VRIPPLE is the acceptable peak-to-peak outputvoltage ripple.
Charge-Pump Rectifier Diodes
To maximize the available output voltage, use Schottky
diodes with a current rating equal to or greater than two
times the average charge-pump input current. If the
loaded charge-pump output voltage is greater than
required, some or all of the Schottky diodes can be
replaced with low-cost silicon switching diodes with an
equivalent current rating. The charge-pump input current is:
ICP _ IN = ICP _ OUT × n
where n is the number of charge-pump stages.
______________________________________________________________________________________
17
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
Power Dissipation
The MAX1542/MAX1543s’ maximum power dissipation
depends on the thermal resistance from the IC die to
the ambient environment and the ambient temperature.
The thermal resistance depends on the IC package, PC
board copper area, other thermal mass, and airflow.
The MAX1542/MAX1543, with their exposed backside
pad soldered to 1in2 of PC board copper, can dissipate
about 1.7W into +70°C still air. More PC board copper,
cooler ambient air, and more airflow increase the possible dissipation while less copper or warmer air
decreases the IC’s dissipation capability. The major
components of power dissipation are the power dissipated in the step-up converter and the power dissipated by the operational amplifiers.
Step-Up Converter
The largest portions of power dissipation in the step-up
converter are the internal MOSFET, inductor, and the
output diode. If the step-up converter has 90% efficiency, about 3% to 5% of the power is lost in the internal
MOSFET, about 3% to 4% in the inductor, and about
1% in the output diode. The rest of the 1% to 3% is distributed among the input and output capacitors and the
PC board traces. If the input power is about 3W, the
power lost in the internal MOSFET is about 90mW to
150mW.
Operational Amplifiers
The power dissipated in the operational amplifiers
depends on their output current, the output voltage,
and the supply voltage:
PDSOURCE = IOUT _(SOURCE) × (VSUP − VOUT _ )
PDSINK = IOUT _(SINK ) × VOUT _
where IOUT_(SOURCE) is the output current sourced by
the operational amplifier, and IOUT_(SINK) is the output
current that the operational amplifier sinks.
In a typical case where the supply voltage is 8V and
the output voltage is 4V with an output source current
of 30mA, the power dissipated is 120mW.
Layout Procedure
Careful PC board layout and routing are required for
high-frequency switching power supplies to achieve
good regulation, high efficiency, and stability. Use the
following guidelines for good PC board layout:
1) Place the input capacitors close enough to the IC to
provide adequate bypassing (within 1.5cm).
Connect the input capacitors to IN with a wide trace.
18
Minimize the area of high-current loops by placing
the inductor, output diode, and output capacitors
near the input capacitors and near LX and PGND.
The high-current input loop goes from the positive
terminal of the input capacitor to the inductor, to the
IC’s LX pin, out PGND, and to the input capacitor
negative terminal. The high-current output loop is
from the positive terminal of the input capacitor to
the inductor, to the catch diode (D1), to the positive
terminal of the output capacitors, reconnecting
between the output capacitor and input capacitor
ground terminals. Connect these loop components
together with short, wide connections. Avoid using
vias in the high-current paths. If vias are unavoidable, use many vias in parallel to reduce resistance
and inductance.
2) Create a power ground island (PGND) consisting of
the input and output capacitor grounds, PGND pin,
and the SRC bypass capacitor and other chargepump components. Connect all of these together
with short, wide traces or a small ground plane.
Maximizing the width of the power ground traces
improves efficiency and reduces output voltage ripple and noise spikes.
Create an analog ground island (AGND) consisting
of the AGND pin, FB divider, the operation amplifier
dividers, the COMP and DEL capacitor ground connections, and the device’s exposed backside pad.
Connect the AGND and PGND islands by connecting the PGND pin directly to the exposed backside
pad. Make no other connections between these
separate ground planes.
3) Place the feedback voltage-divider resistors close to
FB. The divider’s center trace should be kept short.
Placing the resistors far away causes their FB traces
to become antennas that can pick up switching
noise. Avoid running the feedback trace near LX or
the switching nodes in the charge pumps.
4) Minimize the length and maximize the width of the
traces between the output capacitors and the load
for best transient response.
5) Minimize the size of the LX node while keeping it
wide and short. Keep the LX node away from the
feedback node (FB) and analog ground. Use DC
traces as shields if necessary.
Refer to the MAX1543 Evaluation Kit for an example of
proper board layout.
______________________________________________________________________________________
TFT LCD DC-to-DC Converter with
Operational Amplifiers
N.C.
CTL
DEL
COMP
FB
20
19
18
17
16
TOP VIEW
COM
1
15
N.C.
SRC
2
14
IN
N.C.
3
13
LX
PGND
4
12
SUP
AGND
5
11
POS2
7
8
9
10
NEG1
OUT1
OUT2
NEG2
POS1
6
MAX1542
Chip Information
TRANSISTOR COUNT: 2508
PROCESS: BiCMOS
THIN QFN
______________________________________________________________________________________
19
MAX1542/MAX1543
Pin Configurations (continued)
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
D2
0.15 C A
D
b
CL
0.10 M C A B
D2/2
D/2
PIN # 1
I.D.
QFN THIN.EPS
MAX1542/MAX1543
TFT LCD DC-to-DC Converter with
Operational Amplifiers
k
0.15 C B
PIN # 1 I.D.
0.35x45
E/2
E2/2
CL
(NE-1) X e
E
E2
k
L
DETAIL A
e
(ND-1) X e
CL
CL
L
L
e
e
0.10 C
A
C
0.08 C
A1 A3
PROPRIETARY INFORMATION
TITLE:
PACKAGE OUTLINE
16, 20, 28, 32L, QFN THIN, 5x5x0.8 mm
APPROVAL
COMMON DIMENSIONS
DOCUMENT CONTROL NO.
REV.
21-0140
C
1
2
EXPOSED PAD VARIATIONS
NOTES:
1. DIMENSIONING & TOLERANCING CONFORM TO ASME Y14.5M-1994.
2. ALL DIMENSIONS ARE IN MILLIMETERS. ANGLES ARE IN DEGREES.
3. N IS THE TOTAL NUMBER OF TERMINALS.
4. THE TERMINAL #1 IDENTIFIER AND TERMINAL NUMBERING CONVENTION SHALL CONFORM TO JESD 95-1
SPP-012. DETAILS OF TERMINAL #1 IDENTIFIER ARE OPTIONAL, BUT MUST BE LOCATED WITHIN THE
ZONE INDICATED. THE TERMINAL #1 IDENTIFIER MAY BE EITHER A MOLD OR MARKED FEATURE.
5. DIMENSION b APPLIES TO METALLIZED TERMINAL AND IS MEASURED BETWEEN 0.25 mm AND 0.30 mm
FROM TERMINAL TIP.
6. ND AND NE REFER TO THE NUMBER OF TERMINALS ON EACH D AND E SIDE RESPECTIVELY.
7. DEPOPULATION IS POSSIBLE IN A SYMMETRICAL FASHION.
8. COPLANARITY APPLIES TO THE EXPOSED HEAT SINK SLUG AS WELL AS THE TERMINALS.
9. DRAWING CONFORMS TO JEDEC MO220.
10. WARPAGE SHALL NOT EXCEED 0.10 mm.
PROPRIETARY INFORMATION
TITLE:
PACKAGE OUTLINE
16, 20, 28, 32L, QFN THIN, 5x5x0.8 mm
APPROVAL
DOCUMENT CONTROL NO.
REV.
21-0140
C
2
2
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2003 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
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