EVALUATION KIT AVAILABLE
MAX16833/MAX16833B/C/D/G
General Description
The MAX16833, MAX16833B, MAX16833C, MAX16833D,
and MAX16833G are peak current-mode-controlled LED
drivers for boost, buck-boost, SEPIC, flyback, and highside buck topologies. A dimming driver designed to drive
an external p-channel in series with the LED string provides wide-range dimming control. This feature provides
extremely fast PWM current switching to the LEDs with
no transient overvoltage or undervoltage conditions.
In addition to PWM dimming, the ICs provide analog
dimming using a DC input at ICTRL. The ICs sense the
LED current at the high side of the LED string.
A single resistor from RT/SYNC to ground sets the
switching frequency from 100kHz to 1MHz, while an
external clock signal capacitively coupled to RT/SYNC
allows the ICs to synchronize to an external clock. In the
MAX16833/C/G, the switching frequency can be dithered
for spread-spectrum applications. The MAX16833B/D
instead provide a 1.64V reference voltage with a 2%
tolerance.
The ICs operate over a wide 5V to 65V supply range
and include a 3A sink/source gate driver for driving a
power MOSFET in high-power LED driver applications.
Additional features include a fault-indicator output (FLT)
for short or overtemperature conditions and an overvoltage-protection sense input (OVP) for overvoltage protection. High-side current sensing combined with a p-channel
dimming MOSFET allow the positive terminal of the LED
string to be shorted to the positive input terminal or to
the negative input terminal without any damage. This is a
unique feature of the ICs.
Ordering Information appears at end of data sheet.
Applications
●● Automotive Exterior Lighting:
High-Beam/Low-Beam/Signal/Position Lights
Daytime Running Lights (DRLs)
Fog Light and Adaptive Front Light Assemblies
●● Commercial, Industrial, and Architectural Lighting
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Benefits and Features
●● Integration Minimizes BOM for High-Brightness LED
Driver with a Wide Input Range Saving Space and
Cost
• +5V to +65V Wide Input Voltage Range with a
Maximum 65V Boost Output
• Integrated High-Side pMOS Dimming MOSFET
Driver (Allows Single-Wire Connection to LEDs)
• ICTRL Pin for Analog Dimming
• Integrated High-Side Current-Sense Amplifier
• Full-Scale, High-Side, Current-Sense Voltage of
200mV
●● Simple to Optimize for Efficiency, Board Space, and
Input Operating Range
• Boost, SEPIC, and Buck-Boost Single-Channel
LED Drivers
• 2% Accurate 1.64V Reference (MAX16833B/D)
• Programmable Operating Frequency (100kHz to
1MHz) with Synchronization Capability
• Frequency Dithering for Spread-Spectrum
Applications (MAX16833/C/G)
• Thermally Enhanced 5mm x 4.4mm, 16-Pin
TSSOP Package with Exposed Pad
●● Protection Features and Wide Temperature Range
Increase System Reliability
• Short-Circuit, Overvoltage, and Thermal Protection
• Fault-Indicator Output
• -40°C to +125°C Operating Temperature Range
Simplified Operating Circuit
6V TO 18V
WITH LOAD
DUMP UP
TO 70V
IN
NDRV
CS
OVP
ISENSE+
PWMDIM
ISENSEDIMOUT
MAX16833
PWMDIM
LED+
PGND
19-5187; Rev 11; 6/16
LED-
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Absolute Maximum Ratings
IN to PGND............................................................ -0.3V to +70V
ISENSE+, ISENSE-, DIMOUT to PGND................. -0.3V to +80V
DIMOUT to ISENSE+................................................-9V to +0.3V
ISENSE- to ISENSE+.............................................-0.6V to +0.3V
PGND to SGND.....................................................-0.3V to +0.3V
VCC to PGND...........................................................-0.3V to +9V
NDRV to PGND......................................... -0.3V to (VCC + 0.3V)
OVP, PWMDIM, COMP, LFRAMP, REF, ICTRL,
RT/SYNC, FLT to SGND....................................-0.3V to +6.0V
CS to PGND..........................................................-0.3V to +6.0V
Continuous Current on IN.................................................100mA
Peak Current on NDRV......................................................... Q3A
Continuous Current on NDRV........................................ Q100mA
Short-Circuit Duration on VCC....................................Continuous
Continuous Power Dissipation (TA = +70NC)
16-Pin TSSOP (derate 26.1mW/NC above +70NC) ......2089mW
Operating Temperature Range ....................... -40NC to +125NC
Junction Temperature......................................................+150NC
Storage Temperature Range............................. -65NC to +150NC
Lead Temperature (soldering, 10s).................................+300NC
Soldering Temperature (reflow).......................................+260NC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
Package Thermal Characteristics (Note 1)
16 TSSOP
Junction-to-Ambient Thermal Resistance (qJA)........38.3°C/W
Junction-to-Case Thermal Resistance (qJC)..................3°C/W
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer
board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Electrical Characteristics
(VIN = 12V, RRT = 12.4kI, CIN = CVCC = 1µF, CLFRAMP/CREF = 0.1µF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40NC to +125NC, unless otherwise
noted. Typical values are at TA = +25NC.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
65
V
PWMDIM = 0, no switching
1.5
2.5
Switching
2.5
4
SYSTEM SPECIFICATIONS
Operational Supply Voltage
Supply Current
Undervoltage Lockout (UVLO)
VIN
IINQ
5
UVLORIN
VIN rising
4.2
4.55
4.85
UVLOFIN
VIN falling, IVCC = 35mA
4.05
4.3
4.65
UVLO Hysteresis
Startup Delay
UVLO Falling Delay
tSTART_DELAY During power-up
tFALL_DELAY During power-down
mA
V
250
mV
410
Fs
3.3
Fs
VCC LDO REGULATOR
Regulator Output Voltage
VCC
0.1mA P IVCC P 50mA, 9V P VIN P 14V
Dropout Voltage
VDOVCC
14V P VIN P 65V, IVCC = 10mA
IVCC = 50mA, VIN = 5V
Short-Circuit Current
IMAXVCC
VCC = 0V, VIN = 5V
6.75
55
6.95
7.15
V
0.15
0.35
V
100
150
mA
1000
kHz
OSCILLATOR (RT/SYNC)
Switching Frequency Range
fSW
Bias Voltage at RT/SYNC
VRT
100
1
VCS = 0V; MAX16833/MAX16833B only
Maximum Duty Cycle
www.maximintegrated.com
DMAX
VCS = 0V; MAX16833C/MAX16833D/
MAX16833G only
V
87.5
88.5
89.5
93
94
95
%
Maxim Integrated │ 2
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Electrical Characteristics (continued)
(VIN = 12V, RRT = 12.4kI, CIN = CVCC = 1µF, CLFRAMP/CREF = 0.1µF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40NC to +125NC, unless otherwise
noted. Typical values are at TA = +25NC.) (Note 2)
Oscillator Frequency Accuracy
-5
PARAMETER
SYMBOL
Synchronization Logic-High Input
VIH-SYNC
Synchronization Frequency Range
SLOPE COMPENSATION
fSYNCIN
Slope Compensation
Current-Ramp Height
ISLOPE
CONDITIONS
VRT rising
MIN
TYP
%
UNITS
3.8
V
1.1fSW
Ramp peak current added to CS input
per switching cycle
+5
MAX
1.7fSW
46
50
54
FA
DITHERING RAMP GENERATOR (LFRAMP) (MAX16833/MAX16833C/MAX16833G only)
Charging Current
VLFRAMP = 0V
80
100
120
FA
Discharging Current
VLFRAMP = 2.2V
80
100
120
FA
Comparator High Trip Threshold
2
V
Comparator Low Trip Threshold
VRT
V
REFERENCE OUTPUT (REF) (MAX16833B/MAX16833D only)
Reference Output Voltage
VREF
IREF = 0 to 80FA
1.604
1.636
1.669
V
0
35
200
nA
200
400
700
FA
ANALOG DIMMING (ICTRL)
Input-Bias Current
IBICTRL
VICTRL = 0.62V
LED CURRENT-SENSE AMPLIFIER
ISENSE+ Input-Bias Current
ISENSE+ Input-Bias Current with
DIM Low
ISENSE- Input-Bias Current
IBISENSE+
VISENSE+ = 65V, VISENSE- = 64.8V
IBISENSE+OFF
VISENSE+ = 48V, VISENSE- = 48V,
PWMDIM = 0
IBISENSE-
VISENSE+ = 65V, VISENSE- = 64.8V
200
2
Voltage Gain
VSENSE
BW
195
VICTRL = 0.616V
VICTRL = 0.2465V
Bandwidth
8
6.15
VICTRL = 1.4V
Current-Sense Voltage
5
FA
199
V/V
203
100
38.4
AVDC - 3dB
40
FA
mV
41.4
5
MHz
COMP
Transconductance
GMCOMP
2100
Open-Loop DC Gain
AVOTA
COMP Input Leakage
ILCOMP
-300
ISINK
100
ISOURCE
100
COMP Sink Current
COMP Source Current
3500
4900
75
FS
dB
+300
nA
400
700
FA
400
700
FA
PWM COMPARATOR
Input Offset Voltage
VOS-PWM
Leading-Edge Blanking
Propagation Delay to NDRV
tON(MIN)
Includes leading-edge blanking time with
10mV overdrive
2
V
50
ns
55
80
110
ns
406
418
430
mV
CS LIMIT COMPARATOR
Current-Limit Threshold
CS Limit-Comparator
Propagation Delay to NDRV
www.maximintegrated.com
VCS_LIMIT
tCS_PROP
10mV overdrive (excluding leading-edge
blanking time)
30
ns
Maxim Integrated │ 3
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Electrical Characteristics (continued)
(VIN = 12V, RRT = 12.4kI, CIN = CVCC = 1µF, CLFRAMP/CREF = 0.1µF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40NC to +125NC, unless otherwise
noted. Typical values are at TA = +25NC.) (Note 2)
Leading-Edge Blanking
PARAMETER
50
SYMBOL
CONDITIONS
MIN
TYP
ns
MAX
UNITS
GATE DRIVER (NDRV)
Peak Pullup Current
INDRVPU
VCC = 7V, VNDRV = 0V
3
Peak Pulldown Current
INDRVPD
VCC = 7V, VNDRV = 7V
3
A
A
Rise Time
tr
CNDRV = 10nF
30
ns
Fall Time
tf
CNDRV = 10nF
30
ns
I
RDSON Pulldown nMOS
RNDRVON
VCOMP = 0V, ISINK = 100mA
0.25
0.6
1.1
1.19
1.225
1.26
PWM DIMMING (PWMDIM)
ON Threshold
VPWMON
Hysteresis
VPWMHY
Pullup Resistance
RPWMPU
70
1.7
3
V
mV
4.5
MI
PWMDIM to LED Turn-Off Time
PWMDIM falling edge to rising edge on
DIMOUT, CDIMOUT = 7nF
2
Fs
PWMDIM to LED Turn-On Time
PWMDIM rising edge to falling edge on
DIMOUT, CDIMOUT = 7nF
3
Fs
pMOS GATE DRIVER (DIMOUT)
Peak Pullup Current
IDIMOUTPU
VPWMDIM = 0V,
VISENSE+ - VDIMOUT = 7V
25
50
80
mA
Peak Pulldown Current
IDIMOUTPD
VISENSE+ - VDIMOUT = 0V
10
25
45
mA
-8.7
-7.4
-6.3
V
1.19
1.225
1.26
DIMOUT Low Voltage with
Respect to VISENSE+
OVERVOLTAGE PROTECTION (OVP)
Threshold
VOVPOFF
Hysteresis
VOVPHY
Input Leakage
ILOVP
VOVP rising
70
VOVP = 1.235V
-300
V
mV
+300
nA
310
mV
SHORT-CIRCUIT HICCUP MODE (not present in the MAX16833G)
Short-Circuit Threshold
Hiccup Time
VSHORT-HIC (VISENSE+ - VISENSE-) rising
285
tHICCUP
Delay in Short-Circuit Hiccup
Activation
298
8192
Clock
Cycles
1
Fs
BUCK-BOOST SHORT-CIRCUIT DETECT
Buck-Boost Short-Circuit
Threshold
Delay in FLT Assertion from
Buck-Boost Short-Circuit
Condition
Delay in FLT Deassertion After
Buck-Boost Short Circuit is
Removed (Consecutive ClockCycle Count)
www.maximintegrated.com
VSHORT-BB
tDEL-BB-SHRT
(VISENSE+ - VIN) falling, VIN = 12V
1.15
1.55
1.9
V
Counter increments only when
VPWMDIM > VPWMON
8192
Clock
Cycles
Counter increments only when
VPWMDIM > VPWMON
8192
Clock
Cycles
Maxim Integrated │ 4
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Electrical Characteristics (continued)
(VIN = 12V, RRT = 12.4kI, CIN = CVCC = 1µF, CLFRAMP/CREF = 0.1µF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40NC to +125NC, unless otherwise
noted. Typical values are at TA = +25NC.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
40
200
mV
1
FA
OPEN-DRAIN FAULT (FLT)
Output Voltage Low
VOL-FLT
VIN = 4.75V, VOVP = 2V, and ISINK = 5mA
Output Leakage Current
VFLT = 5V
THERMAL SHUTDOWN
Thermal-Shutdown Temperature
Temperature rising
Thermal-Shutdown Hysteresis
+160
NC
10
NC
Note 2: All devices are 100% tested at TA = +25NC. Limits over temperature are guaranteed by design.
Typical Operating Characteristics
(VIN = +12V, CVIN = CVCC = 1FF, CLFRAMP/CREF = 0.1FF, TA = +25NC, unless otherwise noted.)
4.6
4.5
4.4
VPWMDIM = 0V
VIN FALLING
4.3
4.2
3
2
1
10
35
60
85
110 125
1.0
-15
10
35
60
85
1
110 125
100
10
TEMPERATURE (°C)
VIN (V)
VCC vs. IVCC
VCC vs. TEMPERATURE
DIMOUT (WITH RESPECT TO ISENSE+)
vs. TEMPERATURE
7.00
VCC (V)
6.90
6.85
6.95
6.90
6.85
6.80
6.80
6.75
10 15 20 25 30 35 40 45 50
IVCC (mA)
www.maximintegrated.com
MAX16833 toc06
7.05
-6.2
DIMOUT (WITH RESPECT TO ISENSE+) (V)
7.10
MAX16833 toc04
6.95
5
1.5
TEMPERATURE (°C)
7.00
0
VIN ~ 4.6V
2.0
0
-40
MAX16833 toc05
-15
VPWMDIM = 0V
0.5
0
-40
VCC (V)
2.5
QUIESCENT CURRENT (mA)
VIN RISING
QUIESCENT CURRENT vs. VIN
MAX16833 toc02
4.7
4
QUIESCENT CURRENT (mA)
MAX16833 toc01
IN RISING/FALLING UVLO THRESHOLD (V)
4.8
QUIESCENT CURRENT
vs. TEMPERATURE
MAX16833 toc03
IN RISING/FALLING UVLO THRESHOLD
vs. TEMPERATURE
-6.7
-7.2
-7.7
-8.2
-8.7
6.75
-40
-15
10
35
60
TEMPERATURE (°C)
85
110 125
-40
-15
10
35
60
85
110 125
TEMPERATURE (°C)
Maxim Integrated │ 5
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Typical Operating Characteristics (continued)
(VIN = +12V, CVIN = CVCC = 1FF, CLFRAMP/CREF = 0.1FF, TA = +25NC, unless otherwise noted.)
DIMOUT FALL TIME vs. TEMPERATURE
DIMOUT RISE TIME vs. TEMPERATURE
3.5
DIMOUT FALL TIME (µs)
2.2
2.0
1.8
1.6
1.4
MAX16833 toc08
MAX16833 toc07
2.4
DIMOUT RISE TIME (µs)
4.0
3.0
2.5
2.0
1.2
CDIMOUT = 6.8nF
CDIMOUT = 6.8nF
1.0
1.5
-40
-15
10
35
60
85
TEMPERATURE (°C)
110 125
-40
-15
10
VSENSE vs. TEMPERATURE
203
125
1.20
1.40
180
VSENSE (mV)
201
200
199
198
160
140
120
100
80
60
40
20
0
197
196
195
-40
-15
10
35
60
85
0
110 125
0.20
0.40
0.60
0.80
1.00
TEMPERATURE (°C)
VICTRL (V)
OSCILLATOR FREQUENCY vs. TEMPERATURE
(MAX16833/MAX16833B ONLY)
OSCILLATOR FREQUENCY
vs. 1/RRT CONDUCTANCE
(MAX16833/MAX16833B ONLY)
RRT = 24.9kI
308
306
304
302
300
298
296
294
292
1100
1000
OSCILLATOR FREQUENCY (kHz)
MAX16833 toc11
310
MAX16833 toc12
VSENSE (mV)
110
VSENSE vs. VICTRL
202
OSCILLATOR FREQUENCY (kHz)
85
MAX16833 toc10
204
60
240
220
200
MAX16833 toc09
205
35
TEMPERATURE (°C)
900
800
700
600
500
400
300
200
100
290
-40
-15
10
35
60
TEMPERATURE (°C)
www.maximintegrated.com
85
110 125
0
0.005
0.034
0.063
0.092
0.121
0.150
1/RRT (kI-1)
Maxim Integrated │ 6
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Typical Operating Characteristics (continued)
(VIN = +12V, CVIN = CVCC = 1FF, CLFRAMP/CREF = 0.1FF, TA = +25NC, unless otherwise noted.)
NDRV RISE/FALL TIME
vs. TEMPERATURE
600Hz DIMMING OPERATION
MAX16833 toc14
MAX16833 toc13
NDRV RISE/FALL TIME (ns)
60
50
VDIMOUT
50V/div
0V
NDRV FALL TIME
ILED
500mA/div
0mA
40
NDRV RISE TIME
30
0V
0V
CNDRV = 10nF
PWMDIM = 600Hz
20
-40
-15
10
35
60
85
110 125
VCOMP
2V/div
0V
VNDRV
10V/div
0V
400µs/div
TEMPERATURE (°C)
Pin Configuration
TOP VIEW
LFRAMP (REF) 1
RT/SYNC 2
SGND 3
ICTRL 4
COMP 5
+
16 IN
MAX16833
MAX16833B
MAX16833C
MAX16833D
MAX16833G
15 VCC
14 NDRV
13 PGND
12 CS
FLT 6
11 ISENSE+
PWMDIM 7
10 ISENSE-
OVP 8
*EP
9 DIMOUT
TSSOP
*EP = EXPOSED PAD.
( ) FOR MAX16833B/MAX16833D ONLY.
www.maximintegrated.com
Maxim Integrated │ 7
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Pin Description
PIN
NAME
1
LFRAMP
(MAX16833/
MAX16833C/
MAX16833G)
FUNCTION
Low-Frequency Ramp Output. Connect a capacitor from LFRAMP to ground to program the ramp
frequency, or connect to SGND if not used. A resistor can be connected between LFRAMP and
RT/SYNC to dither the PWM switching frequency to achieve spread spectrum.
REF
1.64V Reference Output. Connect a 1FF ceramic capacitor from REF to SGND to provide a stable
(MAX16833B/
reference voltage. Connect a resistive divider from REF to ICTRL for analog dimming.
MAX16833D)
PWM Switching Frequency Programming Input. Connect a resistor (RRT) from RT/SYNC to SGND to
set the internal clock frequency. Frequency = (7.350 x 109)/RRT for the MAX16833/B. Frequency =
(6.929 x109)/RRT for the MAX16833C/D/G. An external pulse can be applied to RT/SYNC through a
coupling capacitor to synchronize the internal clock to the external pulse frequency. The parasitic
capacitance on RT/SYNC should be minimized.
2
RT/SYNC
3
SGND
Signal Ground
4
ICTRL
Analog Dimming-Control Input. The voltage at ICTRL sets the LED current level when VICTRL < 1.2V.
For VICTRL > 1.4V, the internal reference sets the LED current.
5
COMP
Compensation Network Connection. For proper compensation, connect a suitable RC network from
COMP to ground.
6
FLT
7
PWMDIM
PWM Dimming Input. When PWMDIM is pulled low, DIMOUT is pulled high and PWM switching is
disabled. PWMDIM has an internal pullup resistor, defaulting to a high state when left unconnected.
8
OVP
LED String Overvoltage-Protection Input. Connect a resistive divider between ISENSE+, OVP, and
SGND. When the voltage on OVP exceeds 1.23V, a fast-acting comparator immediately stops PWM
switching. This comparator has a hysteresis of 70mV.
9
DIMOUT
Active-Low External Dimming p-Channel MOSFET Gate Driver
10
ISENSE-
Negative LED Current-Sense Input. A 100I resistor is recommended to be connected between
ISENSE- and the negative terminal of the LED current-sense resistor. This preserves the absolute
maximum rating of the ISENSE- pin during LED short circuit.
11
ISENSE+
Positive LED Current-Sense Input. The voltage between ISENSE+ and ISENSE- is proportionally
regulated to the lesser of VICTRL or 1.23V.
12
CS
Switching Regulator Current-Sense Input. Add a resistor from CS to switching MOSFET currentsense resistor terminal for programming slope compensation.
13
PGND
Power Ground
14
NDRV
External n-channel MOSFET Gate-Driver Output
15
VCC
16
IN
Positive Power-Supply Input. Bypass IN to PGND with at least a 1FF ceramic capacitor.
—
EP
Exposed Pad. Connect EP to the ground plane for heat sinking. Do not use EP as the only electrical
connection to ground.
www.maximintegrated.com
Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (FLT) section.
7V Low-Dropout Voltage Regulator Output. Bypass VCC to PGND with a 1FF (min) ceramic capacitor.
Maxim Integrated │ 8
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
MAX16833/MAX16833C Functional Diagram
IN
VCC
UVLO
5V REG
BG
VCC
7V LDO
5.7V
LVSH
5V
UVLO
THERMAL
SHUTDOWN
VBG
NDRV
5V
TSHDN
200kI
PGND
RT/
SYNC
RESET
DOMINANT
RT OSCILLATOR
S
Q
R
SLOPE
COMPENSATION
CS/PWM
BLANKING
CS
MAX
DUTY CYCLE
2V
1.64V (80µA)
REFERENCE
PWM
COMP
0.42V
REF
MAX16833
MAX16833C
VBG
ICTRL
MIN
OUT
LPF
ISENSE+
GM
COMP
6.15
ISENSE-
SYNC
ISENSE+
3.3V
DIMOUT
3MI
PWMDIM
BUCK-BOOST
SHORT DETECTION
FLT
VBG
1µs DELAY
OVP
6.15 x 0.3V
VISENSE+ - 7V
8192 x tOSC
HICCUP TIMER
S
R
Q
TSHDN
SGND
VBG
www.maximintegrated.com
Maxim Integrated │ 9
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
MAX16833B/MAX16833D Functional Diagram
IN
VCC
UVLO
5V REG
BG
VCC
7V LDO
5.7V
LVSH
5V
UVLO
THERMAL
SHUTDOWN
VBG
NDRV
5V
TSHDN
200kI
PGND
RT/
SYNC
RESET
DOMINANT
RT OSCILLATOR
S
Q
R
SLOPE
COMPENSATION
CS/PWM
BLANKING
CS
MAX
DUTY CYCLE
2V
1.64V (80µA)
REFERENCE
PWM
COMP
0.42V
REF
MAX16833B
MAX16833D
VBG
ICTRL
MIN
OUT
LPF
ISENSE+
GM
COMP
6.15
ISENSE-
SYNC
ISENSE+
3.3V
DIMOUT
3MI
PWMDIM
BUCK-BOOST
SHORT DETECTION
FLT
VBG
1µs DELAY
OVP
6.15 x 0.3V
VISENSE+ - 7V
8192 x tOSC
HICCUP TIMER
S
R
Q
TSHDN
SGND
VBG
www.maximintegrated.com
Maxim Integrated │ 10
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
MAX16833G Functional Diagram
IN
VCC
UVLO
5V REG
BG
VCC
7V LDO
5.7V
LVSH
5V
UVLO
THERMAL
SHUTDOWN
VBG
TSHDN
NDRV
5V
200kI
PGND
RT/
SYNC
RESET
DOMINANT
RT OSCILLATOR
S
Q
R
SLOPE
COMPENSATION
CS/PWM
BLANKING
CS
MAX
DUTY CYCLE
2V
RAMP
GENERATION
PWM
COMP
0.42V
LFRAMP
MAX16833G
VBG
ICTRL
MIN
OUT
LPF
ISENSE+
GM
COMP
6.15
ISENSE-
SYNC
ISENSE+
3.3V
DIMOUT
3MI
PWMDIM
VISENSE+ - 7V
BUCK-BOOST
SHORT DETECTION
FLT
VBG
TSHDN
SGND
OVP
VBG
www.maximintegrated.com
Maxim Integrated │ 11
MAX16833/MAX16833B/C/D/G
Detailed Description
The MAX16833, MAX16833B, MAX16833C, MAX16833D,
and MAX16833G are peak current-mode-controlled LED
drivers for boost, buck-boost, SEPIC, flyback, and highside buck topologies. A low-side gate driver capable of
sinking and sourcing 3A can drive a power MOSFET
in the 100kHz to 1MHz frequency range. Constantfrequency peak current-mode control is used to control
the duty cycle of the PWM controller that drives the power
MOSFET. Externally programmable slope compensation
prevents subharmonic oscillations for duty cycles exceeding 50% when the inductor is operating in continuous conduction mode. Most of the power for the internal control
circuitry inside the ICs is provided from an internal 5V regulator. The gate drive for the low-side switching MOSFET
is provided by a separate VCC regulator. A dimming driver
designed to drive an external p-channel in series with the
LED string provides wide-range dimming control. This
dimming driver is powered by a separate unconnected
reference -7V regulator. This feature provides extremely
fast PWM current switching to the LEDs with no transient
overvoltage or undervoltage conditions. In addition to
PWM dimming, the ICs provide analog dimming using a
DC input at the ICTRL input.
A single resistor from RT/SYNC to ground sets the
switching frequency from 100kHz to 1MHz, while an
external clock signal capacitively coupled to RT/SYNC
allows the ICs to synchronize to an external clock. The
switching frequency can be dithered for spread-spectrum
applications by connecting the LFRAMP output to RT/
SYNC through an external resistor in the MAX16833/C/G.
In the MAX16833B/D, the LFRAMP output is replaced by a
REF output, which provides a regulated 1.64V, 2% accurate reference that can be used with a resistive divider
from REF to ICTRL to set the LED current. The maximum
current from the REF output cannot exceed 80FA.
Additional features include a fault-indicator output (FLT)
for short, overvoltage, or overtemperature conditions and
an overvoltage-protection (OVP) sense input for overvoltage protection. In case of LED string short, for a buckboost configuration, the short-circuit current is equal
to the programmed LED current. In the case of boost
configuration, the ICs enter hiccup mode with automatic
recovery from short circuit. In the MAX16833G, the hiccup
mode is disabled. The MAX16833G should not be used in
boost applications.
www.maximintegrated.com
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
UVLO
The ICs feature undervoltage lockout (UVLO) using the
positive power-supply input (IN). The ICs are enabled
when VIN exceeds the 4.6V (typ) threshold and are disabled when VIN drops below the 4.35V (typ) threshold.
The UVLO is internally fixed and cannot be adjusted.
There is a startup delay of 300µs (typ) + 64 switching
clock cycles on power-up after the UVLO threshold is
crossed. There is a 3.3Fs delay on power-down on the
falling edge of the UVLO.
Dimming MOSFET Driver (DIMOUT)
The ICs require an external p-channel MOSFET for PWM
dimming. For normal operation, connect the gate of the
MOSFET to the output of the dimming driver (DIMOUT).
The dimming driver can sink up to 25mA or source up to
50mA of peak current for fast charging and discharging
of the p-MOSFET gate. When the PWMDIM signal is
high, this driver pulls the p-MOSFET gate to 7V below
the ISENSE+ pin to completely turn on the p-channel
dimming MOSFET.
n-Channel MOSFET Switch Driver (NDRV)
The ICs drive an external n-channel switching MOSFET.
NDRV swings between VCC and PGND. NDRV can sink/
source 3A of peak current, allowing the ICs to switch
MOSFETs in high-power applications. The average current demanded from the supply to drive the external
MOSFET depends on the total gate charge (QG) and the
operating frequency of the converter, fSW. Use the following equation to calculate the driver supply current INDRV
required for the switching MOSFET:
INDRV = QG x fSW
Pulse-Dimming Input (PWMDIM)
The ICs offer a dimming input (PWMDIM) for pulse-width
modulating the output current. PWM dimming can be
achieved by driving PWMDIM with a pulsating voltage
source. When the voltage at PWMDIM is greater than
1.23V, the PWM dimming p-channel MOSFET turns on
and the gate drive to the n-channel switching MOSFET is
also enabled. When the voltage on PWMDIM drops 70mV
below 1.23V, the PWM dimming MOSFET turns off and
the n-channel switching MOSFET is also turned off. The
COMP capacitor is also disconnected from the internal
transconductance amplifier when PWMDIM is low. When
left unconnected, a weak internal pullup resistor sets this
input to logic-high.
Maxim Integrated │ 12
MAX16833/MAX16833B/C/D/G
Analog Dimming (ICTRL)
The ICs offer an analog dimming control input (ICTRL).
The voltage at ICTRL sets the LED current level when
VICTRL < 1.2V. The LED current can be linearly adjusted
from zero with the voltage on ICTRL. For VICTRL > 1.4V,
an internal reference sets the LED current. The maximum
withstand voltage of this input is 5.5V.
Low-Side Linear Regulator (VCC)
The ICs feature a 7V low-side linear regulator (VCC).
VCC powers up the switching MOSFET driver with
sourcing capability of up to 50mA. Use a 1FF (min) lowESR ceramic capacitor from VCC to PGND for stable
operation. The VCC regulator goes below 7V if the input
voltage falls below 7V. The dropout voltage for this
regulator at 50mA is 0.2V. This means that for an input voltage of 5V, the VCC voltage is 4.8V. The short-circuit current
on the VCC regulator is 100mA (typ). Connect VCC to IN if
VIN is always less than 7V.
LED Current-Sense Inputs (ISENSE±)
The differential voltage from ISENSE+ to ISENSE- is
fed to an internal current-sense amplifier. This amplified signal is then connected to the negative input of the
transconductance error amplifier. The voltage-gain factor
of this amplifier is 6.15.
The offset voltage for this amplifier is P 1mV.
Internal Transconductance Error Amplifier
The ICs have a built-in transconductance amplifier used
to amplify the error signal inside the feedback loop.
When the dimming signal is low, COMP is disconnected
from the output of the error amplifier and DIMOUT goes
high. When the dimming signal is high, the output of
the error amplifier is connected to COMP and DIMOUT
goes low. This enables the compensation capacitor to
hold the charge when the dimming signal has turned off
the internal switching MOSFET gate drive. To maintain
the charge on the compensation capacitor CCOMP (C4
in the Typical Operating Circuits), the capacitor should
be a low-leakage ceramic type. When the internal dimming signal is enabled, the voltage on the compensation
capacitor forces the converter into steady state almost
instantaneously.
www.maximintegrated.com
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Internal Oscillator (RT/SYNC)
The internal oscillators of the ICs are programmable from
100kHz to 1MHz using a single resistor at RT/SYNC. Use
the following formula to calculate the switching frequency:
fOSC (kHz) =
fOSC (kHz) =
7350 (kΩ)
R RT (kΩ)
6929(kΩ)
R RT (kΩ)
for the MAX16833 B
for the MAX16833C D/G
where RRT is the resistor from RT/SYNC to SGND.
Synchronize the oscillator with an external clock by
AC-coupling the external clock to the RT/SYNC input.
For fOSC between 200kHz and 1MHz, the capacitor used
for the AC-coupling should satisfy the following relation:
C SYNC ≤
9.8624 × 10 -6
− 0.144 × 10 -9 farads
R RT
where RRT is in kω. For fOSC below 200GHz, CSYNC ≤
268nF.
The pulse width for the synchronization pulse should satisfy the following relations:
t PW 0.5
t PW 1.05 × t CLK
<
< 1and
t CLK VS
t CLK
t OSC
t
3.4V < 0.8 - PW VS + VS < 5V
t CLK
where tPW is the synchronization source pulse width,
tCLK is the synchronization clock time period, tOSC is
the free-running oscillator time period, and VS is the
synchronization pulse-voltage level.
Ensure that the external clock signal frequency is at least
1.1 x fOSC, where fOSC is the oscillator frequency set
by RRT. A typical pulse width of 200ns can be used for
proper synchronization of a frequency up to 250kHz. A
rising external clock edge (sync) is interpreted as a synchronization input. If the sync signal is lost, the internal
oscillator takes control of the switching rate returning the
switching frequency to that set by RRT. This maintains
output regulation even with intermittent sync signals.
Maxim Integrated │ 13
MAX16833/MAX16833B/C/D/G
SYNC
C1
680pF
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Voltage-Reference Output (REF/MAX16833B/
MAX16833D)
C2
1000pF
RT PIN
D2
SD103AWS
R2
22I
GND
RRT
24.9I
GND
Figure 1. SYNC Circuit
Figure 1 shows the frequency-synchronization circuit
suitable for applications where a 5V amplitude pulse with
20% to 80% duty cycle is available as the synchronization
source. This circuit can be used for SYNC frequencies
in the 100kHz to 1MHz range. C1 and R2 act as a differentiator that reduces the input pulse width to suit the
ICs’ RT/SYNC input. D2 bypasses the negative current
through C1 at the falling edge of the SYNC source to limit
the minimum voltage at the RT/SYNC pin. The differentiator output is AC-coupled to the RT/SYNC pin through C2.
The output impedance of the SYNC source should be low
enough to drive the current through R2 on the rising edge.
The rise/fall times of the SYNC source should be less than
50ns to avoid excessive voltage drop across C1 during
the rise time. The amplitude of the SYNC source can be
between 4V and 5V. If the SYNC source amplitude is 5V
and the rise time is less than 20ns, then the maximum
peak voltage at RT/SYNC pin can get close to 6V. Under
such conditions, it is desirable to use a resistor in series
with C1 to reduce the maximum voltage at the RT/SYNC
pin. For proper synchronization, the peak SYNC pulse
voltage at RT/SYNC pin should exceed 3.8V.
Frequency Dithering (LFRAMP/MAX16833/
MAX16833C/MAX16833G)
The MAX16833/MAX16833C/MAX16833G feature a lowfrequency ramp output. Connect a capacitor from LFRAMP
to ground to program the ramp frequency. Connect to
SGND if not used. A resistor can be connected between
LFRAMP and RT/SYNC to dither the PWM switching frequency to achieve spread spectrum. A lower value resistor provides a larger amount of frequency dithering. The
LFRAMP voltage is a triangular waveform between 1V
(typ) and 2V (typ). The ramp frequency is given by:
fLFRAMP (Hz) =
www.maximintegrated.com
The MAX16833B/D have a 2% accurate 1.64V reference voltage on the REF output. Connect a 1FF ceramic
capacitor from REF to SGND to provide a stable reference voltage. This reference can supply up to 80µA. This
output can drive a resistive divider to the ICTRL input
for analog dimming. The resistance from REF to ground
should be greater than 20.5kI.
Switching MOSFET Current-Sense Input (CS)
CS is part of the current-mode control loop. The switching control uses the voltage on CS, set by RCS (R4 in the
Typical Operating Circuits) and RSLOPE (R1 in the Typical
Operating Circuits), to terminate the on pulse width of the
switching cycle, thus achieving peak current-mode control.
Internal leading-edge blanking of 50ns is provided to prevent premature turn-off of the switching MOSFET in each
switching cycle. Resistor RCS is connected between the
source of the n-channel switching MOSFET and PGND.
During switching, a current ramp with a slope of 50FA
x fSW is sourced from the CS input. This current ramp,
along with resistor RSLOPE, programs the amount of slope
compensation.
Overvoltage-Protection Input (OVP)
OVP sets the overvoltage-threshold limit across the
LEDs. Use a resistive divider between ISENSE+ to OVP
and SGND to set the overvoltage-threshold limit. An internal overvoltage-protection comparator senses the differential voltage across OVP and SGND. If the differential
voltage is greater than 1.23V, NDRV goes low, DIMOUT
goes high, and FLT asserts. When the differential voltage
drops by 70mV, NDRV is enabled, DIMOUT goes low,
and FLT deasserts.
Fault Indicator (FLT)
The ICs feature an active-low, open-drain fault indicator
(FLT). FLT goes low when one of the following conditions
occur:
U Overvoltage across the LED string
U Short-circuit condition across the LED string
U Overtemperature condition
FLT goes high when the fault condition ends.
50FA
C LFRAMP (F)
Maxim Integrated │ 14
MAX16833/MAX16833B/C/D/G
Thermal Protection
The ICs feature thermal protection. When the junction
temperature exceeds +160NC, the ICs turn off the external
power MOSFETs by pulling the NDRV low and DIMOUT
high. External MOSFETs are enabled again after the
junction temperature has cooled by 10°C. This results in
a cycled output during continuous thermal-overload conditions. Thermal protection protects the ICs in the event of
fault conditions.
Short-Circuit Protection
Boost Configuration (MAX16833/B/C/D only)
In the boost configuration, if the LED string is shorted it
causes the (ISENSE+ to ISENSE-) voltage to exceed
300mV. If this condition occurs for R 1Fs, the ICs activates
the hiccup timer for 8192 clock cycles during which:
U NDRV goes low and DIMOUT goes high.
U The error amplifier is disconnected from COMP.
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Applications Information
Setting the Overvoltage Threshold
The overvoltage threshold is set by resistors R5 and R11
(see the Typical Operating Circuits). The overvoltage
circuit in the ICs is activated when the voltage on OVP
with respect to GND exceeds 1.23V. Use the following
equation to set the desired overvoltage threshold:
VOV = 1.23V (R5 + R11)/R11
Programming the LED Current
Normal sensing of the LED current should be done on
the high side where the LED current-sense resistor is
connected to the boost output. The other side of the LED
current-sense resistor goes to the source of the p-channel
dimming MOSFET if PWM dimming is desired. The LED
current is programmed using R7. When VICTRL > 1.23V,
the internal reference regulates the voltage across R7 to
200mV:
U FLT is pulled to SGND.
After the hiccup time has elapsed, the ICs retry. During
this retry period, FLT is latched and is reset only if there is
no short detected after 20Fs of retrying. The MAX16833G
does not have the hiccup protection and should not be
used for boost applications.
Buck-Boost Configuration
In the case of the buck-boost configuration, once an
LED string short occurs the behavior is different. The ICs
maintain the programmed current across the short. In this
case, the short is detected when the voltage between
ISENSE+ and IN falls below 1.5V. A buck-boost short
fault starts an up counter and FLT is asserted only after
the counter has reached 8192 clock cycles consecutively. If for any reason (VISENSE+ - VIN > 1.5V), the counter
starts down counting, resulting in FLT being deasserted
only after 8192 consecutive clock cycles of (VISENSE+ VIN > 1.5V) condition.
Exposed Pad
The ICs’ package features an exposed thermal pad on
its underside that should be used as a heatsink. This pad
lowers the package’s thermal resistance by providing
a direct heat-conduction path from the die to the PCB.
Connect the exposed pad and GND to the system ground
using a large pad or ground plane, or multiple vias to the
ground plane layer.
www.maximintegrated.com
ILED =
200mV
R7
The LED current can also be programmed using the voltage on ICTRL when VICTRL < 1.2V (analog dimming).
The voltage on ICTRL can be set using a resistive divider
from the REF output in the case of the MAX16833B/D.
The current is given by:
ILED =
VICTRL
R7 × 6.15
where:
V
× R8
VICTRL = REF
(R8 + R9)
where VREF is 1.64V and resistors R8 and R9 are in
ohms. At higher LED currents there can be noticeable
ripple on the voltage across R7. High-ripple voltages can
cause a noticeable difference between the programmed
value of the LED current and the measured value of the
LED current. To minimize this error, the ripple voltage
across R7 should be less than 40mV.
Maxim Integrated │ 15
MAX16833/MAX16833B/C/D/G
Inductor Selection
the input current plus the LED current. Calculate the
maximum duty cycle using the following equation:
Boost Configuration
In the boost converter (see the Typical Operating
Circuits), the average inductor current varies with the
line voltage. The maximum average current occurs at the
lowest line voltage. For the boost converter, the average
inductor current is equal to the input current. Calculate
maximum duty cycle using the following equation:
V
+ VD - VINMIN
D MAX = LED
VLED + VD - VFET
where VLED is the forward voltage of the LED string in
volts, VD is the forward drop of rectifier diode D1 in volts
(approximately 0.6V), VINMIN is the minimum input-supply
voltage in volts, and VFET is the average drain-to-source
voltage of the MOSFET Q1 in volts when it is on. Use an
approximate value of 0.2V initially to calculate DMAX. A
more accurate value of the maximum duty cycle can be
calculated once the power MOSFET is selected based on
the maximum inductor current.
Use the following equations to calculate the maximum average inductor current ILAVG, peak-to-peak
inductor current ripple DIL, and peak inductor current ILP
in amperes:
IL AVG =
ILED
1- D MAX
Allowing the peak-to-peak inductor ripple to be DIL, the
peak inductor current is given by:
=
IL P IL AVG +
∆IL
2
The inductance value (L) of inductor L1 in henries (H) is
calculated as:
-V
(V
) × D MAX
L = INMIN FET
fSW × ∆IL
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and DIL is in amperes.
Choose an inductor that has a minimum inductance
greater than the calculated value. The current rating of
the inductor should be higher than ILP at the operating
temperature.
Buck-Boost Configuration
In the buck-boost LED driver (see the Typical Operating
Circuits), the average inductor current is equal to
www.maximintegrated.com
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
D MAX =
VLED + VD
VLED + VD + VINMIN - VFET
where VLED is the forward voltage of the LED string in
volts, VD is the forward drop of rectifier diode D1 (approximately 0.6V) in volts, VINMIN is the minimum input supply
voltage in volts, and VFET is the average drain-to-source
voltage of the MOSFET Q1 in volts when it is on. Use
an approximate value of 0.2V initially to calculate DMAX.
A more accurate value of maximum duty cycle can be
calculated once the power MOSFET is selected based on
the maximum inductor current.
Use the equations below to calculate the maximum average inductor current ILAVG, peak-to-peak inductor current
ripple DIL, and peak inductor current ILP in amperes:
IL AVG =
ILED
1- D MAX
Allowing the peak-to-peak inductor ripple to be DIL:
=
IL P IL AVG +
∆IL
2
where ILP is the peak inductor current.
The inductance value (L) of inductor L1 in henries is
calculated as:
L=
(VINMIN - VFET ) × D MAX
fSW × ∆IL
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and DIL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value.
Peak Current-Sense Resistor (R4)
The value of the switch current-sense resistor R4 for the
boost and buck-boost configurations is calculated as follows:
=
R4
0.418V - VSC
Ω
IL P
where ILP is the peak inductor current in amperes and
VSC is the peak slope compensation voltage.
Maxim Integrated │ 16
MAX16833/MAX16833B/C/D/G
Slope Compensation
Slope compensation should be added to converters
with peak current-mode control operating in continuousconduction mode with more than 50% duty cycle to avoid
current-loop instability and subharmonic oscillations. The
minimum amount of slope compensation that is required
for stability is:
VSCMIN = 0.5 (inductor current downslope inductor current upslope) x R4
In the ICs, the slope-compensating ramp is added to the
current-sense signal before it is fed to the PWM comparator. Connect a resistor (R1) from CS to the inductor
current-sense resistor terminal to program the amount of
slope compensation.
The ICs generate a current ramp with a slope of 50FA/
tOSC for slope compensation. The current-ramp signal is
forced into the external resistor (R1) connected between
CS and the source of the external MOSFET, thereby
adding a programmable slope compensating voltage
(VSCOMP) at the current-sense input CS. Therefore:
dVSC/dt = (R1 x 50FA)/tOSC in V/s
The minimum value of the slope-compensation voltage
that needs to be added to the current-sense signal at
peak current and at minimum line voltage is:
(D
× (VLED - 2VINMIN ) × R4)
SC MIN = MAX
(V) Boost
2 × L MIN × fSW
SC MIN =
(D MAX × (VLED - VINMIN ) × R4)
(V)Buck-boost
2 × L MIN × fSW
where fSW is the switching frequency, DMAX is the maximum
duty cycle, which occurs at low line, VINMIN is the minimum
input voltage, and LMIN is the minimum value of the selected
inductor. For adequate margin, the slope-compensation voltage is multiplied by a factor of 1.5. Therefore, the actual
slope-compensation voltage is given by:
VSC = 1.5SCMIN
From the previous formulas, it is possible to calculate the
value of R4 as:
For boost configuration:
R4 =
0.418V
− 2V
V
INMIN
IL P + 0.75D MAX LED
L MINfSW
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
For buck-boost configuration:
R4 =
0.418V
− V
V
IL P + 0.75D MAX LED INMIN
L MINfSW
The minimum value of the slope-compensation resistor
(R1) that should be used to ensure stable operation at
minimum input supply voltage can be calculated as:
For boost configuration:
R1 =
(VLED − 2VINMIN ) × R4 × 1.5
2 × L MIN × fSW × 50µA
For buck-boost configuration :
R1 =
(VLED − VINMIN ) × R4 × 1.5
2 × L MIN × fSW × 50µA
where fSW is the switching frequency in hertz, VINMIN
is the minimum input voltage in volts, VLED is the LED
voltage in volts, DMAX is the maximum duty cycle, ILP
is the peak inductor current in amperes, and LMIN is the
minimum value of the selected inductor in henries.
Output Capacitor
The function of the output capacitor is to reduce the output ripple to acceptable levels. The ESR, ESL, and the
bulk capacitance of the output capacitor contribute to the
output ripple. In most applications, the output ESR and
ESL effects can be dramatically reduced by using lowESR ceramic capacitors. To reduce the ESL and ESR
effects, connect multiple ceramic capacitors in parallel
to achieve the required bulk capacitance. To minimize
audible noise generated by the ceramic capacitors during PWM dimming, it could be necessary to minimize
the number of ceramic capacitors on the output. In these
cases, an additional electrolytic or tantalum capacitor
provides most of the bulk capacitance.
Boost and Buck-Boost Configurations
The calculation of the output capacitance is the same for
both boost and buck-boost configurations. The output ripple is caused by the ESR and the bulk capacitance of the
output capacitor if the ESL effect is considered negligible.
For simplicity, assume that the contributions from ESR and
the bulk capacitance are equal, allowing 50% of the ripple
for the bulk capacitance. The capacitance is given by:
C OUT ≥
www.maximintegrated.com
ILED × 2 × D MAX
VOUTRIPPLE × fSW
Maxim Integrated │ 17
MAX16833/MAX16833B/C/D/G
where ILED is in amperes, COUT is in farads, fSW is in
hertz, and VOUTRIPPLE is in volts. The remaining 50%
of allowable ripple is for the ESR of the output capacitor.
Based on this, the ESR of the output capacitor is given by:
V
ESR COUT < OUTRIPPLE (Ω)
(IL P × 2)
where ILP is the peak-inductor current in amperes. Use
the equation below to calculate the RMS current rating of
the output capacitor:
I COUT(RMS) = IL AVG 2 D MAX (1 − D MAX )
Input Capacitor
The input-filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of
high-frequency current conducted to the input supply.
The ESR, ESL, and the bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter. For the boost
configuration, the input current is the same as the
inductor current. For buck-boost configuration, the input
current is the inductor current minus the LED current.
However, for both configurations, the ripple current that
the input filter capacitor has to supply is the same as the
inductor ripple current with the condition that the output
filter capacitor should be connected to ground for buckboost configuration. This reduces the size of the input
capacitor, as the input current is continuous with maximum QDIL/2. Neglecting the effect of LED current ripple,
the calculation of the input capacitor for boost, as well as
buck-boost configurations is the same.
Neglecting the effect of the ESL, the ESR, and the bulk
capacitance at the input contribute to the input-voltage
ripple. For simplicity, assume that the contributions
from the ESR and the bulk capacitance are equal. This
allows 50% of the ripple for the bulk capacitance. The
capacitance is given by:
∆IL
CIN ≥
4 × ∆VIN × fSW
where DIL is in amperes, CIN is in farads, fSW is in hertz,
and DVIN is in volts. The remaining 50% of allowable
ripple is for the ESR of the input capacitor. Based on this,
the ESR of the input capacitor is given by:
ESR CIN <
www.maximintegrated.com
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
where DIL is in amperes, ESRCIN is in ohms, and DVIN
is in volts. Use the equation below to calculate the RMS
current rating of the input capacitor:
I CIN(RMS) =
∆IL
2 3
Selection of Power Semiconductors
Switching MOSFET
The switching MOSFET (Q1) should have a voltage
rating sufficient to withstand the maximum output voltage
together with the diode drop of rectifier diode D1 and any
possible overshoot due to ringing caused by parasitic
inductances and capacitances. Use a MOSFET with a
drain-to-source voltage rating higher than that calculated
by the following equations.
Boost Configuration
VDS = (VLED + VD) x 1.2
where VDS is the drain-to-source voltage in volts and VD
is the forward drop of rectifier diode D1. The factor of 1.2
provides a 20% safety margin.
Buck-Boost Configuration
VDS = (VLED + VINMAX + VD) x 1.2
where VDS is the drain-to-source voltage in volts and VD
is the forward drop of rectifier diode D1. The factor of 1.2
provides a 20% safety margin.
The RMS current rating of the switching MOSFET Q1 is
calculated as follows for boost and buck-boost configurations:
IDRMS =
1.3 × ( (IL AVG) 2 × D MAX )
where IDRMS is the MOSFET Q1’s drain RMS current in
amperes.
The MOSFET Q1 dissipates power due to both switching losses, as well as conduction losses. The conduction
losses in the MOSFET are calculated as follows:
PCOND = (ILAVG)2 x DMAX x RDSON
where RDSON is the on-resistance of Q1 in ohms, PCOND
is in watts, and ILAVG is in amperes. Use the following
equations to calculate the switching losses in the MOSFET.
∆VIN
∆IL × 2
Maxim Integrated │ 18
MAX16833/MAX16833B/C/D/G
Boost Configuration
IL
× VLED 2 × C GD × fSW
PSW = AV G
2
1
1
×
+
IG ON IG OFF
Buck-Boost Configuration
IL
× (VLED + VINMAX ) 2 × C GD × fSW
PSW = AV G
2
1
1
×
+
IG ON IG OFF
where IGON and IGOFF are the gate currents of the
MOSFET Q1 in amperes when it is turned on and turned
off, respectively, VLED and VINMAX are in volts, ILAVG is
in amperes, fSW is in hertz, and CGD is the gate-to-drain
MOSFET capacitance in farads.
Rectifier Diode
Use a Schottky diode as the rectifier (D1) for fast switching and to reduce power dissipation. The selected
Schottky diode must have a voltage rating 20% above
the maximum converter output voltage. The maximum
converter output voltage is VLED in boost configuration
and VLED + VINMAX in buck-boost configuration.
The current rating of the diode should be greater than ID
in the following equation:
ID = ILAVG x (1 - DMAX) x 1.5
Dimming MOSFET
Select a dimming MOSFET (Q2) with continuous current
rating at the operating temperature higher than the LED
current by 30%. The drain-to-source voltage rating of the
dimming MOSFET must be higher than VLED by 20%.
Feedback Compensation
The LED current control loop comprising the switching
converter, the LED current amplifier, and the error amplifier should be compensated for stable control of the LED
current. The switching converter small-signal transfer
function has a right-half-plane (RHP) zero for both boost
and buck-boost configurations as the inductor current is
in continuous conduction mode. The RHP zero adds a
www.maximintegrated.com
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
20dB/decade gain together with a 90-degree phase lag,
which is difficult to compensate. The easiest way to avoid
this zero is to roll off the loop gain to 0dB at a frequency
less than 1/5 the RHP zero frequency with a -20dB/
decade slope.
The worst-case RHP zero frequency (fZRHP) is
calculated as follows:
Boost Configuration
fZRHP =
VLED × (1- D MAX ) 2
2π × L × ILED
Buck-Boost Configuration
VLED × (1- D MAX ) 2
2π × L × ILED × D MAX
fZRHP =
where fZRHP is in hertz, VLED is in volts, L is the
inductance value of L1 in henries, and ILED is in amperes.
The switching converter small-signal transfer function
also has an output pole for both boost and buck-boost
configurations. The effective output impedance that
determines the output pole frequency together with the
output filter capacitance is calculated as follows:
Boost Configuration
R OUT =
(R LED + R7) × VLED
(R LED + R7) × ILED + VLED
Buck-Boost Configuration
R OUT =
(R LED + R7) × VLED
(R LED + R7) × ILED × D MAX + VLED
where RLED is the dynamic impedance of the LED string
at the operating current in ohms, R7 is the LED currentsense resistor in ohms, VLED is in volts, and ILED is in
amperes.
The output pole frequency for both boost and buckboost configurations is calculated as below:
fP2 =
1
2π × C OUT × R OUT
where fP2 is in hertz, COUT is the output filter
capacitance in farads, and ROUT is the effective output
impedance in ohms calculated above.
Maxim Integrated │ 19
MAX16833/MAX16833B/C/D/G
The feedback loop compensation is done by connecting
resistor R10 and capacitor C4 in series from the COMP
pin to GND. R10 is chosen to set the high-frequency gain
of the integrator to set the crossover frequency at fZRHP/5
and C4 is chosen to set the integrator zero frequency
to maintain loop stability. For optimum performance,
choose the components using the following equations:
R10 =
2 × fZRHP × R4
FC × (1 − D MAX ) × R7 × 6.15 × GM COMP
The value of C4 can be calculated as below:
C4 =
25
π × R10 × fZRHP
where R10 is the compensation resistor in ohms, fZRHP
and fP2 are in hertz, R4 is the inductor current-sense
resistor in ohms, R7 is the LED current-sense resistor in
ohms, factor 6.15 is the gain of the LED current-sense
amplifier, and GMCOMP is the transconductance of the
error amplifier in amps/volts.
Layout Recommendations
Typically, there are two sources of noise emission in
a switching power supply: high di/dt loops and high
dV/dt surfaces. For example, traces that carry the drain
current often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dV/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOSFET
power dissipation or shield it. Keep all PCB traces
carrying switching currents as short as possible to minimize current loops. Use ground planes for best results.
www.maximintegrated.com
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer board
whenever possible for better noise immunity and power
dissipation. Follow these guidelines for good PCB layout:
U Use a large contiguous copper plane under the ICs’
package. Ensure that all heat-dissipating components
have adequate cooling.
U Isolate the power components and high-current paths
from the sensitive analog circuitry.
U Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation. Keep switching loops short such that:
a) The anode of D1 must be connected very close to
the drain of the MOSFET Q1.
b) The cathode of D1 must be connected very close to
COUT.
c) COUT and current-sense resistor R4 must be
connected directly to the ground plane.
U Connect PGND and SGND at a single point.
U Keep the power traces and load connections short. This
practice is essential for high efficiency. Use thick copper
PCBs (2oz vs. 1oz) to enhance full-load efficiency.
U Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer for
the PGND and SGND plane as an EMI shield to keep
radiated noise away from the device, feedback dividers,
and analog bypass capacitors.
Maxim Integrated │ 20
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Typical Operating Circuits
L1
VIN
6V TO 18V WITH LOAD
DUMP UP TO 70V
D1
Q1
IN
C1
NDRV
LFRAMP
MAX16833
MAX16833C
PWMDIM
PWMDIM
C2
R1
R5
R7
CS
OVP
ISENSE+
R2
ISENSE-
R3
RT/SYNC
DIMOUT
C3
Q2
FLT
COMP
VCC
C4
R4
LED+
R11
R9
LED-
R8
R10
ICTRL
SGND
PGND
EP
BOOST HEADLAMP DRIVER
LED-
L1
VIN
6V TO 18V WITH LOAD
DUMP UP TO 70V
D1
Q1
IN
C1
NDRV
REF MAX16833B
MAX16833D
PWMDIM
PWMDIM
R5
R7
CS
OVP
ISENSE+
R3
R2
ISENSE-
RT/SYNC
C3
C2
R1
DIMOUT
VCC
Q2
FLT
COMP
R9
C4
R4
R11
LED+
R8
ICTRL
SGND
PGND
EP
R10
BUCK-BOOST HEADLAMP DRIVER
www.maximintegrated.com
Maxim Integrated │ 21
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Typical Operating Circuits (continued)
LED-
L1
VIN
6V TO 18V WITH LOAD
DUMP UP TO 70V
D1
Q1
IN
C1
NDRV
MAX16833G
PWMDIM
R5
R7
CS
LFRAMP
PWMDIM
C2
R1
OVP
ISENSE+
R2
ISENSE-
R3
RT/SYNC
DIMOUT
C3
Q2
FLT
COMP
VCC
C4
R9
R8
R11
LED+
R10
ICTRL
SGND
R4
PGND
EP
BUCK-BOOST HEADLAMP DRIVER
www.maximintegrated.com
Maxim Integrated │ 22
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Ordering Information
TEMP RANGE
PIN-PACKAGE
FUNCTIONALITY
MAX DUTY CYCLE (%)
MAX16833AUE+
PART
-40°C to +125°C
16 TSSOP-EP*
Frequency Dithering
88.5
MAX16833AUE/V+
-40°C to +125°C
16 TSSOP-EP*
Frequency Dithering
88.5
MAX16833BAUE+
-40°C to +125°C
16 TSSOP-EP*
Reference Voltage Output
88.5
MAX16833BAUE/V+
-40°C to +125°C
16 TSSOP-EP*
Reference Voltage Output
88.5
MAX16833CAUE+
-40°C to +125°C
16 TSSOP-EP*
Frequency Dithering
94
MAX16833CAUE/V+
-40°C to +125°C
16 TSSOP-EP*
Frequency Dithering
94
MAX16833DAUE+
-40°C to +125°C
16 TSSOP-EP*
Reference Voltage Output
94
MAX16833DAUE/V+
-40°C to +125°C
16 TSSOP-EP*
Reference Voltage Output
94
MAX16833GAUE/V+
-40°C to +125°C
16 TSSOP-EP*
Frequency Dithering
94
+Denotes a lead(Pb)-free/RoHS-compliant package.
/V denotes an automotive qualified part.
*EP = Exposed pad.
Chip Information
PROCESS: BiCMOS-DMOS
www.maximintegrated.com
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maximintegrated.com/packages. Note
that a “+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but the
drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
16 TSSOP-EP
U16E+3
21-0108
90-0120
Maxim Integrated │ 23
MAX16833/MAX16833B/C/D/G
High-Voltage HB LED Drivers with
Integrated High-Side Current Sense
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
CHANGED
DESCRIPTION
0
6/10
Initial release
1
11/10
Added MAX16833AUE
—
2
12/10
Added MAX16833C and MAX16833D
3
7/11
Added MAX16833E
4
8/12
Removed MAX16833E
1–22
5
4/13
Updated startup delay time and its description
2, 11
6
8/13
Updated Functional Diagrams and the MAX16833B/MAX16833D Typical
Operating Circuit
7
2/15
Updated the Benefits and Features section
8
11/15
Added MAX16833G
9
6/16
Updated MAX16833G Functional Diagram
11
10
6/16
Added MAX16833G to Frequency Dithering (LFRAMP/MAX16833/
MAX16833C/MAX16833G) section
14
11
6/16
Added new MAX16833G Typical Operating Circuit diagram
22
1, 21, 22
22
1–4, 6–14, 20, 21
9, 10, 20
1
1–22
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
© 2016 Maxim Integrated Products, Inc. │ 24
工商网监
湘ICP备2023018690号
