MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
General Description
Features
The MAX16975 is a 1.2A current-mode step-down converter with an integrated high-side switch. The device
operates with input voltages from 3.5V to 28V while using
only 45FA quiescent current at no load. The switching
frequency is adjustable from 220kHz to 1.0MHz by using
an external resistor, and can be synchronized to an external clock. The device’s output voltage is pin-selectable to
a fixed 5V or adjustable from 1V to 10V using external
resistors. The wide input voltage range makes the device
ideal for automotive and industrial applications.
S Wide 3.5V to 28V Input Voltage Range
The device operates in skip mode for reduced current
consumption in light-load conditions. An adjustable
reset threshold helps keep microcontrollers alive down
to the lowest specified input voltage. Protection features
include cycle-by-cycle current limit, soft-start, overvoltage, and thermal shutdown with automatic recovery.
The device also features a power-good monitor to ease
power-supply sequencing.
S Less than 10µA Shutdown Current
S 42V Input Transient Tolerance
S 5V Fixed or 1V to 10V Adjustable Output Voltage
S Integrated 1.2A High-Side Switch
S 220kHz to 1.0MHz Adjustable Switching Frequency
S Frequency Synchronization Input
S Internal Boost Diode
S 45µA Skip-Mode Operating Current
S Adjustable Power-Good Output Level and Timing
S 3.3V Logic Level to 42V Compatible Enable Input
S Current-Limit, Thermal Shutdown, and
Overvoltage Protection
S -40°C to +125°C Automotive Temperature Range
Applications
The device is available in 16-pin QSOP and thermally
enhanced QSOP-EP packages. It operates over the
-40°C to +125°C automotive temperature range.
Automotive
Industrial
Ordering Information appears at end of data sheet.
Typical Application Circuit
3.5V TO 28V
CIN1
47µF
CIN3
0.1µF
CIN2
4.7µF
SUP
SUPSW
BST
LX
EN
FSYNC
CCOMP1
5600pF
RCOMP
12kI
RFB1
25kI
COUT1
47µF
COUT2
47µF
RESETI
FB
BIAS
CCRES
1nF
VOUT = 1.25V AT
1.2A AT 400kHz
VBIAS
FOSC
CBIAS
1µF
D1
OUT
MAX16975
RFOSC
61.9kI
L1
10µH
VOUT
COMP
CCOMP2
OPEN
CBST
0.1µF
CRES
RRES
10kI
RFB2
100kI
RES
GND
PLACE CIN3 (0.1µF) RIGHT NEXT TO SUP.
For related parts and recommended products to use with this part, refer to: www.maximintegrated.com/MAX16975.related
For pricing, delivery, and ordering information, please contact Maxim Direct
at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com.
19-5673; Rev 1; 10/13
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
ABSOLUTE MAXIMUM RATINGS
SUP, SUPSW, LX, EN to GND................................-0.3V to +45V
BST to GND ...........................................................-0.3V to +47V
BST to LX ................................................................-0.3V to +6V
OUT to GND...........................................................-0.3V to +12V
SUP to SUPSW......................................................-0.3V to +0.3V
RESETI, FOSC, COMP, BIAS,
FSYNC, CRES, RES, FB to GND..........................-0.3V to +6V
Output Short-Circuit Duration.....................................Continuous
Continuous Power Dissipation (TA = +70NC)
QSOP (derate 9.6mW/NC above +70NC)...................771.5mW
QSOP-EP (derate 22.7mW/NC above +70NC).......1818.20mW
Operating Temperature Range......................... -40NC to +125NC
Junction Temperature......................................................+150NC
Storage Temperature Range............................. -65NC to +150NC
Lead Temperature (soldering, 10s).................................+300NC
Soldering Temperature (reflow).......................................+260NC
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
PACKAGE THERMAL CHARACTERISTICS (Note 1)
QSOP
Junction-to-Ambient Thermal Resistance (qJA)......103.7°C/W
Junction-to-Case Thermal Resistance (qJC)................37°C/W
QSOP-EP
Junction-to-Ambient Thermal Resistance (qJA)...........44°C/W
Junction-to-Case Thermal Resistance (qJC)..................6°C/W
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
ELECTRICAL CHARACTERISTICS
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.)
PARAMETER
SYMBOL
Supply Voltage
VSUP,
VSUPSW
Supply Current
ISUP
Shutdown Supply Current
BIAS Regulator Voltage
BIAS Undervoltage Lockout
VBIAS
VUVBIAS
CONDITIONS
Normal operation
MIN
TYP
3.5
MAX
UNITS
28
V
Normal operation, no switching
2.9
mA
Skip mode, no load, VOUT = 5V
VEN = 0V
45
FA
9
FA
VSUP = VSUPSW = 6V to 42V, VOUT < 3V or
VOUT > 5.5V, ILOAD = 0A (Note 2)
4.7
5.0
5.3
VBIAS rising
2.95
3.15
3.35
V
V
BIAS Undervoltage Hysteresis
550
mV
Thermal-Shutdown Threshold
+175
NC
Thermal-Shutdown Threshold
Hysteresis
+15
NC
OUTPUT VOLTAGE (OUT)
Output Voltage
Skip-Mode Output Voltage
VOUT
Normal operation, VFB = VBIAS, ILOAD = 1A,
TA = +25°C
4.95
Normal operation, VFB = VBIAS, ILOAD = 1A,
-40°C P TA P +125°C
4.9
5
5.1
4.9
5.05
5.2
VOUT_SKIP No load, VFB = VBIAS (Note 3)
5
5.05
V
V
Load Regulation
VOUT = 5V, VFB = VBIAS, 30mA < ILOAD < 1A
0.3
%
Line Regulation
6V < VSUP < 28V
0.02
%/V
Maxim Integrated
*The parametric values (min, typ, max limits) shown in the Electrical Characteristics table supersede values quoted elsewhere in this data sheet.
2
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
ELECTRICAL CHARACTERISTICS* (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.)
PARAMETER
SYMBOL
BST Input Current
IBST
LX Current Limit
ILX
CONDITIONS
VBST - VLX = 5V
VSUP = 4.5V to 28V, VSUPSW = 14V,
TA = +25°C
VSUP = 4.5V to 28V, VSUPSW = 14V
Skip-Mode Current Threshold
ISKIP_TH
Power-Switch On-Resistance
RON
LX Leakage Current
ILX,LEAK
MIN
TYP
MAX
UNITS
1.7
2.5
mA
1.5
1.8
2.0
1.5
1.8
200
A
mA
RON measured between SUPSW and LX, ILX =
1A, VSUP = 4.5V to 28V, VBST - VLX = 4.5V
300
550
mI
VSUPSW = 28V, VLX = 0V, TA = +25°C
0.01
1
FA
TRANSCONDUCTANCE AMPLIFIER (COMP)
FB Input Current
IFB
FB Regulation Voltage
FB Line Regulation
VFB
nA
FB connected to an external resistive divider,
TA = +25°C
0.99
1.0
1.01
FB connected to an external resistive divider,
-40°C P TA P +125°C
0.985
1.0
1.015
V
4.5V < VSUP < 28V
0.02
%/V
VFB = 1V, VBIAS = 5V
1000
FS
tON
110
ns
DCCC
94
%
DVLINE
Transconductance (from FB to
COMP)
gm
Minimum On-Time
Cold-Crank Event Duty Cycle
20
OSCILLATOR FREQUENCY
1.0
RFOSC = 25.5kI, VSUP = 4.5V to 28V
Oscillator Frequency
Oscillator Frequency Range
fOSC
MHz
RFOSC = 61.9kI, VSUP = 4.5V to 28V
RFOSC = 120kI, VSUP = 4.5V to 28V (Note 3)
348
400
452
kHz
191
220
249
kHz
(Note 3)
220
1000
kHz
EXTERNAL CLOCK INPUT (FSYNC)
External Input Clock Acquisition
Time
1
tFSYNC
External Input Clock Frequency
(Note 3)
External Input Clock High Threshold VFSYNC_HI VFSYNC rising
fOSC +
10%
Hz
1.4
V
External Input Clock Low Threshold VFSYNC_LO VFSYNC falling
FSYNC Pulldown Resistance
Soft-Start Time
RFSYNC
tSS
Cycles
0.4
V
500
kI
fSW = 400kHz
4
ms
fSW = 1.0MHz
1.6
ms
ENABLE INPUT (EN)
Enable On Threshold Voltage Low
VEN_LO
0.8
V
Maxim Integrated
*The parametric values (min, typ, max limits) shown in the Electrical Characteristics table supersede values quoted elsewhere in this data sheet.
3
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
ELECTRICAL CHARACTERISTICS* (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.)
PARAMETER
SYMBOL
Enable On Threshold Voltage High
VEN_HI
Enable Threshold Voltage
Hysteresis
Enable Input Current
CONDITIONS
MIN
TYP
MAX
2.2
UNITS
V
VEN,HYS
0.2
V
IEN
10
nA
RESET
Reset Internal Switching Level
RESETI Threshold Voltage
CRES Threshold Voltage
CRES Threshold Hysteresis
VTH_RISING VFB rising, VRESETI = 0V
93
95
96.5
VTH_FALLING VFB falling, VRESETI = 0V
VRESETI_HI VRESETI falling
91
93
95
1.05
1.25
1.4
V
1.07
1.13
1.19
V
VCRES_HI
VCRES rising
VCRES_HYS
0.05
RESETI Input Current
IRESET
VRESETI = 0V
CRES Source Current
ICRES
VOUT in regulation
CRES Pulldown Current
ICRES_PD
VOUT out of regulation
RES Output Low Voltage
ISINK = 5mA
RES Leakage Current (OpenDrain Output)
VOUT in regulation
Reset Debounce Time
tRES_DEB
VRESETI falling
V
0.02
9.5
10
FA
10.5
1
FA
mA
TA = +25°C
TA = +125°C
%VFB
0.4
V
1
FA
20
nA
25
Fs
Note 2: When 3V < VOUT < 5.5V, the bias regulator is connected to the output to save quiescent current, VBIAS = VOUT.
Note 3: Guaranteed by design; not production tested.
Maxim Integrated
*The parametric values (min, typ, max limits) shown in the Electrical Characteristics table supersede values quoted elsewhere in this data sheet.
4
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Typical Operating Characteristics
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = +25NC, unless otherwise noted.)
STARTUP WITH FULL LOAD
(OUT = 1.25V, fSW = 400kHz)
EFFICIENCY vs. LOAD CURRENT
MAX16975 toc01
OUT
1V/div
ILOAD = 1.2A
90
80
0V
EN
5V/div
0V
EFFICIENCY (%)
0V
RES
5V/div
MAX16975 toc02
100
70
3.3V/400kHz
60
1.25V/400kHz
40
30
20
SUP
5V/div
10
0V
0
2ms/div
0
1200
401.2
400.8
400.4
5V OUTPUT
1000
800
600
400
200
400.0
200
450
700
950
10
1200
32
54
76
98
ILOAD (mA)
RFOSC (kI)
SWITCHING FREQUENCY vs. TEMPERATURE
(1.25V/400kHz, 5V/400kHz)
LOAD-STEP RESPONSE
(1.25V/400kHz)
120
MAX16975 toc06
ILOAD = 1.2A, RFOSC = 64.87kI
430
MAX16975 toc05
450
MAX16975 toc04
401.6
1200
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
PWM MODE
800
SWITCHING FREQUENCY vs. RFOSC
MAX16975 toc03
402.0
400
ILOAD (mA)
SWITCHING FREQUENCY
vs. LOAD CURRENT (1.25V/400kHz)
SWITCHING FREQUENCY (kHz)
8V/400kHz
5V/400kHz
50
0 TO 1.25A LOAD STEP
5V/400kHz
VOUT
AC-COUPLED
410
VOUT
100mV/div
0
390
1.25V/400kHz
ILOAD
1A/div
370
0
350
-40 -25 -10 5 20 35 50 65 80 95 110 125
4ms/div
TEMPERATURE (°C)
Maxim Integrated
5
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Typical Operating Characteristics (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = +25NC, unless otherwise noted.)
COLD-CRANK PULSE (1.25V/400kHz)
DIPS AND DROPS TEST (1.25V/400kHz)
MAX16975 toc07
MAX16975 toc08
VSUPSW
10V/div
0V
VSUPSW
10V/div
0V
VOUT
1V/div
0V
VOUT
1V/div
0V
VLX
10V/div
0V
0V
VRES
5V/div
0V
VLX
10V/div
VRES
5V/div
0V
10ms/div
10ms/div
SLOW VIN RAMP-UP TEST
OUTPUT SHORT-CIRCUIT TEST
(1.25V/400kHz)
MAX16975 toc09
MAX16975 toc10
RLOAD = 0.3I
VSUP/SUPSW
10V/div
0V
VOUT
5V/div
VLX
10V/div
0A
ILOAD
2A/div
0V
VLX
10V/div
1ms/div
QUIESCENT CURRENT
vs. INPUT VOLTAGE
VOUT vs. TEMPERATURE IN PWM MODE
(5V/400kHz)
70
60
50
40
30
5V/400kHz
20
2
OUTPUT VOLTAGE CHANGE (%)
MAX16975 toc11
80
1
ILOAD = 1.2A
5V/400kHz
MAX16975 toc12
10s/div
90
QUIESCENT CURRENT (µA)
VOUT
2V/div
0
-1
10
-2
0
4
8
12
16
20
INPUT VOLTAGE (V)
Maxim Integrated
24
28
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (°C)
6
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Typical Operating Characteristics (continued)
(VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC =
61.9kI, TA = +25NC, unless otherwise noted.)
1
0
-1
2
-2
ILOAD = 0A, SKIP MODE
5V/400kHz
1
0
-1
-2
-40 -25 -10 5 20 35 50 65 80 95 110 125
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (°C)
TEMPERATURE (°C)
VOUT vs. TEMPERATURE IN SKIP MODE
(1.25V/400kHz)
1
5V/400kHz
VOUT (V)
5.05
0
-1
5.00
4.95
-2
-40 -25 -10 5 20 35 50 65 80 95 110 125
TEMPERATURE (°C)
Maxim Integrated
MAX16975 toc16
ILOAD = 0A, SKIP MODE
LINE REGULATION
5.10
MAX16975 toc15
OUTPUT VOLTAGE CHANGE (%)
2
MAX16975 toc14
ILOAD = 1.2A
OUTPUT VOLTAGE CHANGE (%)
OUTPUT VOLTAGE CHANGE (%)
2
VOUT vs. TEMPERATURE IN SKIP MODE
(5V/400kHz)
MAX16975 toc13
VOUT vs. TEMPERATURE IN PWM MODE
(1.25V/400kHz)
4.90
6
8 10 12 14 16 18 20 22 24 26 28
VSUPSW (V)
7
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Pin Configurations
TOP VIEW
CRES 1
TOP VIEW
+
FOSC 2
FSYNC 3
MAX16975A
I.C. 4
CRES 1
15 RES
FOSC 2
14 EN
FSYNC 3
13 SUPSW
COMP 5
12 LX
FB 6
OUT 7
16 RESETI
EP
GND 8
+
16 RESETI
15 RES
MAX16975B
I.C. 4
14 EN
13 SUPSW
COMP 5
12 LX
11 SUP
FB 6
11 SUP
10 BST
OUT 7
10 BST
9 BIAS
GND 8
9 BIAS
QSOP
QSOP
Pin Description
PIN
NAME
FUNCTION
1
CRES
Analog Reset Timer. CRES sources 10FA (typ) of current into an external capacitor to set the reset timeout
period. Reset timeout period is defined as the time between the start of output regulation and RES switching to high impedance. Leave CRES unconnected for minimum delay time.
2
FOSC
Resistor-Programmable Switching Frequency Control Input. Connect a resistor from FOSC to GND to set
the switching frequency (see the Internal Oscillator section).
3
FSYNC
Synchronization Input. The device synchronizes to an external signal applied to FSYNC. The external signal
period must be 10% shorter than the internal clock period for proper operation.
4
I.C.
5
COMP
Error-Amplifier Output. Connect a compensation network from COMP to GND for stable operation. See the
Compensation Network section.
6
FB
Feedback Input. Connect an external resistive divider from FB to OUT and GND to set the output voltage
between 1V and 10V. Connect FB directly to BIAS to set the output voltage to 5V. See the Applications
Information section.
7
OUT
Connect OUT to the output of the converter. OUT provides power to the internal circuitry when the output
voltage of the converter is set between 3V and 5.6V. During shutdown, OUT is pulled to GND with a 50I
resistor.
8
GND
Ground
9
BIAS
Linear Regulator Output. BIAS powers the internal circuitry. Bypass BIAS with a 1FF capacitor to ground as
close as possible to the device. During shutdown, BIAS is actively discharged through a 32kI resistor.
10
BST
High-Side Driver Supply. Connect a 0.1FF capacitor between LX and BST for proper operation.
11
SUP
Voltage Supply Input. SUP powers the internal linear regulator. Connect a 4.7FF capacitor from SUP to
ground. Connect SUP to SUPSW.
12
LX
Maxim Integrated
Internally Connected. Connect to GND.
Inductor Connection. Connect a rectifying Schottky diode between LX and GND. Connect an inductor from
LX to the output.
8
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Pin Description (continued)
PIN
NAME
FUNCTION
Internal High-Side Switch Supply Input. SUPSW provides power to the internal switch. Connect a 4.7FF
capacitor from SUPSW to ground. Connect SUP to SUPSW. See the Input Capacitor section.
13
SUPSW
14
EN
Battery-Compatible Enable Input. Drive EN low to disable the device. Drive EN high to enable the device.
15
RES
16
RESETI
Open-Drain Active-Low Reset Output. RES asserts when VOUT is below the reset threshold set by RESETI.
Reset Threshold Level Input. Connect to a resistive divider to set the reset threshold for RES. Connect
RESETI to GND to enable the internal reset threshold.
—
EP
Exposed Pad (MAX16975A Only). Connect EP to a large-area contiguous copper ground plane for effective power dissipation. Do not use as the only IC ground connection. EP must be connected to GND.
Functional Diagram
SUP
BIAS
BST
FOSC
SUPSW
DRV
ISENSE
LX
LEVEL
SHIFT
FSYNC
OUT
EN
STANDBY
SUPPLY
OSC
SUM
REF
EA
LDO
ILIM
PWM
COMP
LOGIC
MUX
UVLO
COMP
LOGIC FOR
100% DUTY-CYCLE
OPERATION
VBIAS
SOFTSTART
FB
10µA
CRES
RESETI
COMP
B.G.
REF
COMP
RES
MAX16975
GND
Maxim Integrated
9
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
Detailed Description
The MAX16975 is a constant-frequency, current-mode
automotive buck converter with an integrated high-side
switch. The device operates with input voltages from
3.5V to 28V and tolerates input transients up to 42V.
During undervoltage events, such as cold-crank conditions, the internal pass device maintains 94% duty cycle
for a short time.
An open-drain, active-low reset output helps to monitor
the output voltage. The device offers an adjustable reset
threshold that helps to keep microcontrollers alive down
to the lowest specified input voltage and a capacitorprogrammable reset timeout to ensure proper startup.
The switching frequency is resistor-programmable from
220kHz to 1.0MHz to allow optimization for efficiency,
noise, and board space. A clock input, FSYNC, allows
the device to synchronize to an external clock.
During light-load conditions, the device enters skip
mode that reduces the quiescent current down to 45FA
and increases light-load efficiency. The 5V fixed output
voltage eliminates the need for external resistors and
reduces the supply current by up to 50FA.
Linear Regulator Output (BIAS)
The device includes a 5V linear regulator, VBIAS, that
provides power to the internal circuitry. Connect a 1FF
ceramic capacitor from BIAS to GND. When the output
voltage is set between 3V and 5.5V, the internal linear
regulator only provides power until the output is in regulation. The internal linear regulator turns off once the output
is in regulation and allows OUT to provide power to the
device. The internal regulator turns back on once the
external load on the output of the device is higher than
100mA. In addition, the linear regulator turns on anytime
the output voltage is outside the 3V to 5.5V range.
The device also offers a capacitor-programmable reset
timeout period. Connect a capacitor from CRES to GND
to adjust the reset timeout period. When the output voltage goes out of regulation, RES asserts low and the
reset timing capacitor discharges with a 1mA pulldown
current. Once the output is back in regulation the reset
timing capacitor recharges with 10FA (typ) current. RES
stays low until the voltage at CRES reaches 1.13V (typ).
Dropout Operation
The device features an effective maximum duty cycle to
help refresh the BST capacitor when continuously operated in dropout. When the high-side switch is on for three
consecutive clock cycles, the device forces the high-side
switch off during the final 35% of the fourth clock cycle.
When the high-side switch is off, the LX node is pulled low
by the current flowing through the inductor. This increases
the voltage across the BST capacitor. To ensure that the
inductor has enough current to pull LX to ground, an
internal load sinks current from VOUT when the device is
close to dropout and external load is small. Once the input
voltage is increased above the dropout region, the device
continues to regulate at the set output voltage.
The device operates with no load and no external clock
at an effective maximum duty cycle of 94% in deep dropout. This effective maximum duty cycle is influenced by
the external load and by the optional external synchronized clock.
System Enable (EN)
External Clock Input (FSYNC)
An enable-control input (EN) activates the device from
the low-power shutdown mode. EN is compatible with
inputs from the automotive battery level down to 3.3V.
The high-voltage compatibility allows EN to be connected to SUP, KEY/KL30, or the INH inputs of a CAN
transceiver.
Adjustable Reset Level
EN turns on the internal regulator. Once VBIAS is above
the internal lockout level, VUVL = 3.15V (typ), the controller activates and the output voltage ramps up within 2048
cycles of the switching frequency.
The device synchronizes to an external clock signal
applied at FSYNC. The signal at FSYNC must have a frequency of 10% higher than the internal clock frequency
for proper synchronization.
The device features a programmable reset threshold
using a resistive divider between OUT, RESETI, and
GND. Connect RESETI to GND for the internal threshold.
RES asserts low when the output voltage falls to 93% of
the programmed level. RES deasserts when the output
voltage goes above 95% of the set voltage.
Maxim Integrated
Some microprocessors accept a wide input voltage range
(3.3V to 5V, for example) and can operate during dropout
of the device. Use a resistive divider at RESETI to adjust
the reset activation level (RES goes low) to lower levels.
The reference voltage at RESETI is 1.25V (typ).
A logic-low at EN shuts down the device. During shutdown, the internal linear regulator and gate drivers turn
off. Shutdown mode reduces the quiescent current to
9FA (typ). Drive EN high to turn on the device.
10
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
VOUT
RFB1
RESETI
RFB2
MAX16975
FB
RFB3
Figure 1. Output Voltage/Reset Threshold Resistive Divider
Network
Overvoltage Protection
Applications Information
Output Voltage/Reset Threshold
Resistive Divider Network
Although the device’s output voltage and reset threshold
can be set individually, Figure 1 shows a combined resistive divider network to set the desired output voltage and
the reset threshold using three resistors. Use the following formula to determine the RFB3 of the resistive divider
network:
× VREF
R
R FB3 = TOTAL
V OUT
where VREF = 1V, RTOTAL = selected total resistance of
RFB1, RFB2, and RFB3 in ohms, and VOUT is the desired
output voltage in volts.
The device includes overvoltage protection circuitry that
protects the device when there is an overvoltage condiUse the following formula to calculate the value of RFB2
tion at the output. If the output voltage increases by more
of the resistive divider network:
than 12% of its set voltage, the device stops switching.
The device resumes regulation once the overvoltage
condition is removed.
R TOTAL × VREF_RES
=
R FB2
− R FB3
Overload Protection
V RES
The overload protection circuitry is activated when the
device is in current limit and VOUT is below the reset
where VREF_RES is 1.25V (see the Electrical Characteristics
threshold. Under these conditions, the device enters
table) and VRES is the desired reset threshold in volts.
a soft-start mode. When the overcurrent condition is
The precision of the reset threshold function is depenremoved before the soft-start mode is over, the device
dent on the tolerance of the resistors used for the divider.
regulates the output voltage to the set value. Otherwise,
the soft-start cycle repeats until the overcurrent condition
BST Capacitor Selection
is removed.
for Dropout Operation
The
device
includes
an
internal
boost capacitor refresh
Skip Mode
algorithm
for
dropout
operation.
This
is required to ensure
During light-load operation, IINDUCTOR P 200mA, the
proper
boost
capacitor
voltage
that
delivers
power to the
device enters skip-mode operation. Skip mode turns off
gate-drive
circuitry.
When
the
HSFET
is
on
consecutively
the internal switch and allows the output to drop below
for 3.65 clock cycles, the internal counter detects this
regulation voltage before the switch is turned on again.
and turns off the HSFET for 0.35 clock cycles. This is of
The lower the load current, the longer it takes for the
particular concern when VIN is falling and approaching
regulator to initiate a new cycle effectively increasing
VOUT at the minimum switching frequency (220kHz).
light-load efficiency. During skip mode, the device quiescent current drops to as low as 45FA.
Overtemperature Protection
Thermal-overload protection limits the total power dissipation in the device. When the junction temperature
exceeds +175NC (typ), an internal thermal sensor shuts
down the step-down controller, allowing the device to
cool. The thermal sensor turns on the device again after
the junction temperature cools by +15NC.
Maxim Integrated
The worst-case condition for boost capacitor refresh time
is with no load on the output. For the boost capacitor
to recharge completely, the LX node must be pulled to
ground. If there is no current through the inductor then
the LX node does not go to ground. To solve this issue,
an internal load of about 100mA turns on at the sixth
clock cycle, which is determined by a separate counter.
In the worst-case condition with no load, the LX node
does not go below ground during the first detect of the
11
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
3.65 clock cycles. The device waits for the next 3.65
clock cycles to finish. As a result, the soonest the LX
node can go below ground is 4 + 3.65 = 7.65 clock
cycles. This time does not factor in the size of the inductor and the time it takes for the inductor current to build
up to 100mA (internal load).
No load minimum time before refresh is:
∆T (no load) = 7.65 clock cycles = 7.65 x 4.54μs (at
220kHz) = 34.73μs
Assuming a full 100mA is needed to refresh the BST capacitor and depending on the size of the inductor, the time it
takes to build up full 100mA in the inductor is given by:
∆T (inductor) = L x ∆I/∆V (current build-up starts from the
sixth clock cycle)
L = inductor value chosen in the design guide.
∆I is the required current = 100mA.
∆V = voltage across the inductor (assume this to be
0.5V), which means VIN is greater than VOUT by 0.5V.
If ∆T (inductor) < 7.65 – 6 (clock cycles) then the BST
capacitor is sized as follows:
BST_CAP ≥ I_BST(dropout) x ∆T (no load)/∆V (BST
capacitor)
∆T (no load) = 7.65 clock cycles = 34.73μs.
∆V (BST capacitor), for (3.3V to 5V) output = VOUT – 2.7V
(2.7V is the minimum voltage allowed on the BST capacitor).
If ∆T (inductor) > 7.65 - 6 clock cycles then we need to
wait for the next count of 3.65 clock cycles making ∆T (no
load) = 11.65 clock cycles.
Assume ∆T (no load) to be 16 clock cycles when designing the BST capacitor with a typical inductor value for
220kHz operation.
RESET_TIMEOUT =
1.13V × C
(s)
10µA
where C is the capacitor from CRES to GND in Farads.
Internal Oscillator
The device’s internal oscillator is programmable from
220kHz to 1.0MHz using a single resistor at FOSC. Use
the following formula to calculate the switching frequency:
fOSC (Hz) ≈
26.4 × 10 9 (Ω x Hz)
R
where R is the resistor from FOSC to GND in ohms.
For example, a 220kHz switching frequency is set with
RFOSC = 120kI. Higher frequencies allow designs with
lower inductor values and less output capacitance.
Consequently, peak currents and I2R losses are lower
at higher switching frequencies, but core losses, gatecharge currents, and switching losses increase.
Inductor Selection
Three key inductor parameters must be specified for
operation with the device: inductance value (L), inductor
saturation current (ISAT), and DC resistance (RDCR). To
select inductance value, the ratio of inductor peak-topeak AC current to DC average current (LIR) must be
selected first. A good compromise between size and loss
is a 30% peak-to-peak ripple current to average-current
ratio (LIR = 0.3). The switching frequency, input voltage,
output voltage, and selected LIR then determine the
inductor value as follows:
V
(V
-V
)
L = OUT SUPSW OUT
VSUPSW fSWIOUTLIR
where:
where VSUPSW, VOUT, and IOUT are typical values (so
that efficiency is optimum for typical conditions). The
switching frequency is set by RFOSC. The exact inductor
value is not critical and can be adjusted to make tradeoffs among size, cost, efficiency, and transient response
requirements. Table 1 shows a comparison between
small and large inductor sizes.
I_BST (dropout) = 2.5mA (worst case)
Table 1. Inductor Size Comparison
The final BST_CAP equation is:
BST_CAP = I_BST (dropout) x ∆T (no load)/∆V (BST
capacitor)
∆T (no load) = 16 clock cycles
∆V (BST capacitor) = VOUT - 2.7V
Reset Timeout Period
The device offers a capacitor-adjustable reset timeout
period. CRES can source 10FA of current. Use the following formula to set the timeout period.
Maxim Integrated
INDUCTOR SIZE
SMALLER
LARGER
Lower price
Smaller ripple
Smaller form-factor
Higher efficiency
Faster load response
Larger fixed-frequency
range in skip mode
12
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
The inductor value must be chosen so that the maximum
inductor current does not reach the minimum current limit
of the device. The optimum operating point is usually
found between 15% and 35% ripple current. When pulse
skipping (light loads), the inductor value also determines
the load-current value at which PFM/PWM switchover
occurs.
Find a low-loss inductor having the lowest possible
DC resistance that fits in the allotted dimensions. Most
inductor manufacturers provide inductors in standard
values, such as 1.0FH, 1.5FH, 2.2FH, 3.3FH, etc. Also
look for nonstandard values, which can provide a better
compromise in LIR across the input voltage range. If
using a swinging inductor (where the no-load inductance
decreases linearly with increasing current), evaluate
the LIR with properly scaled inductance values. For
the selected inductance value, the actual peak-to-peak
inductor ripple current (DIINDUCTOR) is defined by:
VOUT (VSUPSW - VOUT )
∆IINDUCTOR =
VSUPSW × fSW × L
where DIINDUCTOR is in A, L is in H, and fSW is in Hz.
Ferrite cores are often the best choices, although powdered iron is inexpensive and can work well at 220kHz.
The core must be large enough not to saturate at the
peak inductor current (IPEAK):
=
IPEAK ILOAD(MAX) +
∆IINDUCTOR
2
Input Capacitor
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor RMS current requirement (IRMS) is
defined by the following equation:
IRMS = ILOAD(MAX)
VOUT (VSUPSW - VOUT )
VSUPSW
IRMS is at a maximum value when the input voltage
equals twice the output voltage (VSUPSW = 2VOUT), so
IRMS(MAX) = ILOAD(MAX)/2.
Choose an input capacitor that exhibits less than +10NC
self-heating temperature rise at the RMS input current for
optimal long-term reliability.
Maxim Integrated
The input-voltage ripple comprises DVQ (caused by the
capacitor discharge) and DVESR (caused by the ESR
of the capacitor). Use low-ESR ceramic capacitors with
high ripple-current capability at the input. Assume the
contribution from DVQ and DVESR to be 50%. Calculate
the input capacitance and ESR required for a specified
input-voltage ripple using the following equations:
ESRIN =
∆VESR
∆I
IOUT + L
2
where:
(V
-V
) × VOUT
∆IL = SUPSW OUT
VSUPSW × fSW × L
and
I
× D(1- D)
CIN = OUT
∆VQ × fSW
and
D=
VOUT
VSUPSW
IOUT is the maximum output current and D is the duty
cycle.
Output Capacitor
The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and
load transient requirements, yet have high enough ESR
to satisfy stability requirements. The output capacitance
must be high enough to absorb the inductor energy while
transitioning from full-load to no-load conditions without
tripping the overvoltage fault protection. When using
high-capacitance, low-ESR capacitors, the filter capacitor’s ESR dominates the output voltage ripple. So the size
of the output capacitor depends on the maximum ESR
required to meet the output voltage ripple (VRIPPLE(P-P))
specifications:
VRIPPLE(P −P) =
ESR × ILOAD(MAX) × LIR
The actual capacitance value required relates to the
physical size needed to achieve low ESR, as well as
to the chemistry of the capacitor technology. Thus, the
capacitor is usually selected by ESR and voltage rating
rather than by capacitance value.
13
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
When using low-capacity filter capacitors, such as
ceramic capacitors, size is usually determined by the
capacity needed to prevent VSAG and VSOAR from causing problems during load transients. Generally, once
enough capacitance is added to meet the overshoot
requirement, undershoot at the rising load edge is no
longer a problem. However, low-capacity filter capacitors
typically have high-ESR zeros that can affect the overall
stability.
Rectifier Selection
The device requires an external Schottky diode rectifier as a freewheeling diode. Connect this rectifier close
to the device using short leads and short PCB traces.
Choose a rectifier with a continuous current rating higher
than the highest output current-limit threshold (1.5A) and
with a voltage rating higher than the maximum expected
input voltage, VSUPSW. Use a low forward-voltage-drop
Schottky rectifier to limit the negative voltage at LX. Avoid
higher than necessary reverse-voltage Schottky rectifiers
that have higher forward-voltage drops.
Compensation Network
The device uses an internal transconductance error
amplifier with its inverting input and its output available for
external frequency compensation. The output capacitor
and compensation network determine the loop stability.
The inductor and the output capacitor are chosen based
on performance, size, and cost. Additionally, the compensation network optimizes the control-loop stability.
The controller uses a current-mode control scheme that
regulates the output voltage by forcing the required current through the external inductor, so the device uses
the voltage drop across the high-side MOSFET. Currentmode control eliminates the double pole in the feedback
loop caused by the inductor and output capacitor resulting in a smaller phase shift and requiring less elaborate
error-amplifier compensation than voltage-mode control.
A simple single series resistor (RC) and capacitor (CC)
are all that is required to have a stable, high-bandwidth
loop in applications where ceramic capacitors are used
for output filtering (Figure 2). For other types of capacitors, due to the higher capacitance and ESR, the frequency of the zero created by the capacitance and
ESR is lower than the desired closed-loop crossover frequency. To stabilize a nonceramic output capacitor loop,
add another compensation capacitor (CF) from COMP to
GND to cancel this ESR zero.
The basic regulator loop is modeled as a power modulator, output feedback divider, and an error amplifier. The
power modulator has a DC gain set by gMC O RLOAD,
with a pole and zero pair set by RLOAD, the output
capacitor (COUT), and its ESR. The following equations
allow to approximate the value for the gain of the power
modulator (GAINMOD(DC)), neglecting the effect of the
ramp stabilization. Ramp stabilization is necessary when
the duty cycle is above 50% and is internally done for
the device.
GAINMOD(dc)
= g MC ×
R LOAD × fSW × L
R LOAD + (fSW × L)
where RLOAD = VOUT/ILOUT(MAX) in I, fSW is the switching frequency in MHz, L is the output inductance in FH,
and gMC = 3S.
In a current-mode step-down converter, the output
capacitor, its ESR, and the load resistance introduce a
pole at the following frequency:
fpMOD =
1
R LOAD × fSW × L
2π × C OUT ×
+ ESR
R LOAD + (fSW × L)
The output capacitor and its ESR also introduce a zero at:
fzMOD =
1
2π × ESR × C OUT
When COUT is composed of “n” identical capacitors in
parallel, the resulting COUT = n O COUT(EACH) and ESR
= ESR(EACH)/n. Note that the capacitor zero for a parallel combination of alike capacitors is the same as for an
individual capacitor.
VOUT
R1
R2
COMP
gm
VREF
RC
CF
CC
Figure 2. Compensation Network
Maxim Integrated
14
MAX16975
28V, 1.2A Automotive Step-Down Converter
with Low Operating Current
The feedback voltage-divider has a gain of GAINFB =
VFB/VOUT, where VFB is 1V (typ). The transconductance
error amplifier has a DC gain of GAINEA(DC) = gm,EA O
ROUT,EA, where gm,EA is the error-amplifier transconductance, which is 1000FS (typ), and ROUT,EA is the
output resistance of the 50MI error amplifier.
A dominant pole (fdpEA) is set by the compensation capacitor (CC) and the amplifier output resistance
(ROUT,EA). A zero (fzEA) is set by the compensation resistor (RC) and the compensation capacitor (CC). There is
an optional pole (fpEA) set by CF and RC to cancel the
output capacitor ESR zero if it occurs near the crossover
frequency (fC, where the loop gain equals 1 (0dB)).
Therefore:
GAINMOD(fC) ×
Solving for RC:
RC =
VOUT
g m,EA × VFB × GAINMOD(fC)
Set the error-amplifier compensation zero formed by RC
and CC (fzEA) at the fpMOD. Calculate the value of CC
as follows:
CC =
Thus:
fdpEA =
1
2π × C C × (R OUT,EA + R C )
fzEA =
1
2π × C C × R C
The loop-gain crossover frequency (fC) is set below
1/5th the switching frequency and much higher than the
power-modulator pole (fpMOD):
The total loop gain as the product of the modulator gain,
the feedback voltage-divider gain, and the error-amplifier
gain at fC is equal to 1. So:
For the case where fzMOD is greater than fC:
GAIN
=
MOD(fC ) GAINMOD(dc) ×
fpMOD
fC
fpMOD
fzMOD
The error-amplifier gain at fC is:
f
GAINEA(f =
g m,EA × R C × zMOD
C)
fC
Therefore:
GAINMOD(f ) ×
C
GAINEA(fC)
= g m,EA × R C
Maxim Integrated
1
2π × fzMOD × R C
As the load current decreases, the modulator pole
also decreases; however, the modulator gain increases
accordingly and the crossover frequency remains the
same. For the case where fzMOD is less than fC:
VFB
× GAINEA(f ) =
1
C
VOUT
GAIN
=
MOD(fC) GAINMOD(dc) ×
2π × fpMOD × R C
The power-modulator gain at fC is:
f
fpMOD
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