MAX16976AEE/V+

MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current General Description Features The MAX16976 is a 600mA current-mode step-down converter with an integrated high-side switch. The device operates with input voltages from 3.5V to 28V while using only 45FA quiescent current at no load. The switching frequency is adjustable from 220kHz to 1.0MHz by using an external resistor, and can be synchronized to an external clock. The device’s output voltage is pin-selectable to a fixed 5V or adjustable from 1V to 10V using external resistors. The wide input voltage range makes the device ideal for automotive and industrial applications. S Wide 3.5V to 28V Input Voltage Range The device operates in skip mode for reduced current consumption in light-load conditions. An adjustable reset threshold helps keep microcontrollers alive down to the lowest specified input voltage. Protection features include cycle-by-cycle current limit, soft-start, overvoltage, and thermal shutdown with automatic recovery. The device also features a power-good monitor to ease power-supply sequencing. S Less than 10µA Shutdown Current S 42V Input Transient Tolerance S 5V Fixed or 1V to 10V Adjustable Output Voltage S Integrated 600mA High-Side Switch S 220kHz to 1.0MHz Adjustable Switching Frequency S Frequency Synchronization Input S Internal Boost Diode S 45µA Skip-Mode Operating Current S Adjustable Power-Good Output Level and Timing S 3.3V Logic Level to 42V Compatible Enable Input S Current-Limit, Thermal Shutdown, and Overvoltage Protection S -40°C to +125°C Automotive Temperature Range Applications The device operates over the -40°C to +125°C automotive temperature range, and is available in a 16-pin QSOP package. Automotive Industrial Ordering Information appears at end of data sheet. Typical Application Circuit 3.5V TO 28V CIN1 47µF CIN3 0.1µF CIN2 4.7µF SUP SUPSW BST LX EN FSYNC CCOMP1 5600pF RCOMP 12kI RFB1 25kI COUT1 47µF COUT2 47µF RESETI FB BIAS CCRES 1nF VOUT = 1.25V AT 600mA AT 400kHz VBIAS FOSC CBIAS 1µF D1 OUT MAX16976 RFOSC 61.9kI L1 22µH VOUT COMP CCOMP2 OPEN CBST 0.1µF CRES RRES 10kI RFB2 100kI RES GND PLACE CIN3 (0.1µF) RIGHT NEXT TO SUP. For related parts and recommended products to use with this part, refer to: www.maximintegrated.com/MAX16976.related For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com. 19-5802; Rev 1; 10/14 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current ABSOLUTE MAXIMUM RATINGS SUP, SUPSW, LX, EN to GND................................-0.3V to +45V BST to GND ...........................................................-0.3V to +47V BST to LX ................................................................-0.3V to +6V OUT to GND...........................................................-0.3V to +12V SUP to SUPSW......................................................-0.3V to +0.3V RESETI, FOSC, COMP, BIAS, FSYNC, CRES, RES, FB to GND..........................-0.3V to +6V Output Short-Circuit Duration.....................................Continuous Continuous Power Dissipation (TA = +70NC) QSOP (derate 9.6 mW/NC above +70NC)..................771.5mW Operating Temperature Range......................... -40NC to +125NC Junction Temperature......................................................+150NC Storage Temperature Range............................. -65NC to +150NC Lead Temperature (soldering, 10s).................................+300NC Soldering Temperature (reflow).......................................+260NC Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. PACKAGE THERMAL CHARACTERISTICS (Note 1) QSOP Junction-to-Ambient Thermal Resistance (qJA)......103.7°C/W Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial. ELECTRICAL CHARACTERISTICS (VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.) PARAMETER SYMBOL Supply Voltage VSUP, VSUPSW Supply Current ISUP Shutdown Supply Current BIAS Regulator Voltage BIAS Undervoltage Lockout VBIAS VUVBIAS CONDITIONS Normal operation MIN TYP 3.5 MAX UNITS 28 V Normal operation, no switching 2.9 mA Skip mode, no load, VOUT = 5V 45 FA VEN = 0V 9 FA VSUP = VSUPSW = 6V to 42V, VOUT < 3V or VOUT > 5.5V, ILOAD = 0A (Note 2) 4.7 5.0 5.3 V VBIAS rising 2.95 3.15 3.35 V BIAS Undervoltage Hysteresis 550 mV Thermal-Shutdown Threshold +175 NC Thermal-Shutdown Threshold Hysteresis +15 NC OUTPUT VOLTAGE (OUT) Output Voltage Skip-Mode Output Voltage Load Regulation Line Regulation Maxim Integrated VOUT Normal operation, VFB = VBIAS, ILOAD = 0.6A, TA = +25°C 4.95 5 5.05 Normal operation, VFB = VBIAS, ILOAD = 0.6A, -40°C P TA P +125°C 4.9 5 5.1 4.9 5.05 5.2 VOUT_SKIP No load, VFB = VBIAS (Note 3) VOUT = 5V, VFB = VBIAS, 30mA < ILOAD < 0.6A 6V < VSUP < 28V V V 0.3 % 0.02 %/V   2 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current ELECTRICAL CHARACTERISTICS (continued) (VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.) PARAMETER SYMBOL CONDITIONS BST Input Current IBST VBST - VLX = 5V LX Current Limit ILX VSUP = 4.5V to 28V, VSUPSW = 14V Skip-Mode Current Threshold 0.79 ISKIP_TH Power-Switch On-Resistance LX Leakage Current MIN RON ILX,LEAK TYP MAX UNITS 1.7 2.5 mA 1.0 1.25 A 100 mA RON measured between SUPSW and LX, ILX = 0.6A, VSUP = 4.5V to 28V, VBST - VLX = 4.5V 300 550 mI VSUPSW = 28V, VLX = 0V, TA = +25°C 0.01 1 FA TRANSCONDUCTANCE AMPLIFIER (COMP) FB Input Current IFB FB Regulation Voltage FB Line Regulation VFB Cold-Crank Event Duty Cycle 0.99 FB connected to an external resistive divider, -40°C P TA P +125°C 0.985 1.0 1.01 V 1.0 1.015 0.02 %/V VFB = 1V, VBIAS = 5V 1000 FS tON 110 ns DCCC 93.5 % gm Minimum On-Time FB connected to an external resistive divider, TA = +25°C nA 4.5V < VSUP < 28V DVLINE Transconductance (from FB to COMP) 20 OSCILLATOR FREQUENCY RFOSC = 25.5kI, VSUP = 4.5V to 28V Oscillator Frequency Oscillator Frequency Range fOSC 1.0 RFOSC = 61.9kI, VSUP = 4.5V to 28V RFOSC = 120kI, VSUP = 4.5V to 28V (Note 3) 348 400 191 220 (Note 3) 220 MHz 452 kHz 249 kHz 1000 kHz EXTERNAL CLOCK INPUT (FSYNC) External Input Clock Acquisition Time tFSYNC External Input Clock Frequency 1 (Note 3) External Input Clock High Threshold VFSYNC_HI VFSYNC rising External Input Clock Low Threshold VFSYNC_LO VFSYNC falling FSYNC Pulldown Resistance Soft-Start Time fOSC + 10% Hz 1.4 V 0.4 RFSYNC tSS Cycles V 500 kI fSW = 400kHz 4 ms fSW = 1.0MHz 1.6 ms ENABLE INPUT (EN) Enable On Threshold Voltage Low VEN_LO Enable On Threshold Voltage High VEN_HI Maxim Integrated 0.8 2.2 V V   3 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current ELECTRICAL CHARACTERISTICS (continued) (VSUP = VSUPSW = 14V, VEN = 14V, L1 = 22FH, CIN = 4.7FF, COUT = 100FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 61.9kI, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = +25NC.) PARAMETER Enable Threshold Voltage Hysteresis Enable Input Current SYMBOL CONDITIONS MIN TYP MAX UNITS VEN,HYS 0.2 V IEN 10 nA RESET Reset Internal Switching Level RESETI Threshold Voltage VTH_RISING VFB rising, VRESETI = 0V VTH_FALLING VFB falling, VRESETI = 0V CRES Threshold Voltage VRESETI_HI VRESETI falling VCRES_HI VCRES rising CRES Threshold Hysteresis VCRES_HYS RESETI Input Current CRES Source Current CRES Pulldown Current IRESET VRESETI = 0V ICRES VOUT in regulation ICRES_PD RES Output Low Voltage 95 96.5 91 93 95 1.05 1.25 1.4 V 1.07 1.13 1.19 V 9.5 VOUT out of regulation VOUT in regulation V 0.02 FA 10 tRES_DEB VRESETI falling 10.5 1 FA mA TA = +25°C TA = +125°C %VFB 0.05 ISINK = 5mA RES Leakage Current (OpenDrain Output) Reset Debounce Time 93 0.4 V 1 FA 20 nA 25 Fs Note 2: When 3V < VOUT < 5.5V, the bias regulator is connected to the output to save quiescent current, VBIAS = VOUT. Note 3: Guaranteed by design; not production tested. Maxim Integrated   4 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current Typical Operating Characteristics (VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 61.9kI, TA = +25NC, unless otherwise noted.) STARTUP WITH FULL LOAD (VOUT = 1.25V, fSW = 400kHz) EFFICIENCY vs. LOAD CURRENT (BUCK) MAX16976 toc01 90 VEN 5V/div VRES 5V/div 80 EFFICIENCY (%) VOUT 1V/div MAX16976 toc02 100 70 60 VOUT = 3.3V VOUT = 8V 50 40 VOUT = 5V VOUT = 1.25V 30 20 10 ILOAD = 600mA 0 0 2ms/div 400 407 406 405 404 403 402 500 600 5V OUTPUT 1000 MAX16976 toc04 1200 SWITCHING FREQUENCY (kHz) 408 300 SWITCHING FREQUENCY vs. RFOSC MAX16976 toc03 SWITCHING FREQUENCY (kHz) 409 200 LOAD CURRENT (mA) SWITCHING FREQUENCY vs. LOAD CURRENT 410 100 800 600 400 401 400 200 0 200 300 400 500 600 10 32 54 76 98 LOAD CURRENT (mA) RFOSC (kI) SWITCHING FREQUENCY vs. TEMPERATURE (1.25V/400kHz, 5V/400kHz) LOAD-STEP RESPONSE (1.25V/400kHz) 120 MAX16976 toc06 ILOAD = 600mA, RFOSC = 64.87kI 430 MAX16976 toc05 450 SWITCHING FREQUENCY (kHz) 100 ILOAD 500mA/div 5V/400kHz 410 390 1.25V/400kHz VOUT 100mV/div 370 350 -40 -25 -10 5 20 35 50 65 80 95 110 125 1ms/div TEMPERATURE (°C) Maxim Integrated   5 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current Typical Operating Characteristics (continued) (VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 61.9kI, TA = +25NC, unless otherwise noted.) COLD-CRANK PULSE (1.25V/400kHz) DIPS AND DROPS TEST (1.25V/400kHz) MAX16976 toc07 MAX16976 toc08 VSUPSW 10V/div 0V VSUPSW 10V/div 0V VOUT 1V/div 0V VOUT 1V/div 0V VLX 10V/div 0V 0V VRES 5V/div 0V VLX 10V/div VRES 5V/div 0V 10ms/div 10ms/div SLOW VIN RAMP-UP TEST OUTPUT SHORT-CIRCUIT TEST (1.25V/400kHz) MAX16976 toc09 MAX16976 toc10 VSUP/SUPSW 10V/div VOUT 10V/div VOUT 5V/div ILX 1A/div VLX 10V/div VLX 10V/div 2ms/div QUIESCENT CURRENT vs. INPUT VOLTAGE OUTPUT VOLTAGE ERROR vs. TEMPERATURE MAX16976 toc11 2.0 80 70 1.5 1.0 ERROR (%) 60 50 40 30 0 -0.5 -1.5 10 0 4 8 12 16 20 INPUT VOLTAGE (V) Maxim Integrated 0.5 -1.0 5V/400kHz 20 VOUT = 5V ILOAD = 600mA PWM MODE MAX16976 toc12 10s/div 90 QUIESCENT CURRENT (µA) VRES 5V/div 24 28 -2.0 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C)   6 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current Typical Operating Characteristics (continued) (VSUP = VSUPSW = 14V, VEN = 14V, L1 = 4.7FH, CIN = 4.7FF, COUT = 22FF, CBIAS = 1FF, CBST = 0.1FF, CCRES = 1nF, RFOSC = 61.9kI, TA = +25NC, unless otherwise noted.) ERROR (%) 1.0 0.5 0 -0.5 -1.0 2 OUTPUT VOLTAGE CHANGE (%) 1.5 VOUT = 1.25V ILOAD = 600mA PWM MODE MAX16976 toc13 2.0 ILOAD = 0A, SKIP MODE 5V/400kHz 1 MAX16976 toc14 VOUT vs. TEMPERATURE IN SKIP MODE (5V/400kHz) OUTPUT VOLTAGE ERROR vs. TEMPERATURE 0 -1 -1.5 -2.0 -2 -40 -25 -10 5 20 35 50 65 80 95 110 125 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) TEMPERATURE (°C) VOUT vs. TEMPERATURE IN SKIP MODE (1.25V/400kHz) 0 5.05 5.00 4.95 -1 4.90 -2 -40 -25 -10 5 20 35 50 65 80 95 110 125 TEMPERATURE (°C) Maxim Integrated 5V/400kHz VOUT (V) 1 MAX16976 toc16 ILOAD = 0A, SKIP MODE LINE REGULATION 5.10 MAX16976 toc15 OUTPUT VOLTAGE CHANGE (%) 2 6 8 10 12 14 16 18 20 22 24 26 28 VSUPSW (V)   7 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current Pin Configuration TOP VIEW CRES 1 + 16 RESETI FOSC 2 FSYNC 3 15 RES MAX16976 I.C. 4 14 EN 13 SUPSW COMP 5 12 LX FB 6 11 SUP OUT 7 10 BST GND 8 9 BIAS QSOP Pin Description PIN NAME FUNCTION 1 CRES Analog Reset Timer. CRES sources 10FA (typ) of current into an external capacitor to set the reset timeout period. Reset timeout period is defined as the time between the start of output regulation and RES switching to high impedance. Leave CRES unconnected for minimum delay time. 2 FOSC Resistor-Programmable Switching Frequency Control Input. Connect a resistor from FOSC to GND to set the switching frequency (see the Internal Oscillator section). 3 FSYNC Synchronization Input. The device synchronizes to an external signal applied to FSYNC. The external signal period must be 10% shorter than the internal clock period for proper operation. 4 I.C. 5 COMP Error-Amplifier Output. Connect a compensation network from COMP to GND for stable operation. See the Compensation Network section. 6 FB Feedback Input. Connect an external resistive divider from FB to OUT and GND to set the output voltage between 1V and 10V. Connect FB directly to BIAS to set the output voltage to 5V. See the Applications Information section. 7 OUT Connect OUT to the output of the converter. OUT provides power to the internal circuitry when the output voltage of the converter is set between 3V and 5.6V. During shutdown, OUT is pulled to GND with a 50I resistor. 8 GND Ground 9 BIAS Linear Regulator Output. BIAS powers the internal circuitry. Bypass BIAS with a 1FF capacitor to ground as close as possible to the device. During shutdown, BIAS is actively discharged through a 32kI resistor. 10 BST High-Side Driver Supply. Connect a 0.1FF capacitor between LX and BST for proper operation. 11 SUP Voltage Supply Input. SUP powers the internal linear regulator. Connect a 4.7FF capacitor from SUP to ground. Connect SUP to SUPSW. 12 LX Maxim Integrated Internally Connected. Connect to GND. Inductor Connection. Connect a rectifying Schottky diode between LX and GND. Connect an inductor from LX to the output.   8 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current Pin Description (continued) PIN NAME FUNCTION Internal High-Side Switch Supply Input. SUPSW provides power to the internal switch. Connect a 4.7FF capacitor from SUPSW to ground. Connect SUP to SUPSW. See the Input Capacitor section. 13 SUPSW 14 EN Battery-Compatible Enable Input. Drive EN low to disable the device. Drive EN high to enable the device. 15 RES 16 RESETI Open-Drain Active-Low Reset Output. RES asserts when VOUT is below the reset threshold set by RESETI. Reset Threshold Level Input. Connect to a resistive divider to set the reset threshold for RES. Connect RESETI to GND to enable the internal reset threshold. Functional Diagram SUP BIAS FOSC BST SUPSW DRV ISENSE LX LEVEL SHIFT FSYNC OUT EN STANDBY SUPPLY OSC SUM REF EA LDO ILIM PWM COMP LOGIC MUX UVLO COMP LOGIC FOR 100% DUTY-CYCLE OPERATION VBIAS SOFTSTART FB 10µA CRES RESETI COMP B.G. REF COMP RES MAX16976 GND Maxim Integrated   9 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current Detailed Description The MAX16976 is a constant-frequency, current-mode automotive buck converter with an integrated high-side switch. The device operates with input voltages from 3.5V to 28V and tolerates input transients up to 42V. During undervoltage events, such as cold-crank conditions, the internal pass device maintains 93.5% duty cycle for a short time. An open-drain, active-low reset output helps to monitor the output voltage. The device offers an adjustable reset threshold that helps to keep microcontrollers alive down to the lowest specified input voltage and a capacitorprogrammable reset timeout to ensure proper startup. The switching frequency is resistor-programmable from 220kHz to 1.0MHz to allow optimization for efficiency, noise, and board space. A clock input, FSYNC, allows the device to synchronize to an external clock. During light-load conditions, the device enters skip mode that reduces the quiescent current down to 45FA and increases light-load efficiency. The 5V fixed output voltage eliminates the need for external resistors and reduces the supply current by up to 50FA. Linear Regulator Output (BIAS) The device includes a 5V linear regulator, VBIAS, that provides power to the internal circuitry. Connect a 1FF ceramic capacitor from BIAS to GND. When the output voltage is set between 3V and 5.5V, the internal linear regulator only provides power until the output is in regulation. The internal linear regulator turns off once the output is in regulation and allows OUT to provide power to the device. The internal regulator turns back on once the external load on the output of the device is higher than 100mA. In addition, the linear regulator turns on anytime the output voltage is outside the 3V to 5.5V range. The device also offers a capacitor-programmable reset timeout period. Connect a capacitor from CRES to GND to adjust the reset timeout period. When the output voltage goes out of regulation, RES asserts low and the reset timing capacitor discharges with a 1mA pulldown current. Once the output is back in regulation the reset timing capacitor recharges with 10FA (typ) current. RES stays low until the voltage at CRES reaches 1.13V (typ). Dropout Operation The device features an effective maximum duty cycle to help refresh the BST capacitor when continuously operated in dropout. When the high-side switch is on for three consecutive clock cycles, the device forces the high-side switch off during the final 35% of the fourth clock cycle. When the high-side switch is off, the LX node is pulled low by the current flowing through the inductor. This increases the voltage across the BST capacitor. To ensure that the inductor has enough current to pull LX to ground, an internal load sinks current from VOUT when the device is close to dropout and external load is small. Once the input voltage is increased above the dropout region, the device continues to regulate at the set output voltage. The device operates with no load and no external clock at an effective maximum duty cycle of 93.5% in deep dropout. This effective maximum duty cycle is influenced by the external load and by the optional external synchronized clock. System Enable (EN) External Clock Input (FSYNC) An enable-control input (EN) activates the device from the low-power shutdown mode. EN is compatible with inputs from the automotive battery level down to 3.3V. The high-voltage compatibility allows EN to be connected to SUP, KEY/KL30, or the INH inputs of a CAN transceiver. Adjustable Reset Level EN turns on the internal regulator. Once VBIAS is above the internal lockout level, VUVL = 3.15V (typ), the controller activates and the output voltage ramps up within 2048 cycles of the switching frequency. The device synchronizes to an external clock signal applied at FSYNC. The signal at FSYNC must have a frequency of 10% higher than the internal clock frequency for proper synchronization. The device features a programmable reset threshold using a resistive divider between OUT, RESETI, and GND. Connect RESETI to GND for the internal threshold. RES asserts low when the output voltage falls to 93% of the programmed level. RES deasserts when the output voltage goes above 95% of the set voltage. Maxim Integrated Some microprocessors accept a wide input voltage range (3.3V to 5V, for example) and can operate during dropout of the device. Use a resistive divider at RESETI to adjust the reset activation level (RES goes low) to lower levels. The reference voltage at RESETI is 1.25V (typ). A logic-low at EN shuts down the device. During shutdown, the internal linear regulator and gate drivers turn off. Shutdown mode reduces the quiescent current to 9FA (typ). Drive EN high to turn on the device.   10 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current VOUT RFB1 RESETI RFB2 MAX16976 FB RFB3 Figure 1. Output Voltage/Reset Threshold Resistive Divider Network Overvoltage Protection Applications Information Output Voltage/Reset Threshold Resistive Divider Network Although the device’s output voltage and reset threshold can be set individually, Figure 1 shows a combined resistive divider network to set the desired output voltage and the reset threshold using three resistors. Use the following formula to determine the RFB3 of the resistive divider network: × VREF R R FB3 = TOTAL V OUT where VREF = 1V, RTOTAL = selected total resistance of RFB1, RFB2, and RFB3 in ohms, and VOUT is the desired output voltage in volts. The device includes overvoltage protection circuitry that protects the device when there is an overvoltage condiUse the following formula to calculate the value of RFB2 tion at the output. If the output voltage increases by more of the resistive divider network: than 12% of its set voltage, the device stops switching. The device resumes regulation once the overvoltage condition is removed. R TOTAL × VREF_RES = R FB2 − R FB3 V RES Overload Protection The overload protection circuitry is activated when the where VREF_RES is 1.25V (see the Electrical Characteristics device is in current limit and VOUT is below the reset table) and VRES is the desired reset threshold in volts. threshold. Under these conditions, the device enters a soft-start mode. When the overcurrent condition is The precision of the reset threshold function is depenremoved before the soft-start mode is over, the device dent on the tolerance of the resistors used for the divider. regulates the output voltage to the set value. Otherwise, BST Capacitor Selection the soft-start cycle repeats until the overcurrent condition for Dropout Operation is removed. The device includes an internal boost capacitor refresh Skip Mode algorithm for dropout operation. This is required to ensure During light-load operation, IINDUCTOR P 100mA, the proper boost capacitor voltage that delivers power to the device enters skip-mode operation. Skip mode turns off gate-drive circuitry. When the HSFET is on consecutively the internal switch and allows the output to drop below for 3.65 clock cycles, the internal counter detects this regulation voltage before the switch is turned on again. and turns off the HSFET for 0.35 clock cycles. This is of The lower the load current, the longer it takes for the particular concern when VIN is falling and approaching regulator to initiate a new cycle effectively increasing VOUT at the minimum switching frequency (220kHz). light-load efficiency. During skip mode, the device quiThe worst-case condition for boost capacitor refresh time escent current drops to as low as 45FA. is with no load on the output. For the boost capacitor to recharge completely, the LX node must be pulled to Overtemperature Protection ground. If there is no current through the inductor then Thermal-overload protection limits the total power disthe LX node does not go to ground. To solve this issue, sipation in the device. When the junction temperature an internal load of about 100mA turns on at the sixth exceeds +175NC (typ), an internal thermal sensor shuts clock cycle, which is determined by a separate counter. down the step-down controller, allowing the device to cool. The thermal sensor turns on the device again after In the worst-case condition with no load, the LX node the junction temperature cools by +15NC. does not go below ground during the first detect of the Maxim Integrated   11 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current 3.65 clock cycles. The device waits for the next 3.65 clock cycles to finish. As a result, the soonest the LX node can go below ground is 4 + 3.65 = 7.65 clock cycles. This time does not factor in the size of the inductor and the time it takes for the inductor current to build up to 100mA (internal load). No load minimum time before refresh is: ∆T (no load) = 7.65 clock cycles = 7.65 x 4.54μs (at 220kHz) = 34.73μs Assuming a full 100mA is needed to refresh the BST capacitor and depending on the size of the inductor, the time it takes to build up full 100mA in the inductor is given by: ∆T (inductor) = L x ∆I/∆V (current build-up starts from the sixth clock cycle) L = inductor value chosen in the design guide. ∆I is the required current = 100mA. ∆V = voltage across the inductor (assume this to be 0.5V), which means VIN is greater than VOUT by 0.5V. If ∆T (inductor) < 7.65 – 6 (clock cycles) then the BST capacitor is sized as follows: BST_CAP ≥ I_BST(dropout) x ∆T (no load)/∆V (BST capacitor) ∆T (no load) = 7.65 clock cycles = 34.73μs. ∆V (BST capacitor), for (3.3V to 5V) output = VOUT – 2.7V (2.7V is the minimum voltage allowed on the BST capacitor). If ∆T (inductor) > 7.65 - 6 clock cycles then we need to wait for the next count of 3.65 clock cycles making ∆T (no load) = 11.65 clock cycles. Assume ∆T (no load) to be 16 clock cycles when designing the BST capacitor with a typical inductor value for 220kHz operation. RESET_TIMEOUT = 1.13V × C (s) 10µA where C is the capacitor from CRES to GND in Farads. Internal Oscillator The device’s internal oscillator is programmable from 220kHz to 1.0MHz using a single resistor at FOSC. Use the following formula to calculate the switching frequency: fOSC (Hz) ≈ 26.4 × 10 9 (Ω x Hz) R where R is the resistor from FOSC to GND in ohms. For example, a 220kHz switching frequency is set with RFOSC = 120kI. Higher frequencies allow designs with lower inductor values and less output capacitance. Consequently, peak currents and I2R losses are lower at higher switching frequencies, but core losses, gatecharge currents, and switching losses increase. Inductor Selection Three key inductor parameters must be specified for operation with the device: inductance value (L), inductor saturation current (ISAT), and DC resistance (RDCR). To select inductance value, the ratio of inductor peak-topeak AC current to DC average current (LIR) must be selected first. A good compromise between size and loss is a 30% peak-to-peak ripple current to average-current ratio (LIR = 0.3). The switching frequency, input voltage, output voltage, and selected LIR then determine the inductor value as follows: V (V -V ) L = OUT SUPSW OUT VSUPSW fSWIOUTLIR where: where VSUPSW, VOUT, and IOUT are typical values (so that efficiency is optimum for typical conditions). The switching frequency is set by RFOSC. The exact inductor value is not critical and can be adjusted to make tradeoffs among size, cost, efficiency, and transient response requirements. Table 1 shows a comparison between small and large inductor sizes. I_BST (dropout) = 2.5mA (worst case) Table 1. Inductor Size Comparison The final BST_CAP equation is: BST_CAP = I_BST (dropout) x ∆T (no load)/∆V (BST capacitor) ∆T (no load) = 16 clock cycles ∆V (BST capacitor) = VOUT - 2.7V Reset Timeout Period The device offers a capacitor-adjustable reset timeout period. CRES can source 10FA of current. Use the following formula to set the timeout period. Maxim Integrated INDUCTOR SIZE SMALLER LARGER Lower price Smaller ripple Smaller form-factor Higher efficiency Faster load response Larger fixed-frequency range in skip mode   12 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current The inductor value must be chosen so that the maximum inductor current does not reach the minimum current limit of the device. The optimum operating point is usually found between 15% and 35% ripple current. When pulse skipping (light loads), the inductor value also determines the load-current value at which PFM/PWM switchover occurs. Find a low-loss inductor having the lowest possible DC resistance that fits in the allotted dimensions. Most inductor manufacturers provide inductors in standard values, such as 1.0FH, 1.5FH, 2.2FH, 3.3FH, etc. Also look for nonstandard values, which can provide a better compromise in LIR across the input voltage range. If using a swinging inductor (where the no-load inductance decreases linearly with increasing current), evaluate the LIR with properly scaled inductance values. For the selected inductance value, the actual peak-to-peak inductor ripple current (DIINDUCTOR) is defined by: VOUT (VSUPSW - VOUT ) ∆IINDUCTOR = VSUPSW × fSW × L where DIINDUCTOR is in A, L is in H, and fSW is in Hz. Ferrite cores are often the best choices, although powdered iron is inexpensive and can work well at 220kHz. The core must be large enough not to saturate at the peak inductor current (IPEAK): = IPEAK ILOAD(MAX) + ∆IINDUCTOR 2 Input Capacitor The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit’s switching. The input capacitor RMS current requirement (IRMS) is defined by the following equation: IRMS = ILOAD(MAX) VOUT (VSUPSW - VOUT ) VSUPSW IRMS is at a maximum value when the input voltage equals twice the output voltage (VSUPSW = 2VOUT), so IRMS(MAX) = ILOAD(MAX)/2. Choose an input capacitor that exhibits less than +10NC self-heating temperature rise at the RMS input current for optimal long-term reliability. Maxim Integrated The input-voltage ripple comprises DVQ (caused by the capacitor discharge) and DVESR (caused by the ESR of the capacitor). Use low-ESR ceramic capacitors with high ripple-current capability at the input. Assume the contribution from DVQ and DVESR to be 50%. Calculate the input capacitance and ESR required for a specified input-voltage ripple using the following equations: ESRIN = ∆VESR ∆I IOUT + L 2 where: (V -V ) × VOUT ∆IL = SUPSW OUT VSUPSW × fSW × L and I × D(1- D) CIN = OUT ∆VQ × fSW and D= VOUT VSUPSW IOUT is the maximum output current and D is the duty cycle. Output Capacitor The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and load transient requirements, yet have high enough ESR to satisfy stability requirements. The output capacitance must be high enough to absorb the inductor energy while transitioning from full-load to no-load conditions without tripping the overvoltage fault protection. When using high-capacitance, low-ESR capacitors, the filter capacitor’s ESR dominates the output voltage ripple. So the size of the output capacitor depends on the maximum ESR required to meet the output voltage ripple (VRIPPLE(P-P)) specifications: VRIPPLE(P −P) = ESR × ILOAD(MAX) × LIR The actual capacitance value required relates to the physical size needed to achieve low ESR, as well as to the chemistry of the capacitor technology. Thus, the capacitor is usually selected by ESR and voltage rating rather than by capacitance value.   13 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current When using low-capacity filter capacitors, such as ceramic capacitors, size is usually determined by the capacity needed to prevent VSAG and VSOAR from causing problems during load transients. Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem. However, low-capacity filter capacitors typically have high-ESR zeros that can affect the overall stability. Rectifier Selection The device requires an external Schottky diode rectifier as a freewheeling diode. Connect this rectifier close to the device using short leads and short PCB traces. Choose a rectifier with a continuous current rating higher than the highest output current-limit threshold (1.25A) and with a voltage rating higher than the maximum expected input voltage, VSUPSW. Use a low forward-voltage-drop Schottky rectifier to limit the negative voltage at LX. Avoid higher than necessary reverse-voltage Schottky rectifiers that have higher forward-voltage drops. Compensation Network The device uses an internal transconductance error amplifier with its inverting input and its output available for external frequency compensation. The output capacitor and compensation network determine the loop stability. The inductor and the output capacitor are chosen based on performance, size, and cost. Additionally, the compensation network optimizes the control-loop stability. The controller uses a current-mode control scheme that regulates the output voltage by forcing the required current through the external inductor, so the device uses the voltage drop across the high-side MOSFET. Currentmode control eliminates the double pole in the feedback loop caused by the inductor and output capacitor resulting in a smaller phase shift and requiring less elaborate error-amplifier compensation than voltage-mode control. A simple single series resistor (RC) and capacitor (CC) are all that is required to have a stable, high-bandwidth loop in applications where ceramic capacitors are used for output filtering (Figure 2). For other types of capacitors, due to the higher capacitance and ESR, the frequency of the zero created by the capacitance and ESR is lower than the desired closed-loop crossover frequency. To stabilize a nonceramic output capacitor loop, add another compensation capacitor (CF) from COMP to GND to cancel this ESR zero. The basic regulator loop is modeled as a power modulator, output feedback divider, and an error amplifier. The power modulator has a DC gain set by gmc O RLOAD, with a pole and zero pair set by RLOAD, the output capacitor (COUT), and its ESR. The following equations allow to approximate the value for the gain of the power modulator (GAINMOD(dc)), neglecting the effect of the ramp stabilization. Ramp stabilization is necessary when the duty cycle is above 50% and is internally done for the device. GAINMOD(dc) = g mc × R LOAD × fSW × L R LOAD + (fSW × L) where RLOAD = VOUT/ILOUT(MAX) in I, fSW is the switching frequency in MHz, L is the output inductance in FH, and gmc = 3S. In a current-mode step-down converter, the output capacitor, its ESR, and the load resistance introduce a pole at the following frequency: fpMOD = 1  R LOAD × fSW × L  2π × C OUT ×  + ESR  R LOAD + (fSW × L)  The output capacitor and its ESR also introduce a zero at: fzMOD = 1 2π × ESR × C OUT When COUT is composed of “n” identical capacitors in parallel, the resulting COUT = n O COUT(EACH) and ESR = ESR(EACH)/n. Note that the capacitor zero for a parallel combination of alike capacitors is the same as for an individual capacitor. VOUT R1 COMP gm R2 VREF RC CF CC Figure 2. Compensation Network Maxim Integrated   14 MAX16976 28V, 600mA Automotive Step-Down Converter with Low Operating Current The feedback voltage-divider has a gain of GAINFB = VFB/VOUT, where VFB is 1V (typ). The transconductance error amplifier has a DC gain of GAINEA(DC) = gm,EA O ROUT,EA, where gm,EA is the error-amplifier transconductance, which is 1000FS (typ), and ROUT,EA is the output resistance of the 50MI error amplifier. A dominant pole (fdpEA) is set by the compensation capacitor (CC) and the amplifier output resistance (ROUT,EA). A zero (fzEA) is set by the compensation resistor (RC) and the compensation capacitor (CC). There is an optional pole (fpEA) set by CF and RC to cancel the output capacitor ESR zero if it occurs near the crossover frequency (fC, where the loop gain equals 1 (0dB)). Therefore: GAINMOD(fC) × Solving for RC: RC = VOUT g m,EA × VFB × GAINMOD(fC) Set the error-amplifier compensation zero formed by RC and CC (fzEA) at the fpMOD. Calculate the value of CC as follows: CC = Thus: fdpEA = 1 2π × C C × (R OUT,EA + R C ) fzEA = 1 2π × C C × R C The loop-gain crossover frequency (fC) is set below 1/5th the switching frequency and much higher than the power-modulator pole (fpMOD): The total loop gain as the product of the modulator gain, the feedback voltage-divider gain, and the error-amplifier gain at fC is equal to 1. So: For the case where fzMOD is greater than fC: GAIN = MOD(fC ) GAINMOD(dc) × fpMOD fC fpMOD fzMOD The error-amplifier gain at fC is: f GAINEA(f = g m,EA × R C × zMOD C) fC Therefore: GAINMOD(f ) × C GAINEA(fC) = g m,EA × R C Maxim Integrated 1 2π × fzMOD × R C As the load current decreases, the modulator pole also decreases; however, the modulator gain increases accordingly and the crossover frequency remains the same. For the case where fzMOD is less than fC: VFB × GAINEA(f ) = 1 C VOUT GAIN = MOD(fC) GAINMOD(dc) × 2π × fpMOD × R C The power-modulator gain at fC is: f fpMOD