EVALUATION KIT AVAILABLE
MAX17016
General Description
The MAX17016 pulse-width-modulation (PWM) controller
provides high efficiency, excellent transient response, and
high DC-output accuracy needed for stepping down highvoltage batteries to generate low-voltage core or chipset/
RAM bias supplies in notebook computers. Combined
with low on-resistance MOSFETs (6mΩ low-side MOSFET
and 12mΩ high-side MOSFET), the MAX17016 provides
a highly efficient and compact solution for small form factor applications that need a high-power density.
Maxim’s proprietary Quick-PWM™ quick-response, constant-on-time PWM control scheme handles wide input/
output voltage ratios (low-duty-cycle applications) with
ease and provides 100ns “instant-on” response to load
transients while maintaining a relatively constant switching frequency. The output voltage can be dynamically
controlled using the dynamic REFIN, which supports
input voltages between 0 to 2V. The REFIN adjustability
combined with a resistive voltage-divider on the feedback
input allows the MAX17016 to be configured for any output voltage between 0 to 0.9VIN.
The controller senses the current across the 6mΩ synchronous rectifier to achieve a low-cost and highly efficient valley current-limit protection. External current-limit
control is still provided to allow higher current-limit settings for applications with heatsinks and air flow, or for
lower current applications that need lower current-limit
settings to avoid overdesigning the application circuit. The
adjustable current limit provides a high degree of flexibility, allowing thermally compensated protection or foldback
current-limit protection using a voltage-divider partially
derived from the output.
The MAX17016 includes a voltage-controlled soft-start
and soft-shutdown in order to limit the input surge
current, provide a monotonic power-up (even into a
precharged output), and provide a predictable powerup time. The controller also includes output fault protection—undervoltage and overvoltage protection—as well
as thermal-fault protection.
The MAX17016 is available in a small 40-pin, 6mm x
6mm, 2W TQFN package.
Quick-PWM is a trademark of Maxim Integrated Products, Inc.
19-2771; Rev 1; 4/17
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Applications
●●
●●
●●
●●
●●
●●
Notebook Computers
I/O and Chipset Supplies
GPU Core Supply
DDR Memory—VDDQ or VTT
Point-of-Load Applications
Step-Down Power Supply
Benefits and Features
●● Quick-PWM with Fast Transient Response
●● 6mΩ, 26V Low-Side MOSFET
●● 12mΩ, 26V High-Side MOSFET
●● Supports Any Output Capacitor
• No Compensation Required with
Polymers/Tantalum
• Stable with Ceramic Output Capacitors Using
External Compensation
●● Precision 2V ±10mV Reference
●● Dynamically Adjustable Output Voltage
(0 to 0.9 VIN Range)
• Feedback Input Regulates from 0 to 2V REFIN
Voltage
• 0.5% VOUT Accuracy Over Line and Load
●● 26V Maximum Input Voltage Rating
●● Adjustable Valley Current-Limit Protection
• Thermal Compensation with NTC
• Supports Foldback Current Limit
●● Resistively Programmable Switching Frequency
●● Overvoltage Protection
●● Undervoltage/Thermal Protection
●● Voltage Soft-Start and Soft-Shutdown
●● Monotonic Power-Up with Precharged Output
●● Power-Good Window Comparator
Ordering Information appears at end of data sheet.
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Absolute Maximum Ratings
IN to PGND............................................................-0.3V to +28V
TON to AGND.........................................................-0.3V to +28V
VDD to AGND...........................................................-0.3V to +6V
VCC to AGND............................................ -0.3V to (VDD + 0.3V)
EN, SKIP, PGOOD to AGND...................................-0.3V to +6V
REF, REFIN to AGND............................... -0.3V to (VCC + 0.3V)
ILIM, FB to AGND..................................... -0.3V to (VCC + 0.3V)
GND to PGND.......................................................-0.3V to +0.3V
LX to PGND..............................................................-1V to +28V
BST to PGND............................................. (VDD - 0.3V) to +34V
BST to LX.................................................................-0.3V to +6V
BST to VDD............................................................-0.3V to +28V
REF Short Circuit to AGND........................................Continuous
IN RMS Current Rating (continuous).....................................15A
PGND RMS Current Rating (continuous)..............................20A
Continuous Power Dissipation (TA = +70°C)
40-Pin, 6mm x 6mm TQFN (T4066-MCM)
(derated 27mW/°C above +70°C)..............................2162mW
Operating Temperature Range (extended)......... -40°C to +85°C
Junction Temperature Range...........................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Package Information
PACKAGE TYPE: 40 TQFN
Package Code
T4066M+1
Outline Number
21-0177
Land Pattern
90-0085
For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”,
“#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing
pertains to the package regardless of RoHS status.
Electrical Characteristics
(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise
specified. Typical values are at TA = +25°C.) (Note 1)
PARAMETER
PWM CONTROLLER
Input Voltage Range
SYMBOL
CONDITIONS
TYP
MAX
26
V
IDD + ICC
FB forced above REFIN
0.7
1.2
mA
Shutdown Supply Current (VDD)
ISHDN
EN = GND, TA = +25°C
0.1
2
µA
On-Time
Minimum Off-Time
2
UNITS
Quiescent Supply Current (VDD)
VDD-to-VCC Resistance
VIN
MIN
RCC
tON
tOFF(MIN)
TON Shutdown Supply Current
20
RTON = 97.5kΩ
123
164
205
RTON = 200kΩ
275
303
331
RTON = 302.5kΩ
379
442
505
(Note 2)
225
350
ns
EN = GND, VTON = 26V,
VCC = 0V or 5V, TA = +25°C
0.01
1
µA
VIN = 12V,
VFB = 1.0V
(Note 2)
REFIN Voltage Range
VREFIN
(Note 3)
REFIN Input Current
IREFIN
TA = +25°C, REFIN = 0.5V to 2V
FB Voltage Range
www.maximintegrated.com
VFB
Ω
(Note 3)
ns
0
VREF
V
-50
+50
mA
0
VREF
V
Maxim Integrated │ 2
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Electrical Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise
specified. Typical values are at TA = +25°C.) (Note 1)
PARAMETER
FB Voltage Accuracy
SYMBOL
VFB
CONDITIONS
VREFIN = 0.5V,
measured at FB,
VIN = 2V to 26V,
SKIP = VDD
VREFIN = 1.0V
VREFIN = 2.0V
FB Input Bias Current
IFB
MIN
TYP
MAX
TA = +25°C
0.495
0.5
0.505
TA = 0°C to +85°C
0.493
TA = +25°C
0.995
TA = 0°C to +85°C
0.993
TA = 0°C to +85°C
1.990
VFB = 0.5V to 2.0V, TA = +25°C
0.507
1.0
UNITS
V
1.005
1.007
2.0
2.010
-0.1
+0.1
FB Output Low Voltage
ISINK = 3mA
Load-Regulation Error
SKIP = VDD, ILOAD = 0.1A to 10A
0.1
%
Line-Regulation Error
VCC = 4.5V to 5.5V, VIN = 2V to 26V
0.2
%
Soft-Start/Soft-Stop Slew Rate
Dynamic REFIN Slew Rate
SSSR
DYNSR
0.4
µA
Rising/falling edge on EN
Rising edge on REFIN
V
0.4
1.2
2.2
mV/µs
3
9.45
18
mV/µs
REFERENCE
Reference Voltage
No load
1.990
IREF = -10µA to +50µA
1.98
With respect to the internal target voltage
(error comparator threshold); rising edge;
hysteresis = 50mV
250
VREF
VCC = 4.5V
to 5.5V
OVP
2.00
2.010
2.02
V
FAULT DETECTION
Output Overvoltage-Protection
Trip Threshold
Dynamic transition
300
350
VREF + 0.30
Minimum OVP threshold
mV
V
0.7
Output Overvoltage
Fault-Propagation Delay
tOVP
FB forced 25mV above trip threshold
Output Undervoltage-Protection
Trip Threshold
UVP
With respect to the internal target voltage
(error comparator threshold) falling edge;
hysteresis = 50mV
-240
-200
-160
mV
Output Undervoltage
Fault-Propagation Delay
tUVP
FB forced 25mV below trip threshold
100
200
350
µs
PGOOD Propagation Delay
tPGOOD
UVP falling edge, 25mV overdrive
5
OVP rising edge, 25mV overdrive
5
Startup delay
PGOOD Output-Low Voltage
PGOOD Leakage Current
Dynamic REFIN Transition FaultBlanking Threshold
www.maximintegrated.com
5
100
200
ISINK = 3mA
IPGOOD
FB = REFIN (PGOOD high impedance),
PGOOD forced to 5V, TA = +25°C
Fault blanking initiated; REFIN deviation
from the internal target voltage (error
comparator threshold); hysteresis = 10mV
±50
µs
µs
350
0.4
V
1
µA
mV
Maxim Integrated │ 3
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Electrical Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise
specified. Typical values are at TA = +25°C.) (Note 1)
PARAMETER
Thermal-Shutdown Threshold
VCC Undervoltage Lockout
Threshold
SYMBOL
TSHDN
CONDITIONS
MIN
Hysteresis = 15°C
TYP
MAX
160
Rising edge, PWM disabled below this
VUVLO(VCC)
level; hysteresis = 100mV
3.95
4.2
UNITS
°C
4.45
V
0.4
VREF
V
-0.1
+0.1
µA
CURRENT LIMIT
ILIM Input Range
ILIM Input Bias Current
TA = +25°C, ILIM = 0.4V to 2V
Current-Limit Threshold
VILIMIT
Current-Limit Threshold
(Negative)
VINEG
Current-Limit Threshold
(Zero Crossing)
VZX
VILIM = 0.4V, VGND - VLX
18
20
22
ILIM = REF (2.0V), VGND - VLX
92
100
108
VILIM = 0.4V, VGND - VLX
VILIM = 0.4V,
VGND - VLX, SKIP = GND or open
Ultrasonic Frequency
SKIP = open (3.3V); VFB = VREFIN + 50mV
Ultrasonic Current-Limit
Threshold
SKIP = open (3.3V); VFB = VREFIN + 50mV,
VGND - VLX
18
mV
-24
mV
1
mV
30
kHz
-35
mV
POWER MOSFETS
Low-side MOSFET
enabled; VDD = 5V,
VFB = VREFIN + 50mV
Low-Side MOSFET
On-Resistance
6
7.5
TA = +85°C
6
7.5
High-side MOSFET enabled, VDD = 5V,
TA = +25°C
High-Side MOSFET
On-Resistance
Internal BST Switch
On-Resistance
TA = +25°C
RBST
mΩ
10
mΩ
IBST = 10mA, VDD = 5V
12
16
mΩ
IBST = 10mA, VDD = 5V
4
7
Ω
INPUTS AND OUTPUTS
EN Logic-Input Threshold
VEN
EN rising edge, hysteresis = 450mV (typ)
1.20
EN Logic-Input Current
IEN
EN forced to GND or VDD, TA = +25°C
-0.5
High (5V VDD)
SKIP Quad-Level Input
Logic Levels
VSKIP
www.maximintegrated.com
ISKIP
2.20
V
+0.5
µA
VCC - 0.4
Open (3.3V)
3.0
3.6
Ref (2.0V)
1.7
2.3
Low (GND)
SKIP Logic-Input Current
1.7
SKIP forced to GND or VDD, TA = +25°C
V
0.4
-2
+2
µA
Maxim Integrated │ 4
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Electrical Characteristics
(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = -40°C to +85°C, unless otherwise
specified.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
MAX
UNITS
2
26
V
1.2
mA
PWM CONTROLLER
Input Voltage Range
Quiescent Supply Current (VDD)
On-Time
Minimum Off-Time
VIN
IDD + ICC
tON
FB forced above REFIN
VIN = 12V,
VFB = 1.0V
(Note 2)
RTON = 97.5kΩ
115
213
RTON = 200kΩ
270
336
RTON = 302.5kΩ
368
516
tOFF(MIN)
(Note 2)
VREFIN
(Note 3)
VFB
(Note 3)
VFB
Measured at FB, VREFIN = 0.5V
VIN = 2V to 26V, VREFIN = 1.0V
SKIP = VDD
V
= 2.0V
VREF
VDD = 4.5V to 5.5V
Output Overvoltage-Protection
Trip Threshold
OVP
Output Undervoltage-Protection
Trip Threshold
Output Undervoltage
Fault-Propagation Delay
REFIN Voltage Range
FB Voltage Range
FB Voltage Accuracy
ns
400
ns
0
VREF
V
V
0
VREF
0.49
0.51
0.99
1.01
1.985
2.015
1.985
2.015
V
With respect to the internal target voltage
(error comparator threshold) rising edge;
hysteresis = 50mV
250
350
mV
UVP
With respect to the internal target voltage
(error comparator threshold);
falling edge; hysteresis = 50mV
-240
-160
mV
tUVP
FB forced 25mV below trip threshold
80
400
µs
0.4
V
4.45
V
REFIN
V
REFERENCE
Reference Voltage
FAULT DETECTION
PGOOD Output-Low Voltage
VCC Undervoltage Lockout
Threshold
www.maximintegrated.com
ISINK = 3mA
VUVLO(VCC)
Rising edge, PWM disabled below this level,
hysteresis = 100mV
3.95
Maxim Integrated │ 5
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Electrical Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = -40°C to +85°C, unless otherwise
specified.) (Note 1)
PARAMETER
SYMBOL
CONDITIONS
MIN
MAX
UNITS
0.4
VREF
V
VILIM = 0.4V, VGND = VLX
17
23
ILIM = REF (2.0V), VGND = VLX
90
110
SKIP = open (3.3V), VFB = VREFIN + 50mV
17
CURRENT LIMIT
ILIM Input Range
Current-Limit Threshold
VILIMIT
Ultrasonic Frequency
mV
kHz
INPUTS AND OUTPUTS
EN Logic-Input Threshold
VEN
EN rising edge hysteresis = 450mV (typ)
High (5V VDD)
SKIP Quad-Level Input
Logic Levels
VSKIP
1.20
2.20
VCC - 0.4
Mid (3.3V)
3.0
3.6
Ref (2.0V)
1.7
2.3
Low (GND)
V
V
0.4
Note 1: Limits are 100% production tested at TA = +25°C. Maximum and minimum limits over temperature are guaranteed by
design and characterization.
Note 2: On-time and off-time specifications are measured from the 50% point to the 50% point at the unloaded LX node. The typical 25ns dead time that occurs between the high-side driver falling edge (high-side MOSFET turn-off) and the low-side
MOSFET turn-on) is included in the on-time measurement. Similarly, the typical 25ns dead time that occurs between the
low-side driver falling edge (low-side MOSFET turn-off) and the high-side driver rising edge (high-side MOSFET turn-on) is
included in the off-time measurement.
Note 3: The 0 to 0.5V range is guaranteed by design, not production tested.
www.maximintegrated.com
Maxim Integrated │ 6
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Typical Operating Characteristics
(MAX17016 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)
1.5V OUTPUT EFFICIENCY
vs. LOAD CURRENT
100
toc01
100
7V
EFFICIENCY (%)
20V
60
50
30
0.01
0.1
1
ULTRASONIC
MODE
20
10
0.01
0.1
PWM MODE
1
toc04
100
toc05
40
70
PWM MODE
60
50
ULTRASONIC
MODE
40
30
1
20
10
1.05
0.01
0.1
1
1.04
10
0
2
4
6
8
LOAD CURRENT (A)
SWITCHING FREQUENCY
vs. LOAD CURRENT
PWM MODE SWITCHING FREQUENCY
vs. INPUT VOLTAGE
SWITCHING FREQUENCY
vs. TEMPERATURE
SWITCHING FREQUENCY (kHz)
250
200
150
ULTRASONIC
MODE
0.1
SKIP MODE
1
LOAD CURRENT (A)
www.maximintegrated.com
10
380
370
360
350
340
330
320
310
300
290
280
270
260
250
240
toc08
ILOAD = 5A
NO LOAD
6
8
10
12
14
16
18
INPUT VOLTAGE (V)
toc06
PWM MODE
LOAD CURRENT (A)
toc07
10
SKIP MODE
LOAD CURRENT (A)
PWM MODE
0.01
8
ULTRASONIC
MODE
30
SKIP MODE
PWM MODE
0.1
OUTPUT VOLTAGE (V)
EFFICIENCY (%)
20V
50
50
6
1.06
SKIP MODE
90
60
100
4
1.05V OUTPUT VOLTAGE
vs. LOAD CURRENT
70
300
2
1.05V OUTPUT EFFICIENCY
vs. LOAD CURRENT
80
350
0
1.05V OUTPUT EFFICIENCY
vs. LOAD CURRENT
12V
400
1.50
10
LOAD CURRENT (A)
80
0.01
SKIP MODE
LOAD CURRENT (A)
7V
toc03
ULTRASONIC
MODE
1.51
LOAD CURRENT (A)
90
EFFICIENCY (%)
PWM MODE
50
30
SKIP MODE
PWM MODE
100
SWITCHING FREQUENCY (kHz)
60
40
40
0
70
390
SWITCHING FREQUENCY (kHz)
EFFICIENCY (%)
70
12V
1.5V OUTPUT VOLTAGE
vs. LOAD CURRENT
1.52
OUTPUT VOLTAGE (V)
80
80
20
toc02
SKIP MODE
90
90
20
1.5V OUTPUT EFFICIENCY
vs. LOAD CURRENT
20
22
24
10
toc09
ILOAD = 10A
380
370
360
350
ILOAD = 5A
-40
-20
0
20
40
60
80
100
TEMPERATURE (°C)
Maxim Integrated │ 7
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Typical Operating Characteristics (continued)
(MAX17016 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)
16.00
toc10
16
MAXIMUM OUTPUT CURRENT (A)
MAXIMUM OUTPUT CURRENT (A)
15.80
15.60
15.40
15.20
15.00
14.80
14.60
14.40
MAXIMUM OUTPUT CURRENT
vs. TEMPERATURE
toc11
6
9
12
18
21
24
PWM MODE
10
8
14
6
4
13
ULTRASONIC MODE
2
12
-40
-20
0
20
40
60
80
100
0
SKIP MODE
6
8
10
12
18
20
INPUT VOLTAGE (V)
NO-LOAD SUPPLY CURRENT (IIN)
vs. INPUT VOLTAGE
REF OUTPUT VOLTAGE
vs. LOAD CURRENT
SOFT-START WAVEFORM
(HEAVY LOAD)
ULTRASONIC MODE
1
SKIP MODE
0.1
toc14
2.005
REF OUTPUT VOLTAGE (V)
PWM MODE
5V
2.004
8
10
12
14
16
18
INPUT VOLTAGE (V)
www.maximintegrated.com
20
22
24
22
24
toc15
A
0
5V
2.003
0
1.5V
B
2.002
0
C
2.001
8A
D
0
6
16
TEMPERATURE (°C)
10
0.01
14
INPUT VOLTAGE (V)
toc13
100
IIN (mA)
15
toc12
12
15
14.20
14.00
NO-LOAD SUPPLY CURRENT (IBIAS)
vs. INPUT VOLTAGE
IBIAS (mA)
MAXIMUM OUTPUT CURRENT
vs. INPUT VOLTAGE
2.000
-10
0
10
20
30
LOAD CURRENT (µA)
40
50
A. EN, 5V/div
B. PGOOD, 5V/div
IOUT = 8A
200µs/div
C. VOUT, 1V/div
D. INDUCTOR CURRENT,
10A/div
Maxim Integrated │ 8
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Typical Operating Characteristics (continued)
(MAX17016 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)
SOFT-START WAVEFORM
(LIGHT LOAD)
SHUTDOWN WAVEFORM
toc16
5V
A
0
5V
0
toc17
5V
A
0
5V
B
B
0
1.5V
1.5V
C
0
C
0
8A
D
1A
0
A. EN, 5V/div
B. PGOOD, 5V/div
IOUT = 1A
8A
1A
D
0
200µs/div
C. VOUT, 1V/div
D. INDUCTOR CURRENT,
10A/div
LOAD-TRANSIENT RESPONSE
(SKIP MODE)
OUTPUT OVERLOAD WAVEFORM
B
8A
A
1A
1.5V
B
8A
C
200µs/div
C. VOUT, 1V/div
D. INDUCTOR CURRENT,
5A/div
20µs/div
A. IOUT, 10A/div
B. VOUT, 20mV/div
IOUT = 1A to 8A to 1A
C. INDUCTOR CURRENT,
5A/div
OUTPUT OVERVOLTAGE WAVEFORM
toc21
toc20
20A
A
1.5V
0
1.5V
toc18
0
A. EN, 5V/div
B. PGOOD, 5V/div
IOUT = 6A
toc19
A
LOAD-TRANSIENT RESPONSE
(PWM MODE)
A
0
1.5V
B
8A
C
0
0
5V
5V
C
0
20µs/div
A. IOUT, 10A/div
B. VOUT, 20mV/div
IOUT = 1A TO 8A to 1A C. INDUCTOR CURRENT,
5A/div
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200µs/div
B. VOUT, 1V/div
A. INDUCTOR CURRENT,
10A/div
C. PGOOD, 5V/div
IOUT = 2A to 20A
B
0
200µs/div
B. PGOOD, 5V/div
A. VOUT, 1V/div
IOUT = 2A to 20A
Maxim Integrated │ 9
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Typical Operating Characteristics (continued)
(MAX17016 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)
DYNAMIC OUTPUT-VOLTAGE TRANSITION
(PWM MODE)
DYNAMIC OUTPUT-VOLTAGE TRANSITION
(SKIP MODE)
toc23
toc22
1.5V
A
1.05V
1.5V
1.5V
1.05V
A
1.5V
B
1.05V
B
1.05V
0
C
-6A
12V
D
0
40µs/div
C. INDUCTOR CURRENT,
A. REFIN, 500mV/div
10A/div
B. VOUT, 200mV/div,
D. LX, 10V/div
IOUT = 2A
www.maximintegrated.com
10A
0
12V
0
C
D
40µs/div
C. INDUCTOR CURRENT,
A. REFIN, 500mV/div
10A/div
B. VOUT, 200mV/div
D. LX, 10V/div
IOUT = 2A
Maxim Integrated │ 10
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
IN
IN
IN
IN
IN
N.C.
IN
AGND
BST
TOP VIEW
TON
Pin Configuration
30 29 28 27 26 25 24 23 22 21
N.C. 31
EP1
FB 32
ILIM 33
EP3
19 IN
18 IN
MAX17016
AGND
REFIN 34
20 IN
IN
17 N.C.
REF 35
16 LX
EP2
SKIP 36
15 PGND
VCC 37
14 PGND
LX
PGOOD 38
13 PGND
7
PGND
8
9
10
PGND
6
PGND
5
PGND
4
PGND
3
LX
2
VDD
1
AGND
11 PGND
+
EN
12 PGND
N.C. 40
N.C.
N.C. 39
TQFN
(5mm x 5mm)
Pin Description
PIN
NAME
1, 17, 27, 31,
39, 40
N.C.
FUNCTION
No Connection. Not internally connected.
Shutdown Control Input. Connect to VDD for normal operation. Pull EN low to put the controller into its
2µA (max) shutdown state. The MAX17016 slowly ramps down the target/output voltage to ground and
after the target voltage reaches 0.1V, the controller forces LX into a high-impedance state and enters
the low-power shutdown state. Toggle EN to clear the fault-protection latch.
2
EN
3, 28
AGND
4
VDD
Supply Voltage Input for the DL Gate Driver. Connect to the system supply voltage (+4.5V to +5.5V).
Bypass VDD to power ground with a 1µF or greater ceramic capacitor.
5, 16
LX
Inductor Connection. Internally connected to EP2. Connect LX to the switched side of the inductor as
shown in Figure 1.
6–15
PGND
18–26
IN
Analog Ground. Internally connected to EP1.
Power Ground
Power MOSFET Input Power Source. Internally connected to EP3.
Switching Frequency-Setting Input. An external resistor between the input power source and TON sets
the switching period (tSW = 1/fSW) according to the following equation:
29
TON
where CTON = 16.26pF and VFB = VREFIN under normal operating conditions. If the TON current
drops below 10µA, the MAX17016 shuts down and enters a high-impedance state.
TON is high impedance in shutdown.
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Maxim Integrated │ 11
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Pin Description (continued)
PIN
NAME
30
BST
32
FB
33
ILIM
FUNCTION
Boost Flying Capacitor Connection. Connect to an external 0.1µF, 6V capacitor as shown in
Figure 1. The MAX17016 contains an internal boost switch/diode (Figure 2).
Feedback Voltage Sense Connection. Connect directly to the positive terminal of the output capacitors
for output voltages less than 2V as shown in the Standard Application Circuit (Figure 1). For fixed-output
voltages greater than 2V, connect REFIN to REF and use a resistive divider to set the output voltage
(Figure 6). FB senses the output voltage to determine the on-time for the high-side switching MOSFET.
Current-Limit Threshold Adjustment. The current-limit threshold is 0.05 times (1/20) the voltage at
ILIM. Connect ILIM to a resistive divider (from REF) to set the current-limit threshold between 20mV
and 100mV (with 0.4V to 2V at ILIM).
34
REFIN
External Reference Input. REFIN sets the feedback regulation voltage (VFB = VREFIN) of the
MAX17016 using a resistor-divider connected between REF and GND. The MAX17016 includes
an internal window comparator to detect REFIN voltage transitions, allowing the controller to blank
PGOOD and the fault protection.
35
REF
2V Reference Voltage. Bypass to analog ground using a 1nF ceramic capacitor. The reference can
source up to 50µA for external loads.
36
SKIP
Pulse-Skipping Control Input. This four-level input determines the mode of operation under normal
steady-state conditions and dynamic output-voltage transitions:
VDD (5V) = Forced-PWM operation
REF (2V) = Pulse-skipping mode with forced-PWM during TRANSITIONS
Open (3.3V) = Ultrasonic mode (without forced-PWM during transitions)
GND = Pulse-skipping mode (without forced-PWM during transitions)
37
VCC
5V Analog Supply Voltage. Internally connected to VDD through an internal 20Ω resistor. Bypass VCC
to analog ground using a 1µF ceramic capacitor.
38
PGOOD
Open-Drain Power-Good Output. PGOOD is low when the output voltage is more than 200mV (typ)
below or 300mV (typ) above the target voltage (VREFIN), during soft-start, and soft-shutdown. After
the soft-start circuit has terminated, PGOOD becomes high impedance if the output is in regulation.
PGOOD is blanked—forced high-impedance state—when a dynamic REFIN transition is detected.
EP1
(41)
AGND
Exposed Pad 1/Analog Ground. Internally connected to the controller’s ground plane and substrate.
Connect directly to ground.
EP2
(42)
LX
Exposed Pad 2/Inductor Connection. Internally connected to drain of the low-side MOSFET and
source of the high-side MOSFET (Figure 2). Connect LX to the switched side of the inductor as shown
in Figure 1.
EP3
(43)
IN
Exposed Pad 3/Power MOSFET Input Power Source. Internally connected to drain of the high-side
MOSFET (Figure 2).
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Maxim Integrated │ 12
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
4
5V BIAS
SUPPLY
C1
1µF
C2
1µF
AGND
PWR
36
C3
1000pF
35
LO
CBST
0.1µF
LX
PGOOD
L1
EN
FB
SKIP
COUT
PWR
32
RT
60.4kΩ
REF
REFIN
PWR
OUTPUT
1.05V/1.50V
10A (MAX)
PWR
ILIM
AGND
AGND
CIN
5, 16, EP2
PGND
3, 28, EP1
HI
INPUT
7V TO 24V
PWR
6–15
R1
49.9kΩ
R2
54.9kΩ
R3
97.6kΩ
30
MAX17016
34
AGND
VCC
R10
100kΩ
OFF
AGND
18–26, EP3
BST
37
2
GND/OPEN/REF/VCC
TON
IN
38
ON
VDD
RTON
200kΩ
29
33
R4
40.2kΩ
R5
49.4kΩ
NTC
10kΩ
B = 3435
SEE TABLE 1 FOR COMPONENT SELECTION.
AGND
Figure 1. MAX17016 Standard Application Circuit
Table 1. Component Selection for Standard Applications
VOUT = 1.5V/1.05V AT 10A
(Figure 1)
VOUT = 3.3V AT 6A
(Figure 6)
VOUT = 1.5V/1.05V AT 10A
(Figure 1)
VIN = 7V TO 20V
TON = 200kΩ (300kHz)
VIN = 7V TO 20V
TON = 332kΩ (200kHz)
VIN = 5V TO 12V
TON = 96kΩ (600kHz)
Input Capacitor
(3x) 10µF, 25V
Taiyo Yuden TMK432BJ106KM
(2x) 10µF, 25V
Taiyo Yuden TMK432BJ106KM
(3x) 10µF, 25V
Taiyo Yuden TMK432BJ106KM
Output Capacitor
(2x) 330µF, 6mΩ, 2V
Panasonic EEFSX0D331XR
(1x) 330µF, 18mΩ, 4V
SANYO 4TPE330MI
(1x) 470µF, 7mΩ, 2.5V
SANYO 2R5TPLF470M7
Inductor
1.0µH, 3.25mΩ, 20A
Wurth 744 3552 100
1.5µH, 14mΩ, 9A
NEC Tokin MPLC1040L3R3
0.47µH, 3.7mΩ, 15A
Coiltronics FP3-R47-R
COMPONENT
Table 2. Component Suppliers
MANUFACTURER
WEBSITE
MANUFACTURER
WEBSITE
AVX Corp.
www.avxcorp.com
Pulse Engineering
www.pulseeng.com
BI Technologies
www.bitechnologies.com
SANYO Electric Co., Ltd.
www.sanyodevice.com
Coiltronics
www.cooperet.com
Sumida Corp
www.sumida.com
KEMET Corp.
www.kemet.com
Taiyo Yuden
www.t-yuden.com
Murata Electronics North
America, Inc.
www.murata-northamerica.com
TDK Corp.
www.component.tdk.com
NEC Tokin America, Inc.
www.nec-tokinamerica.com
TOKO America, Inc.
www.tokoam.com
Panasonic Corp.
www.panasonic.com
Würth Electronik GmbH and Co. KG
www.we-online.com
www.maximintegrated.com
Maxim Integrated │ 13
MAX17016
Standard Application Circuit
The MAX17016 (Figure 1) generates a 1.5V or 1.05V output rail for general-purpose use in a notebook computer.
See Table 1 for component selections. Table 2 lists the
component manufacturers.
Detailed Description
The MAX17016 step-down controller is ideal for the
low-duty-cycle (high-input voltage to low-output voltage)
applications required by notebook computers. Maxim’s
proprietary Quick-PWM pulse-width modulator in the
MAX17016 is specifically designed for handling fast load
steps while maintaining a relatively constant operating
frequency and inductor operating point over a wide range
of input voltages. The Quick-PWM architecture circumvents the poor load-transient timing problems of fixedfrequency, current-mode PWMs while also avoiding the
problems caused by widely varying switching frequencies
in conventional constant-on-time (regardless of input voltage) pulse-frequency modulation (PFM) control schemes.
+5V Bias Supply (VCC/VDD)
The MAX17016 requires an external 5V bias supply in
addition to the battery. Typically, this 5V bias supply is the
notebook’s main 95% efficient 5V system supply. Keeping
the bias supply external to the IC improves efficiency and
eliminates the cost associated with the 5V linear regulator
that would otherwise be needed to supply the PWM circuit
and gate drivers. If stand-alone capability is needed, the
5V supply can be generated with an external linear regulator such as the MAX1615.
The 5V bias supply powers both the PWM controller and
internal gate-drive power, so the maximum current drawn
is determined by:
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Another one-shot sets a minimum off-time (200ns typ).
The on-time one-shot is triggered if the error comparator is low, the low-side switch current is below the valley
current-limit threshold, and the minimum off-time one-shot
has timed out.
On-Time One-Shot
The heart of the PWM core is the one-shot that sets
the high-side switch on-time. This fast, low-jitter, adjustable one-shot includes circuitry that varies the on-time
in response to input and output voltage. The high-side
switch on-time is inversely proportional to the input voltage as sensed by the TON input, and proportional to the
feedback voltage as sensed by the FB input:
On-Time (tON) = tSW (VFB/VIN)
where tSW (switching period) is set by the resistance
(RTON) between TON and VIN. This algorithm results in
a nearly constant switching frequency despite the lack
of a fixed-frequency clock generator. Connect a resistor
(RTON) between TON and VIN to set the switching period
tSW = 1/fSW:
V
=
t SW C TON (R TON + 6.5kΩ) FB
VOUT
where CTON = 16.26pF. When used with unity-gain feedback (VOUT = VFB), a 96kΩ to 301kΩ corresponds to
switching periods of 1.67µs (600kHz) to 5µs (200kHz),
respectively. High-frequency (600kHz) operation optimizes the application for the smallest component size,
trading off efficiency due to higher switching losses. This
might be acceptable in ultra-portable devices where the
load currents are lower and the controller is powered from
a lower voltage supply. Low-frequency (200kHz) operation offers the best overall efficiency at the expense of
component size and board space.
IBIAS = IQ + fSWQG = 2mA to 20mA (typ)
The MAX17016 includes a 20Ω resistor between VDD and
VCC, simplifying the printed-circuit board (PCB) layout
requirement.
For continuous conduction operation, the actual switching
frequency can be estimated by:
Free-Running Constant-On-Time PWM
Controller with Input Feed-Forward
where VDIS is the sum of the parasitic voltage drops in the
inductor discharge path, including synchronous rectifier,
inductor, and PCB resistances; VCHG is the sum of the
resistances in the charging path, including the high-side
switch, inductor, and PCB resistances; and tON is the ontime calculated by the MAX17016.
The Quick-PWM control architecture is a pseudo-fixedfrequency, constant on-time, current-mode regulator with
voltage feed-forward (Figure 2). This architecture relies
on the output filter capacitor’s ESR to act as a currentsense resistor, so the output ripple voltage provides
the PWM ramp signal. The control algorithm is simple:
the high-side switch on-time is determined solely by a
one-shot whose pulse width is inversely proportional to
input voltage and directly proportional to output voltage.
www.maximintegrated.com
f SW =
VFB + VDIS
t ON ( VIN − VCHG )
Power-Up Sequence (POR, UVLO)
The MAX17016 is enabled when EN is driven high and
the 5V bias supply (VDD) is present. The reference powers up first. Once the reference exceeds its UVLO threshold, the internal analog blocks are turned on and masked
Maxim Integrated │ 14
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
TON
IN
ON-TIME
COMPUTE
tOFF(MIN)
FB
ONE-SHOT
S
tON
TRIG
BST
TRIG
Q
IN
Q
R
Q
LX
ONE-SHOT
INTEGRATOR
(CCV)
ERROR
AMPLIFIER
VDD
S
R
Q
PGND
FB
FAULT
BLANK
EA + 0.3V
QUADLEVEL
DECODE
SKIP
ZERO CROSSING
PGOOD
AND FAULT
PROTECTION
VALLEY CURRENT LIMIT
ILIM
REF
EA - 0.2V
EN
SOFTSTART/-STOP
PGOOD
VCC
2V
REF
REFIN
EA
BLANK
MAX17016
DYNAMIC OUTPUT
TRANSITION DETECTION
Figure 2. MAX17016 Block Diagram
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Maxim Integrated │ 15
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
by a 50μs one-shot delay in order to allow the bias
circuitry and analog blocks enough time to settle to their
proper states. With the control circuitry reliably powered
up, the PWM controller can begin switching.
Power-on reset (POR) occurs when VCC rises above
approximately 3V, resetting the fault latch and preparing
the controller for operation. The VCC UVLO circuitry inhibits switching until VCC rises above 4.25V. The controller
powers up the reference once the system enables the
controller, VCC exceeds 4.25V, and EN is driven high.
With the reference in regulation, the controller ramps the
output voltage to the target REFIN voltage with a 1.2mV/
μs slew rate:
=
t START
VFB
VFB
=
1.2mV/ µs
1.2V/ms
The soft-start circuitry does not use a variable current
limit, so full output current is available immediately.
PGOOD becomes high impedance approximately 200μs
after the target REFIN voltage has been reached. The
MAX17016 automatically uses pulse-skipping mode during soft-start and uses forced-PWM mode during softshutdown, regardless of the SKIP configuration.
For automatic startup, the battery voltage should be present before VCC. If the controller attempts to bring the
output into regulation without the battery voltage present,
the fault latch trips. The controller remains shut down until
the fault latch is cleared by toggling EN or cycling the VCC
power supply below 0.5V.
If the VCC voltage drops below 4.25V, the controller
assumes that there is not enough supply voltage to
make valid decisions. To protect the output from overvoltage faults, the controller shuts down immediately and
forces a high impedance on LX.
need for the Schottky diode normally connected between
the output and ground to clamp the negative outputvoltage excursion. After the controller reaches the zero
target, the MAX17016 shuts down completely—the drivers
are disabled (high impedance on LX)—the reference turns
off, and the supply currents drop to about 0.1μA (typ).
When a fault condition—output UVP or thermal shutdown—activates the shutdown sequence, the protection
circuitry sets the fault latch to prevent the controller from
restarting. To clear the fault latch and reactivate the controller, toggle EN or cycle VCC power below 0.5V.
The MAX17016 automatically uses pulse-skipping mode
during soft-start and uses forced-PWM mode during softshutdown, regardless of the SKIP configuration.
Modes of Operation
Ultrasonic Mode (SKIP = Open = 3.3V)
Leaving SKIP unconnected activates a unique pulse-skipping mode with a minimum switching frequency of 18kHz.
This ultrasonic pulse-skipping mode eliminates audio-frequency modulation that would otherwise be present when
a lightly loaded controller automatically skips pulses. In
ultrasonic mode, the controller automatically transitions
to fixed-frequency PWM operation when the load reaches
the same critical conduction point (ILOAD(SKIP)) that
occurs when normally pulse skipping.
An ultrasonic pulse occurs when the controller detects
that no switching has occurred within the last 33μs. Once
triggered, the ultrasonic controller turns on the low-side
33s (typ)
INDUCTOR
CURRENT
Shutdown
When the system pulls EN low, the MAX17016 enters
low-power shutdown mode. PGOOD is pulled low immediately, and the output voltage ramps down with a
1.2mV/μs slew rate:
ZERO-CROSSING
DETECTION
VFB
VFB
=
t SHDN
=
1.2mV/µs 1.2V/ms
Slowly discharging the output capacitors by slewing the
output over a long period of time (typically 0.5ms to 2ms)
keeps the average negative inductor current low (damped
response), thereby preventing the negative output-voltage excursion that occurs when the controller discharges
the output quickly by permanently turning on the low-side
MOSFET (underdamped response). This eliminates the
www.maximintegrated.com
0
ISONIC
ON-TIME (tON)
Figure 3. Ultrasonic Waveform
Maxim Integrated │ 16
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
MOSFET to induce a negative inductor current (Figure 3).
After the inductor current reaches the negative ultrasonic
current threshold, the controller turns off the low-side
MOSFET and triggers a constant on-time.
When the on-time has expired, the controller reenables
the low-side MOSFET until the controller detects that the
inductor current dropped below the zero-crossing threshold. Starting with a negative inductor current pulse greatly
reduces the peak output voltage when compared to starting with a positive inductor current pulse.
The output voltage at the beginning of the ultrasonic pulse
determines the negative ultrasonic current threshold,
resulting in the following equation:
VISONIC= IL R CS=
(VREFIN − VFB ) × 0.7
where VFB > VREFIN and RCS is 6mΩ low-side on-resistance seen across GND to LX.
Forced-PWM Mode (SKIP = VDD)
The low-noise, forced-PWM mode (SKIP = VDD) disables
the zero-crossing comparator, which controls the low-side
switch on-time. This forces the low-side gate-drive waveform to constantly be the complement of the high-side
gate-drive waveform, so the inductor current reverses at
light loads while LX maintains a duty factor of VOUT/VIN.
The benefit of forced-PWM mode is to keep the switching
frequency fairly constant. However, forced-PWM operation comes at a cost: the no-load 5V bias current remains
between 10mA to 50mA, depending on the switching
frequency.
The MAX17016 automatically always uses forced-PWM
operation during shutdown, regardless of the SKIP
configuration.
Automatic Pulse-Skipping Mode
(SKIP = GND or REF)
In skip mode (SKIP = GND or 3.3V), an inherent automatic switchover to PFM takes place at light loads. This
switchover is affected by a comparator that truncates
the low-side switch on-time at the inductor current’s zero
crossing. The zero-crossing comparator threshold is set
by the differential across LX to GND.
DC output-accuracy specifications refer to the threshold
of the error comparator. When the inductor is in continuous conduction, the MAX17016 regulates the valley of the
output ripple, so the actual DC output voltage is higher
than the trip level by 50% of the output ripple voltage.
In discontinuous conduction (SKIP = GND and IOUT <
ILOAD(SKIP)), the output voltage has a DC regulation level
higher than the error-comparator threshold by approximately 1.5% due to slope compensation.
When SKIP is pulled to GND, the MAX17016 remains in
pulse-skipping mode. Since the output is not able to sink
current, the timing for negative dynamic output-voltage
transitions depends on the load current and output capacitance. Letting the output voltage drift down is typically
recommended in order to reduce the potential for audible
noise since this eliminates the input current surge during
negative output-voltage transitions. See Figure 4 and
Figure 5.
DYNAMIC REFIN WINDOW
REFIN
OUTPUT
VOLTAGE
INTERNAL
PWM CONTROL
LX
PGOOD
OVP
ACTUAL VOUT
INTERNAL TARGET
SKIP
NO PULSES: VOUT > VTARGET
BLANK HIGH-Z
SET TO REF + 300mV
BLANK HIGH-Z
EA TARGET + 300mV
DYNAMIC TRANSITION WHEN SKIP# = GND
Figure 4. Dynamic Transition when SKIP = GND
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Maxim Integrated │ 17
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
DYNAMIC REFIN WINDOW
REFIN
OUTPUT
VOLTAGE
INTERNAL EA TARGET = ACTUAL VOUT
INTERNAL
PWM CONTROL
PWM
PWM
SKIP
SKIP
LX
PGOOD
BLANK HIGH-Z
OVP
SET TO REF +
300mV
BLANK HIGH-Z
EA TARGET + 300mV
EA TARGET + 300mV
DYNAMIC TRANSITION WHEN SKIP = REF
Figure 5. Dynamic Transition when SKIP = REF
Valley Current-Limit Protection
The current-limit circuit employs a unique “valley”
current-sensing algorithm that senses the inductor current through the low-side MOSFET. If the current through
the low-side MOSFET exceeds the valley current-limit
threshold, the PWM controller is not allowed to initiate
a new cycle. The actual peak current is greater than the
valley current-limit threshold by an amount equal to the
inductor ripple current. Therefore, the exact current-limit
characteristic and maximum load capability are a function
of the inductor value and input voltage. When combined
with the undervoltage protection circuit, this current-limit
method is effective in almost every circumstance.
In forced-PWM mode, the MAX17016 also implements a
negative current limit to prevent excessive reverse inductor currents when VOUT is sinking current. The negative
current-limit threshold is set to approximately 120% of the
positive current limit.
Integrated Output Voltage
The MAX17016 regulates the valley of the output ripple,
so the actual DC output voltage is higher than the slopecompensated target by 50% of the output ripple voltage.
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Under steady-state conditions, the MAX17016’s internal
integrator corrects for this 50% output ripple-voltage error,
resulting in an output voltage that is dependent only on
the offset voltage of the integrator amplifier provided in
the Electrical Characteristics table.
Dynamic Output Voltages
The MAX17016 regulates FB to the voltage set at
REFIN. By changing the voltage at REFIN (Figure 1),
the MAX17016 can be used in applications that require
dynamic output-voltage changes between two set points.
For a step-voltage change at REFIN, the rate of change of
the output voltage is limited either by the internal 9.45mV/
μs slew-rate circuit or by the component selection—inductor current ramp, the total output capacitance, the current limit, and the load during the transition—whichever
is slower. The total output capacitance determines how
much current is needed to change the output voltage,
while the inductor limits the current ramp rate. Additional
load current could slow down the output voltage change
during a positive REFIN voltage change, and could speed
up the output voltage change during a negative REFIN
voltage change.
Maxim Integrated │ 18
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
4
5V BIAS
SUPPLY
C1
1µF
C2
1µF
AGND
TON
IN
PWR
BST
37
VCC
LX
R10
100kΩ
38
ON
VDD
2
OFF
36
GND/OPEN/REF/VCC
C3
1000pF
PGOOD
PGND
EN
FB
SKIP
RTON
332kΩ
29
18–26, EP3
CIN
30
5, 16, EP2
CBST
0.1µF
6–15
32
PWR
L1
OUTPUT
3.3V
PWR
PWR
R7
20.0kΩ
REF
AGND
AGND
34
COUT
R6
13.0kΩ
MAX17016
35
INPUT
7V TO 24V
REFIN
ILIM
33
AGND
3, 28, EP1
R4
49.9kΩ
REF
R5
49.4kΩ
AGND
PWR
AGND
SEE TABLE 1 FOR COMPONENT SELECTION.
Figure 6. High Output-Voltage Application Using a Feedback Divider
Output Voltages Greater than 2V
Although REFIN is limited to a 0 to 2V range, the outputvoltage range is unlimited since the MAX17016 utilizes a
high-impedance feedback input (FB). By adding a resistive voltage-divider from the output to FB to analog
ground (Figure 6), the MAX17016 supports output voltages above 2V. However, the controller also uses FB to
determine the on-time, so the voltage-divider influences
the actual switching frequency, as detailed in the On-Time
One-Shot section.
Internal Integration
An integrator amplifier forces the DC average of the FB
voltage to equal the target voltage. This internal amplifier integrates the feedback voltage and provides a fine
adjustment to the regulation voltage (Figure 2), allowing accurate DC output-voltage regulation regardless of
the compensated feedback ripple voltage and internal
slope-compensation variation. The integrator amplifier
has the ability to shift the output voltage by ±55mV (typ).
www.maximintegrated.com
The MAX17016 disables the integrator by connecting the
amplifier inputs together at the beginning of all downward
REFIN transitions done in pulse-skipping mode. The integrator remains disabled until 20μs after the transition is
completed (the internal target settles) and the output is in
regulation (edge detected on the error comparator).
Power-Good Outputs (PGOOD)
and Fault Protection
PGOOD is the open-drain output that continuously
monitors the output voltage for undervoltage and overvoltage conditions. PGOOD is actively held low in shutdown
(EN = GND), and during soft-start and soft-shutdown.
Approximately 200μs (typ) after the soft-start terminates,
PGOOD becomes high impedance as long as the feedback voltage is above the UVP threshold (REFIN - 200mV)
and below the OVP threshold (REFIN + 300mV). PGOOD
goes low if the feedback voltage drops 200mV below the
target voltage (REFIN) or rises 300mV above the target
voltage (REFIN), or the SMPS controller is shutdown. For
Maxim Integrated │ 19
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
TARGET
+ 300mV
TARGET
- 200mV
POWER-GOOD AND FAULT PROTECTION
FB
EN
OVP
SOFT-START
COMPLETE
UVP
OVP ENABLED
ONESHOT
200µs
FAULT
LATCH
FAULT
POWER-GOOD
IN
OUT
CLK
Figure 7. Power-Good and Fault Protection
a logic-level PGOOD output voltage, connect an external pullup resistor between PGOOD and VDD. A 100kΩ
pullup resistor works well in most applications. Figure 7
shows the power-good and fault-protection circuitry.
shuts down the controller, and forces a high impedance
on LX. Toggle EN or cycle VCC power below VCC POR
to reactivate the controller after the junction temperature
cools by 15°C.
Overvoltage Protection (OVP)
Quick-PWM Design Procedure
When the internal feedback voltage rises 300mV above
the target voltage and OVP is enabled, the OVP comparator immediately forces LX low, pulls PGOOD low, sets the
fault latch, and disables the SMPS controller. Toggle EN
or cycle VCC power below the VCC POR to clear the fault
latch and restart the controller.
Undervoltage Protection (UVP)
When the feedback voltage drops 200mV below the target
voltage (REFIN), the controller immediately pulls PGOOD
low and triggers a 200μs one-shot timer. If the feedback
voltage remains below the undervoltage fault threshold
for the entire 200μs, then the undervoltage fault latch
is set and the SMPS begins the shutdown sequence.
When the internal target voltage drops below 0.1V, the
MAX17016 forces a high impedance on LX. Toggle EN or
cycle VCC power below VCC POR to clear the fault latch
and restart the controller.
Thermal-Fault Protection (TSHDN)
The MAX17016 features a thermal-fault protection circuit.
When the junction temperature rises above +160°C, a
thermal sensor activates the fault latch, pulls PGOOD low,
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Firmly establish the input voltage range and maximum
load current before choosing a switching frequency and
inductor operating point (ripple-current ratio). The primary design trade-off lies in choosing a good switching
frequency and inductor operating point, and the following
four factors dictate the rest of the design:
●● Input voltage range: The maximum value (VIN(MAX))
must accommodate the worst-case input supply voltage allowed by the notebook’s AC adapter voltage. The
minimum value (VIN(MIN)) must account for the lowest
input voltage after drops due to connectors, fuses, and
battery selector switches. If there is a choice at all,
lower input voltages result in better efficiency.
●● Maximum load current: There are two values to consider. The peak load current (ILOAD(MAX)) determines
the instantaneous component stresses and filtering
requirements, and thus drives output capacitor selection, inductor saturation rating, and the design of
the current-limit circuit. The continuous load current
(ILOAD) determines the thermal stresses and thus
drives the selection of input capacitors, MOSFETs,
Maxim Integrated │ 20
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
and other critical heat-contributing components. Most
notebook loads generally exhibit ILOAD = ILOAD(MAX)
x 80%.
●● Switching frequency: This choice determines the basic
trade-off between size and efficiency. The optimal frequency is largely a function of maximum input voltage
due to MOSFET switching losses that are proportional
to frequency and VIN2. The optimum frequency is
also a moving target, due to rapid improvements in
MOSFET technology that are making higher frequencies more practical.
●● Inductor operating point: This choice provides tradeoffs between size vs. efficiency and transient response
vs. output noise. Low inductor values provide better
transient response and smaller physical size, but also
result in lower efficiency and higher output noise due
to increased ripple current. The minimum practical
inductor value is one that causes the circuit to operate
at the edge of critical conduction (where the inductor
current just touches zero with every cycle at maximum
load). Inductor values lower than this grant no further
size-reduction benefit. The optimum operating point is
usually found between 20% and 50% ripple current.
Inductor Selection
The switching frequency and operating point (% ripple
current or LIR) determine the inductor value as follows:
VOUT
VIN − VOUT
L=
f SW ILOAD ( MAX ) LIR VIN
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice, although powdered iron
is inexpensive and can work well at 200kHz. The core
must be large enough not to saturate at the peak inductor
current (IPEAK):
=
IPEAK ILOAD ( MAX) +
Transient Response
∆IL
2
The inductor ripple current impacts transient-response
performance, especially at low VIN - VOUT differentials.
Low inductor values allow the inductor current to slew
faster, replenishing charge removed from the output filter
capacitors by a sudden load step. The amount of output
sag is also a function of the maximum duty factor, which
can be calculated from the on-time and minimum offtime. The worst-case output sag voltage can be determined by:
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(
L ∆ILOAD(MAX)
VSAG =
)
2 VOUT TSW
+ t OFF ( MIN )
VIN
(VIN − VOUT )TSW
2C OUT VOUT
VIN
− t OFF ( MIN )
where tOFF(MIN) is the minimum off-time (see the Electrical
Characteristics table).
The amount of overshoot due to stored inductor energy
when the load is removed can be calculated as:
VSOAR
(∆ILOAD(MAX))
≈
2
L
2C OUT VOUT
Setting the Valley Current Limit
The minimum current-limit threshold must be high enough
to support the maximum load current when the current
limit is at the minimum tolerance value. The valley of the
inductor current occurs at ILOAD(MAX) minus half the
inductor ripple current (ΔIL); therefore:
ILIMIT ( LOW ) > ILOAD ( MAX ) −
∆IL
2
where ILIMIT(LOW) equals the minimum current-limit
threshold voltage divided by the low-side MOSFETs onresistance (RDS(ON)).
The valley current-limit threshold is precisely 1/20 the
voltage seen at ILIM. Connect a resistive divider from
REF to ILIM to analog ground (GND) in order to set a
fixed valley current-limit threshold. The external 400mV
to 2V adjustment range corresponds to a 20mV to 100mV
valley current-limit threshold. When adjusting the currentlimit threshold, use 1% tolerance resistors and a divider
current of approximately 5μA to 10μA to prevent significant inaccuracy in the valley current-limit tolerance.
The MAX17016 uses the low-side MOSFET’s onresistance as the current-sense element (RSENSE =
RDS(ON)). Therefore, special attention must be made to
the tolerance and thermal variation of the on-resistance.
Use the worst-case maximum value for RDS(ON) from the
MOSFET data sheet, and add some margin for the rise in
RDS(ON) with temperature. A good general rule is to allow
0.5% additional resistance for each °C of temperature
rise, which must be included in the design margin unless
the design includes an NTC thermistor in the ILIM resistive voltage-divider to thermally compensate the currentlimit threshold.
Maxim Integrated │ 21
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
4
5V BIAS
SUPPLY
C1
1µF
C2
1µF
AGND
BST
PWR
37
2
36
C3
1000pF
AGND
35
AGND
LO
18–26, EP3
30
LX
PGOOD
PGND
FB
SKIP
R2
54.9kΩ
COUT
OUTPUT
1.50V 10A
1.05V 7A
PWR
REF
R8
100kΩ
REFIN
ILIM
33
AGND
R4
49.9kΩ
REF
R5
49.4kΩ
AGND
AGND
PWR
32
3, 28, EP1
HI
L1
PWR
MAX17016
INPUT
7V TO 24V
CIN
5, 16, EP2
6–15
R1
49.9kΩ
34
R3
97.6kΩ
VCC
EN
RTON
200kΩ
CBST
0.1µF
R10
100kΩ
OFF
GND/OPEN/REF/VCC
TON
IN
38
ON
VDD
29
SEE TABLE 1 FOR COMPONENT SELECTION.
PWR AGND
Figure 8. Standard Application with Foldback Current-Limit Protection
Foldback Current Limit
Including an additional resistor between ILIM and the
output automatically creates a current-limit threshold that
folds back as the output voltage drops (see Figure 8).
The foldback current limit helps limit the inductor current
under fault conditions, but must be carefully designed in
order to provide reliable performance under normal conditions. The current-limit threshold must not be set too low,
or the controller will not reliably power up. To ensure the
controller powers up properly, the minimum current-limit
threshold (when VOUT = 0V) must always be greater than
the maximum load during startup (which at least consists
of leakage currents), plus the maximum current required
to charge the output capacitors:
ISTART = COUT x 1mV/μs + ILOAD(START)
Output Capacitor Selection
The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple
and load-transient requirements. Additionally, the ESR
impacts stability requirements. Capacitors with a high
ESR value (polymers/tantalums) do not need additional
external compensation components.
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In core and chipset converters and other applications
where the output is subject to large-load transients, the
output capacitor’s size typically depends on how much
ESR is needed to prevent the output from dipping too
low under a load transient. Ignoring the sag due to finite
capacitance:
(R ESR + R PCB ) ≤
VSTEP
∆ILOAD(MAX)
In low-power applications, the output capacitor’s size
often depends on how much ESR is needed to maintain
an acceptable level of output ripple voltage. The output
ripple voltage of a step-down controller equals the total
inductor ripple current multiplied by the output capacitor’s
ESR. The maximum ESR to meet ripple requirements is:
VIN × f SW × L
R ESR ≤
VRIPPLE
(VIN − VOUT )VOUT
where fSW is the switching frequency.
Maxim Integrated │ 22
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
With most chemistries (polymer, tantalum, aluminum
electrolytic), the actual capacitance value required relates
to the physical size needed to achieve low ESR and the
chemistry limits of the selected capacitor technology.
Ceramic capacitors provide low ESR, but the capacitance and voltage rating (after derating) are determined
by the capacity needed to prevent VSAG and VSOAR
from causing problems during load transients. Generally,
once enough capacitance is added to meet the overshoot
requirement, undershoot at the rising load edge is no longer a problem (see the VSAG and VSOAR equations in the
Transient Response section). Thus, the output capacitor
selection requires carefully balancing capacitor chemistry
limitations (capacitance vs. ESR vs. voltage rating) and
cost. See Figure 9.
For a standard 300kHz application, the effective zero
frequency must be well below 95kHz, preferably below
50kHz. With these frequency requirements, standard
tantalum and polymer capacitors already commonly used
have typical ESR zero frequencies below 50kHz, allowing the stability requirements to be achieved without any
additional current-sense compensation. In Figure 1, the
ESR needed to support a 15mVP-P ripple is 15mV/(10A x
0.3) = 5mΩ. Two 330μF, 9mΩ polymer capacitors in parallel provide 4.5mΩ (max) ESR and 1/(2π x 330μF x 9mΩ)
= 53kHz ESR zero frequency. See Figure 10.
Output Capacitor Stability Considerations
IN
For Quick-PWM controllers, stability is determined by the
in-phase feedback ripple relative to the switching frequency, which is typically dominated by the output ESR. The
boundary of instability is given by the following equation:
f SW
≥
π
BST
PWR
L1
LX
1
MAX17016
R EFF = R ESR + R PCB + R COMP
IN
CIN
BST
PWR
PWR
FB
where COUT is the total output capacitance, RESR is the
total ESR of the output capacitors, RPCB is the parasitic
board resistance between the output capacitors and feedback sense point, and RCOMP is the effective resistance
of the DC- or AC-coupled current-sense compensation
(Figure 11).
STABILITY REQUIREMENT
1
RESRCOUT
2fSW
AGND
AGND
Figure 9. Standard Application with Output Polymer or Tantalum
INPUT
PCB PARASITIC RESISTANCE-SENSE
RESISTANCE FOR EVALUATION
PWR
DH
L1
LX
OUTPUT
C OUT
PGND
FB
OUTPUT
COUT
PGND
2 πR EFF C OUT
MAX17016
INPUT
CIN
CCOMP
0.1µF
PWR
PWR
CLOAD
PWR
RCOMP
100Ω
OUTPUT VOLTAGE REMOTELY
SENSED NEAR POINT OF LOAD
GND
AGND
PWR
STABILITY REQUIREMENT
1
1
R ESR COUT ≥
AND RCOMPCCOMP ≥
2fSW
fSW
FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT
Figure 10. Remote-Sense Compensation for Stability and Noise Immunity
www.maximintegrated.com
Maxim Integrated │ 23
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Ceramic capacitors have a high-ESR zero frequency, but
applications with sufficient current-sense compensation
can still take advantage of the small size, low ESR, and
high reliability of the ceramic chemistry. Using the inductor
DCR, applications using ceramic output capacitors can
be compensated using either a DC compensation or AC
compensation method (Figure 11).
The DC-coupling requires fewer external compensation capacitors, but this also creates an output load line
that depends on the inductor’s DCR (parasitic resistance). Alternatively, the current-sense information can
be AC-coupled, allowing stability to be dependent only on
the inductance value and compensation components and
eliminating the DC load line.
OPTION A: DC-COUPLED CURRENT-SENSE COMPENSATION
IN
CIN
BST
DC COMPENSATION
FEWER COMPENSATION COMPONENTS
CREATES OUTPUT LOAD LINE
LESS OUTPUT CAPACITANCE REQUIRED
FOR TRANSIENT RESPONSE
INPUT
PWR
L
LX
RSENA
PGND
MAX17016
OUTPUT
COUT
RSENB
PWR
PWR
CSEN
FB
GND
AGND
STABILITY REQUIREMENT
PWR
L
COUT ≥
(RSENA|| R SENB) CSEN
R SENBR DCR
1
AND LOAD LINE =
2fSW
R SENA + RSENB
FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT
OPTION B: AC-COUPLED CURRENT-SENSE COMPENSATION
IN
CIN
BST
PWR
L
LX
PWR
OUTPUT
COUT
RSEN
PGND
MAX17016
AC COMPENSATION
NOT DEPENDENT ON ACTUAL DCR VALUE
NO OUTPUT LOAD LINE
INPUT
CSEN
PWR
CCOMP
FB
RCOMP
GND
AGND
STABILITY REQUIREMENT
PWR
L
RSEN CSEN
COUT ≥
1
1
AND RCOMPCCOMP ≥
2fSW
fSW
FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT
Figure 11. Feedback Compensation for Ceramic Output Capacitors
www.maximintegrated.com
Maxim Integrated │ 24
MAX17016
When only using ceramic output capacitors, output overshoot (VSOAR) typically determines the minimum output
capacitance requirement. Their relatively low capacitance
value could allow significant output overshoot when stepping from full-load to no-load conditions, unless designed
with a small inductance value and high switching frequency to minimize the energy transferred from the inductor to
the capacitor during load-step recovery.
Unstable operation manifests itself in two related but
distinctly different ways: double pulsing and feedbackloop instability. Double pulsing occurs due to noise on
the output or because the ESR is so low that there is
not enough voltage ramp in the output voltage signal.
This “fools” the error comparator into triggering a new
cycle immediately after the minimum off-time period has
expired. Double pulsing is more annoying than harmful, resulting in nothing worse than increased output
ripple. However, it can indicate the possible presence of
loop instability due to insufficient ESR. Loop instability
can result in oscillations at the output after line or load
steps. Such perturbations are usually damped, but can
cause the output voltage to rise above or fall below the
tolerance limits.
The easiest method for checking stability is to apply a
very fast zero-to-max load transient and carefully observe
the output voltage-ripple envelope for overshoot and ringing. It can help to simultaneously monitor the inductor
current with an AC current probe. Do not allow more than
one cycle of ringing after the initial step-response under/
overshoot.
Input Capacitor Selection
The input capacitor must meet the ripple current requirement (IRMS) imposed by the switching currents. The
IRMS requirements can be determined by the following
equation:
I
=
IRMS LOAD VOUT (VIN − VOUT )
V
IN
The worst-case RMS current requirement occurs when
operating with VIN = 2VOUT. At this point, the above
equation simplifies to IRMS = 0.5 x ILOAD.
For most applications, nontantalum chemistries (ceramic,
aluminum, or OS-CON) are preferred due to their resistance to inrush surge currents typical of systems with a
mechanical switch or connector in series with the input.
If the Quick-PWM controller is operated as the second
www.maximintegrated.com
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
stage of a two-stage power-conversion system, tantalum
input capacitors are acceptable. In either configuration,
choose an input capacitor that exhibits less than +10°C
temperature rise at the RMS input current for optimal
circuit longevity.
Minimum
Input-Voltage
and Dropout Performance
Requirements
The output voltage-adjustable range for continuousconduction operation is restricted by the nonadjustable
minimum off-time one-shot. For best dropout performance, use the slower (200kHz) on-time settings. When
working with low-input voltages, the duty-factor limit must
be calculated using worst-case values for on- and offtimes. Manufacturing tolerances and internal propagation
delays introduce an error to the on-times. This error is
greater at higher frequencies. Also, keep in mind that
transient response performance of buck regulators operated too close to dropout is poor, and bulk output capacitance must often be added (see the VSAG equation in the
Quick-PWM Design Procedure section).
The absolute point of dropout is when the inductor current
ramps down during the minimum off-time (ΔIDOWN) as
much as it ramps up during the on-time (ΔIUP). The ratio
h = ΔIUP/ΔIDOWN is an indicator of the ability to slew the
inductor current higher in response to increased load,
and must always be greater than 1. As h approaches 1,
the absolute minimum dropout point, the inductor current
cannot increase as much during each switching cycle and
VSAG greatly increases unless additional output capacitance is used.
A reasonable minimum value for h is 1.5, but adjusting this
up or down allows trade-offs between VSAG, output capacitance, and minimum operating voltage. For a given value
of h, the minimum operating voltage can be calculated as:
V
− VDROOP + VCHG
VIN( MIN ) = FB
1 − h × t OFF ( MIN ) f SW
(
)
where VFB is the voltage-positioning droop, VCHG is the
parasitic voltage drop in the charge path, and tOFF(MIN)
is from the Electrical Characteristics table. The absolute
minimum input voltage is calculated with h = 1.
If the calculated VIN(MIN) is greater than the required minimum input voltage, then reduce the operating frequency
or add output capacitance to obtain an acceptable VSAG.
If operation near dropout is anticipated, calculate VSAG to
be sure of adequate transient response.
Maxim Integrated │ 25
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Dropout design example:
tor, REF bypass capacitors, REFIN components, and
feedback compensation/dividers.
VOUT = 1.5V
fSW = 300kHz
tOFF(MIN) = 350ns
No droop/load line (VDROOP = 0V)
VDROPCHG = 150mV (10A load)
h = 1.5
VIN( MIN )
1.5 V − 0 V + 150mV
=
1.96 V
1 − (1.5 × 350ns × 300kHz )
Calculating again with h = 1 gives the absolute limit of
dropout:
VIN( MIN )
1.5 V − 0 V + 150mV
=
1.84 V
1 − (1.0 × 350ns × 300kHz )
Therefore, VIN must be greater than 1.84V, even with very
large output capacitance, and a practical input voltage
with reasonable output capacitance would be 2.0V.
Applications Information
PCB Layout Guidelines
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. The switching power
stage requires particular attention. If possible, mount all
the power components on the top side of the board with
their ground terminals flush against one another. Follow
these guidelines for good PCB layout:
●● Keep the high-current paths short, especially at the
ground terminals. This is essential for stable, jitter-free
operation.
●● Connect all analog grounds to a separate solid copper
plane, which connects to the GND pin of the QuickPWM controller. This includes the VCC bypass capaci-
Ordering Information
●● Keep the power traces and load connections short.
This is essential for high efficiency. The use of thick
copper PCB (2oz vs. 1oz) can enhance full-load efficiency by 1% or more. Correctly routing PCB traces
is a difficult task that must be approached in terms of
fractions of centimeters, where a single mΩ of excess
trace resistance causes a measurable efficiency
penalty.
●● Keep the power plane—especially LX—away from
sensitive analog areas (REF, REFIN, FB, ILIM).
Layout Procedure
1) Place the power components first, with ground terminals adjacent (CIN and COUT). If possible, make all
these connections on the top layer with wide, copperfilled areas.
2) Make the DC-DC controller ground connections as
shown in Figure 1. This diagram can be viewed as
having four separate ground planes: input/output
ground, where all the high-power components go; the
power ground plane, where the PGND pin and VDD
bypass capacitor go; the controller’s analog ground
plane where sensitive analog components, the controller’s GND pin, and VCC bypass capacitor go. The
controller’s GND plane must meet the PGND plane
only at a single point directly beneath the IC. This point
must also be very close to the output capacitor ground
terminal.
3) Connect the output power planes (VCORE and system
ground planes) directly to the output filter capacitor
positive and negative terminals with multiple vias.
Place the entire DC-DC converter circuit as close to
the load as is practical.
Chip Information
PART
TEMP RANGE
PIN-PACKAGE
MAX17016ETL+T
-40°C to +85°C
40 TQFN-EP*
TRANSISTOR COUNT: 7169
PROCESS: BiCMOS
+Denotes a lead-free package.
*EP = Exposed pad.
www.maximintegrated.com
Maxim Integrated │ 26
MAX17016
Single Quick-PWM Step-Down
Controller with Internal 26V MOSFETs
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
CHANGED
0
11/07
Initial release
1
4/17
Updated Package Information table, Electrical Characteristics table, and Table 1.
Updated the On-Time One-Shot and Minimum Input-Voltage Requirements and
Dropout Performance sections.
DESCRIPTION
—
2–3, 5 ,
13–14, 25
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
© 2017 Maxim Integrated Products, Inc. │ 27
