MAX17292ETCC+T

EVALUATION KIT AVAILABLE MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI General Description The MAX17290/MAX17292 high-efficiency, synchronous step-up DC-DC controllers operate over a 4.5V to 36V input voltage range with 42V input transient protection. The input operating range can be extended to as low as 2.5V in Bootstrapped mode. The MAX17290 and MAX17292 use a constant-frequency, pulse-width modulating (PWM), peak current-mode control architecture. There are multiple versions of the devices offering one or more of the following functions: a synchronization output (SYNCO) for 180° out-of-phase operation, an overvoltage protection function using a separate input pin (OVP), and a reference input pin (REFIN) to allow on-the-fly output voltage adjustment. The MAX17290 and MAX17292 operate in different frequency ranges. All versions can be synchronized to an external master clock using the FSET/SYNC input. The devices are available in a compact 12-pin (3mm x 3mm) TQFN and 10-pin µMAX packages. Both packages have exposed pads. -40°C to +85°C Operation. Applications ●● ●● ●● ●● Distributed Supply Regulation Offline Power Supplies Telecom Hardware General-Purpose Point-of-Load Benefits and Features ●● Reduces Solution Size and Cost • All-Ceramic Capacitor Solution Allows Ultra-Compact Solution Size • 100kHz to 1MHz (MAX17290) and 1MHz to 2.5MHz (MAX17292) Switching-Frequency with External Synchronization ●● Increases Design Flexibility • Bootstrapped Mode Allows Input Voltage to be 2.5V • Adjustable Slope Compensation ●● Reduces Power Dissipation • >90% Peak Efficiency • Low 4μA (typ.) Shutdown Current ●● Operates Reliably • 42V Input Voltage Transient Protection • Fixed 9ms Internal Software Start Reduces Input Inrush Current • PGOOD Output and Hiccup Mode for Enhanced System Protection • Overtemperature Shutdown • Reduced EMI Emission with Spread-Spectrum Control Typical Application Circuit BOOTSTRAPPED 2.2MHz APPLICATION WITH LOW OPERATING VOLTAGE 22µF 0.47µH BATTERY INPUT 2.5V to 40V 47µF CERAMIC PVL SUP 10kΩ DRV PGOOD 91kΩ ISNS 22mΩ PVL 2.2µF N 1kΩ MAX17292EUBA/B FB COMP FSET/SYNC 13kΩ 12kΩ GND Ordering Information appears at end of data sheet. µMAX is a registered trademark of Maxim Integrated Products, Inc. 19-8544; Rev 0; 8/16 EN ENABLE SW_OUT 8V/2A MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Absolute Maximum Ratings EN, SUP, OVP, FB to GND.....................................-0.3V to +42V DRV, SYNCO, FSET/SYNC, COMP, PGOOD, ISNS, REFIN to GND............. -0.3V to (VPVL + 0.3V) PVL to GND................................................................ -0.3V to 6V Continuous Power Dissipation (TA = +70NC) μMAX on SLB (derate 10.3mW/NC above +70NC).......825mW μMAX on MLB (derate 12.9mW/NC above +70NC).....1031mW TQFN on SLB (derate 13.2mW/NC above +70NC)......1053mW TQFN on MLB (derate 14.7mW/NC above +70NC).....1176mW Operating Temperature Range........................... -40NC to +85NC Maximum Junction Temperature......................................+150NC Storage Temperature Range............................. -65NC to +150NC Lead Temperature (soldering, 10s).................................+300NC Soldering Temperature (reflow).......................................+260NC Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Package Thermal Characteristics (Note 1) μMAX (Single-Layer Board) Junction-to-Ambient Thermal Resistance (BJA)...........97NC/W Junction-to-Case Thermal Resistance (BJC)..................5NC/W μMAX (Four-Layer Board) Junction-to-Ambient Thermal Resistance (BJA)...........78NC/W Junction-to-Case Thermal Resistance (BJC)..................5NC/W TQFN (Single-Layer Board) Junction-to-Ambient Thermal Resistance (BJA)...........76NC/W Junction-to-Case Thermal Resistance (BJC)................11NC/W TQFN (Four-Layer Board) Junction-to-Ambient Thermal Resistance (BJA)...........68NC/W Junction-to-Case Thermal Resistance (BJC)................11NC/W Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial. Electrical Characteristics (VSUP = 14V, TA = TJ = -40NC to +85NC, unless otherwise noted. Typical values are at TA =+25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 36 V POWER SUPPLY SUP Operating Supply Range VSUP 4.5 VFB = 1.1V, no switching SUP Supply Current in Operation ICC SUP Supply Current in Shutdown ISHDN VEN = 0V OVP Threshold Voltage VOVP OVP rising OVP Threshold Voltage Hysteresis VOVPH OVP Input Current MAX17290 0.75 1.3 MAX17292 1.25 2 4 7 FA 110 115 % of VFB 105 % of VFB 2.5 IOVP -1 mA +1 FA PVL REGULATOR PVL Output Voltage VPVL PVL Undervoltage Lockout VUV PVL Undervoltage-Lockout Hysteresis VUVH www.maximintegrated.com SUP rising 4.7 5 5.45 V 3.8 4 4.3 V 0.4 V Maxim Integrated │  2 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Electrical Characteristics (continued) (VSUP = 14V, TA = TJ = -40NC to +85NC, unless otherwise noted. Typical values are at TA =+25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX RFSET = 69kI 360 400 440 RFSET = 12kI 2000 2200 2400 UNITS OSCILLATOR Switching Frequency fSW Spread-Spectrum Spreading Factor SS Switching Frequency Range fSWR When set with resistor on pin MAX17290 100 1000 MAX17292 1000 2500 FSET/SYNC Frequency Range fSYNC Using external SYNC signal MAX17290 220 1000 MAX17292 1000 2500 FSET Regulation Voltage VFSET 12kI < RFSET < 69kI Soft-Start Time tSS Hiccup Period tHICCUP Maximum Duty Cycle DCMAX Minimum On-Time B, D, and F versions % of fSW Q6 0.9 Internally set 6 9 93 MAX17292, RFSET = 12kI 85 tON 50 kHz kHz V 12 55 MAX17290, RFSET = 69kI kHz ms ms % 80 110 ns THERMAL SHUTDOWN Thermal-Shutdown Temperature TS Thermal-Shutdown Hysteresis TH Temperature rising 165 NC 10 NC GATE DRIVERS DRV Pullup Resistance RDRVH IDRV = 100mA 3 5.5 I DRV Pulldown Resistance RDRVL IDRV = -100mA 1.4 2.5 I DRV Output Peak Current IDRV Sourcing, CDRV = 10nF 0.75 Sinking, CDRV = 10nF A 1 REGULATION/CURRENT SENSE VREFIN = VPVL FB Regulation Voltage VFB VREFIN = 2V VREFIN = 0.5V FB Input Current Across full line, load, and temperature range IFB Current-Sense Gain PGOOD Threshold www.maximintegrated.com 1.98 2 2.02 0.5 0.505 60 MAX16992 40 Percentage of final value V +0.5 FA 288 mV ns 8 Added to ISNS input VPG 250 MAX16990 AVI Peak Slope Compensation Current-Ramp Magnitude 1.01 0.495 212 tBLANK 1 -0.5 ISNS Threshold ISNS Leading-Edge Blanking Time 0.99 V/V 40 50 60 Rising 85 90 95 Falling 80 85 90 FA % Maxim Integrated │  3 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Electrical Characteristics (continued) (VSUP = 14V, TA = TJ = -40NC to +85NC, unless otherwise noted. Typical values are at TA =+25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 2 V VPVL 0.1 V ERROR AMPLIFIER REFIN Input Voltage Range 0.5 REFIN Threshold for 1V FB Regulation VPVL 0.8 VPVL 0.4 Error-Amplifier gm AVEA 700 FS Error-Amplifier Output Impedance ROEA 50 MI COMP Output Current ICOMP 140 μA COMP Clamp Voltage 2.7 3 3.3 V LOGIC-LEVEL INPUTS/OUTPUTS PGOOD/SYNCO Output Leakage Current VPGOOD/VSYNCO = 5V PGOOD/SYNCO Output Low Level Sinking 1mA EN High Input Threshold EN rising 0.5 FA 0.4 1.7 V EN Low Input Threshold FSET/SYNC High Input Threshold 1.2 2.5 V V FSET/SYNC Low Input Threshold EN and REFIN Input Current V -1 1 V +1 FA Note 2: All devices 100% production tested at TA = +25NC. Limits over temperature are guaranteed by design. www.maximintegrated.com Maxim Integrated │  4 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Typical Operating Characteristics (VSUP = 14V, TA = +25NC, unless otherwise noted.) SHUTDOWN SUPPLY CURRENT vs. SUPPLY VOLTAGE SUPPLY CURRENT vs. SUPPLY VOLTAGE toc01 2.2MHz 0.8 0.6 400kHz 0.4 VEN = VSUP VFB = 1.1V 0 4 9 8 7 6 5 4 3 3 2 1 12 20 VEN = 0V 0 36 28 4 4.8 4.6 4.4 4.2 4.0 3.8 VEN = 0V 20 28 -40 -20 36 0 20 40 60 80 100 120 SUPPLY VOLTAGE (V) SUPPLY VOLTAGE (V) TEMPERATURE (°C) PVL VOLTAGE vs. SUPPLY VOLTAGE PVL VOLTAGE vs. SUPPLY VOLTAGE MAX17290 INTERNAL OSCILLATOR FREQUENCY vs. SUPPLY VOLTAGE toc04 5.2 toc05 5.2 IPVL = 1mA 5.1 5.0 5.0 IPVL = 1mA 4.8 4.9 4.8 4.7 IPVL = 10mA 4.6 IPVL = 10mA 4.6 4.5 4.4 4.4 4.2 4.0 3.8 3.6 4.3 4.2 4.1 4.0 3.4 3.2 20 36 28 3 SUPPLY VOLTAGE (V) 410 405 400 395 390 385 RSET = 68.1kI 0 20 40 60 80 100 120 TEMPERATURE (°C) www.maximintegrated.com 4 5 402 400 398 396 394 392 RSET = 68.1kI 4 7 6 12 2350 2300 2250 2200 2150 2100 2050 RSET = 12.1kI 4 12 20 SUPPLY VOLTAGE (V) 36 28 MAX17292 INTERNAL OSCILLATOR FREQUENCY vs. TEMPERATURE toc08 2000 20 SUPPLY VOLTAGE (V) 2400 INTERNAL OSCILLATOR FREQUENCY (kHz) 415 -40 -20 404 MAX17292 INTERNAL OSCILLATOR FREQUENCY vs. SUPPLY VOLTAGE toc07 380 406 SUPPLY VOLTAGE (V) MAX17290 INTERNAL OSCILLATOR FREQUENCY vs. TEMPERATURE 420 408 390 28 toc09 2200 INTERNAL OSCILLATOR FREQUENCY (kHz) 12 toc06 410 3.0 4 INTERNAL OSCILLATOR FREQUENCY (kHz) 5.0 3.6 12 toc03 5.2 INTERNAL OSCILLATOR FREQUENCY (kHz) 0.2 PVL VOLTAGE (V) SHUTDOWN SUPPLY CURRENT (µA) 1.0 PVL VOLTAGE (V) SUPPLY CURRENT (mA) 1.2 toc02 10 SHUTDOWN SUPPLY CURRENT (µA) 1.4 SHUTDOWN SUPPLY CURRENT vs. TEMPERATURE 2190 2180 2170 2160 2150 2140 2130 2120 2110 RSET = 12.1kI 2100 36 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) Maxim Integrated │  5 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Typical Operating Characteristics (continued) (VSUP = 14V, TA = +25NC, unless otherwise noted.) POWER-UP RESPONSE POWER-UP RESPONSE toc10 toc11 5V/div 0V VSUP 5V/div 0V VSUP 5V/div VOUT 0V 5V/div VOUT 0V VDRV 5V/div 0V VPGOOD 5V/div 0V 5V/div VPVL 0V 5V/div 0V VPGOOD 2ms/div 2ms/div STARTUP RESPONSE STARTUP RESPONSE toc12 toc13 5V/div 0V VSUP 5V/div VOUT 0V VPVL 5V/div 0V 5V/div 0V VEN 5V/div 0V VPGOOD 5V/div VOUT 0V 5V/div 0V VDRV 5V/div 0V VEN 2ms/div 2ms/div STARTUP RESPONSE (WITH SWITCHED OUTPUT) OUTPUT LOAD TRANSIENT toc15 toc14 5V/div 0V VPGOOD 5V/div VOUT 0V 5V/div 0V VSW_OUT 5V/div 0V VEN 2ms/div www.maximintegrated.com 5V/div 0V VSUP 5V/div VOUT 0V 500mV/div (AC-COUPLED) VOUT 1A/div ILOAD 0A 50ms/div Maxim Integrated │  6 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Typical Operating Characteristics (continued) (VSUP = 14V, TA = +25NC, unless otherwise noted.) LINE TRANSIENT MAX17292 VSYNC vs. VSYNCO toc16 toc17 5V/div VSUP 2V/div 0V 5V/div VOUT VSYNC 0V 0V 500mV/div (AC-COUPLED) VOUT 1A/div ILOAD 0V 0A 20ms/div 200ns/div SWITCHING WAVEFORM OUTPUT VOLTAGE vs. REFIN VOLTAGE toc19 toc18 30 25 OUTPUT VOLTAGE (V) 2V/div VSYNCO 5V/div VOUT 20 VIN 0V 5V/div 0V VLX 5V/div 0V ILOAD 1A/div 0A 15 10 5 IOUT = 0A 0 0.5 1.0 1.5 2.0 2.5 500ns/div 3.0 REFIN VOLTAGE (V) OVP SHUTDOWN HICCUP MODE toc20 toc21 VOUT 5V/div 0V VOUT 5V/div 0V 1V/div VOVP 0V VDRV 5V/div 0V 5V/div 0V VPGOOD 5V/div 0V 5V/div 0V VPGOOD 1s/div www.maximintegrated.com VDRV 20ms/div Maxim Integrated │  7 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Typical Operating Characteristics (continued) (VSUP = 14V, TA = +25NC, unless otherwise noted.) MAX17292 EFFICIENCY toc22 100 100 IOUT = 1A 95 95 85 IOUT = 2A 80 IOUT = 100mA 75 70 85 80 70 65 60 60 55 55 50 50 5 8 6 7 SUPPLY VOLTAGE (V) IOUT = 100mA 4 5 700 600 500 400 300 200 252 250 248 246 244 242 240 0 100 200 300 1400 1200 1000 15 20 25 30 91.0 toc28 90.0 89.5 89.0 88.5 88.0 87.5 -40 -20 RSET(kΩ) 1600 90.5 254 0 1800 MAX17292 MAXIMUM DUTY CYCLE vs. TEMPERATURE toc27 256 100 2000 10 MAXIMUM DUTY CYCLE (%) CURRENT-LIMIT THRESHOLD (mV) 800 2200 800 258 900 2400 RSET (kI) 260 toc25 1000 toc24 2600 8 6 7 SUPPLY VOLTAGE (V) CURRENT-LIMIT THRESHOLD vs. TEMPERATURE MAX17290 INTERNAL OSCILLATOR FREQUENCY vs. RSET 1100 IOUT = 1A 75 65 4 INTERNAL OSCILLATOR FREQUQNCY (kHz) IOUT = 2A 90 EFFICIENCY (%) EFFICIENCY (%) 90 MAX17292 INTERNAL OSCILLATOR FREQUENCY vs. RSET toc23 INTERNAL OSCILLATOR FREQUQNCY (kHz) MAX17290 EFFICIENCY 0 20 40 60 80 100 120 RSET = 12.1kI 87.0 -40 -20 TEMPERATURE (°C) MAX17290 MAXIMUM DUTY CYCLE vs. TEMPERATURE 0 20 40 60 80 100 120 TEMPERATURE (°C) INPUT VOLTAGE TRANSIENT toc29 toc30 MAXIMUM DUTY CYCLE (%) 95.9 95.7 5V/div 95.5 VIN 95.3 VOUT 0V 5V/div 95.1 0V 1A /div 0A 5V/div 0V ILOAD 94.9 VPGOOD 94.7 RSET = 68.1kI 94.5 -40 -20 0 20 40 60 80 100 120 20ms/div TEMPERATURE (°C) www.maximintegrated.com Maxim Integrated │  8 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI 3 FSET/SYNC DRV 4 PVL 5 MAX17292EUBA/B EP 8 7 PGOOD 6 ISNS MAX17290ETCC/D MAX17292ETCC/D COMP 11 FB 12 µMAX EP + 6 ISNS 5 PVL COMP 11 4 DRV FB 12 PGOOD REFIN SYNCO 7 FSET/SYNC 10 MAX17290ETCE/F MAX17292ETCE/F EP + 1 2 3 1 2 TQFN (3mm x 3mm) 6 ISNS 5 PVL 4 DRV 3 EN GND FSET/SYNC 10 8 GND COMP 9 SUP FB 7 EN 10 8 GND 1 2 MAX17290EUBA/B 9 EN 9 TOP VIEW SUP SUP + OVP TOP VIEW REFIN TOP VIEW PGOOD Pin Configurations TQFN (3mm x 3mm) Pin Descriptions MAX17290EUBA/B, MAX17292EUBA/B MAX17290ETCC/D, MAX17292ETCC/D MAX17290ETCE/F, MAX17292ETCE/F μMAX-EP TQFN-EP TQFN-EP 1 1 1 SUP Power-Supply Input. Place a bypass capacitor of at least 1FF between this pin and ground. 2 3 3 EN Active-High Enable Input. This input is high-voltage capable or can alternatively be driven from a logiclevel signal. 3 2 2 GND Ground Connection 4 4 4 DRV Drive Output for Gate of nMOS Boost Switch. The nominal voltage swing of this output is between PVL and GND. 5 5 5 PVL Output of 5V Internal Regulator. Connect a ceramic capacitor of at least 2.2FF from this pin to ground, placing it as close as possible to the pin. ISNS Current-Sense Input to Regulator. Connect a sense resistor between the source of the external switching FET and GND. Then connect another resistor between ISNS and the source of the FET for slope compensation adjustment. SYNCO Open-Drain Synchronization Output. SYNCO outputs a square-wave signal which is 180N out-of-phase with the device’s operational clock. Connect a pullup resistor from this pin to PVL or to a 5V or lower supply when used. 6 — www.maximintegrated.com 6 — 6 7 NAME FUNCTION Maxim Integrated │  9 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Pin Descriptions (continued) MAX17290EUBA/B, MAX17292EUBA/B MAX17290ETCC/D, MAX17292ETCC/D MAX17290ETCE/F, MAX17292ETCE/F μMAX-EP TQFN-EP TQFN-EP — — 7 7 8 9 — 8 9 NAME FUNCTION OVP Overvoltage Protection Input. When this pin goes above 110% of the FB regulation voltage, all switching is disabled. Operation resumes normally when OVP drops below 107.5% of the FB regulation point. Connect a resistor-divider between the output, OVP, and GND to set the overvoltage protection level. REFIN Reference Input. When using the internal reference connect REFIN to PVL. Otherwise, drive this pin with an external voltage between 0.5V and 2V to set the boost output voltage. PGOOD Open-Drain Power-Good Output. Connect a resistor from this pin to PVL or to another voltage less than or equal to 5V. PGOOD goes high after soft-start when the output exceeds 90% of its final value. When EN is low PGOOD is also low. After soft-start is complete, if PGOOD goes low and 16 consecutive current-limit cycles occur, the devices enter hiccup mode and a new soft-start is initiated after a delay of 44ms. 8 10 10 FSET/ SYNC Frequency Set/Synchronization. To set a switching frequency between 100kHz and 1000kHz (MAX16990) or between 1000kHz and 2500kHz (MAX16992), connect a resistor from this pin to GND. To synchronize the converter, connect a logic signal in the range 220kHz to 1000kHz (MAX16990) or 1000kHz to 2500kHz (MAX16992) to this input. The external nMOSFET is turned on (i.e., DRV goes high) after a short delay (60ns for 2.2MHz operation, 125ns for 400kHz) when SYNC transitions low. 9 11 11 COMP Output of Error Amplifier. Connect the compensation network between COMP and GND. FB Boost Converter Feedback. This pin is regulated to 1V when REFIN is tied to PVL or otherwise regulated to REFIN during boost operation. Connect a resistordivider between the boost output, the FB pin and GND to set the boost output voltage. In a two-phase converter connect the FB pin of the slave IC to PVL. EP Exposed Pad. Internally connected to GND. Connect to a large ground plane to maximize thermal performance. Not intended as an electrical connection point. 10 — www.maximintegrated.com 12 — 12 — Maxim Integrated │  10 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Functional Diagram 5V REGULATOR + REFERENCE SUP PVL (OVP) UVLO REF. EN EN THERMAL DRV THERMAL 50µA x fSW GND 250mV FSET/SYNC OSCILLATOR BLANKING TIME CONTROL LOGIC ISNS 8 (SYNCO) AGND PGOOD COMP PGOOD COMPARATOR AGND MAX17290 MAX17292 FB OTA VPVL - 0.4V 1V (REFIN) Detailed Description The MAX17290/MAX17292 are high-performance, current-mode PWM controllers for wide input voltage range boost converters. The input operating voltage range of 4.5V to 36V makes these devices ideal in battery operated harsh environment applications such as for front-end “preboost” for the first boost stage in high-power LED lighting applications. An internal low-dropout regulator (PVL regulator) with an output voltage of 5V enables the devices to operate directly from an automotive battery www.maximintegrated.com input. The input operating range can be as low as 2.5V when the converter output supplies the SUP input. The input undervoltage lockout (UVLO) circuit monitors the PVL voltage and turns off the converter when the voltage drops below 3.6V (typ). An external resistor programs the switching frequency in two ranges from 100kHz to 1000kHz (MAX17290) or between 1000kHz and 2500kHz (MAX17292). The FSET/SYNC input can also be used for synchronization to an external clock. The SYNC pulse width should be greater than 70ns. Maxim Integrated │  11 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Inductor current information is obtained by means of an external sense resistor connected from the source of the external nMOSFET to GND. The devices include an internal transconductance error amplifier with 1% accurate reference. At startup, the internal reference is ramped in a time of 9ms to obtain soft-start. The devices also include protection features such as hiccup mode and thermal shutdown as well as an optional overvoltage-detection circuit (OVP pin, C and D versions). Current-Mode Control Loop The MAX17290/MAX17292 offers peak current-mode control operation for best load step performance and simpler compensation. The inherent feed-forward characteristic is useful especially in applications where the input voltage changes quickly. While the current-mode architecture offers many advantages, there are some shortcomings. In high duty-cycle operation, subharmonic oscillations can occur. To avoid this, the device offers programmable slope compensation using a single resistor between the ISNS pin and the current-sense resistor. To avoid premature turn-off at the beginning of the on-cycle the current-limit and PWM comparator inputs have leading-edge blanking. Startup Operation/UVLO/EN The devices feature undervoltage lockout on the PVLregulator and turn on the converter once PVL rises above 4V. The internal UVLO circuit has about 400mV hysteresis to avoid chattering during turn-on. Once the converter is operating and if SUP is fed from the output, the converter input voltage can drop below 4.5V. This feature allows operation at voltages as low as 2.5V or even lower with careful selection of external components. The EN input can be used to disable the device and reduce the standby current to less than 4μA (typ). Soft-Start The devices are provided with an internal soft-start time of 9ms. At startup, after voltage is applied and the UVLO threshold is reached, the device enters soft-start. During soft-start, the reference voltage ramps linearly to its final value in 9ms. www.maximintegrated.com Oscillator Frequency/External Synchronization/ Spread Spectrum Use an external resistor at FSET/SYNC to program the MAX17290 internal oscillator frequency from 100kHz to 1MHz and the MAX17292 frequency between 1MHz and 2.5MHz. See TOCs 24 and 25 in the Typical Operating Characteristics section for resistor selection. The SYNCO output is a 180N phase-shifted version of the internal clock and can be used to synchronize other converters in the system or to implement a two-phase boost converter with a second MAX17290/MAX17292. The advantages of a two-phase boost topology are lower input and output ripple and simpler thermal management as the power dissipation is spread over more components. See the Multiphase Operation section for further details. The devices can be synchronized using an external clock at the FSET/SYNC input. A falling clock edge on FSET/ SYNC turns on the external MOSFET by driving DRV high after a short delay. The B, D, and F versions of the devices have spreadspectrum oscillators. In these parts the internal oscillator frequency is varied dynamically ±6% around the switching frequency. Spread spectrum can improve system EMI performance by reducing the height of peaks due to the switching frequency and its harmonics in the spectrum. The SYNCO output includes spread-spectrum modulation when the internal oscillator is used on the B, D, and F versions. Spread spectrum is not active when an external clock is applied to the FSET/SYNC pin. nMOSFET Driver DRV drives the gate of an external nMOSFET. The driver is powered by the internal regulator (PVL), which provides approximately 5V. This makes both the devices suitable for use with logic-level MOSFETs. DRV can source 750mA and sink 1000mA peak current. The average current sourced by DRV depends on the switching frequency and total gate charge of the external MOSFET (see the Power Dissipation section). Maxim Integrated │  12 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Error Amplifier The devices include an internal transconductance error amplifier. The noninverting input of the error amplifier is connected to the internal 1V reference and feedback is provided at the inverting input. High 700μS open-loop transconductance and 50MΩ output impedance allow good closed-loop bandwidth and transient response. Moreover, the source and sink current capability of 140μA provides fast error correction during output load transients. Slope Compensation The devices use an internal current-ramp generator for slope compensation. The internal ramp signal resets at the beginning of each cycle and slews at a typical rate of 50μA x fSW. The amount of slope compensation needed depends on the slope of the current ramp in the inductor. See the Current-Sense Resistor Selection and Setting Slope Compensation section for further information. Current Limit The current-sense resistor (RCS) connected between the source of the MOSFET and ground sets the current limit. The ISNS input has a voltage trip level (VCS) of 250mV. When the voltage produced by the current in the inductor exceeds the current-limit comparator threshold, the MOSFET driver (DRV) quickly terminates the on-cycle. In some cases, a short time-constant RC filter could be required to filter out the leading-edge spike on the sense waveform in addition to the internal blanking time. The amplitude and width of the leading edge spike depends on the gate capacitance, drain capacitance, and switching speed (MOSFET turn-on time). Hiccup Operation The devices incorporate a hiccup mode in an effort to protect the external power components when there is an output short-circuit. If PGOOD is low (i.e., the output voltage is less than 85% of its set value) and there are 16 consecutive current-limit events, switching is stopped. There is then a waiting period of 44ms before the device tries to restart by initiating a soft-start. Note that a short-circuit on the output places considerable stress on all the power components even with hiccup mode, so that careful component selection is important if this condition is encountered. For more complete protection against output short-circuits, a series pMOS switch driven from PGOOD through a level-shifter can be employed. www.maximintegrated.com Applications Information Inductor Selection Using the following equation, calculate the minimum inductor value so that the converter remains in continuous mode operation at minimum output current (IOMIN): LMIN = (VIN2 x D x E)/(2 x fSW x VOUT x IOMIN) where: D = (VOUT + VD - VIN)/(VOUT + VD - VDS) and: IOMIN is between 10% and 25% of IOUT A higher value of IOMIN reduces the required inductance; however, it increases the peak and RMS currents in the switching MOSFET and inductor. Select IOMIN between 10% to 25% of the full load current. VD is the forward voltage drop of the external Schottky diode, D is the duty cycle, and VDS is the voltage drop across the external switch. Select an inductor with low DC resistance and with a saturation current (ISAT) rating higher than the peak switch current limit of the converter. Input and Output Capacitors The input current to a boost converter is almost continuous and the RMS ripple current at the input capacitor is low. Calculate the minimum input capacitor value and maximum ESR using the following equations: CIN = DIL x D/(4 x fSW x DVQ) ESRMAX = DVESR/DIL where DIL = ((VIN - VDS) x D)/(L x fSW). VDS is the total voltage drop across the external MOSFET plus the voltage drop across the inductor ESR. DIL is peak-to-peak inductor ripple current as calculated above. DVQ is the portion of input ripple due to the capacitor discharge and DVESR is the contribution due to ESR of the capacitor. Assume the input capacitor ripple contribution due to ESR (DVESR) and capacitor discharge (DVQ) are equal when using a combination of ceramic and aluminium capacitors. During the converter turn-on, a large current is drawn from the input source especially at high output to input differential. The devices have an internal soft-start, however, a larger input capacitor than calculated above could be necessary to avoid chattering due to finite hysteresis during turn-on. Maxim Integrated │  13 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI In a boost converter, the output capacitor supplies the load current when the main switch is on. The required output capacitance is high, especially at lower duty cycles. Also, the output capacitor ESR needs to be low enough to minimize the voltage drop due to the ESR while supporting the load current. Use the following equations to calculate the output capacitor, for a specified output ripple. All ripple values are peak-to-peak. ESR = DVESR/IOUT COUT = (IOUT x DMAX)/(DVQ x fSW) where IOUT is the output current, DVQ is the portion of the ripple due to the capacitor discharge, and DVESR is the ripple contribution due to the ESR of the capacitor. DMAX is the maximum duty cycle (i.e., the duty cycle at the minimum input voltage). Use a combination of low-ESR ceramic and high-value, low-cost aluminium capacitors for lower output ripple and noise. Current-Sense Resistor Selection and Setting Slope Compensation Set the current-limit threshold 20% higher than the peak switch current at the rated output power and minimum input voltage. Use the following equation to calculate an initial value for RCS: RCS = 0.2/{1.2 x [((VOUT x IOUT)/E)/VINMIN + 0.5 x ((VOUT – VINMIN)/VOUT) x (VINMIN/(fSW x L))]} where E is the estimated efficiency of the converter (use 0.85 as an initial value or consult the graph in the Typical Operating Characteristics section); VOUT and IOUT are the output voltage and current, respectively; VINMIN is the minimum value of the input voltage; fSW is the switching frequency; and L is the minimum value of the chosen inductor. The devices use an internal ramp generator for slope compensation to stabilize the current loop when operating at duty cycles above 50%. The amount of slope compensation required depends on the down-slope of the inductor current when the main switch is off. The inductor down-slope in turn depends on the input to output voltage differential of the converter and the inductor value. Theoretically, the compensation slope should be equal to 50% of the inductor downslope; however, a little higher than 50% slope is advised. Use the following equation to calculate the required compensating slope (mc) for the boost converter: The internal ramp signal resets at the beginning of each cycle and slews at the rate of 50μA x fSW. Adjust the amount of slope compensation by choosing RSCOMP to satisfy the following equation: RSCOMP = (mc x RCS)/(50e-6 x fSW) In some applications a filter could be needed between the current-sense resistor and the ISNS pin to augment the internal blanking time. Set the RC time constant just long enough to suppress the leading-edge spike of the MOSFET current. For a given design, measure the leading spike at the lowest input and rated output load to determine the value of the RC filter which can be formed from the slope-compensation resistor and an added capacitor from ISNS to GND. MOSFET Selection The devices drive a wide variety of logic-level n-channel power MOSFETs. The best performance is achieved with low-threshold nMOSFETs that specify on-resistance with a gate-source voltage (VGS) of 5V or less. When selecting the MOSFET, key parameters can include: 1) Total gate charge (Qg). 2) Reverse-transfer capacitance or charge (CRSS). 3) On-resistance (RDS(ON)). 4) Maximum drain-to-source voltage (VDS(MAX)). 5) Maximum gate frequencies threshold voltage (VTH(MAX)). Non-Synchronous Diode Selection The average diode current for a Boost converter is equal to the output load current. The peak diode current depends on how much ripple current is implemented in the design. Therefore at minimum, choose a diode with average forward current rating that is higher than the output current and ensure the peak forward current rating is higher than the output current plus one half the ripple current. As a rule of thumb, choose I_AVG_DIODE at least equal to two times IOUT for minimum power loss and proper component thermal dissipation. Once that is met the diode’s peak specification will be more than enough. I_AVG_DIODE = IOUT x 2 V_DIODE >> VOUT mc = 0.5 x (VOUT – VIN)/L A/s www.maximintegrated.com Maxim Integrated │  14 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI At high switching frequencies, dynamic characteristics (parameters 1 and 2 of the above list) that predict switching losses have more impact on efficiency than RDS(ON), which predicts DC losses. Qg includes all capacitances associated with charging the gate. The VDS(MAX) of the selected MOSFET must be greater than the maximum output voltage setting plus a diode drop (or the maximum input voltage if greater) plus an additional margin to allow for spikes at the MOSFET drain due to the inductance in the rectifier diode and output capacitor path. In addition, Qg determines the current needed to drive the gate at the selected operating frequency via the PVL linear regulator and thus determines the power dissipation of the IC (see the Power Dissipation section). Low-Voltage Operation The devices operate down to a voltage of 4.5V or less on their SUP pins. If the system input voltage is lower than this the circuit can be operated from its own output as shown in the Typical Application Circuit. At very low input voltages it is important to remember that input current will be high and the power components (inductor, MOSFET, and diode) must be specified for this higher input current. VOUT INPUT In addition, the current-limit of the devices must be set high enough so that the limit is not reached during the ontime of the MOSFET which would result in output power limitation and eventually entering hiccup mode. Estimate the maximum input current using the following equation: IINMAX = ((VOUT x IOUT)/E)/VINMIN + 0.5 x ((VOUT – VINMIN)/VOUT) x (VINMIN/(fSW x L)) where IINMAX is the maximum input current; VOUT and IOUT are the output voltage and current, respectively; E is the estimated efficiency (which is lower at low input voltages due to higher resistive losses); VINMIN is the minimum value of the input voltage; fSW is the switching frequency; and L is the minimum value of the chosen inductor. Boost Converter Compensation Refer to Application Note 5587. Overvoltage Protection The “C” and “D” variants of the devices include the overvoltage protection input. When the OVP pin goes above 110% of the FB regulation voltage, all switching is disabled. For an example application circuit, see Figure 2. INPUT VOUT SUP SUP EN DRV DRV N N ISNS REFIN ISNS MAX17290 MAX17292EUBA PVL PVL MAX17290ETCC/D MAX17292ETCC/D PVL OVP PGOOD FB FB SYNCO FSET/SYNC COMP COMP FSET/SYNC GND EN ENABLE GND Figure 1. Standard Boost Application Circuit. www.maximintegrated.com Figure 2. Application with Independent Output Overvoltage Protection Maxim Integrated │  15 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI 1µF 10µH VIN 22µF SUP DRV 2x47µF CERAMIC N 50V/1A FSET/SYNC 2200Ω 69kΩ ISNS MAX17290ETCE/F 20mΩ PGOOD PVL 2.2µF COMP FB 10kΩ SYNCO EN GND 10µH 22µF 1µF FSET/ SYNC SUP DRV N 75kΩ 2200Ω ISNS REFIN 20mΩ MAX17290ETCE/F PGOOD PVL 2.2µF 1500Ω COMP SYNCO GND FB EN N ENABLE Figure 3. Two-Phase 400kHz Boost Application with Minimum Component Count Multiphase Operation Two boost phases can be implemented with no extra components using two ICs as shown in Figure 3. In this circuit the SYNCO output of the master device drives the SYNC input of the slave forcing it to operate 180N outof-phase. The FB pin of the slave device is connected to PVL, thus disabling its error amplifier. In this way the error amplifier of the master controls both devices by means of the COMP signal and good current-sharing is attained between the two phases. When designing the PCB for a www.maximintegrated.com multiphase converter it is important to protect the COMP trace in the layout from noisy signals by placing it on an inner layer and surrounding it with ground traces. Using REFIN to Adjust the Output Voltage The REFIN pin can be used to directly adjust the reference voltage of the boost converter, thus altering the output voltage. When not used, REFIN should be connected to PVL. Because REFIN is a high-impedance pin, it is simple to drive it by means of an external digitalto-analog converter (DAC) or a filtered PWM signal. Maxim Integrated │  16 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Power Dissipation The power dissipation of the IC comes from two sources: the current consumption of the IC itself and the current required to drive the external MOSFET, of which the latter is usually dominant. The total power dissipation can be estimated using the following equation: where VSUP is the voltage at the SUP pin of the IC, ICC is the IC quiescent current consumption or typically 0.75mA (MAX17290) or 1.25mA (MAX17292), Qg is the total gate charge of the chosen MOSFET at 5V, and fSW is the switching frequency. PIC reaches it maximum at maximum VSUP. PIC = VSUP x ICC + (VSUP – 5) x (Qg x fSW) Ordering Information FREQUENCY RANGE OVP/ SYNCO SPREAD SPECTRUM TEMP RANGE PIN-PACKAGE MAX17290EUBA+ 220kHz to 1MHz None Off -40NC to +85NC 10 FMAX-EP* MAX17290EUBB+ 220kHz to 1MHz None On -40NC to +85NC 10 FMAX-EP* MAX17290ETCC+ 220kHz to 1MHz OVP Off -40NC to +85NC 12 TQFN-EP* MAX17290ETCD+ 220kHz to 1MHz OVP On -40NC to +85NC 12 TQFN-EP* MAX17290ETCE+ 220kHz to 1MHz SYNCO Off -40NC to +85NC 12 TQFN-EP* MAX17290ETCF+ 220kHz to 1MHz SYNCO On -40NC to +85NC 12 TQFN-EP* MAX17292EUBA+ 1MHz to 2.5MHz None Off -40NC to +85NC 10 FMAX-EP* MAX17292EUBB+ 1MHz to 2.5MHz None On -40NC to +85NC 10 FMAX-EP* MAX17292ETCC+ 1MHz to 2.5MHz OVP Off -40NC to +85NC 12 TQFN-EP* MAX17292ETCD+ 1MHz to 2.5MHz OVP On -40NC to +85NC 12 TQFN-EP* MAX17292ETCE+ 1MHz to 2.5MHz SYNCO Off -40NC to +85NC 12 TQFN-EP* MAX17292ETCF+ 1MHz to 2.5MHz SYNCO On -40NC to +85NC 12 TQFN-EP* PART +Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad. Chip Information PROCESS: BiCMOS www.maximintegrated.com Package Information For the latest package outline information and land patterns (footprints), go to www.maximintegrated.com/packages. Note that a “+”, “#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE PACKAGE CODE OUTLINE NO. LAND PATTERN NO. 12 TQFN-EP T1233+4 21-0136 90-0019 10 μMAX-EP U10E+3 21-0109 90-0148 Maxim Integrated │  17 MAX17290/MAX17292 2.5V to 36V, 2.5MHz, PWM Boost Controller with 4µA Shutdown Current and Reduced EMI Revision History REVISION NUMBER REVISION DATE 0 8/16 DESCRIPTION Initial release PAGES CHANGED — For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com. Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance. Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc. © 2016 Maxim Integrated Products, Inc. │  18