EVALUATION KIT AVAILABLE
MAX20090/MAX20090B
General Description
The MAX20090 is a single-channel high-brightness LED
(HB LED) driver for automotive front-light applications such
as high beam, low beam, daytime running lights (DRLs),
turn indicators, fog lights, and other LED lights. It can take
an input voltage from 5V to 65V and drive a string of LEDs
with a maximum output voltage of 65V.
The device senses output current at the high side of the
LED string. High-side current sensing is required to protect for shorts from the output to the ground or battery
input. It is also the most flexible scheme for driving LEDs,
allowing boost, high-side buck, SEPIC mode, or buckboost-mode configurations. The PWM input provides LED
dimming ratios of up to 1000:1, and the ICTRL input provides
additional analog dimming capability in the controller. The
device also includes a fault flag (FLT) that indicates open
string, shorted string, and thermal shutdown. The device
has built-in spread-spectrum modulation for improved
electromagnetic-compatibility performance. The device
can also be used in zeta and Ćuk converter configurations, if necessary in some applications. The MAX20090 is
available in a space-saving (4mm x 4mm), 20-pin TQFN,
20-pin side-wettable TQFN, or a 20-pin TSSOP package and is specified to operate over the -40°C to +125°C
automotive temperature range
Applications
●
●
●
●
●
●
Automotive Exterior Lighting
High-Beam/Low-Beam/Signal/Position Lights
Daytime Running Lights (DRLs)
Fog Lights and Adaptive Front-Light Assemblies
Head-Up Displays
Commercial, Industrial, and Architectural Lighting
Automotive High-Voltage,
High-Brightness LED Controller
Benefits and Features
● High-Brightness LED Driver with a Wide Input Range
Saves Space and Cost Through Integration
• +5V to +65V Wide Input Voltage Range
• +65V Maximum Boost Output Voltage
• ICTRL Pin for Analog Dimming
• Integrated High-Side Current-Sense Amplifier
• 200Hz On-Board Ramp Simplifies PWM Dimming
● Flexible Architecture Enables Easy Design
Optimization
• Configurable as Boost, High-Side Buck, SEPIC,
Buck-Boost, Zeta, and Ćuk
• Programmable Switching Frequency (200kHz to
2.2MHz)
• Spread-Spectrum Modulation to Reduce EMI Noise
● Automotive Features and Robustness Improve
System Reliability
• Fault Diagnosis Through Fault Flag
• Short Circuit, Overvoltage, and Thermal Protection
• -40°C to +125°C Operating Temperature Range
Simplified Typical Operating Circuit
L1
INPUT
BST
IN
RUVEN 1
MAX20090
UVEN
PWMDIM
PWMDIM
RRT
RT
C2
V7V
RUVEN 2
C1
Ordering Information appears at end of data sheet.
19-8747; Rev 5; 5/19
N1
CS
OVP
COUT
ROVP1
RSC
RCS_LED
ISENSE+
ISENSE100Ω
DIMOUT
P1
FLT
COMP
VCC
ANALOG DIM
D1
CBST
NDRV
CIN
RCOMP
ICTRL
SGND
EP PGND
CCOMP
RCS_FET
ROVP2
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Simplified Schematic
L1
INPUT
D1
BST
IN
CBST
COUT
NDRV
CIN
N1
ROVP1
CS
RUVEN1
OVP
UVEN
PWMDIM
RCS_LED
ISENSE+
PWMDIM
RRT
RSC
ISENSE100Ω
RT
MAX20090
RUVEN2
DIMOUT
C2
P1
FLT
V7V
RCS_FET
C1
ROVP2
COMP
VCC
R1
RCOMP
ICTRL
R2
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SGND
PGND
EP
CCOMP
Maxim Integrated │ 2
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Absolute Maximum Ratings
IN to PGND............................................................-0.3V to +70V
ISENSE+, ISENSE-, DIMOUT to PGND................-0.3V to +70V
DIMOUT to ISENSE+............................................-8.5V to +0.3V
ISENSE- to ISENSE+............................................-0.3V to +0.3V
PGND to SGND.....................................................-0.6V to +0.3V
VCC, UVEN to PGND...............................................-0.3V to +6V
V7V to PGND...........................................................-0.3V to +9V
BST to PGND..................................................-0.3V to V7V + 5V
BST to NDRV...........................................................-0.3V to +6V
NDRV to PGND.....................................................-0.3V to +7.3V
OVP, PWMDIM, ICTRL, FLT to PGND.....................-0.3V to +6V
COMP, RT, CS to PGND....................................... -0.3V to +VCC
Continuous Current on IN.................................................100mA
Peak Current on NDRV..........................................................+2A
Continuous Current on NDRV...........................................+50mA
Continuous Power Dissipation (TA = +70°C) (Note 1)
20-pin TSSOP
(derate 26.1mW/°C above +70°C)............................2089 mW
20-pin TQFN
(derate 25.6mW/°C above +70°C).............................2051mW
Operating Temperature Range.......................... -40°C to +125°C
Junction Temperature.......................................................+150°C
Storage Temperature Range............................. -65°C to +150°C
Lead Temperature (soldering, 10s).................................. +300°C
Soldering Temperature (reflow)........................................+260°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Package Thermal Characteristics (Note 1)
TSSOP
Junction-to-Ambient Thermal Resistance (θJA)...........37°C/W
Junction-to-Case Thermal Resistance (θJC)..................2°C/W
TQFN
Junction-to-Ambient Thermal Resistance (θJA)...........33°C/W
Junction-to-Case Thermal Resistance (θJC)..................2°C/W
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-layer
board. For detailed information on package thermal considerations, refer to www.maximintegrated.com/thermal-tutorial.
Electrical Characteristics
VIN = 12V, RRT = 85.4kΩ, CIN = CVCC = 1μF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40°C to +125°C, unless otherwise
noted. Typical values are at TA = +25°C.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
65
V
1.8
5.0
mA
1.24
1.37
V
SUPPLY VOLTAGE
Input Voltage Range
VIN
Supply Current
IINQ
5.0
VOVP = 1.5V, no switching
UNDERVOLTAGE LOCKOUT
Undervoltage-Lockout Rising
VUVEN_THUP
VUVEN rising
1.12
Hysteresis
Shutdown Current
106
ISHTDN
VUVEN = 0V, VIN = 12V
mV
15
26
μA
5
5.125
V
0.16
V
VCC REGULATOR
Load 0.1mA to 15mA,
6.0V < VIN < 16V
Output Voltage
VCC
Dropout Voltage
VCC DROP
VIN = 4.5V, IVCC = 5mA
0.07
VCC UVLO Rising
VCC UVLOR
Rising
4.0
V
0.4
V
Hysteresis
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4.875
Maxim Integrated │ 3
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Electrical Characteristics (continued)
VIN = 12V, RRT = 85.4kΩ, CIN = CVCC = 1μF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40°C to +125°C, unless otherwise
noted. Typical values are at TA = +25°C.) (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
0.1mA ≤ IVCC ≤ 50mA, 9V ≤
VIN ≤ 12V
6.72
7.0
7.28
12V ≤ VIN ≤ 65V, IVCC = 10mA
6.72
UNITS
V7V REGULATOR
Output Voltage
Dropout Voltage
V7V UVLO Rising
V7V
V7VDROPOUT
V7VUVLO_R
V
7.0
7.28
VIN = 5.0V, IV7V = 50mA
0.175
0.42
V
Rising
4.33
4.7
V
Hysteresis
Short-Circuit Current Limit
0.36
IV7VSC
V7V shorted to GND, VIN = 5V
V
55
mA
BOOTSTRAP SUPPLY
BST Input Current
IBST_OFF
0.02
mA
OSCILLATOR (RT)
Switching-Frequency Range
fSW
Bias Voltage at RT
VRT
Minimum Off-Time
tOFF_MIN
200
VCOMP = high, VCS = 0V
Oscillator Frequency Accuracy
Dither disabled
Frequency Dither
fDITH
Dither enabled, fSW = 200kHz to
2.2MHz
ISLOPE
Peak current ramp added to CS
input per switching cycle
2200
kHz
1.25
V
85
ns
-10
+10
±6
%
%
SLOPE COMPENSATION
Slope-Compensation Current-Ramp
Height
42.5
50
57.5
μA
1.2
V
ANALOG DIMMING
ICTRL Input Control-Voltage Range
ICTRL Zero-Current Threshold
ICTRL Clamp Voltage
ICTRL Input Bias Current
0.2
ICTRLRNG
(VISENSE+ - VISENSE-) < 5mV
0.16
0.18
0.200
V
ICTRLCLMP
ICTRL sink = 1μA
1.25
1.30
1.35
V
ICTRLIIN
VICTRL = 0 to 5.5V
20
500
nA
-0.2
+65
V
0
225
mV
ICTRLZC_VTH
LED CURRENT-SENSE AMPLIFIER
Common-Mode Input Range
Differential Signal Range
ISENSE+/- Input Bias Current
Input Offset Voltage
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IBISENSE+
VISENSE+ - VISENSE- = 200mV,
VISENSE+ = 60V
350
550
IBISENSE-
VISENSE+ - VISENSE- = 200mV,
VISENSE+ = 60V
22
60
μA
TJ = +25°C, VISENSE+,
VISENSE- = 3V to 60V
-0.1
3V < VISENSE+, VISENSE- < 60V
-0.1
mV
Maxim Integrated │ 4
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Electrical Characteristics (continued)
VIN = 12V, RRT = 85.4kΩ, CIN = CVCC = 1μF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40°C to +125°C, unless otherwise
noted. Typical values are at TA = +25°C.) (Note 2)
PARAMETER
SYMBOL
Voltage Gain
LED Current-Sense Regulation
Voltage
VSENSE
LED Current-Sense Regulation
Voltage (Low Range)
Common-Mode Input Range
Selector
RNGSEL
MIN
TYP
MAX
UNITS
(VISENSE+ - VISENSE-) = 200mV,
3V < VISENSE+, VISENSE- < 60V
CONDITIONS
4.90
5.0
5.1
V/V
VICTRL = 1.3V,
3V < (VISENSE+, VISENSE-) < 60V
213.8
220
226.2
VICTRL = 1.2V,
3V < (VISENSE+, VISENSE-) < 60V
194.0
200
206.0
VICTRL = 0.4V,
3V < (VISENSE+, VISENSE-) < 60V
37.0
40
43.0
VICTRL = 1.2V,
0V < (VISENSE+, VISENSE-) < 3V
192
200
208
VICTRL = 0.4V,
0V < (VISENSE+, VISENSE-) < 3V
35
40
45
VISENSE+ rising
2.72
2.85
2.98
VISENSE+ falling
2.48
2.6
2.72
VISENSE+ - VISENSE- = 200mV
1170
1800
2430
mV
mV
V
ERROR AMP
Transconductance
gM
μS
COMP Sink Current
COMPISINK
VCOMP = 5V
300
μA
COMP Source Current
COMPISRC
VCOMP = 0V
300
μA
1
V
90
ns
PWM COMPARATOR
Input Offset Voltage
Includes leading-edge blanking
time
PWM-to-NDRV Propagation Delay
CS LIMIT COMPARATOR
Current-Limit Threshold
VCS_LIMIT
388
418
448
mV
RDS(ON) Pullup nMOS
RNDRVON
0.3
0.6
1.3
Ω
RDS(ON) Pulldown nMOS
RNDRVOFF
0.3
0.6
1.3
GATE DRIVER (NDRV)
VCOMP = 0V, ISINK = 100mA
Ω
Rise Time
tR
CNDRV = 10nF
40
ns
Fall Time
tF
CNDRV = 10nF
40
ns
PWM DIMMING
Internal Ramp Frequency
fRAMP
160
External Sync-Frequency Range
fDIM
60
External Sync Low-Level Voltage
VLTH
External Sync High-Level Voltage
DIM Comparator Offset Voltage
DIM Voltage for 100% Duty Cycle
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VHTH
2
VDIMOFS
170
3.3
200
240
Hz
2000
Hz
0.4
V
230
mV
V
200
V
Maxim Integrated │ 5
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Electrical Characteristics (continued)
VIN = 12V, RRT = 85.4kΩ, CIN = CVCC = 1μF, NDRV = COMP = DIMOUT = PWMDIM = FLT = unconnected,
VOVP = VCS = VPGND = VSGND = 0V, VISENSE+ = VISENSE- = 45V, VICTRL = 1.40V, TA = TJ = -40ºC to +125ºC, unless otherwise
noted. Typical values are at TA = +25C (Note 2)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
PWMDIM-Low to NDRV-Low Delay
70
ns
PWMDIM-High to NDRV-High Delay
40
ns
PWMDIM-to-LED Turn-Off Time
PWMDIM falling edge to rising edge
on DIMOUT, CDIMOUT = 10nF
2
μs
PWMDIM-to-LED Turn-On Time
PWMDIM rising edge to falling edge
on DIMOUT, CDIMOUT = 10nF
3
μs
pMOS GATE DRIVER (DIMOUT)
Peak Pullup Current
IDIMOUTPU
PWMDIM = 0V,
(VISENSE+ - VDIMOUT) = 7V
40
73
120
mA
Peak Pulldown Current
IDIMOUTPD
(VISENSE+ - VDIMOUT) = 0V
15
35
65
mA
-8.4
-7.4
-6.1
V
1.17
1.23
1.29
V
DIMOUT Low Voltage with Respect
to ISENSE+
OVERVOLTAGE PROTECTION (OVP)
OVP Threshold Rising
VOVP
Output rising
Hysteresis
Input Bias Current
70
IBOVP
VOVP = 1.235V
-500
(VISENSE+ - VISENSE-),
VOVP < 0.15V
369
mV
+500
nA
427
mV
SHORT-CIRCUIT HICCUP MODE
Short-Circuit Threshold
Hiccup Time
VSHORT-HIC
tHICCUP
398
Clock
Cycle
8192
SHORT-CIRCUIT VOLTAGE DETECT
Short-Circuit Voltage Detect
Threshold (MAX20090 only)
VSHORT-VOUT
(VISENSE+ - VIN) falling, VIN = 12V
1.15
1.55
1.95
V
68.6
200
mV
1
μA
OPEN-DRAIN FAULT (FLT)
Output-Voltage Low
VOL-FLT
VIN = 4.75V, VOVP = 2V, and
ISINK = 5mA
VFLT = 5V
Output Leakage Current
THERMAL SHUTDOWN
Thermal-Shutdown Threshold
TSHUTDOWN
Thermal-Shutdown Hysteresis
THYS
Temperature rising
165
°C
10
°C
Note 2: All devices are 100% tested at TA = TJ = +125°C, Limits over temperature are guaranteed by design.
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Maxim Integrated │ 6
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Operating Characteristics
(VIN = VEN = 12V, TA = +25°C, unless otherwise noted.)
QUIESCENT CURRENT
vs. SUPPLY VOLTAGE
UVEN vs. TEMPERATURE
toc01
1.50
4.0
1.30
1.25
1.20
1.15
3.0
2.5
2.0
1.5
1.10
1.0
1.05
0.5
1.00
-40
-10
20
50
80
0.0
110
0.10
0.08
0.06
0.04
0.02
0
10
20
30
40
0.00
50
0
10
SUPPLY VOLTAGE (V)
AMBIENT TEMPERATURE (°C)
SWITCHING FREQUENCY vs. RT
20
30
40
50
SUPPLY VOLTAGE (V)
VCC vs. SUPPLY VOLTAGE
VISENSE vs. VICTRL
toc04
1.0E+07
toc03
-40 °C
25 °C
125 °C
0.12
3.5
SUPPLY CURRENT (mA)
FALLING
1.35
0.14
-40 °C
25 °C
125 °C
4.5
RISING
1.40
UVEN VOLTAGE (V)
5.0
SUPPLY CURRENT (mA)
1.45
SHUTDOWN SUPPLY CURRENT
vs. SUPPLY VOLTAGE
toc02
toc05
0.30
toc06
5.04
5.03
5.02
5.01
0.20
5.00
VCC (V)
1.0E+06
VISENSE (V)
SWITCHING FREQUENCY (Hz)
0.25
0.15
0.10
1.0E+05
1.0E+05
RT (Ω)
0.00
1.0E+06
4.96
0
0.2
0.4
0.6
0.8
1
1.2
0
10
5.10
-40 °C
25 °C
125 °C
7.03
7.02
VCC VOLTAGE (V)
6.99
6.98
4.98
4.94
6.95
4.92
SUPPLY VOLTAGE (V)
toc08
5.00
4.96
40
50
5.02
6.96
30
40
5.04
6.97
20
30
5.06
7.00
10
20
5.08
7.01
V7V (V)
4.94
VCC vs. TEMPERATURE
toc07
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1.4
SUPPLY VOLTAGE (V)
V7V vs. SUPPLY VOLTAGE
0
-40 °C
25 °C
125 °C
4.95
ICTRL VOLTAGE (V)
7.04
6.94
4.98
4.97
0.05
1.0E+04
1.0E+04
4.99
50
4.90
-40
50
-10
20
80
AMBIENT TEMPERATURE (°C)
110
Maxim Integrated │ 7
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Operating Characteristics (continued)
(VIN = VEN = 12V, TA = +25°C, unless otherwise noted.)
7.08
45.0
7.06
40.0
7.04
35.0
7.02
30.0
7.00
6.98
6.94
10.0
6.92
5.0
-10
20
50
80
AMBIENT TEMPERATURE (°C)
RISING
0.30
0
0.15
2
4
6
8
0.00
10
0
10
Cndrv (nF)
20
30
V7V CURRENT (mA)
VDIMOUTB - VISENSE+
vs. TEMPERATURE
toc12
4.0E-06
-7.00
40
50
toc13
-7.05
3.5E-06
RISING
3.0E-06
-7.10
VDIMOUT - VISENSE+ (V)
FALLING
-7.15
2.5E-06
TiME (s)
0.20
0.05
DIMOUT RISE AND FALL TIMES
vs. NDRV TEMPERATURE
-7.20
2.0E-06
-7.25
1.5E-06
-7.30
1.0E-06
-7.35
5.0E-07
0.0E+00
0.25
0.10
0.0
110
toc11
TA = 125°C
TA = 25°C
TA = -40°C
0.35
FALLING
20.0
15.0
-40
0.40
25.0
6.96
6.90
V7V DROPOUT VOLTAGE
vs. V7V LOAD CURRENT
toc10
50.0
TiME (ns)
V7V VOLTAGE (V)
7.10
NDRV RISE AND FALL TIMES
vs. NDRV CAPACITANCE
toc09
VIN - VV7V (V)
V7V vs. TEMPERATURE
-7.40
-40
-10
20
50
80
110
-40
-10
20
50
TEMPERATURE (°C)
VISENSE vs. TEMPERATURE
110
600Hz DIMMING OPERATION
toc14
0.210
80
TEMPERATURE (°C)
toc15
0.208
5V/div
0.206
VPWMDIM
0.204
VISENSE (V)
0.202
1A/div
0.200
ILED
0.198
0.196
5V/div
VNDRV
0.194
0.192
0.190
-40
-10
20
50
80
110
400mS/div
TEMPERATURE (°C)
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Maxim Integrated │ 8
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
INSENSE-
DIMOUT
N.C.
IN
TOP VIEW
INSENSE +
Pin Configurations
15
14
13
12
11
TOP VIEW
N.C 16
10
V7V
UVEN 17
9
BST
MAX20090/
MAX20090B
VCC 18
8
NDRV
RT 19
7
PGND
PWMDIM 20
6
CS
EP
1
2
3
4
5
OVP
SGND
COMP
FLT
ICTRL
+
+
20
ISENSE+
2
19
ISENSE-
RT
3
18
DIMOUT
PWMDIM
4
17
N.C
16
IN
15
N.C.
UVEN
1
VCC
MAX20090/
MAX20090B
OVP
5
SGND
6
COMP
7
14
V7V
FLT
8
13
BST
ICTRL
9
12
NDRV
CS
10
11
PGND
TSSOP
TQFN
(4mm x 4mm)
Pin Description
PIN
TSSOP
TQFN
NAME
FUNCTION
Undervoltage-Lockout (UVEN) Threshold/Enable Input. UVEN is a dual-function
adjustable UVLO threshold input with an enable feature. Connect UVEN to VIN through a
resistive voltage-divider to program the UVLO threshold. Observe the absolute maximum
value for this pin.
1
17
UVEN
2
18
VCC
3
19
RT
PWM Switching-Frequency Programming. Connect a resistor (RRT) from RT to SGND to
set the internal clock frequency. fOSC (kHz) = 34200/RRT (kΩ).
5V Supply
4
20
PWMDIM
Dimming-Control Input. Connect PWMDIM to an external PWM signal for PWM dimming.
For analog voltage-controlled PWM dimming, connect PWMDIM to VCC through a
resistive voltage-divider. The dimming frequency is 200Hz under these conditions.
Connect PWMDIM to SGND to turn off the LEDs.
5
1
OVP
Overvoltage-Protection Input for the LED String. Connect a resistive divider between the
boost output, OVP, and PGND. When the voltage on OVP exceeds 1.23V, a fast-acting
comparator immediately stops PWM switching. This comparator has hysteresis of 70mV.
6
2
SGND
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Signal Ground
Maxim Integrated │ 9
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Pin Description (continued)
PIN
NAME
FUNCTION
TSSOP
TQFN
7
3
COMP
Compensation-Network Connection. For proper compensation, connect a suitable RC
network from COMP to SGND.
8
4
FLT
Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (FLT) section.
9
5
ICTRL
Analog Dimming Control Input. Connect an analog voltage from 0 to 1.2V for analog
dimming of LED current.
10
6
CS
11
7
PGND
Power Ground
12
8
NDRV
External n-Channel Gate-Driver Output
13
9
BST
Connect a minimum of 0.01μF capacitor from BST to NDRV to provide power supply for
the gate driver.
14
10
V7V
7V Low-Dropout Voltage-Regulator Output. Bypass V7V to PGND with a 1µF (min)
ceramic capacitor.
15, 17
12, 16
N.C
No Connection
Current-Sense Amplifier Positive Input for the Switching Regulator. Add a resistor from
CS to the switching-MOSFET current-sense resistor terminal for programming the slope
compensation.
16
11
IN
18
13
DIMOUT
19
14
ISENSE-
Negative LED Current-Sense Input
Positive LED Current-Sense Input. The voltage between ISENSE+ and ISENSE- is
proportionally regulated to the lesser of (ICTRL, 1.3V).
20
15
ISENSE+
—
—
EP
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Positive Power-Supply Input. Bypass IN to PGND with at least a 1µF ceramic capacitor.
External Dimming p-Channel MOSFET Gate Driver
Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective
power dissipation. Do not use as the main IC ground connection. EP must be connected
to GND.
Maxim Integrated │ 10
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Block Diagram
IN
MAX20090
5V
UVEN
BG
EN
5V REG
1.24V
V7V
7V LDO
5V V CC
BST
VCC (5V)
VICTRLC LMP
VCC UVLO
7V7 UVLO
NDRV
5V
THERMAL
SHUTDOW N
TSHDN
PGND
PWMDIM
RT
RT OSCILLATOR
SYNC TO RISING
EDGE OF PWM
RESET
DOMINANT
S
SLOPE
COMPENSATI ON
R
CS/PWM
BLANKING
CS
PWMDIM
1.0V
RAMP GENERA TION
SYNC TO RISING
EDGE OF PWMDIM
VICTRLC LMP
ICTRL
Q
MAX DUTY
CYCLE
PWM
COMP
0.418V
MI N
OUT
LPF
gM
ISENSE+
COMP
5X
ISENSE-
SYNC
0.2V
PWMDIM
ISENSE+
DIMOUT
0.2V
BUCK-BOOST
SHORT DE TECTION
(MAX20090 ONLY)
200Hz
0.32V
2.2V
OVP
ISENSE+ - 7V
S
8192 x TOSC
HICCUP TIMER
Q
FLT
R
TSHDN
SGND
1.23V
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Maxim Integrated │ 11
MAX20090/MAX20090B
Detailed Description
The MAX20090 is a single-channel HB LED driver for
automotive front-light applications such as high beam,
low beam, daytime-running lights (DRLs), turn indicators, fog lights, and other LED lights. It can take an input
voltage from 5V to 65V and drive a string of LEDs with a
maximum output voltage of 65V.
The device senses output current at the high side of the
LED string. High-side current sensing is required to protect for shorts from the output to the ground or battery
input. It is also the most flexible scheme for driving LEDs,
allowing boost, high-side buck, SEPIC mode, or buckboost-mode configurations. The PWM input provides
LED dimming ratios of up to 1000:1, and the ICTRL
input provides additional analog-dimming capability in the
device. The device also includes a fault flag (FLT) that
indicates open string, shorted string, and thermal shutdown. The device has built-in spread-spectrum modulation
for improved electromagnetic-compatibility performance.
The device can also be used in zeta and Ćuk converter
configurations, if necessary in some applications.
Functional Operation of the MAX20090
The operation of the device is best understood by
referring to the Block Diagram. The device is enabled when
the UVEN pin goes above 1.24V (typ). In addition to the
UVEN input, the 5V regulator and the 7V regulator inputs
also need to be above their respective UVLO limits, before
switching on NDRV can begin. The device is a constantfrequency, current-mode controller with a low-side nMOS
gate driver. The nMOS gate-drive voltage is enhanced to
7V by the V7V pin. The control circuitry inside the device
uses a 5V supply, but the gate driver has a 7V output. This
can be seen from the Block Diagram. When PWMDIM
goes high, switching is initiated. The RT oscillator can be
programmed from 200kHz to 2.2MHz by the resistor on
the RT pin (RRT). Additional spread-spectrum dithering is
added to the oscillator to alleviate EMI problems in the LED
driver. The RT oscillator is synchronized to the positivegoing edge of the PWM pulse. This means that the NDRV
pulse goes high at the same instant as the positive-going
pulse on PWMDIM. Synchronizing the RT oscillator to
the PWMDIM pulse also guarantees that the switchingfrequency variation over a period of a PWMDIM pulse
is the same from one PWMDIM pulse to the next. This
prevents flicker during PWM dimming when spread
spectrum is added to the RT oscillator.
Once PWMDIM goes high, the external switching MOSFET is turned on. A current flows through the
external switching MOSFET and this current is sensed
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Automotive High-Voltage,
High-Brightness LED Controller
by the voltage across the current-sense resistor from the
source of the external MOSFET to PGND. The source
of the external MOSFET is connected to the CS pin of
the device through a slope-compensation resistor (RSC)
(see the Simplified Schematic). The slope-compensation
current flows out of the CS pin into the RSC resistor.
The voltage on CS is the voltage across the currentsense resistor (RCS_FET) + slope-compensation current
x RSC. The slope compensation prevents subharmonic
oscillation when the duty cycle exceeds 50%. The
current in the external inductor increases steadily when
the external MOSFET is on. The voltage on CS is fed to
a current-limit comparator. This current-limit comparator
is used to protect the external switch from overcurrents,
and causes switching to stop for that particular cycle if the
CS voltage exceeds 0.418V. An offset of 1.0V is added
to the CS voltage, and this voltage is fed to the positive
input of a PWM comparator. The negative input of this
comparator is a control voltage from the error amplifier
that regulates the LED current. When the positive input
of the PWM comparator exceeds the control voltage
from the error amplifier, the switching is stopped for that
particular cycle and the external nMOS stays off until the
next switching cycle. When the external MOSFET is turned
off, the inductor current decays. When the next switching
cycle starts and the external MOSFET is turned on, the
inductor current starts ramping back up. Through this
repetitive action, the PWM-control algorithm establishes
a switching duty cycle to regulate current to the LED load.
When PWMDIM goes high, the external dimming MOSFET
driven by DIMOUT is also tuned on. This external dimming MOSFET is a p-channel MOSFET and is connected
on the high side. The source of this pMOS is connected
to ISENSE- and the gate is connected to DIMOUT.
The drain of this MOSFET is connected to the anode of
the external LED string. In certain applications, it is not
necessary to use this dimming MOSFET and in these
cases, the DIMOUT pin is left open. The external pMOS
is turned on when PWMDIM is high and is turned off
when PWMDIM is low. During normal operation when
PWMDIM is high, the voltage across the resistor from
ISENSE+ to ISENSE- is regulated to a programmed voltage. This programmed voltage is 0.2 x (V(ICTRL) - 0.2).
The external pMOS switch is also used for fault protection
as well. Once a fault condition is detected, the DIMOUT
pin is pulled high to turn off the pMOS switch. This
isolates the LED string from the fault condition and
prevents excessive voltage or current from damaging the
LEDs.
Maxim Integrated │ 12
MAX20090/MAX20090B
Input Voltage (IN)
The input supply pin (IN) must be locally bypassed with
a minimum of 1μF capacitance close to the pin. All the
input current drawn by the device goes through this pin.
The positive terminal of the bypass capacitor must be
placed as close as possible to this pin and the negative
terminal of the bypass capacitor must be placed as close as
possible to the PGND pin.
Undervoltage Lockout (UVLO)
The device features adjustable UVLO using the enable
input (UVEN). Connect UVEN to VIN through a resistive
divider to set the UVLO threshold. The device is enabled
when VUVEN exceeds the 1.24V (typ) threshold. UVEN
also functions as an enable/disable input to the device.
Drive UVEN low to disable the output and high to enable
the output.
VCC Regulator
The VCC supply is the low-voltage analog supply for the
device and derives power from the input voltage from IN to
PGND. Use a 1μF low-ESR ceramic capacitor from VCC
to PGND for stable operation. The VCC regulator provides
power to all the internal logic and control circuitry inside
the device.
7V Linear Regulator (V7V)
The device features a 7V low-side linear regulator (V7V).
V7V powers up the switching MOSFET driver with
sourcing capability of up to 50mA. Use a 1μF (min) lowESR ceramic capacitor from V7V to PGND for stable
operation. The V7V regulator goes below 7V if the input
voltage falls below 7V. The dropout voltage for this regulator at 50mA is 0.2V. This means that for an input voltage of
5V, the V7V voltage is 4.8V. The short-circuit current on the
V7V regulator is 100mA (typ). It is also possible to apply an
external voltage on the V7V regulator output and save its
power dissipation. The maximum externally applied voltage
on V7V should not exceed its absolute maximum rating.
BST Capacitor Node (BST)
Use the BST pin to provide a drive voltage to the low-side
switching MOSFET that is higher than VCC. An internal
diode is connected from BST to VCC. Connect a 0.01μF
(min) ceramic capacitor from this pin to the NDRV pin.
Place the capacitor as close as possible to this pin.
Dimming MOSFET Driver (DIMOUT)
The device requires an external p-channel MOSFET
for PWM dimming. For normal operation, connect the
gate of the MOSFET to the output of the dimming driver
(DIMOUT). The dimming driver can sink up to 35mA
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Automotive High-Voltage,
High-Brightness LED Controller
or source up to 77mA of peak current for fast charging
and discharging of the pMOS gate. When the PWMDIM
signal is high, this driver pulls the pMOS gate to 7.4V
below the ISENSE+ pin to completely turn on the
p-channel dimming MOSFET. The DIMOUT pin inverts
and level shifts the signal on PWMDIM to drive the gate
of the external pMOS. In some applications, the pMOS
dimming MOSFET is not used. In these cases, the
DIMOUT pin can be left open.
LED Current-Sense Inputs (ISENSE+/ISENSE-)
The differential voltage from ISENSE+ to ISENSEis fed to an internal current-sense amplifier. This
amplified signal is then connected to the negative input
of the transconductance error amplifier. The voltage-gain
factor of this amplifier is 5. The resistor is connected
between ISENSE+ and ISENSE- to program the maximum LED current. The full-scale signal is 200mV. The
ISENSE+ pin should be connected to the positive terminal
of the current-sense resistor and the ISENSE- pin should
be connected to the negative terminal of the currentsense resistor (LED string anode side).
Internal Oscillator (RT)
The internal oscillators of the MAX20090 are
programmable from 200kHz to 2.2MHz using a single
resistor at RT. Use the equation below to calculate the
switching frequency:
fOSC (kHz) = 34,200/RRT (kΩ)
where RRT is the resistor from RT to SGND.
The frequency calculated from the above formula may
not be totally accurate, and some final trimming might
be needed. The resistor values for a frequency of
200kHz is 188kΩ, 1MHz is 34.2kΩ, and 2.2MHz is 14.7kΩ.
The switching-frequency oscillator in the device is
synchronized to the leading edge of the PWM
dimming pulse on input PWMDIM. The device has built-in
frequency dithering of ±6% of the programmed frequency
to alleviate EMI problems.
Spread-Spectrum Option
The device has an internal spread-spectrum option to
optimize EMI performance. This is factory set and the
S-version of the device should be ordered. For spreadspectrum-enabled devices, the operating frequency is
varied ±6%, centered on the oscillator frequency (fOSC).
The modulation signal is a triangular wave with a period
of 190μs at 2.2MHz. Therefore, fOSC ramps down 6% and
back to 2.2MHz in 190μs and also ramps up 6% and back
to 2.2MHz in 190μs. The cycle then repeats.
Maxim Integrated │ 13
MAX20090/MAX20090B
For operations at fOSC values other than 2.2MHz, the
modulation signal scales proportionally (e.g., at 400kHz,
the 100μs modulation period increases to 190μs x
2.2MHz/400kHz = 1045μs).
n-Channel Switching-MOSFET Driver (NDRV)
The device drives an external n-channel switching
MOSFET (NDRV). NDRV swings between V7V and
PGND. NDRV can sink/source 2A of peak current, allowing the ICs to switch MOSFETs in high-power applications.
The average current demanded from the supply to drive
the external MOSFET depends on the total gate charge
(Qg) and the operating frequency of the converter (fSW).
Use the following equation to calculate the driver supply
current (INDRV) required for the switching MOSFET:
INDRV = Qg x fSW
Switching-MOSFET Current-Sense Input (CS)
CS is part of the current-mode-control loop. The switching control uses the voltage on CS, set by RCS_FET and
RSC to terminate the on-pulse width of the switching
cycle, thus achieving peak current-mode control. Internal
leading-edge blanking of 66ns is provided to prevent
premature turn-off of the switching MOSFET in each
switching cycle. Resistor RCS_FET is connected between
the source of the n-channel switching MOSFET and
PGND. During switching, a current ramp with a slope of
50μA x fSW is sourced from the CS input. This current
ramp, along with resistor RSC, programs the amount of
slope compensation.
Overvoltage Protection (OVP)
OVP sets the overvoltage-threshold limit across the
LEDs. Use a resistive divider between ISENSE+ to OVP
and SGND to set the overvoltage-threshold limit. An
internal overvoltage-protection comparator senses the
differential voltage across OVP and SGND. If the
differential voltage is greater than 1.23V, NDRV goes
low, DIMOUT goes high, and FLT asserts. When the
differential voltage drops by 70mV, NDRV is enabled if
PWMDIM is high and DIMOUT goes low. FLT deasserts
only if PWMDIM is high and V(ISENSE+ - ISENSE-) is >
20mV.
Output Short-Circuit Protection
The MAX20090/MAX20090B feature output short-circuit
protection. This feature is most useful when the LEDs are
connected to the LED driver by long cables and there is
the possibility of a short occurring when connectors are
exposed.
For the MAX20090, a short circuit is detected when the
following two conditions are met:
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Automotive High-Voltage,
High-Brightness LED Controller
● (VISENSE+ - VIN) falls below the VSHORT-VOUT
threshold, 1.55V (typ).
● The current-sense voltage across (VISENSE+ VISENSE-) exceeds the VSHORT-HIC threshold,
398mV (typ).
The VSHORT-VOUT threshold flag in MAX20090B is disabled for applications in which (VISENSE+ - VIN) is expected to be less than 1.55V (typ) during normal operation. In
this case, the VSHORT-HIC threshold is the only criteria for
detecting a short circuit.
The MAX20090/MAX20090B respond to a short circuit
by entering hiccup mode, which stops NDRV and pulls
DIMOUT high to turn off the DIM FET, disconnecting the
output of the LED driver from the shorted LEDs. The device
waits 8192 clock cycles before attempting to drive the
LEDs again.
Internal Transconductance Amplifier
The device has a built-in transconductance amplifier used
to amplify the error signal inside the feedback loop. The
typical transconductance is 1800µS. For proper operation
of this transconductance amplifier, it is necessary to add
a 500kΩ resistor from the COMP pin to ground. Without
this resistor, the performance during PWM dimming is
compromised.
Analog Dimming
The device offers an analog dimming-control input pin
(ICTRL). The voltage at ICTRL sets the LED current
level when VICTRL < 1.2V. The LED current can be
linearly adjusted from zero with the voltage on ICTRL.
For VICTRL > 1.4V, an internal reference sets the LED
current. The maximum withstand voltage of this input is
6V. The LED current is guaranteed to be at zero when the
ICTRL voltage is at or below 0.18V. The LED current can
be linearly adjusted from zero to full scale for the ICTRL
voltage in the range of 0.2V to 1.2V.
Pulsed-Dimming Input (PWMDIM)
PWMDIM functions with either analog or PWM control
signals. Once the internal pulse detector detects three
successive edges of a PWM signal with a frequency
between 60Hz and 2kHz, the device synchronizes to
the external signal and pulse-width modulates the LED
current at the external PWMDIM input frequency, with
the same duty cycle as the PWMDIM input. If an analog
control signal is applied to PWMDIM, the device
compares the DC input to an internally generated 200Hz
ramp to pulse-width-modulate the LED current (fDIM
= 200Hz). The output-current duty cycle is linearly
adjustable from 0% to 100% (0.2V < VPWMDIM < 3.2V).
Maxim Integrated │ 14
MAX20090/MAX20090B
Use the following formula to calculate the voltage
(VPWMDIM), necessary for a given output-current duty
cycle (D):
VPWMDIM = (D x 3.0) + 0.2V
where VPWMDIM is the voltage applied to PWMDIM in
volts.
Power Ground (PGND)
This pin is the power ground for the LED driver circuitry.
Place the negative terminal of the input bypass capacitor
as close as possible to the PGND pin.
Signal Ground (SGND)
This is the analog ground pin for all the LED driver
control circuitry. Connect PGND (power ground) and
SGND together at the negative terminal of the input bypass
capacitor.
Thermal Shutdown
Internal thermal-shutdown circuitry is provided to protect
the device in the event the maximum junction temperature is exceeded. The threshold for thermal shutdown is
165°C with 10°C hysteresis (both values typ). The part
returns to regulation mode once the junction temperature
goes below +155°C. This results in a cycled output during
continuous thermal-overload conditions.
Fault Indicator (FLT)
The device features an active-low, open-drain fault
indicator (FLT). FLT asserts when one of the following
conditions occur:
1)
Overvoltage or open across the LED string
2)
Short-circuit condition across the LED string
3)
Overtemperature condition
For overvoltage or open across the LED string, the FLT
indicator asserts only when an overvoltage occurs with
the PWMDIM in the high state. Once asserted, FLT stays
low and only changes state if PWMDIM is high, the overvoltage condition is removed, and the voltage across
the LED current-sense resistor is above 20mV. The FLT
signal never changes state when PWMDIM is low.
Exposed Pad
The MAX20090 package features an exposed thermal
pad on its underside that should be used as a heat sink.
This pad lowers the package’s thermal resistance by
providing a direct heat-conduction path from the die to the
PCB. Connect the exposed pad and GND to the system
ground using a large pad or ground plane, or multiple vias
to the ground plane layer.
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Automotive High-Voltage,
High-Brightness LED Controller
Applications Information
VCC Regulator
The internal 5V regulator is used to power the internal control circuitry inside the device, except for the output gate
driver. This regulator can provide a load of 10mA to external circuitry. The 5V regulator requires an external ceramic
capacitor for stable operation. A 1FF ceramic capacitor is
adequate for most applications. Place the ceramic capacitor close to the IC to minimize trace length to the internal
VCC pin and also to the IC ground. Choose a 10V rated
low-ESR, X7R ceramic capacitor for optimal performance.
7V Regulator
The 7V regulator also requires a capacitor on the output
for stable operation. Place the capacitor close to the IC to
minimize trace length to the V7V pin and to the PGND pin.
Use a 10V or higher low-ESR, X7R ceramic capacitor for
best performance. A 2.2FF ceramic capacitor should be
adequate in most applications. This capacitor is used to
provide the peak switching currents required to drive the
external MOSFET on NDRV. The maximum current that
can be delivered by the 7V regulator is 50mA. The current
from the 7V regulator is given by:
I7V = Qg x fSW
where Qg is the gate charge of the external MOSFET at
7V VGS and fSW is the switching frequency. This current
should not exceed 50mA. The 7V regulator has UVLO at
4.33V that causes the gate drive to be disabled if the input
voltage falls below the UVLO threshold.
Programming the UVLO Threshold
The UVLO threshold is set by resistors RUVEN1 and
RUVEN2 (see the Simplified Schematic). The device turns
on when the voltage across RUVEN2 exceeds 1.24V, the
UVLO threshold. Use the following equation to set the
desired UVLO threshold:
VUVEN = 1.24 x (RUVEN1 + RUVEN2)/RUVEN2
The UVEN pin can also be used as a separate enable
pin where an external logic signal can turn on and off the
device.
Programming the LED Current
Normal sensing of the LED current should be done on
the high side where the LED current-sense resistor is
connected to the anode of the LED string. The LED
current is programmed using resistor RCS_LED (see the
Simplified Schematic).
Maxim Integrated │ 15
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
The LED current can also be programmed adjusting the
voltage on ICTRL when VICTRL ≤ 1.2V (analog dimming).
The current is given by:
current (ILAVG), peak-to-peak inductor current ripple (∆IL),
and the peak inductor current (ILP) in amperes:
ILED = (VICTRL - 0.2)/(5 x RCS_LED)
Allowing the peak-to-peak inductor ripple to be ∆IL, the
peak inductor current is given by:
Programming the Switching Frequency
The internal oscillator of the device is programmable from
200kHz to 2.2MHz using a single resistor at RT.
Use the following equation to calculate the value of the
resistor (RRT):
RRT(kΩ) = 34,200/fOSC(kHz)
where fOSC(kHz) is the desired switching frequency in kHz.
The frequency calculated from the above formula may
not be totally accurate, and some final trimming might be
needed. The resistor values for a frequency of 200kHz is
88kΩ, 1MHz is 34.2kΩ, and 2.2MHz is 14.7kΩ.
Additional ±6% spread spectrum is added internally to the
oscillator to improve EMI performance.
Setting the Overvoltage Threshold
The overvoltage threshold is set by resistors ROVP1 and
ROVP2 (see the Simplified Schematic). The overvoltage
circuit in the device is activated when the voltage on OVP
with respect to GND exceeds 1.23V. Use the following
equation to set the desired overvoltage threshold:
VOVP = 1.23 x (ROVP1 + ROVP2)/ROVP2
Inductor Selection
Boost Configuration
In the boost converter, the average inductor current varies
with the line voltage. The maximum average current occurs
at the lowest line voltage.
For the boost converter, the average inductor current is
equal to the input current. Calculate maximum duty cycle
using the equation below:
DMAX = (VLED - VD - VINMIN)/(VLED + VD - VFET)
where VLED is the forward voltage of the LED string in
volts, VD is the forward drop of rectifier diode D1 in volts
(approximately 0.6V), VINMIN is the minimum input supply
voltage in volts, and VFET is the average drain-to source
voltage of MOSFET N1 in volts when it is on. Use an
approximate value of 0.2V initially to calculate DMAX. A
more accurate value of the maximum duty cycle can be
calculated once the power MOSFET is selected based
on the maximum inductor current. Use the following
equations to calculate the maximum average inductor
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ILAVG = ILED/(1 - DMAX)
ILP = ILAVG + 0.5 x ∆IL
The inductance value (L) of inductor L1 in henries (H) is
calculated as:
L = (VINMIN - VFET) x DMAX/(fSW x ∆IL)
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value. The current rating of the inductor should
be higher than ILP at the operating temperature.
Buck-Boost Configuration
In the buck-boost LED driver, the average inductor
current is equal to the input current plus the LED current.
Calculate the maximum duty cycle using the following
equation:
DMAX = (VLED + VD)/(VLED + VD + VINMIN - VFET)
where VLED is the forward voltage of the LED string in
volts, VD is the forward drop of rectifier diode D1 (~ 0.6V)
in volts, VINMIN is the minimum input supply voltage in
volts, and VFET is the average drain-to-source voltage of
MOSFET N1 in volts when it is on. Use an approximate
value of 0.2V initially to calculate DMAX. A more accurate
value of the maximum duty cycle can be calculated once
the power MOSFET is selected based on the maximum
inductor current.
Use the equations below to calculate the maximum average inductor current (ILAVG), peak-to-peak inductor current
ripple (∆IL), and the peak inductor current (ILP) in amperes:
ILAVG = ILED/(1 - DMAX)
Allowing the peak-to-peak inductor ripple to be ∆IL:
ILP = ILAVG + 0.5 x ∆IL
where ILP is the peak inductor current.
The inductance value (L) of inductor L1 in henries is
calculated as:
L = (VINMIN - VFET) x DMAX/(fSW x ∆IL)
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value.
Maxim Integrated │ 16
MAX20090/MAX20090B
High-Side Buck Configuration
In the high-side buck LED driver, the average inductor
current is the same as the LED current. The peak inductor
current occurs at the maximum input line voltage where the
duty cycle is at the minimum:
DMIN = (VLED + VD)/(VINMAX - VFET)
where VLED is the forward voltage of the LED string
in volts, VD is the forward drop of rectifier diode
D1 (~ 0.6V) in volts, VINMAX is the maximum input
supply voltage in volts, and VFET is the average drainto-source voltage of MOSFET N1 in volts when it
is on. Use an approximate value of 0.2V initially to
calculate DMIN. The maximum peak-to-peak inductor ripple
(∆IL) occurs at the maximum input line. The peak inductor
current is given by:
ILP = ILED + 0.5 x ∆IL
The inductance value (L) of inductor L1 in henries is
calculated as:
L = (VINMAX - VFET - VLED) x DMIN/(fSW x ∆IL)
where fSW is the switching frequency in hertz, VINMAX
and VFET are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than the
calculated value.
SEPIC, Zeta, and Ćuk Configurations
In the SEPIC, zeta, and Ćuk converters, there are
separate inductors for L1 and L2. Neglecting the drops in
the switching MOSFET and diode, the maximum duty cycle
(DMAX) occurs at low line and is given by:
DMAX = VLED/(VINMIN + VLED)
where VLED is the LED string voltage and VINMIN is the
minimum input voltage. If the desired maximum input
current ripple is ∆ILIN, then the inductor value of L1 is given
by:
L1 = VINMIN x DMAX/(∆ILIN x fSW)
The peak inductor current in L1 is ILINP and is given by:
ILINP = (ILED x DMAX/(1 - DMAX)) + 0.5 x ∆ILIN
To account for current transients, the peak saturation
rating of the inductor should be 1.2 times the calculated
value above. The average output current in inductor L2 is
the same as the LED current. The desired maximum peakto-peak output current ripple is ∆ILOUT. The value of the
inductor L2 is given by:
L2 = VINMIN x DMAX/(∆ILOUT x fSW)
The peak inductor current in L2 is ILOUTP and is given by:
ILOUTP = ILED + 0.5 x ∆ILOUT
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Automotive High-Voltage,
High-Brightness LED Controller
Selecting Slope Compensation and
MOSFET Current-Sense Resistor
Slope compensation should be added to converters
with peak current-mode-control operating in continuousconduction mode with more than 50% duty cycle to avoid
current-loop instability and subharmonic oscillations. The
minimum amount of slope compensation required for
stability is:
min slope = 0.5 x (inductor current downslope - inductor
current upslope) x RCS_FET
In the MAX20090, the slope-compensating ramp is added
to the current-sense signal before it is fed to the PWM
comparator. Connect a resistor (RSC) from CS to the
switch current-sense resistor terminal for programming the
amount of slope compensation.
The device generates a current ramp with a slope of 50μA/
tOSC for slope compensation. The current-ramp signal is
forced into an external resistor (RSC) connected between
CS and the source of the external MOSFET, thereby adding
a programmable slope-compensating voltage (VSCOMP) at
the current-sense input CS. Therefore:
dv(VSCOMP)/dt = (RSC x 50μA)/tOSC in V/s
The minimum required value of the slope-compensation
voltage that needs to be added to the current signal at peak
of the current signal at minimum line voltage is:
For boost LED driver:
VSLOPE (MIN) = (DMAX x (VLED - 2VINMIN) x RCS_FET)/
(2 x LMIN x fSW) volts
For buck-boost LED driver:
VSLOPE (MIN) = (DMAX x (VLED - VINMIN) x RCS_FET)/
(2 x LMIN x fSW) volts
For high-side buck LED driver:
VSLOPE (MIN) = (DMAX x (2VLED - VINMIN) x RCS_FET)/
(2 x LMIN x fSW) volts
For SEPIC LED driver:
VSLOPE (MIN) = (DMAX x (VLED - VINMIN) x RCS_FET)/
(2 x LMIN x fSW) volts
where LMIN = SQRT (L1MIN x L2MIN)
where L1 and L2 are the two inductors in the SEPIC
configuration, fSW is the switching frequency, DMAX is the
maximum duty cycle that occurs at minimum input voltage
VINMIN, and LMIN is the minimum value of the selected
inductor. For adequate margin, use a slope compensation
that is 1.5 times the minimum required value of the slope
compensation.
Maxim Integrated │ 17
MAX20090/MAX20090B
The minimum value of the peak current-limit comparator is
0.388V. The current-sense resistor value is given by:
RCS_FET = (0.388-slope compensation voltage)/ILP
where ILP is the peak inductor current that occurs at low
line in the boost, SEPIC, and buck-boost configuration.
For boost configuration:
R CS_FET =
0.388
VLED - 2VINMIN
ILP + 0.75D MAX
L MIN f SW
For buck-boost configuration:
0.388
V
- VINMIN
ILP + 0.75D MAX LED
L MIN f SW
For SEPIC configuration:
R CS_FET =
R CS_FET =
0.388
ILP1 + ILP 2 + 0.75D MAX VLED - VINMIN
f SW L 1MIN L 2MIN
Input Capacitor
The input-filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of highfrequency current conducted to the input supply.
The ESR, ESL, and bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter. For the boost
configuration, the input current is the same as the
inductor current. For buck-boost configuration, the input
current is the inductor current minus the LED current.
However, for both configurations, the ripple current that
the input filter capacitor has to supply is the same as the
inductor ripple current with the condition that the output
filter capacitor should be connected to ground for buckboost configuration. Neglecting the effect of LED current
ripple, the calculation of the input capacitor for boost, as
well as buck-boost configurations is the same. Neglecting
the effect of the ESL, ESR, and bulk capacitance at the
input contributes to the input-voltage ripple. For simplicity,
assume that the contribution from the ESR and the bulk
capacitance is equal. This allows 50% of the ripple for the
bulk capacitance. The capacitance is given by:
Automotive High-Voltage,
High-Brightness LED Controller
Use X7R ceramic capacitors for optimal performance.
The selected capacitor should have the minimum required
capacitance at the operating voltage.
In the buck mode, the input capacitor has large pulsed
currents due to the current flowing in the freewheeling diode when the switching MOSFET is off. It is very
important to consider the ripple-current rating of the input
capacitor in this application.
Output Capacitor Selection
The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
bulk capacitance of the output capacitor contribute to the
output ripple. In most applications, the output ESR and
ESL effects can be dramatically reduced by using lowESR ceramic capacitors. To reduce the ESL and ESR
effects, connect multiple ceramic capacitors in parallel to
achieve the required bulk capacitance. To minimize audible
noise generated by the ceramic capacitors during PWM
dimming, it may be necessary to minimize the number
of ceramic capacitors on the output. In these cases, an
additional electrolytic or tantalum capacitor provides most
of the bulk capacitance.
Boost and Buck-Boost Configurations
The calculation of the output capacitance is the same
for both boost and buck-boost configurations. The output
ripple is caused by the ESR and bulk capacitance of the
output capacitor if the ESL effect is considered negligible.
For simplicity, assume that the contributions from ESR and
bulk capacitance are equal, allowing 50% of the ripple for
the bulk capacitance. The capacitance is given by:
C OUT ≥
ILED × 2 × D MAX
VOUTRIPPLE × f SW
where ILED is in amperes, COUT is in farads, fSW is in
hertz, and VOUTRIPPLE is in volts. The remaining 50%
of allowable ripple is for the ESR of the output capacitor.
Based on this, the ESR of the output capacitor is given by:
ESR COUT <
VOUTRIPPLE (Ω )
(IL P × 2)
where ILP is the peak inductor current in amperes.
CIN > ∆IL/(4 x ∆VIN x fSW)
where ∆IL is in amperes, CIN is in farads, fSW is in hertz,
and ∆VIN is in volts. The remaining 50% of allowable ripple
is for the ESR of the output capacitor.
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Maxim Integrated │ 18
MAX20090/MAX20090B
Rectifier Diode Selection
Use a Schottky diode as the rectifier (D1) for fast switching
and to reduce power dissipation. The selected Schottky
diode must have a voltage rating 20% above the maximum
converter output voltage. The maximum converter output
voltage is VLED in the boost configuration and VLED +
VINMAX in the buck-boost configuration.
The current rating of the diode should be greater than ID in
the following equation:
Automotive High-Voltage,
High-Brightness LED Controller
The worst-case RHP zero frequency (fZRHP) is calculated
as follows:
Boost configuration:
f ZRHP =
Buck-boost configuration:
f ZRHP =
ID = ILAVG (1 - DMAX)1.5
where ILAVG is the average inductor current.
Switching MOSFET Selection
The switching MOSFET (N1) should have a voltage
rating sufficient to withstand the maximum output voltage together with the diode drop of rectifier diode D1,
and any possible overshoot due to ringing caused by
parasitic inductances and capacitances. Use a MOSFET
with a drain-to-source voltage rating higher than that
calculated by the following equations:
Boost configuration:
VDS = (VLED + VD) x 1.2
Buck-boost configuration:
VDS = (VLED + VINMAX + VD) x 1.2
where VLED is the LED string voltage, VINMAX is the
maximum input voltage, and VD is the forward drop of the
rectifier diode. The factor 1.2 provides 20% safety margin.
Dimming MOSFET Selection
Select a dimming MOSFET (P1) with continuous current
rating at the operating temperature higher than the LED
current by 30%. The drain-to-source voltage rating of the
dimming MOSFET must be higher than VLED by 20%.
Feedback Compensation
The LED current-control loop comprising the switching converter, LED current amplifier, and the error
amplifier should be compensated for stable control of the
LED current. The switching converter small-signal transfer
function has a right half-plane (RHP) zero for both boost
and buck-boost configurations, as the inductor current is in
continuous-conduction mode. The RHP zero adds a 20dB/
decade gain together with a 90° phase lag, which is difficult
to compensate. The easiest way to avoid this zero is to roll
off the loop gain to 0dB at a frequency less than 1/5 of the
RHP zero frequency with a -20dB/decade slope.
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VLED × (1 - D MAX ) 2
2π × L × ILED
VLED × (1 − D MAX ) 2
2π × L × ILED
where fZRHP is in hertz, VLED is in volts, L is the
inductance value of L1 in henries (H), and ILED is in
amperes.
The switching converter small-signal transfer function
also has an output pole for both boost and buck-boost
configurations. The effective output impedance that
determines the output pole frequency together with the
output filter capacitance is calculated as:
Boost configuration:
R OUT =
(R LED + R CS_LED ) × VLED
(R LED + R CS_LED ) × ILED + VLED
Buck-boost configuration:
R OUT =
(R LED + R CS_LED ) × VLED
(R LED + R CS_LED ) × ILED × D MAX + VLED
where RLED is the dynamic impedance of the LED string
at the operating current in ohms, RCS_LED is the LED
current-sense resistor in ohms, VLED is in volts, and ILED
is in amperes.
The output pole frequency for both boost and buck-boost
configurations is calculated as follows:
fP =
1
2π × C OUT × R OUT
where fP is in hertz, COUT is the output filter capacitance
in farads, and ROUT is the effective output impedance in
ohms calculated above.
The feedback-loop compensation is done by connecting a resistor (RCOMP) and capacitor (CCOMP) in series
from COMP to GND. RCOMP is chosen to set the highfrequency integrator gain for fast transient response, while
CCOMP is chosen to set the integrator zero to maintain
Maxim Integrated │ 19
MAX20090/MAX20090B
loop stability. For optimum performance, choose the
components using the following equations:
f=
C 0.2 × f ZRHP
R COMP =
Automotive High-Voltage,
High-Brightness LED Controller
3)
a) The anode of D1 must be connected very close
to the drain of MOSFET N1.
2 × f ZRHP × R CS_FET
f C × (1 - D MAX ) × R CS_LED × 5 × G M
b) The cathode of D1 must be connected very close
to COUT.
The value of CCOMP can be calculated as:
C COMP =
25
π × f ZRHP × R COMP
PCB Layout
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/
dt surfaces. For example, traces that carry the drain
current often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dv/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOSFET
power dissipation, or shield it. Keep all PCB traces
carrying switching currents as short as possible to minimize current loops. Use ground planes for best results.
Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation. Keep switching loops short:
c) COUT and current-sense resistor R4 must be
connected directly to the ground plane.
4)
Connect PGND
configuration.
and
SGND
to
a
star-point
5)
Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick-copper PCBs (2oz vs. 1oz) to enhance full-load
efficiency.
6)
Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer
for the PGND and SGND plane as an EMI shield to
keep radiated noise away from the device, feedback
dividers, and analog bypass capacitors.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer board
whenever possible for better noise immunity and power
dissipation. Follow these guidelines for good PCB layout:
1)
Use a large contiguous copper plane under the IC
package. Ensure that all heat-dissipating components
have adequate cooling.
2)
Isolate the power components and high-current path
from the sensitive analog circuitry.
www.maximintegrated.com
Maxim Integrated │ 20
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Operating Circuit
B3100
10uH
INPUT
L1
BST
IN
RUVEN1
40.2kΩ
CIN
4.7μF
NDRV
CS
OVP
UVEN
PWMDIM
PWMDIM
ROVP1
474kΩ
2.49kΩ
RCS_LED
0.2Ω
100Ω
MAX20090
86.6kΩ
1.0μF
N1
IRLR3110
RSC
ISENSE-
RT
C2
COUT
4x4.7μF
ISENSE+
RRT
RUVEN2
12.4kΩ
D1
CBST
0.1μF
DIMOUT
P1
FDC3535
VCC
LED+
10kΩ
V7V
FLT
C1
RCS_FET
COMP
VCC
0.025Ω
ROVP2
10kΩ
RL
1.0μF
R1
RCOMP
18Ω
499kΩ
ICTRL
R2
www.maximintegrated.com
SGND
PGND
EP
CCOMP
1.0μF
LED-
Maxim Integrated │ 21
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Application Circuits
Buck-Boost LED Driver Using the MAX20090
L1
INPUT
D1
BST
IN
COUT
NDRV
CIN
N1
ROVP1
RCS_LED
CS
RUVEN1
OVP
UVEN
PWMDIM
RSC
ISENSE+
PWMDIM
ISENSE-
RRT
100Ω
RT
RUVEN2
CBST
MAX20090
DIMOUT
C2
P1
FLT
V7V
C1
RCS_FET
COMP
VCC
R1
ROVP2
RCOMP
ICTRL
R2
SGND
PGND
EP
CCOMP
INPUT
www.maximintegrated.com
Maxim Integrated │ 22
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Application Circuits (continued)
Boost LED Driver Using the MAX20090
L1
INPUT
D1
BST
IN
CBST
COUT
NDRV
CIN
N1
ROVP1
CS
RUVEN1
OVP
UVEN
PWMDIM
RUVEN2
RCS_LED
ISENSE+
PWMDIM
RRT
RSC
ISENSE100Ω
RT
MAX20090
DIMOUT
C2
P1
FLT
V7V
C1
RRT
RCS_FET
COMP
VCC
R1
ROVP2
RCOMP
ICTRL
R2
www.maximintegrated.com
SGND
PGND
EP
CCOMP
Maxim Integrated │ 23
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Application Circuits (continued)
High-Side Buck LED Driver Using the MAX20090
RCS_LED
COUT
ROVP1
D1
INPUT
L1
BST
IN
NDRV
CIN
N2
CBST
N1
CS
RUVEN1
OVP
UVEN
PWMDIM
ISENSE+
PWMDIM
RRT
RUVEN2
RSC
ISENSE100Ω
RT
MAX20090B
DIMOUT
C2
FLT
V7V
C1
RCS_FET
COMP
VCC
R1
ROVP2
RCOMP
ICTRL
R2
www.maximintegrated.com
SGND
PGND
EP
CCOMP
Maxim Integrated │ 24
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Application Circuits (continued)
SEPIC LED Driver Using the MAX20090
BST
IN
L2
CBST
NDRV
CIN
D1
CS
L1
INPUT
N1
COUT
ROVP1
CS
RUVEN1
OVP
UVEN
PWMDIM
RUVEN2
RCS_LED
ISENSE+
ISENSE-
PWMDIM
RRT
RSC
RT
MAX20090B
DIMOUT
C2
P1
FLT
V7V
C1
COMP
VCC
R1
RCS_FET
ROVP2
RCOMP
ICTRL
R2
www.maximintegrated.com
SGND
PGND
EP
CCOMP
Maxim Integrated │ 25
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Application Circuits (continued)
Zeta Converter LED Driver using the MAX20090
ROVP1
RCS_LED
D1
COUT
L1
C6
INPUT
C1
C2
NDRV
CS
RUVEN1
R5
N2
N1
OVP
UVEN
ISENSE+
PWMDIM
ISENSE100Ω
R5
RT
RUVEN2
C5
0.1μF
BST
IN
L2
MAX20090B
DIMOUT
C3
FLT
V7V
C4
RCS_FET
COMP
VCC
ROVP2
R1
RCOMP
ICTRL
R2
www.maximintegrated.com
SGND
PGND
EP
CCOMP
Maxim Integrated │ 26
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Typical Application Circuits (continued)
Ćuk Converter LED Driver using the MAX20090
CS
L1
INPUT
L2
D1
BST
IN
NDRV
CIN
VCC
CBST
N1
CS
RUVEN1
OVP
UVEN
PWMDIM
ISENSE+
RUVEN2
ROVP1
100Ω
RT
MAX20090B
DIMOUT
C2
FLT
V7V
VCC
C1
ROVP2
ISENSE-
PWMDIM
RRT
RSC
COMP
VCC
RCS_FET
COUT
R1
RCOMP
ICTRL
R2
SGND
PGND
EP
CCOMP
RCS_LED
www.maximintegrated.com
Maxim Integrated │ 27
MAX20090/MAX20090B
Ordering Information
PART
Chip Information
PIN-PACKAGE
MAX20090ATP/V+
20 TQFN-EP*
MAX20090ATP/VY+
20 TQFN-EP (SW)*
MAX20090AUP/V+
20 TSSOP-EP*
MAX20090BATP/V+
20 TQFN-EP*
MAX20090BATP/VY+
20 TQFN-EP (SW)*
MAX20090BAUP/V+
20 TSSOP-EP*
Note: All parts operate over the -40°C to +125°C automotive
temperature range.
/V denotes an automotive-qualified part.
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
(SW) = Side wettable.
www.maximintegrated.com
Automotive High-Voltage,
High-Brightness LED Controller
PROCESS: BiCMOS
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maximintegrated.com/packages. Note
that a “+”, “#”, or “-” in the package code indicates RoHS status
only. Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
PACKAGE TYPE
20 TQFN-EP
PACKAGE
CODE
OUTLINE
NO.
T2044+4C 21-100172
20 TQFN-EP (SW) T2044Y+4C 21-100068
20 TSSOP-EP
U20E+3C
21-100132
LAND
PATTERN NO.
90-0409
90-0409
90-100049
Maxim Integrated │ 28
MAX20090/MAX20090B
Automotive High-Voltage,
High-Brightness LED Controller
Revision History
REVISION
NUMBER
REVISION
DATE
PAGES
CHANGED
0
2/17
Initial release
1
6/17
Changed data sheet title and deleted tape-and-reel variants from Ordering
Information
2
7/17
Removed future product designation in Ordering Information from the
MAX20090ATP/VY+
28
3
4/18
Removed future product designation in Ordering Information from the
MAX20090AUP/V+
28
4
1/19
Added MAX20090B to data sheet title, updated Simplified Typical Operating
Circuit, Simplified Schematic, Electrical Characteristics, Typical Operating
Characteristics, Pin Configurations, Block Diagram, Detailed Description,
Applications Information, Typical Operating Circuits, and Ordering Information
5
5/19
Added MAX20090BATP/V+ to Ordering Information
DESCRIPTION
—
1―29
1―29
28
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
© 2017 Maxim Integrated Products, Inc. │ 29
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