19-0793; Rev 5; 5/12
KIT
ATION
EVALU
E
L
B
A
IL
AVA
TFT-LCD DC-DC Converter with
Operational Amplifiers
Features
The MAX8795A includes a high-performance step-up
regulator, two linear-regulator controllers, and five highcurrent operational amplifiers for active-matrix, thin-film
transistor (TFT), liquid-crystal displays (LCDs). Also
included is a logic-controlled, high-voltage switch with
adjustable delay.
The step-up DC-DC converter provides the regulated
supply voltage for the panel source driver ICs. The converter is a high-frequency (1.2MHz) current-mode regulator with an integrated 20V n-channel MOSFET that
allows the use of ultra-small inductors and ceramic
capacitors. It provides fast transient response to pulsed
loads while achieving efficiencies over 85%.
The gate-on and gate-off linear-regulator controllers
provide regulated TFT gate-on and gate-off supplies
using external charge pumps attached to the switching
node. The MAX8795A includes five high-performance
operational amplifiers. These amplifiers are designed to
drive the LCD backplane (VCOM) and/or the gammacorrection divider string. The device features high output current (±130mA), fast slew rate (45V/µs), wide
bandwidth (20MHz), and rail-to-rail inputs and outputs.
The MAX8795A is available in a lead-free, 32-pin, thin
QFN package with a maximum thickness of 0.8mm for
ultra-thin LCD panels, as well as in a 32-pin LQFP
package with 0.8mm pin pitch.
o 2.5V to 5.5V Input Supply Range
o 1.2MHz Current-Mode Step-Up Regulator
Fast Transient Response to Pulsed Load
High-Accuracy Output Voltage (1%)
Built-In 20V, 3A, 0.16Ω n-Channel MOSFET
High Efficiency (85%)
o Linear-Regulator Controllers for VGON and VGOFF
o High-Performance Operational Amplifiers
±130mA Output Short-Circuit Current
45V/µs Slew Rate
20MHz, -3dB Bandwidth
Rail-to-Rail Inputs/Outputs
o Logic-Controlled, High-Voltage Switch with
Adjustable Delay
o Timer-Delay Fault Latch for All Regulator Outputs
o Thermal-Overload Protection
o 0.6mA Quiescent Current
Applications
Minimal Operating Circuit
VCN
VCP
VIN
VMAIN
LX
IN
FB
STEP-UP
CONTROLLER
PGND
COMP
AGND
Notebook Computer Displays
LCD Monitor Panels
VCP
MAX8795A
DRVP
GATE-ON
CONTROLLER
FBP
VGON
Automotive Displays
SRC
DEL
COM
Ordering Information
PART
MAX8795AETJ+
TEMP RANGE
PIN-PACKAGE
-40°C to +85°C
32 Thin QFN
SWITCH
CONTROL
CTL
VCN
DRN
DRVN
GATE-OFF
CONTROLLER
SUP
VGOFF
NEG1
MAX8795AGCJ+
-40°C to +105°C
32 LQFP
FBN
OUT1
MAX8795AGCJ/V+
-40°C to +105°C
32 LQFP
MAX8795AGTJ+
-40°C to +105°C
32 TQFN
MAX8795AGTJ/V+
-40°C to +105°C
32 TQFN
+Denotes a lead(Pb)-free/RoHS-compliant package.
/V denotes an automotive-qualified part.
OP1
POS1
REF
OUT2
REF
NEG4
NEG2
OP2
OP4
OUT4
POS4
POS2
NEG5
OUT3
OP5
OP3
OUT5
POS5
POS3
EP
BGND
Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
1
MAX8795A
General Description
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
ABSOLUTE MAXIMUM RATINGS
IN, CTL to AGND ...................................................-0.3V to +7.5V
COMP, FB, FBP, FBN, DEL, REF to AGND ....-0.3V to (VIN + 0.3V)
PGND, BGND to AGND ......................................................±0.3V
LX to PGND ............................................................-0.3V to +20V
SUP to AGND .........................................................-0.3V to +20V
DRVP to AGND.......................................................-0.3V to +36V
POS_, NEG_, OUT_ to AGND ...................-0.3V to (VSUP + 0.3V)
DRVN to AGND ...................................(VIN - 30V) to (VIN + 0.3V)
SRC to AGND .........................................................-0.3V to +40V
COM, DRN to AGND ................................-0.3V to (VSRC + 0.3V)
DRN to COM............................................................-30V to +30V
POS_ to NEG_ RMS Current ...................................5mA (Note 1)
Note 1: See Figure 2 for the op amp clamp structure.
OUT_ Maximum Continuous Output Current....................±75mA
LX Switch Maximum Continuous RMS Current .....................1.6A
Continuous Power Dissipation (TA = +70°C)
32-Pin Thin QFN (derate 34.5mW/°C above +70°C) 2758mW
32-Pin LQFP (derate 48.4mW/°C above +70°C)....1652.9mW
Operating Temperature Range, E Grade ............-40°C to +85°C
Operating Temperature Range, G Grade .........-40°C to +105°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow) .......................................+260°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = 0°C to +85°C. Typical values are at TA = +25°C,
unless otherwise noted.)
PARAMETER
IN Supply Range
IN Undervoltage-Lockout
Threshold
SYMBOL
VIN
VUVLO
CONDITIONS
MIN
(Note 2)
2.5
VIN rising, typical hysteresis = 50mV
2.05
VFB = VFBP = 1.3V, VFBN = 0V,
LX not switching
IN Quiescent Current
IIN
MAX
UNITS
6.0
V
2.25
2.45
V
0.6
1.0
2
3
mA
VFB = 1.2V, VFBP = 1.4V, VFBN = 0V,
LX switching
Duration-to-Trigger Fault
Condition
FB or FBP below threshold or FBN above
threshold
REF Output Voltage
No external load
REF Load Regulation
0 < ILOAD < 50µA
REF Sink Current
In regulation
REF Undervoltage Lockout
Threshold
Rising edge; typical hysteresis = 160mV
200
ms
TA = +25°C to +85°C
1.238
1.250
1.262
TA = 0°C to +85°C
1.232
1.250
1.266
10
10
+160
Hysteresis
V
mV
µA
1.15
Temperature rising
Thermal Shutdown
TYP
V
°C
15
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
VIN
18
V
Operating Frequency
fOSC
1000
1200
1400
kHz
86
90
93
%
Oscillator Maximum Duty Cycle
FB Regulation Voltage
FB Fault Trip Level
VFB
No load
TA = +25°C to +85°C
1.221
1.233
1.245
TA = 0°C to +85°C
1.212
1.233
1.248
1.10
1.14
1.17
V
VFB falling
V
FB Load Regulation
0 < IMAIN < full load, transient only
-1
FB Line Regulation
VIN = 2.5V to 6V
0.1
±0.4
%/ V
FB Input Bias Current
VFB = 1.233V
+100
+200
nA
2
_______________________________________________________________________________________
%
TFT-LCD DC-DC Converter with
Operational Amplifiers
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = 0°C to +85°C. Typical values are at TA = +25°C,
unless otherwise noted.)
PARAMETER
SYMBOL
FB Voltage Gain
LX On-Resistance
CONDITIONS
∆ICOMP = ±2.5µA
FB Transconductance
RLX(ON)
TYP
MAX
UNITS
75
160
280
µS
260
mΩ
µA
From FB to COMP
700
ILX = 200mA
160
LX Leakage Current
ILX
VLX = 19V
LX Current Limit
ILIM
VFB = 1.2V, duty cycle = 75%
Current-Sense Transresistance
Soft-Start Period
MIN
V/ V
10
20
2.5
3.0
3.5
A
0.1
0.2
0.3
V/A
tSS
Soft-Start Step Size
14
ms
VREF /
128
V
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
6.0
SUP Overvoltage Fault Threshold
18.0
ISUP
Buffer configuration, VPOS_ = VSUP / 2,
no load
Input Offset Voltage
VOS
(VNEG_, VPOS_, VOUT_) ≅ VSUP / 2
Input Bias Current
IBIAS
(VNEG_ , VPOS_, VOUT_) ≅ VSUP / 2
Input Common-Mode Voltage
Range
VCM
SUP Supply Current
Common-Mode Rejection Ratio
CMRR
-50
0 ≤ (VNEG_, VPOS_) ≤ VSUP
Output Voltage Swing, High
VOH
IOUT_ = 5mA
Output Voltage Swing, Low
45
19
19.9
V
3.5
5.0
mA
0
12
mV
0
+50
nA
VSUP
V
80
dB
125
dB
VSUP 100
VSUP 50
mV
VOL
IOUT_ = -5mA
130
50
Short-Circuit Current
To VSUP / 2, source or sink
75
Power-Supply Rejection Ratio
DC, 6V ≤ VSUP ≤ 18V,
(VNEG_, VPOS_) ≅ VSUP / 2
60
Slew Rate
-3dB Bandwidth
V
0
Open-Loop Gain
PSRR
18.0
RL = 10kΩ, CL = 10pF, buffer configuration
100
mV
mA
dB
45
V/µs
20
MHz
GATE-ON LINEAR-REGULATOR CONTROLLER
FBP Regulation Voltage
VFBP
FBP Fault Trip Level
FBP Input Bias Current
IFBP
FBP Effective Load-Regulation
Error (Transconductance)
Soft-Start Step Size
1.250
1.269
VFBP falling
0.96
1.00
1.04
V
VFBP = 1.25V
-50
+50
nA
-0.7
-1.5
%
±1
±10
mV
IDRVP = 100µA, 2.5V < VIN < 6V
IDRVP
DRVP Off-Leakage Current
Soft-Start Period
1.231
VDRVP = 10V, IDRVP = 50µA to 1mA
FBP Line (IN) Regulation Error
DRVP Sink Current
IDRVN = 100µA
VFBP = 1.1V, VDRVP = 10V
VFBP = 1.4V, VDRVP = 34V
tSS
1
5
0.01
V
mA
10
µA
14
ms
VREF /
128
V
_______________________________________________________________________________________
3
MAX8795A
ELECTRICAL CHARACTERISTICS (continued)
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = 0°C to +85°C. Typical values are at TA = +25°C,
unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
GATE-OFF LINEAR-REGULATOR CONTROLLER
FBN Regulation Voltage
VFBN
FBN Fault Trip Level
FBN Input Bias Current
IFBN
FBN Effective Load-Regulation
Error (Transconductance)
1
1.015
V
370
420
470
mV
VFBN = 0.25V
-50
+50
nA
11
25
mV
±0.7
±5
mV
IDRVN = 0.1mA, 2.5V < VIN < 6V
IDRVN
DRVN Off-Leakage Current
Soft-Start Period
0.984
VFBN rising
VDRVN = -10V, IDRVN = 50µA to 1mA
FBN Line (IN) Regulation Error
DRVN Source Current
IDRVN = 100µA, VREF - VFBN
VFBN = 300mV, VDRVN = -10V
1
VFBN = 0V, VDRVN = -25V
5
-0.01
tSS
Soft-Start Step Size
mA
-10
µA
14
ms
(VREF VFBN) /
128
V
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1V
VTH(DEL)
4
5
6
µA
1.19
1.25
1.31
V
DEL Discharge Switch
On-Resistance
During UVLO, VIN = 2.0V
CTL Input Low Voltage
VIN = 2.5V to 5.5V
CTL Input High Voltage
VIN = 2.5V to 5.5V
2
CTL Input Leakage Current
CTL = AGND or IN
-1
CTL-to-SRC Propagation Delay
0.6
+1
100
µA
ns
36
ISRC
V
V
SRC Input Voltage Range
SRC Input Current
Ω
20
VDEL = 1.5V, CTL = IN
200
300
VDEL = 1.5V, CTL = AGND
115
200
V
µA
SRC-to-COM Switch On-Resistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
5
10
Ω
DRN-to-COM Switch
On-Resistance
RDRN(ON)
VDEL = 1.5V, CTL = AGND
30
60
Ω
4
_______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = -40°C to +85°C, unless otherwise noted.) (Note 3)
PARAMETER
IN Supply Range
SYMBOL
VIN
IN Undervoltage-Lockout
Threshold
IN Quiescent Current
VUVLO
IIN
REF Output Voltage
REF Undervoltage-Lockout
Threshold
MAIN STEP-UP REGULATOR
CONDITIONS
MIN
MAX
UNITS
(Note 2)
2.5
6.0
V
VIN rising, typical hysteresis = 150mV
2.05
2.45
V
VFB = VFBP = 1.3V, VFBN = 0V,
LX not switching
1.0
mA
VFB = 1.2V, VFBP = 1.4V, VFBN = 0V,
LX switching
No external load
3
1.218
1.277
V
1.15
V
18
V
900
1400
kHz
1.198
1.260
V
±0.4
%/ V
Rising edge; typical hysteresis = 160mV
Output Voltage Range
VMAIN
Operating Frequency
fOSC
FB Regulation Voltage
VFB
FB Line Regulation
VIN
No load
VIN = 2.5V to 6V
∆ICOMP = ±2.5µA
FB Transconductance
LX On-Resistance
RLX(ON)
LX Current Limit
ILIM
75
280
µS
260
mΩ
2.5
3.5
A
6
18
V
18.0
ILX = 200mA
VFB = 1.2V, duty cycle = 75%
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
SUP Overvoltage Fault Threshold
SUP Supply Current
ISUP
Buffer configuration, VPOS_ = VSUP / 2,
no load
Input Offset Voltage
VOS
(VNEG_, VPOS_, IOUT_) = ≅ VSUP / 2
Input Common-Mode Voltage
Range
VCM
Output Voltage Swing, High
VOH
IOUT_ = 5mA
Output Voltage Swing Low
VOL
IOUT_ = -5mA
Short-Circuit Current
0
To VSUP / 2
19.9
V
5
mA
12
mV
VSUP
V
VSUP 100
mV
100
Source
75
Sink
75
mA
GATE-ON LINEAR-REGULATOR CONTROLLER
FBP Regulation Voltage
VFBP
FBP Effective Load-Regulation
Error (Transconductance)
1.210
VDRVP = 10V, IDRVP = 50µA to 1mA
FBP Line (IN) Regulation Error
DRVP Sink Current
IDRVP = 100µA
IDRVP = 100µA, 2.5V < VIN < 6V
IDRVP
VFBP = 1.1V, VDRVP = 10V
1
1.280
V
-1.5
%
10
mV
mA
_______________________________________________________________________________________
5
MAX8795A
ELECTRICAL CHARACTERISTICS
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = -40°C to +85°C, unless otherwise noted.) (Note 3)
PARAMETER
SYMBOL
CONDITIONS
MIN
MAX
UNITS
0.972
1.022
V
25
mV
±5
mV
GATE-OFF LINEAR-REGULATOR CONTROLLER
FBN Regulation Voltage
VFBN
FBN Effective Load-Regulation
Error (Transconductance)
VDRVN = -10V, IDRVN = 50µA to 1mA
FBN Line (IN) Regulation Error
DRVN Source Current
IDRVN = 100µA, VREF - VFBN
IDRVN = 0.1mA, 2.5V < VIN < 6V
IDRVN
VFBN = 300mV, VDRVN = -10V
1
mA
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1V
VTH(DEL)
CTL Input Low Voltage
VIN = 2.5V to 5.5V
CTL Input High Voltage
VIN = 2.5V to 5.5V
4
6
1.19
1.31
V
0.6
V
2
V
SRC Input Voltage Range
SRC Input Current
36
ISRC
µA
VDEL = 1.5V, CTL = IN
300
VDEL = 1.5V, CTL = AGND
200
V
µA
SRC-to-COM Switch On-Resistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
10
Ω
DRN-to-COM Switch On-Resistance
RDRN(ON)
VDEL = 1.5V, CTL = AGND
60
Ω
ELECTRICAL CHARACTERISTICS
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = 0°C to +105°C. Typical values are at TA = +25°C,
unless otherwise noted.)
PARAMETER
IN Supply Range
IN Undervoltage-Lockout
Threshold
SYMBOL
VIN
VUVLO
CONDITIONS
MIN
(Note 2)
2.5
VIN rising, typical hysteresis = 50mV
2.05
VFB = VFBP = 1.3V, VFBN = 0V,
LX not switching
IN Quiescent Current
IIN
VFB or FBP below threshold or FBN above
threshold
REF Output Voltage
No external load
UNITS
6.0
V
2.25
2.45
V
0.6
1.0
200
ms
1.250
1.262
TA = 0°C to +105°C
1.232
1.250
1.266
0 < ILOAD < 50µA
In regulation
REF Undervoltage-Lockout
Threshold
Rising edge, typical hysteresis = 160mV
Hysteresis
3
1.238
REF Sink Current
Temperature rising
2
TA = +25°C to +105°C
REF Load Regulation
6
MAX
mA
VFB = 1.2V, VFBP = 1.4V, VFBN = 0V,
LX switching
Duration-to-Trigger Fault
Condition
Thermal Shutdown
TYP
10
10
V
mV
µA
1.15
+160
15
_______________________________________________________________________________________
V
°C
TFT-LCD DC-DC Converter with
Operational Amplifiers
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = 0°C to +105°C. Typical values are at TA = +25°C,
unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
18
V
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
VIN
Operating Frequency
fOSC
1000
1200
1400
kHz
86
90
93
%
TA = +25°C to +105°C
1.221
1.233
1.245
TA = 0°C to +105°C
1.212
1.233
1.248
1.10
1.14
1.17
V
%/V
Oscillator Maximum Duty Cycle
FB Regulation Voltage
VFB
No load
FB Fault Trip Level
VFB falling
FB Load Regulation
0 < IMAIN < full load, transient only
FB Line Regulation
VIN = 2.5V to 6V
FB Input Bias Current
VFB = 1.233V
FB Transconductance
∆ICOMP = ±2.5µA
FB Voltage Gain
LX On-Resistance
RLX(ON)
-1
75
0.1
±0.4
+100
+200
nA
160
280
µS
From FB to COMP
700
ILX = 200mA
160
300
mΩ
10
20
µA
2.5
3.0
3.5
A
0.1
0.2
0.3
V/A
LX Leakage Current
ILX
VLX = 19V
LX Current Limit
ILIM
VFB = 1.2V, duty cycle = 75%
Current-Sense Transresistance
Soft-Start Period
V
tSS
Soft-Start Step Size
V/V
14
ms
VREF /
128
V
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
6.0
SUP Overvoltage Fault Threshold
18.0
SUP Supply Current
ISUP
Buffer configuration, VPOS_ = VSUP / 2,
no load
Input Offset Voltage
VOS
(VNEG_, VPOS_, VOUT_) ≅ VSUP / 2
Input Bias Current
IBIAS
(VNEG_, VPOS_, VOUT_) ≅ VSUP / 2
Input Common-Mode Voltage
Range
VCM
Common-Mode Rejection Ratio
CMRR
-50
0 ≤ (VNEG_, VPOS_) ≤ VSUP
45
mA
0
12
mV
0
+50
nA
VSUP
V
dB
dB
VSUP 100
VSUP 50
mV
To VSUP / 2, source or sink
75
130
DC, 6V ≤ VSUP ≤ 18V,
(VNEG_, VPOS_) ≅ VSUP / 2
60
IOUT_ = 5mA
Output Voltage Swing, Low
VOL
IOUT_ = -5mA
Slew Rate
-3dB Bandwidth
5.0
80
VOH
PSRR
3.5
125
Output Voltage Swing, High
Power-Supply Rejection Ratio
V
19.9
0
Open-Loop Gain
Short-Circuit Current
18.0
19
RL = 10kΩ, CL = 10pF, buffer configuration
50
100
mV
mA
dB
45
V/µs
20
MHz
_______________________________________________________________________________________
7
MAX8795A
ELECTRICAL CHARACTERISTICS (continued)
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = 0°C to +105°C. Typical values are at TA = +25°C,
unless otherwise noted.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
V
GATE-ON LINEAR-REGULATOR CONTROLLER
FBP Regulation Voltage
VFBP
FBP Fault Trip Level
FBP Input Bias Current
IFBP
IDRVP = 100µA
1.231
1.250
1.269
VFBP falling
0.96
1.00
1.04
V
VFBP = 1.25V
-50
+50
nA
FBP Effective Load-Regulation
Error (Transconductance)
VDRVP = 10V, IDRVP = 50µA to 1mA
-0.7
-1.5
%
FBP Line (IN) Regulation Error
IDRVP = 100µA, 2.5V < VIN < 6V
±1
±10
mV
mA
DRVP Sink Current
IDRVP
DRVP Off-Leakage Current
Soft-Start Period
VFBP = 1.1V, VDRVP = 10V
1
VFBP = 1.4V, VDRVP = 34V
5
0.01
tSS
Soft-Start Step Size
10
µA
14
ms
VREF /
128
V
GATE-OFF LINEAR-REGULATOR CONTROLLER
FBN Regulation Voltage
VFBN
FBN Fault Trip Level
FBN Input Bias Current
IFBN
IDRVN = 100µA, VREF - VFBN
0.984
1
1.015
V
VFBN rising
340
420
510
mV
VFBN = 0.25V
-50
+50
nA
11
25
mV
±0.7
±5
mV
FBN Effective Load-Regulation
Error (Transconductance)
VDRVN = -10V, IDRVN = 50µA to 1mA
FBN Line (IN) Regulation Error
IDRVN = 0.1mA, 2.5V < VIN < 6V
DRVN Source Current
IDRVN
DRVN Off-Leakage Current
Soft-Start Period
VFBN = 300mV, VDRVN = -10V
1
VFBN = 0V, VDRVN = -25V
5
-0.01
tSS
Soft-Start Step Size
mA
-10
µA
14
ms
(VREF VFBN) /
128
V
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1V
VTH(DEL)
4
5
6
µA
1.19
1.25
1.31
V
DEL Discharge Switch
On-Resistance
During UVLO, VIN = 2.0V
CTL Input Low Voltage
VIN = 2.5V to 5.5V
CTL Input High Voltage
VIN = 2.5V to 5.5V
2
CTL Input Leakage Current
CTL = AGND or IN
-1
CTL-to-SRC Propagation Delay
0.6
V
+1
µA
V
100
SRC Input Voltage Range
SRC Input Current
Ω
20
ns
36
ISRC
VDEL = 1.5V, CTL = IN
200
300
VDEL = 1.5V, CTL = AGND
115
200
V
µA
SRC-to-COM Switch On-Resistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
5
12
Ω
DRN-to-COM Switch On-Resistance
RDRN(ON)
VDEL = 1.5V, CTL = AGND
30
70
Ω
8
_______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = -40°C to +105°C, unless otherwise noted.) (Note3)
PARAMETER
IN Supply Range
SYMBOL
VIN
IN Undervoltage-Lockout
Threshold
VUVLO
CONDITIONS
MIN
MAX
UNITS
(Note 2)
2.5
6.0
V
VIN rising, typical hysteresis = 150mV
2.05
2.45
V
VFB = VFBP = 1.3V, VFBN = 0V,
LX not switching
IN Quiescent Current
IIN
1.0
mA
VFB = 1.2V, VFBP = 1.4V, VFBN = 0V,
LX switching
REF Output Voltage
No external load
REF Undervoltage-Lockout
Threshold
Rising edge, typical hysteresis = 160mV
3
1.218
1.277
V
1.15
V
18
V
900
1400
kHz
1.198
1.260
V
±0.4
%/ V
MAIN STEP-UP REGULATOR
Output Voltage Range
VMAIN
Operating Frequency
fOSC
FB Regulation Voltage
VFB
FB Line Regulation
VIN
No load
VIN = 2.5V to 6V
∆ICOMP = ±2.5µA
FB Transconductance
LX On-Resistance
LX Current Limit
RLX(ON)
ILIM
75
280
µS
300
mΩ
3.5
A
6
18
V
18.0
19.9
V
ILX = 200mA
VFB = 1.2V, duty cycle = 75%
2.5
OPERATIONAL AMPLIFIERS
SUP Supply Range
VSUP
SUP Overvoltage Fault Threshold
SUP Supply Current
ISUP
Buffer configuration, VPOS_ = VSUP / 2,
no load
5
mA
Input Offset Voltage
VOS
(VNEG_, VPOS_, VOUT_) ≅ VSUP / 2
12
mV
Input Common-Mode Voltage
Range
VCM
VSUP
V
Output Voltage Swing, High
VOH
IOUT_ = 5mA
Output Voltage Swing, Low
VOL
IOUT_ = -5mA
Short-Circuit Current
0
To VSUP / 2
VSUP 100
mV
100
Source
75
Sink
75
mV
mA
_______________________________________________________________________________________
9
MAX8795A
ELECTRICAL CHARACTERISTICS
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 3V, VMAIN = VSUP = 14V, VPGND = VAGND = VBGND = 0V, IREF = 25µA, TA = -40°C to +105°C, unless otherwise noted.) (Note3)
PARAMETER
SYMBOL
CONDITIONS
MIN
MAX
UNITS
1.210
1.280
V
-1.5
%
10
mV
GATE-ON LINEAR-REGULATOR CONTROLLER
FBP Regulation Voltage
VFBP
FBP Effective Load-Regulation
Error (Transconductance)
VDRVP = 10V, IDRVP = 50µA to 1mA
FBP Line (IN) Regulation Error
DRVP Sink Current
IDRVP = 100µA
IDRVP = 100µA, 2.5V < VIN < 6V
IDRVP
VFBP = 1.1V, VDRVP = 10V
1
mA
GATE-OFF LINEAR-REGULATOR CONTROLLER
FBN Regulation Voltage
VFBN
FBN Effective Load-Regulation
Error (Transconductance)
0.972
VDRVN = -10V, IDRVN = 50µA to 1mA
FBN Line (IN) Regulation Error
DRVN Source Current
IDRVN = 100µA, VREF - VFBN
IDRVN = 0.1mA, 2.5V < VIN < 6V
IDRVN
VFBN = 300mV, VDRVN = -10V
1.022
V
25
mV
±5
mV
1
mA
POSITIVE GATE-DRIVER TIMING AND CONTROL SWITCHES
DEL Capacitor Charge Current
DEL Turn-On Threshold
During startup, VDEL = 1V
VTH(DEL)
CTL Input Low Voltage
VIN = 2.5V to 5.5V
CTL Input High Voltage
VIN = 2.5V to 5.5V
4
6
µA
1.19
1.31
V
0.6
2
SRC Input Voltage Range
SRC Input Current
36
ISRC
V
V
VDEL = 1.5V, CTL = IN
300
VDEL = 1.5V, CTL = AGND
200
V
µA
SRC-to-COM Switch
On-Resistance
RSRC(ON)
VDEL = 1.5V, CTL = IN
12
Ω
DRN-to-COM Switch
On-Resistance
RDRN(ON)
VDEL = 1.5V, CTL = AGND
70
Ω
Note 2: For 5.5V < VIN < 6.0V, use MAX8795A for no longer than 1% of IC lifetime. For continuous operation, input voltage should
not exceed 5.5V.
Note 3: Specifications to -40°C and +105°C are guaranteed by design, not production tested.
10
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
80
75
MAX9795A toc02
18
1.3
1.2
9
6
CURRENT INTO IN PIN
0
1.0
2.5
1000
100
CURRENT INTO INDUCTOR
3
70
10
12
1.1
VIN = 5V
VMAIN = 13.9V
1
NO LOAD, SUP DISCONNECTED,
R1 = 221kΩ, R2 = 21.5kΩ
15
SUPPLY CURRENT (mA)
85
1.4
SWITCHING FREQUENCY (MHz)
MAX9795A toc01
90
EFFICIENCY (%)
STEP-UP SUPPLY CURRENT
vs. SUPPLY VOLTAGE
SWITCHING FREQUENCY
vs. INPUT VOLTAGE
MAX8795A toc03
STEP-UP EFFICIENCY
vs. LOAD CURRENT
3.0
3.5
4.0
4.5
5.0
2.5
5.5
3.0
3.5
STEP-UP REGULATOR PULSED
LOAD-TRANSIENT RESPONSE
STEP-UP REGULATOR SOFT-START
(HEAVY LOAD)
4.0
4.5
5.0
6.0
TIMER-DELAYED OVERLOAD PROTECTION
MAX8795A toc05
MAX8795A toc04
5.5
SUPPLY VOLTAGE (V)
INPUT VOLTAGE (V)
LOAD CURRENT (mA)
MAX8795A toc06
A
A
0V
B
50mA
A
13.9V
B
0V
C
B
0A
0A
C
0U
0A
10µs/div
A: VIN, 5V/div
B: VMAIN, 5V/div
C: INDUCTOR CURRENT, 1A/div
40ms/div
A: VMAIN, 2V/div
B: INDUCTOR CURRENT, 1A/div
A: LOAD CURRENT, 1A/div
B: VMAIN, 200mV/div, AC-COUPLED
C: INDUCTOR CURRENT, 1A/div
REF VOLTAGE LOAD REGULATION
MAX8795A toc08
1.2495
VOLTAGE ERROR (%)
1.2490
1.2485
1.2480
-0.1
VOLTAGE ERROR (%)
-0.2
REF VOLTAGE (V)
GATE-ON REGULATOR LOAD REGULATION
GATE-ON REGULATOR LINE REGULATION
0
MAX8795A toc07
1.2500
-0.4
MAX8795A toc09
2ms/div
-0.3
-0.6
1.2475
1.2470
IPOS = 20mA
-0.8
0
10
20
30
LOAD CURRENT (µA)
40
50
25
26
27
28
INPUT VOLTAGE (V)
29
IBOOST = 200mA
-0.5
30
0
5
10
15
20
LOAD CURRENT (mA)
______________________________________________________________________________________
11
MAX8795A
Typical Operating Characteristics
(Circuit of Figure 1, VIN = 5V, VMAIN = 14V, VGON = 25V, VGOFF = -10V, TA = +25°C, unless otherwise noted.)
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 5V, VMAIN = 14V, VGON = 25V, VGOFF = -10V, TA = +25°C, unless otherwise noted.)
GATE-OFF REGULATOR LOAD REGULATION
GATE-OFF REGULATOR LINE REGULATION
0
VOLTAGE ERROR (%)
0.2
0
-0.2
-0.4
-0.6
POWER-UP SEQUENCE
MAX8795A toc12
MAX8795A toc11
0.2
MAX8795A toc10
0.4
VOLTAGE ERROR (%)
A
0V
-0.2
B
-0.4
0V
0V
-0.6
C
D
-0.8
-0.8
INEG = 50mA
-16
IBOOST = 0mA
-1.0
-1.0
-14
-12
0V
0
-10
10
20
30
40
50
4ms/div
A: VMAIN, 10V/div
B: VPOS, 20V/div
LOAD CURRENT (mA)
INPUT VOLTAGE (V)
C: VNEG, 10V/div
D: VCOM, 20V/div
OPERATIONAL-AMPLIFIER
RAIL-TO-RAIL INPUT/OUTPUT
SUP SUPPLY CURRENT
vs. SUP VOLTAGE
MAX8795A toc14
MAX8795A toc13
3.6
3.5
VSUP = 15V
3.4
ISUP (mA)
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
A
3.3
3.2
0V
3.1
B
3.0
2.9
0V
2.8
6
8
10
12
VSUP (V)
12
14
16
18
4µs/div
A: INPUT SIGNAL, 5V/div
B: OUTPUT SIGNAL, 5V/div
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
OPERATIONAL-AMPLIFIER
LARGE-SIGNAL RESPONSE
OPERATIONAL-AMPLIFIER
LOAD-TRANSIENT RESPONSE
MAX8795A toc15
OPERATIONAL-AMPLIFIER
SMALL-SIGNAL RESPONSE
MAX8795A toc16
MAX8795A toc17
VSUP = 15V
A
0V
A
A
0V
0V
+50mA
B
0mA
B
B
0V
-50mA
0V
1µs/div
400ns/div
A: OUTPUT VOLTAGE, 1V/div, AC-COUPLED
B: OUTPUT CURRENT, 50mA/div
400ns/div
A: INPUT SIGNAL, 5V/div
B: OUTPUT SIGNAL, 5V/div
A: INPUT SIGNAL, 100mV/div
B: OUTPUT SIGNAL, 100mV/div
Pin Description
PIN
NAME
FUNCTION
1
SRC
Switch Input. Source of the internal high-voltage p-channel MOSFET. Bypass SRC to PGND with a
minimum 0.1µF capacitor close to the pins.
2
REF
Reference Bypass Terminal. Bypass REF to AGND with a minimum of 0.22µF close to the pins.
3
AGND
Analog Ground for Step-Up Regulator and Linear Regulators. Connect to power ground (PGND)
underneath the IC.
4
PGND
Power Ground. PGND is the source of the main step-up n-channel power MOSFET. Connect PGND to
the output-capacitor ground terminals through a short, wide PCB trace. Connect to analog ground
(AGND) underneath the IC.
5
OUT1
Operational-Amplifier 1 Output
6
NEG1
Operational-Amplifier 1 Inverting Input
7
POS1
Operational-Amplifier 1 Noninverting Input
8
OUT2
Operational-Amplifier 2 Output
9
NEG2
Operational-Amplifier 2 Inverting Input
10
POS2
Operational-Amplifier 2 Noninverting Input
11
BGND
Analog Ground for Operational Amplifiers. Connect to power ground (PGND) underneath the IC.
12
POS3
Operational-Amplifier 3 Noninverting Input
13
OUT3
Operational-Amplifier 3 Output
14
SUP
Operational-Amplifier Power Input. Positive supply rail for the operational amplifiers. Typically
connected to VMAIN. Bypass SUP to BGND with a 0.1µF capacitor.
15
POS4
Operational-Amplifier 4 Noninverting Input
16
NEG4
Operational-Amplifier 4 Inverting Input
______________________________________________________________________________________
13
MAX8795A
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 5V, VMAIN = 14V, VGON = 25V, VGOFF = -10V, TA = +25°C, unless otherwise noted.)
TFT-LCD DC-DC Converter with
Operational Amplifiers
MAX8795A
Pin Description (continued)
14
PIN
NAME
FUNCTION
17
OUT4
Operational-Amplifier 4 Output
18
POS5
Operational-Amplifier 5 Noninverting Input
19
NEG5
Operational-Amplifier 5 Inverting Input
20
OUT5
Operational-Amplifier 5 Output
21
LX
n-Channel Power MOSFET Drain and Switching Node. Connect the inductor and Schottky diode to LX
and minimize the trace area for lowest EMI.
22
IN
Supply Voltage Input. IN can range from 2.5V to 6V.
23
FB
Step-Up Regulator Feedback Input. Regulates to 1.233V (nominal). Connect a resistive voltage-divider
from the output (VMAIN) to FB to analog ground (AGND). Place the divider within 5mm of FB.
24
COMP
25
FBP
Gate-On Linear-Regulator Feedback Input. FBP regulates to 1.25V (nominal). Connect FBP to the
center of a resistive voltage-divider between the regulator output and AGND to set the gate-on linearregulator output voltage. Place the resistive voltage-divider within 5mm of FBP.
26
DRVP
Gate-On Linear-Regulator Base Drive. Open drain of an internal n-channel MOSFET. Connect DRVP to
the base of an external pnp pass transistor. See the Pass-Transistor Selection section.
27
FBN
28
DRVN
29
DEL
High-Voltage Switch Delay Input. Connect a capacitor from DEL to AGND to set the high-voltage
switch startup delay.
30
CTL
High-Voltage Switch Control Input. When CTL is high, the high-voltage switch between COM and SRC
is on and the high-voltage switch between COM and DRN is off. When CTL is low, the high-voltage
switch between COM and SRC is off and the high-voltage switch between COM and DRN is on. CTL is
inhibited by the undervoltage lockout or when the voltage on DEL is less than 1.25V.
31
DRN
Switch Input. Drain of the internal high-voltage back-to-back p-channel MOSFETs connected to COM.
32
COM
Internal High-Voltage MOSFET Switch Common Terminal. Do not allow the voltage on COM to exceed
VSRC.
—
EP
Step-Up Regulator Error-Amplifier Compensation Point. Connect a series RC from COMP to AGND.
See the Loop Compensation section for component selection guidelines.
Gate-Off Linear-Regulator Feedback Input. FBN regulates to 250mV (nominal). Connect FBN to the
center of a resistive voltage-divider between the regulator output and REF to set the gate-off linearregulator output voltage. Place the resistive voltage-divider within 5mm of FBN.
Gate-Off Linear-Regulator Base Drive. Open drain of an internal p-channel MOSFET. Connect DRVN to
the base of an external npn pass transistor. See the Pass-Transistor Selection section.
Exposed Paddle. Must be connected to AGND. Do not use as the only ground connection.
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
-10V gate-driver supplies. The input voltage range for the
IC is from +2.5V to +5.5V. The listed load currents in
Figure 1 are available from a +4.5V to +5.5V supply.
Table 1 lists some recommended components, and Table
2 lists the contact information of component suppliers.
The MAX8795A typical operating circuit (Figure 1) is a
complete power-supply system for TFT LCDs. The circuit
generates a +14V source-driver supply and +25V and
LX
L1
3.0µH
VIN
4.5V TO 5.5V
C1
22µF
D1
R10
10Ω
R1
137kΩ
1%
LX
IN
COMP
C10
0.1µF
VGOFF
-10V/50mA
C14
68pF
R3
6.8kΩ
MAX8795A
DRVP
Q2
R7
324kΩ
1%
FBP
R4
191kΩ
1%
R5
10.0kΩ
1%
R8
31.6kΩ
1%
C5
0.47µF
SRC
COM
REF
C8
0.22µF
VGON
25V/20mA
DRN
R6
1kΩ
CTL
DEL
SUP
C7
0.033µF
C6
0.1µF
BGND
NEG1
R19
100kΩ
OUT1
NEG2
TO VCOM
BACKPLANE
Q1
DRVN
FBN
C9
0.22µF
D2
PGND
R9
6.8kΩ
D3
C4
0.1µF
AGND
C12
220µF
C11
0.1µF
C3
0.1µF
R2
13.3kΩ
1%
180kΩ
LX
LX
FB
C13
0.1µF
VMAIN
14V/500mA
C2
22µF
OUT2
POS1
OUT3
POS2
NEG4
POS3
OUT4
POS4
R17
100kΩ
OUT5
EP
R13
100kΩ
R11
100kΩ
R12
100kΩ
POS5
NEG5
R15
100kΩ
R20
100kΩ
R18
100kΩ
R16
100kΩ
R14
100kΩ
Figure 1. Typical Operating Circuit
Table 1. Component List
DESIGNATION
DESCRIPTION
DESIGNATION
DESCRIPTION
C1
22µF, 6.3V X5R ceramic capacitor (1210)
TDK C3225X5R0J227M
L1
3.0µH, 3A inductor
Sumida CDRH6D28-3R0
C2
22µF, 16V X5R ceramic capacitor (1812)
TDK C4532X5X1C226M
Q1
200mA, 40V pnp bipolar transistor (SOT23)
Fairchild MMBT3906
D1
3A, 30V Schottky diode (M-flat)
Toshiba CMS02
Q2
200mA, 40V npn bipolar transistor (SOT23)
Fairchild MMBT3904
D2, D3
200mA, 100V, dual ultra-fast diodes (SOT23)
Fairchild MMBD4148SE
______________________________________________________________________________________
15
MAX8795A
Typical Operating Circuit
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
Table 2. Component Suppliers
SUPPLIER
Fairchild
Sumida
TDK
Toshiba
PHONE
408-822-2000
847-545-6700
847-803-6100
949-455-2000
VCN
FAX
408-822-2102
847-545-6720
847-390-4405
949-859-3963
Detailed Description
VCP
VIN
The MAX8795A contains a high-performance step-up
switching regulator, two low-cost linear-regulator controllers, multiple high-current operational amplifiers,
and startup timing and level-shifting functionality useful
for active-matrix TFT LCDs. Figure 2 shows the
MAX8795A functional diagram.
VMAIN
LX
IN
WEBSITE
www.fairchildsemi.com
www.sumida.com
www.component.tdk.com
www.toshiba.com/taec
FB
STEP-UP
CONTROLLER
Main Step-Up Regulator
PGND
COMP
AGND
VCP
MAX8795A
DRVP
GATE-ON
CONTROLLER
FBP
VGON
SRC
DEL
COM
SWITCH
CONTROL
CTL
VCN
DRN
DRVN
GATE-OFF
CONTROLLER
SUP
VGOFF
NEG1
FBN
OUT1
REF
NEG2
REF
NEG4
OP2
OP4
POS2
OUT4
POS4
NEG5
OUT3
OP5
OP3
POS3
OUT5
POS5
EP
BGND
Figure 2. MAX8795A Functional Diagram
16
V
−V
D ≈ MAIN IN
VMAIN
OP1
POS1
OUT2
The main step-up regulator employs a current-mode,
fixed-frequency PWM architecture to maximize loop
bandwidth and provide fast transient response to
pulsed loads typical of TFT-LCD panel source drivers.
The 1.2MHz switching frequency allows the use of lowprofile inductors and ceramic capacitors to minimize
the thickness of LCD panel designs. The integrated
high-efficiency MOSFET and the IC’s built-in digital
soft-start functions reduce the number of external components required while controlling inrush currents. The
output voltage can be set from VIN to 18V with an external resistive voltage-divider.
The regulator controls the output voltage and the power
delivered to the output by modulating the duty cycle (D)
of the internal power MOSFET in each switching cycle.
The duty cycle of the MOSFET is approximated by:
Figure 3 shows the functional diagram of the step-up
regulator. An error amplifier compares the signal at FB
to 1.233V and changes the COMP output. The voltage
at COMP sets the peak inductor current. As the load
varies, the error amplifier sources or sinks current to the
COMP output accordingly to produce the inductor peak
current necessary to service the load. To maintain stability at high duty cycles, a slope-compensation signal
is summed with the current-sense signal.
On the rising edge of the internal clock, the controller sets
a flip-flop, turning on the n-channel MOSFET and applying
the input voltage across the inductor. The current through
the inductor ramps up linearly, storing energy in its
magnetic field. Once the sum of the current-feedback
signal and the slope compensation exceeds the COMP
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
RESET DOMINANT
CLOCK
S
R
pnp PASS
TRANSISTOR
DRVP
PGND
Q
npn CASCODE
TRANSISTOR
ILIM
COMPARATOR
VGON
MAX8795A
SOFTSTART
VLIMIT
FBP
SLOPE COMP
PWM
COMPARATOR
Σ
CURRENT
SENSE
Figure 4. Using Cascoded npn for Charge-Pump Output
Voltages > 36V
OSCILLATOR
FAULT
COMPARATOR
LX
0.1µF
TO FAULT LATCH
1.14V
VMAIN
14V
ERROR AMP
FB
0.1µF
68pF
1.233V
COMP
6.8kΩ
Q1
DRVP
VGON
35V
Figure 3. Step-Up Regulator Functional Diagram
voltage, the controller resets the flip-flop and turns off
the MOSFET. Since the inductor current is continuous,
a transverse potential develops across the inductor that
turns on the diode (D1). The voltage across the inductor then becomes the difference between the output
voltage and the input voltage. This discharge condition
forces the current through the inductor to ramp back
down, transferring the energy stored in the magnetic
field to the output capacitor and the load. The MOSFET
remains off for the rest of the clock cycle.
Gate-On Linear-Regulator Controller, REG P
The gate-on linear-regulator controller (REG P) is an
analog gain block with an open-drain n-channel output.
It drives an external pnp pass transistor with a 6.8kΩ
base-to-emitter resistor (Figure 1). Its guaranteed basedrive sink current is at least 1mA. The regulator including
Q1 in Figure 1 uses a 0.47µF ceramic output capacitor
and is designed to deliver 20mA at 25V. Other output
voltages and currents are possible with the proper pass
transistor and output capacitor. See the Pass-Transistor
Selection and Stability Requirements sections.
MAX8795A
0.22µF
0.47µF
47pF
274kΩ
1%
150pF
10.2kΩ
1%
FBP
Figure 5. The linear regulator controls the intermediate chargepump stage.
REG P is typically used to provide the TFT-LCD gate
drivers’ gate-on voltage. Use a charge pump with as
many stages as necessary to obtain a voltage exceeding the required gate-on voltage (see the Selecting the
Number of Charge-Pump Stages section). Note the
voltage rating of DRVP is 36V. If the charge-pump output voltage can exceed 36V, an external cascode npn
transistor should be added as shown in Figure 4.
Alternately, the linear regulator can control an intermediate charge-pump stage while regulating the final
charge-pump output (Figure 5).
______________________________________________________________________________________
17
MAX8795A
FROM CHARGE-PUMP
OUTPUT
VMAIN
LX
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
REG P is enabled after the REF voltage exceeds 1.0V.
Each time it is enabled, the controller goes through a
soft-start routine that ramps up its internal reference
DAC in 128 steps.
Gate-Off Linear-Regulator Controller, REG N
The gate-off linear-regulator controller (REG N) is an
analog gain block with an open-drain p-channel output.
It drives an external npn pass transistor with a 6.8kΩ
base-to-emitter resistor (Figure 1). Its guaranteed basedrive source current is at least 1mA. The regulator
including Q2 in Figure 1 uses a 0.47µF ceramic output
capacitor and is designed to deliver 50mA at -10V. Other
output voltages and currents are possible with the proper
pass transistor and output capacitor (see the PassTransistor Selection and Stability Requirements sections).
REG N is typically used to provide the TFT-LCD gate
drivers’ gate-off voltage. A negative voltage can be
produced using a charge-pump circuit as shown in
Figure 1. REG N is enabled after the voltage on REF
exceeds 1.0V. Each time it is enabled, the control goes
through a soft-start routine that ramps down its internal
reference DAC from VREF to 250mV in 128 steps.
Operational Amplifiers
The MAX8795A has five operational amplifiers. The operational amplifiers are typically used to drive the LCD backplane (VCOM) or the gamma-correction divider string.
They feature ±130mA output short-circuit current, 45V/µs
slew rate, and 20MHz/3dB bandwidth. The rail-to-rail input
and output capability maximizes system flexibility.
Short-Circuit Current Limit and Input Clamp
The operational amplifiers limit short-circuit current to
approximately ±130mA if the output is directly shorted to
SUP or to BGND. If the short-circuit condition persists, the
junction temperature of the IC rises until it reaches the
thermal-shutdown threshold (+160°C typ). Once the junction temperature reaches the thermal-shutdown threshold,
an internal thermal sensor immediately sets the thermal
fault latch, shutting off all the IC’s outputs. The device
remains inactive until the input voltage is cycled.
The operational amplifiers have 4V input clamp structures
in series with a 500Ω resistance and a diode (Figure 2).
Driving Pure Capacitive Load
The operational amplifiers are typically used to drive
the LCD backplane (VCOM) or the gamma-correction
divider string. The LCD backplane consists of a distributed series capacitance and resistance, a load that can
be easily driven by the operational amplifier. However,
if the operational amplifier is used in an application with
a pure capacitive load, steps must be taken to ensure
stable operation.
18
As the operational amplifier’s capacitive load increases,
the amplifier’s bandwidth decreases and gain peaking
increases. A 5Ω to 50Ω small resistor placed between
OUT_ and the capacitive load reduces peaking, but also
reduces the gain. An alternative method of reducing
peaking is to place a series RC network (snubber) in parallel with the capacitive load. The RC network does not
continuously load the output or reduce the gain. Typical
values of the resistor are between 100Ω and 200Ω, and
the typical value of the capacitor is 10nF.
Undervoltage Lockout (UVLO)
The UVLO circuit compares the input voltage at IN with the
UVLO threshold (2.25V rising, 2.20V falling, typ) to ensure
the input voltage is high enough for reliable operation. The
50mV (typ) hysteresis prevents supply transients from
causing a restart. Once the input voltage exceeds the
UVLO rising threshold, startup begins. When the input voltage falls below the UVLO falling threshold, the controller
turns off the main step-up regulator, turns off the linearregulator outputs, and disables the switch control block;
the operational-amplifier outputs are high impedance.
Reference Voltage (REF)
The reference output is nominally 1.25V and can
source at least 50µA (see the Typical Operating
Characteristics). Bypass REF with a 0.22µF ceramic
capacitor connected between REF and AGND.
Power-Up Sequence and Soft-Start
Once the voltage on IN exceeds approximately 2.25V, the
reference turns on. With a 0.22µF REF bypass capacitor,
the reference reaches its regulation voltage of 1.25V in
approximately 1ms. When the reference voltage exceeds
1.0V, the IC enables the main step-up regulator, the gateon linear-regulator controller, and the gate-off linear-regulator controller simultaneously.
The IC employs soft-start for each regulator to minimize
inrush current and voltage overshoot and to ensure a welldefined startup behavior. Each output uses a 7-bit soft-start
DAC. For the step-up and the gate-on linear regulator, the
DAC output is stepped in 128 steps from zero up to the reference voltage. For the gate-off linear regulator, the DAC
output steps from the reference down to 250mV in 128
steps from zero up to the reference voltage. For the gateoff linear regulator’s voltage ramp soft-start, the DAC output
steps from the reference down to 250mV in 128 steps. The
soft-start duration is 14ms (typ) for all three regulators, and
DEL remains pulled down to AGND during the soft start
period.
Once the main step-up regulator, the gate-on linear-regulator controller, and the gate-off linear-regulator controller
reach regulation, a 5µA current source starts charging
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
VREF
1.05V
VMAIN
VGON
CDEL = DELAY _ TIME ×
I6µA
1.25V
Switch-Control Block
12ms
1.25V
INPUT
SOFT- SOFTVOLTAGE START START
OK
BEGINS ENDS
SWITCH
CONTROL
ENABLED
VGOFF
VDEL
The switch-control input (CTL) is not activated until all
four of the following conditions are satisfied: the input
voltage exceeds VUVLO, the soft-start routine of all the
regulators is complete, there is no fault condition detected, and VDEL exceeds its turn-on threshold. Once activated and if CTL is high, the 5Ω internal p-channel
switch (Q1) between COM and SRC turns on and the
Figure 6. Power-Up Sequence
IN
MAX8795A
5µA
2.25V
FB OK
FBP OK
FBN OK
Q1
SRC
DEL
REF
COM
Q2
CTL
DRN
Figure 7. Switch-Control Block
______________________________________________________________________________________
19
MAX8795A
VIN
2.25V
CDEL. Once the CDEL capacitor voltage exceeds 1.25V
(typ), the switch-control block is and op amps are enabled
as shown in Figure 6. After the switch-control block is
enabled, COM can be connected to SRC or DRN through
the internal p-channel switches, depending upon the state of
CTL. Before startup and when IN is less than VUVLO, DEL is
internally connected to AGND to discharge CDEL. Select
CDEL to set the initial start-up delay and the switch-control
block startup delay times using the following equation:
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
30Ω p-channel switch (Q2) between DRN and COM
turns off. If CTL is low, Q1 turns off and Q2 turns on.
Fault Protection
During steady-state operation, if the output of the main
regulator or any of the linear-regulator outputs does not
exceed its respective fault-detection threshold, the
MAX8795A activates an internal fault timer. If any condition or combination of conditions indicates a continuous
fault for the fault-timer duration (200ms typ), the
MAX8795A sets the fault latch to shut down all the outputs
except the reference. Once the fault condition is removed,
cycle the input voltage (below the UVLO falling threshold)
to clear the fault latch and reactivate the device. The faultdetection circuit is disabled during the soft-start time.
Thermal-Overload Protection
Thermal-overload protection prevents excessive power dissipation from overheating the MAX8795A. When the junction temperature exceeds +160°C, a thermal sensor
immediately activates the fault protection, which shuts
down all outputs except the reference, allowing the device
to cool down. Once the device cools down by approximately 15°C, cycle the input voltage (below the UVLO falling
threshold) to clear the fault latch and reactivate the device.
The thermal-overload protection protects the controller
in the event of fault conditions. For continuous operation, do not exceed the absolute maximum junction
temperature rating of +150°C.
Design Procedure
Main Step-Up Regulator
Inductor Selection
The minimum inductance value, peak current rating,
and series resistance are factors to consider when
selecting the inductor. These factors influence the converter’s efficiency, maximum output load capability,
transient-response time, and output voltage ripple. Size
and cost are also important factors to consider.
The maximum output current, input voltage, output voltage, and switching frequency determine the inductor
value. Very high inductance values minimize the current
ripple, and therefore, reduce the peak current, which
decreases core losses in the inductor and conduction
losses in the entire power path. However, large inductor
values also require more energy storage and more turns of
wire, which increase size and can increase conduction
losses in the inductor. Low inductance values decrease
the size, but increase the current ripple and peak current.
Finding the best inductor involves choosing the best compromise between circuit efficiency, inductor size, and cost.
The equations used here include a constant LIR, which is
the ratio of the inductor peak-to-peak ripple current to the
20
average DC inductor current at the full load current. The
best trade-off between inductor size and circuit efficiency
for step-up regulators generally has an LIR between 0.3
and 0.6. However, depending on the AC characteristics of
the inductor core material and ratio of inductor resistance
to other power-path resistances, the best LIR can shift up
or down. If the inductor resistance is relatively high, more
ripple can be accepted to reduce the number of turns
required and increase the wire diameter. If the inductor
resistance is relatively low, increasing inductance to lower
the peak current can decrease losses throughout the
power path. If extremely thin high-resistance inductors are
used, as is common for LCD-panel applications, the best
LIR can increase to between 0.5 and 1.0.
Once a physical inductor is chosen, higher and lower
values of the inductor should be evaluated for efficiency improvements in typical operating regions.
Calculate the approximate inductor value using the typical input voltage (VIN), the maximum output current
(IMAIN(MAX)), the expected efficiency (ηTYP) taken from
an appropriate curve in the Typical Operating
Characteristics section, and an estimate of LIR based
on the above discussion:
⎞⎛ η
⎛ V
⎞ 2⎛
VMAIN − VIN
TYP ⎞
L = ⎜ IN ⎟ ⎜
⎟
⎟ ⎜⎝
⎠
V
I
f
LIR
×
⎝ MAIN ⎠ ⎝ MAIN(MAX) OSC ⎠
Choose an available inductor value from an appropriate
inductor family. Calculate the maximum DC input current at the minimum input voltage (VIN(MIN)) using conservation of energy and the expected efficiency at that
operating point (ηMIN) taken from the appropriate curve
in the Typical Operating Characteristics:
IIN(DCMAX
,
)=
IMAIN(MAX) × VMAIN
VIN(MIN) × ηMIN
Calculate the ripple current at that operating point and
the peak current required for the inductor:
IRIPPLE =
VIN(MIN) × (VMAIN − VIN(MIN) )
L × VMAIN × fOSC
IRIPPLE
IPEAK = IIN(DCMAX
,
)+
2
The inductor’s saturation current rating and the
MAX8795A’s LX current limit (ILIM) should exceed IPEAK,
and the inductor’s DC current rating should exceed
IIN(DC,MAX). For good efficiency, choose an inductor with
less than 0.1Ω series resistance.
Considering the typical operating circuit, the maximum
load current (IMAIN(MAX)) is 500mA with a 14V output and
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
be tolerated on CIN if IN is decoupled from CIN using an
RC lowpass filter (see R10 and C13 in Figure 1).
⎛ 5V ⎞ 2 ⎛ 14V − 5V ⎞ ⎛ 0.85 ⎞
L=⎜
⎟ ⎜
⎟⎜
⎟ ≈ 3.3µH
⎝ 14V ⎠ ⎝ 0.5A × 1.2 MHz ⎠ ⎝ 0.5 ⎠
Rectifier Diode
The MAX8795A’s high switching frequency demands a
high-speed rectifier. Schottky diodes are recommended
for most applications because of their fast recovery time
and low forward voltage. In general, a 2A Schottky
diode complements the internal MOSFET well.
Using the circuit’s minimum input voltage (4.5V) and
estimating efficiency of 80% at that operating point:
0.5A × 14V
I IN(DCMAX
≈ 1.94A
,
)=
4.5V × 0.8
The ripple current and the peak current are:
4.5V × (14V − 4.5V)
I RIPPLE =
≈ 0.77A
3.3µH × 14V × 1.2 MHz
0.77A
I PEAK = 1.94A +
≈ 2.33A
2
Output-Capacitor Selection
The total output voltage ripple has two components: the
capacitive ripple caused by the charging and discharging
of the output capacitance, and the ohmic ripple due to the
capacitor’s equivalent series resistance (ESR):
VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR)
⎛V
I
− VIN ⎞
VRIPPLE(C) ≈ MAIN ⎜ MAIN
COUT ⎝ VMAINfOSC ⎟⎠
and :
VRIPPLE(ESR) ≈ IPEAKR ESR(COUT)
where IRIPPLE is the RIPPLE inductor current (see the
Inductor Selection section). For ceramic capacitors, the
output voltage ripple is typically dominated by
VRIPPLE(C). The voltage rating and temperature characteristics of the output capacitor must also be considered.
Input-Capacitor Selection
The input capacitor (CIN) reduces the current peaks
drawn from the input supply and reduces noise injection
into the IC. A 22µF ceramic capacitor is used in the typical applications circuit (Figure 1) because of the high
source impedance seen in typical lab setups. Actual
applications usually have much lower source impedance
since the step-up regulator often runs directly from the
output of another regulated supply. Typically, CIN can
be reduced below the values used in the typical applications circuit. Ensure a low-noise supply at IN by using
adequate CIN. Alternately, greater voltage variation can
Output-Voltage Selection
The output voltage of the main step-up regulator can be
adjusted by connecting a resistive voltage-divider from the
output (VMAIN) to AGND with the center tap connected to
FB (see Figure 1). Select R2 in the 10kΩ to 50kΩ range.
Calculate R1 with the following equation:
⎛V
⎞
R1 = R2 × ⎜ MAIN − 1⎟
⎝ VFB
⎠
where VFB, the step-up regulator’s feedback set point,
is 1.233V. Place R1 and R2 close to the IC.
Loop Compensation
Choose RCOMP to set the high-frequency integrator
gain for fast transient response. Choose CCOMP to set
the integrator zero to maintain loop stability.
For low-ESR output capacitors, use the following equations to obtain stable performance and good transient
response:
RCOMP ≈
253 × VIN × VOUT × COUT
L × I MAIN(MAX)
CCOMP ≈
VOUT × COUT
10 × I MAIN(MAX) × RCOMP
To further optimize transient response, vary RCOMP in
20% steps and CCOMP in 50% steps while observing
transient-response waveforms.
Charge Pumps
Selecting the Number of Charge-Pump Stages
For highest efficiency, always choose the lowest number of charge-pump stages that meet the output
requirement. Figures 8 and 9 show the positive and
negative charge-pump output voltages for a given
VMAIN for one-, two-, and three-stage charge pumps.
______________________________________________________________________________________
21
MAX8795A
a typical input voltage of 5V. Choosing an LIR of 0.5 and
estimating efficiency of 85% at this operating point:
The number of positive charge-pump stages is given by:
V
+V
−V
nPOS = GON DROPOUT MAIN
VMAIN − 2 × VD
where nPOS is the number of positive charge-pump
stages, VGON is the gate-on linear-regulator REG P output, VMAIN is the main step-up regulator output, VD is
60
VD = 0.3V TO 1V
3-STAGE CHARGE PUMP
50
G_ON (V)
40
2-STAGE CHARGE PUMP
30
20
10
1-STAGE CHARGE PUMP
0
2
4
6
8
10
12
14
VMAIN (V)
Figure 8. Positive Charge-Pump Output Voltage vs. VMAIN
NEGATIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. VMAIN
-0
1-STAGE
CHARGE PUMP
-5
-10
-15
-20
2-STAGE
CHARGE PUMP
-25
-30
3-STAGE
CHARGE PUMP
-35
-40
VD = 0.3V TO 1V
4
6
nNEG =
−VGOFF + VDROPOUT
VMAIN − 2 × VD
V
+V
+V
nPOS = GON DROPOUT IN
VMAIN − 2 × VD
−V
+ VDROPOUT + VIN
nNEG = GOFF
VMAIN − 2 × VD
Flying Capacitors
Increasing the flying-capacitor (CX) value lowers the
effective source impedance and increases the outputcurrent capability. Increasing the capacitance indefinitely
has a negligible effect on output-current capability
because the internal switch resistance and the diode
impedance place a lower limit on the source impedance. A 0.1µF ceramic capacitor works well in most
low-current applications. The flying capacitor’s voltage
rating must exceed the following:
VCX > n × VMAIN
-45
2
the forward-voltage drop of the charge-pump diode,
and VDROPOUT is the dropout margin for the linear regulator. Use VDROPOUT = 0.3V.
The number of negative charge-pump stages is given by:
where nNEG is the number of negative charge-pump
stages, VGOFF is the gate-off linear-regulator REG N
output, VMAIN is the main step-up regulator output, VD
is the forward-voltage drop of the charge-pump diode,
and VDROPOUT is the dropout margin for the linear regulator. Use VDROPOUT = 0.3V.
The above equations are derived based on the
assumption that the first stage of the positive charge
pump is connected to VMAIN and the first stage of the
negative charge pump is connected to ground.
Sometimes fractional stages are more desirable for better efficiency. This can be done by connecting the first
stage to VIN or another available supply. If the first
charge-pump stage is powered from VIN, the above
equations become:
POSITIVE CHARGE-PUMP
OUTPUT VOLTAGE vs. VMAIN
G_OFF (V)
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
8
10
12
14
where n is the stage number in which the flying capacitor appears, and VMAIN is the output voltage of the
main step-up regulator.
VMAIN (V)
Figure 9. Negative Charge-Pump Output Voltage vs. VMAIN
22
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
COUT _ CP ≥
ILOAD _ CP
2fOSC VRIPPLE _ CP
where COUT_CP is the output capacitor of the charge
pump, I LOAD_CP is the load current of the charge
pump, and VRIPPLE_CP is the peak-to-peak value of the
output ripple.
Charge-Pump Rectifier Diodes
Use low-cost silicon switching diodes with a current rating equal to or greater than two times the average
charge-pump input current. If it helps avoid an extra
stage, some or all of the diodes can be replaced with
Schottky diodes with an equivalent current rating.
Linear-Regulator Controllers
Output-Voltage Selection
Adjust the gate-on linear-regulator (REG P) output voltage by connecting a resistive voltage-divider from the
REG P output to AGND with the center tap connected
to FBP (Figure 1). Select the lower resistor of the divider
R5 in the range of 10kΩ to 30kΩ. Calculate the upper
resistor R4 with the following equation:
⎛V
⎞
R4 = R5 × ⎜ GON − 1⎟
V
⎝ FBP
⎠
where VFBP = 1.25V (typ).
Adjust the gate-off linear-regulator REG N output voltage by connecting a resistive voltage-divider from
VGOFF to REF with the center tap connected to FBN
(Figure 1). Select R8 in the 20kΩ to 50kΩ range.
Calculate R7 with the following equation:
voltage, and power dissipation. The transistor’s current
gain limits the guaranteed maximum output current to:
⎛
V ⎞
ILOAD(MAX) = ⎜ IDRV − BE ⎟ × hFE(MIN)
RBE ⎠
⎝
where IDRV is the minimum guaranteed base-drive current, VBE is the transistor’s base-to-emitter forward voltage drop, and RBE is the pullup resistor connected
between the transistor’s base and emitter. Furthermore,
the transistor’s current gain increases the linear regulator’s DC loop gain (see the Stability Requirements section), so excessive gain destabilizes the output.
Therefore, transistors with current gain over 100 at the
maximum output current can be difficult to stabilize and
are not recommended unless the high gain is needed to
meet the load-current requirements.
The transistor’s saturation voltage at the maximum output current determines the minimum input-to-output
voltage differential that the linear regulator can support.
Also, the package’s power dissipation limits the usable
maximum input-to-output voltage differential. The maximum power-dissipation capability of the transistor’s
package and mounting must exceed the actual power
dissipated in the device. The power dissipated equals
the maximum load current (ILOAD(MAX)_LR) multiplied
by the maximum input-to-output voltage differential:
P = ILOAD(MAX)_ LR × (VIN(MAX)_ LR − VOUT _ LR )
where VIN(MAX)_LR is the maximum input voltage of the
linear regulator, and VOUT_LR is the output voltage of
the linear regulator.
Stability Requirements
The MAX8795A linear-regulator controllers use an internal transconductance amplifier to drive an external
pass transistor. The transconductance amplifier, the
pass transistor, the base-emitter resistor, and the output capacitor determine the loop stability. The following
applies to both linear-regulator controllers in the
MAX8795A.
V
−V
R7 = R8 × FBN GOFF
VREF − VFBN
The transconductance amplifier regulates the output
voltage by controlling the pass transistor’s base current. The total DC loop gain is approximately:
where VFBN = 250mV, VREF = 1.25V. Note that REF can
only source up to 50µA; using a resistor less than 20kΩ
for R8 results in higher bias current than REF can supply.
⎛ 10 ⎞ ⎡ ⎛ I
×h ⎞⎤
A V _ LR ≅ ⎜ ⎟ × ⎢1 + ⎜ BIAS FE ⎟ ⎥ × VREF
⎝ VT ⎠ ⎢⎣ ⎝ ILOAD _ LR ⎠ ⎥⎦
Pass-Transistor Selection
The pass transistor must meet specifications for current
gain (hFE), input capacitance, collector-emitter saturation
where VT is 26mV at room temperature, and IBIAS is the
current through the base-to-emitter resistor (RBE). For
the MAX8795A, the bias currents for both the gate-on
______________________________________________________________________________________
23
MAX8795A
Charge-Pump Output Capacitor
Increasing the output capacitance or decreasing the
ESR reduces the output ripple voltage and the peak-topeak transient voltage. With ceramic capacitors, the
output voltage ripple is dominated by the capacitance
value. Use the following equation to approximate the
required capacitor value:
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
and gate-off linear-regulator controllers are 0.1mA.
Therefore, the base-to-emitter resistor for both linear
regulators should be chosen to set 0.1mA bias current:
RBE =
0.7V
VBE
=
≈ 6.8kΩ
0.1mA 0.1mA
The output capacitor and the load resistance create the
dominant pole in the system. However, the internal
amplifier delay, pass transistor’s input capacitance,
and the stray capacitance at the feedback node create
additional poles in the system, and the output capacitor’s ESR generates a zero. For proper operation, use
the following equations to verify the linear regulator is
properly compensated:
1) First, determine the dominant pole set by the linear
regulator’s output capacitor and the load resistor:
I LOAD(MAX)_ LR
fPOLE _ LR =
2π × COUT _ LR × VOUT _ LR
The unity-gain crossover of the linear regulator is:
fCROSSOVER = AV_LR fPOLE_LR
2) The pole created by the internal amplifier delay is
approximately 1MHz:
fPOLE_AMP = 1MHz
3) Next, calculate the pole set by the transistor’s input
capacitance, the transistor’s input resistance, and
the base-to-emitter pullup resistor:
fPOLE _ IN =
1
2π × CIN × (RBE || RIN )
fPOLE _ FB =
1
2π × CFB × (RUPPER || RLOWER )
where CFB is the capacitance between FB_ and
AGND, RUPPER is the upper resistor of the linear regulator’s feedback divider, and RLOWER is the lower resistor of the divider.
5) Next, calculate the zero caused by the output
capacitor’s ESR:
fPOLE _ ESR =
1
2π × COUT _ LR × RESR
where RESR is the equivalent series resistance of
COUT_LR.
To ensure stability, choose COUT_LR large enough so
the crossover occurs well before the poles and zero
calculated in steps 2 to 5. The poles in steps 3 and 4
generally occur at several megahertz, and using
ceramic capacitors ensures the ESR zero occurs at
several megahertz as well. Placing the crossover below
500kHz is sufficient to avoid the amplifier-delay pole
and generally works well, unless unusual component
choices or extra capacitances move one of the other
poles or the zero below 1MHz.
Applications Information
Power Dissipation
where :
g
h
CIN = m , RIN = FE
2πfT
gm
gm is the transconductance of the pass transistor, and fT
is the transition frequency. Both parameters can be found
in the transistor’s data sheet. Because RBE is much
greater than RIN, the above equation can be simplified:
fPOLE _ IN =
1
2π × CIN × RIN
Substituting for CIN and RIN yields:
f
fPOLE _ IN = T
hFE
24
4) Next, calculate the pole set by the linear regulator’s
feedback resistance and the capacitance between
FB_ and AGND (including stray capacitance):
An IC’s maximum power dissipation depends on the
thermal resistance from the die to the ambient environment and the ambient temperature. The thermal resistance depends on the IC package, PCB copper area,
other thermal mass, and airflow.
The MAX8795A, with its exposed backside paddle soldered to 1in2 of PCB copper and a large internal ground
plane layer, can dissipate approximately 2.76W into
+70°C still air. More PCB copper, cooler ambient air,
and more airflow increase the possible dissipation, while
less copper or warmer air decreases the IC’s dissipation
capability. The major components of power dissipation
are the power dissipated in the step-up regulator and
the power dissipated by the operational amplifiers.
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
Operational Amplifier
The power dissipated in the operational amplifiers
depends on their output current, the output voltage,
and the supply voltage:
PDSOURCE = IOUT _(SOURCE) × (VSUP − VOUT _ )
PDSINK = IOUT _(SINK) × VOUT _
where IOUT_(SOURCE) is the output current sourced by
the operational amplifier, and IOUT_(SINK) is the output
current that the operational amplifier sinks.
In a typical case where the supply voltage is 13V and
the output voltage is 6V with an output source current
of 30mA, the power dissipated is 180mW.
PCB Layout and Grounding
Careful PCB layout is important for proper operation.
Use the following guidelines for good PCB layout:
• Minimize the area of high-current loops by placing
the inductor, the output diode, and the output
capacitors near the input capacitors and near the
LX and PGND pins. The high-current input loop
goes from the positive terminal of the input capacitor
to the inductor, to the IC’s LX pin, out of PGND, and
to the input capacitor’s negative terminal. The highcurrent output loop is from the positive terminal of
the input capacitor to the inductor, to the output
diode (D1), and to the positive terminal of the output
capacitors, reconnecting between the output capacitor and input capacitor ground terminals. Connect
these loop components with short, wide connections. Avoid using vias in the high-current paths. If
vias are unavoidable, use many vias in parallel to
reduce resistance and inductance.
• Create a power-ground island (PGND) consisting of
the input and output capacitor grounds, PGND pin,
and any charge-pump components. Connect all of
these together with short, wide traces or a small
ground plane. Maximizing the width of the powerground traces improves efficiency and reduces output voltage ripple and noise spikes. Create an
analog ground plane (AGND) consisting of the
AGND pin, all the feedback-divider ground connections, the operational-amplifier divider ground connections, the COMP and DEL capacitor ground
connections, and the device’s exposed backside
paddle. Connect the AGND and PGND islands by
connecting the PGND pin directly to the exposed
backside paddle. Make no other connections
between these separate ground planes.
• Place all feedback voltage-divider resistors within
5mm of their respective feedback pins. The divider’s
center trace should be kept short. Placing the resistors far away causes their FB traces to become
antennas that can pick up switching noise. Take
care to avoid running any feedback trace near LX or
the switching nodes in the charge pumps, or provide a ground shield.
• Place the IN pin and REF pin bypass capacitors as
close as possible to the device. The ground connection of the IN bypass capacitor should be connected
directly to the AGND pin with a wide trace.
• Minimize the length and maximize the width of the
traces between the output capacitors and the load
for best transient responses.
• Minimize the size of the LX node while keeping it
wide and short. Keep the LX node away from feedback nodes (FB, FBP, and FBN) and analog ground.
Use DC traces to shield if necessary.
Refer to the MAX8795A evaluation kit for an example of
proper PCB layout.
Chip Information
TRANSISTOR COUNT: 6595
PROCESS: BiCMOS
______________________________________________________________________________________
25
MAX8795A
Step-Up Regulator
The largest portions of power dissipation in the step-up
regulator are the internal MOSFET, the inductor, and the
output diode. If the step-up regulator has 90% efficiency,
approximately 3% to 5% of the power is lost in the internal
MOSFET, approximately 3% to 4% in the inductor, and
approximately 1% in the output diode. The remaining 1%
to 3% is distributed among the input and output capacitors
and the PCB traces. If the input power is about 5W, the
power lost in the internal MOSFET is approximately 150mW
to 250mW.
TFT-LCD DC-DC Converter with
Operational Amplifiers
COMP
FB
IN
LX
OUT5
NEG5
POS5
OUT4
TOP VIEW
24
23
22
21
20
19
18
17
FBP
25
16
NEG4
DRVP
26
15
POS4
FBN
27
14
SUP
DRVN
28
13
OUT3
DEL
29
12
POS3
CTL
30
11
BGND
DRN
31
10
POS2
COM
32
9
NEG2
FBP 25
16
NEG4
DRVP 26
15
POS4
FBN 27
14
SUP
DRVN 28
13
OUT3
6
7
8
OUT2
5
POS1
REF
4
NEG1
3
OUT1
2
PGND
1
AGND
MAX8795A
SRC
MAX8795A
Pin Configurations
FB
IN
LX
OUT5
NEG5
POS5
OUT4
TOP VIEW
COMP
THIN QFN
5mm x 5mm
24
23
22
21
20
19
18
17
MAX8795A
12
POS3
CTL 30
11
BGND
DRN 31
10
POS2
9
NEG2
DEL 29
COM 32
1
2
3
4
5
6
7
8
SRC
REF
AGND
PGND
OUT1
NEG1
POS1
OUT2
+
LQFP
7mm x 7mm
26
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a "+", "#", or
"-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND PATTERN
NO.
32 TQFN
T3255+3
21-0140
90-0025
32 LQFP
C32+2
21-0054
90-0111
______________________________________________________________________________________
27
MAX8795A
Package Information
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
Package Information (continued)
For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a "+", "#", or
"-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status.
28
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a "+", "#", or
"-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status.
______________________________________________________________________________________
29
MAX8795A
Package Information (continued)
MAX8795A
TFT-LCD DC-DC Converter with
Operational Amplifiers
Package Information (continued)
For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a "+", "#", or
"-" in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status.
30
______________________________________________________________________________________
TFT-LCD DC-DC Converter with
Operational Amplifiers
REVISION
NUMBER
REVISION
DATE
DESCRIPTION
PAGES
CHANGED
0
4/07
Initial release
1
6/07
Added LQFP package and G temperature grade versions
0
2
12/10
Added TQFN version
3
3/11
Added automotive-qualified part
1
4
6/11
Corrected automotive /V temperature range
1
5
5/12
Added automotive-qualified part
1
1, 2, 6–30
1–10, 27–30
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in
the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
31 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2012 Maxim Integrated Products
Maxim is a registered trademark of Maxim Integrated Products, Inc.
MAX8795A
Revision History
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