PD-97472
IR3871MPBF
SupIRBuck
TM
8A HIGHLY INTEGRATED
WIDE-INPUT VOLTAGE, SYNCHRONOUS BUCK REGULATOR
Features
Description
The IR3871 SupIRBuckTM is an easy-to-use,
fully integrated and highly efficient DC/DC
voltage regulator. The onboard constant on time
hysteretic controller and MOSFETs make
IR3871 a space-efficient solution that delivers up
to 8A of precisely controlled output voltage in
60°C ambient temperature applications without
airflow.
Input Voltage Range: 3V to 26V
Output Voltage Range: 0.5V to 12V
Continuous 8A Load Capability
Constant On-Time control
Excellent Efficiency at very low output current levels
Compensation Loop not required
Programmable switching frequency, soft start, and
over current protection
Power Good Output
Precision Voltage Reference (0.5V, +/-1%)
Pre-bias Start Up
Under/Over Voltage Fault Protection
Ultra small, low profile 5 x 6mm QFN Package
Applications
Programmable switching frequency, soft start,
and over current protection allows for a very
flexible solution suitable for many different
applications and an ideal choice for battery
powered applications.
Additional features include pre-bias startup, very
precise 0.5V reference, over/under voltage shut
down, power good output, and enable input with
voltage monitoring capability.
Notebook and desktop computers
Game consoles
Consumer electronics – STB, LCD, TV, printers
General purpose POL DC-DC converters
1
IR3871MPBF
ABSOLUTE MAXIMUM RATINGS
(Voltages referenced to GND unless otherwise specified)
•
VIN. FF …………………………………………..……. -0.3V to 30V
•
VCC, PGood, EN ………………….…....….…..….… -0.3V to 8.0V
•
Boot ……………………………………..……..…...…. -0.3V to 40V
•
PHASE ……………………………………………....... -0.3V to 30V(DC), -5V(100ns)
•
Boot to PHASE …..………………………………..….. -0.3V to 8V
•
ISET …………………………………………..….……. -0.3V to 30V
•
PGND to GND ……………...……………..………….. -0.3V to +0.3V
•
All other pins ……………...………………….……….. -0.3V to 3.9V
•
Storage Temperature Range ................................... -65°C To 150°C
•
Junction Temperature Range ................................... -40°C To 150°C
•
ESD Classification …………………………….……… JEDEC Class 1C
•
Moisture sensitivity level ……….……...…………..… JEDEC Level 2 @ 260°C (Note 2)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the
device. These are stress ratings only and functional operation of the device at these or any other
conditions beyond those indicated in the operational sections of the specifications are not implied.
PACKAGE INFORMATION
5mm x 6mm POWER QFN
θJA = 35 o C / W
θJ -PCB = 2 o C / W
ORDERING INFORMATION
PKG DESIG
PACKAGE
DESCRIPTION
PIN COUNT
PARTS PER
REEL
M
IR3871MTRPbF
17
4000
M
IR3871MTR1PbF
17
750
2
IR3871MPBF
Simplified Block Diagram
3
IR3871MPBF
Pin Description
NAME
NUMBER
NC
1
ISET
2
PGOOD
3
GND
4,17
I/O
LEVEL
DESCRIPTION
-----
No connection.
Connecting resistor to PHASE pin sets over current trip point.
5V
Power good open drain output – pull up with a resistor to 3.3V.
Reference
Bias return and signal reference.
FB
5
3.3V
Inverting input to PWM comparator, OVP / PGood sense.
SS
6
3.3V
Soft start/shutdown. This pin provides user programmable softstart function. Connect an external capacitor from this pin to
GND to set the startup time of the output voltage. The converter
can be shutdown by pulling this pin below 0.3V.
NC
7
3VCBP
8
NC
9
VCC
10
5V
Gate drive supply. A minimum of 1.0µF ceramic capacitor must
be connected from this pin to the power return (PGND).
PGND
11
Reference
Power return.
PHASE
12
VIN
Phase node (or switching node) of MOSFET half bridge.
VIN
13
VIN
Input voltage for the system.
BOOT
14
VIN +VCC
Bootstrapped gate drive supply – connect a capacitor to
PHASE.
FF
15
VIN
Input voltage feed forward – sets on-time with a resistor to VIN.
EN
16
5V
Enable pin to turn on and off the device. Use two external
resistors to set the turn on threshold (see Electrical
Specifications) for input voltage monitoring.
----3.3V
-----
No connection.
LDO output. A minimum of 1.0 µF ceramic capacitor is required
from 3VCBP to GND.
No connection.
4
IR3871MPBF
Recommended Operating Conditions
Min
Max
Unit
VIN
VCC
Symbol
Input Voltage
Supply Voltage
Definition
3
4.5
26*
7.5
V
VOUT
Output Voltage
0.5
12
IOUT
Output Current
0
8
A
Fs
Switching Frequency
N/A
1000
kHz
TJ
Junction Temperature
0
125
oC
* PHASE pin must not exceed 30V.
Electrical Specifications
Unless otherwise specified, these specification apply over VIN = 12V, VCC = 5V, 0oC ≤ TJ ≤ 125oC.
PARAMETER
BIAS SUPPLIES
VCC Turn-on Threshold
VCC Turn-off Threshold
VCC Threshold Hysteresis
VCC Operating Current
NOTE
TEST CONDITION
C3VCBP = 1µF
Output Current
CONTROL LOOP
Reference Accuracy, VREF
VREF
On-Time Accuracy
RFF = 180K, TJ = 65oC
1
PGOOD Delay Threshold
(VSS)
EN = HIGH
Measure at VPHASE
Falling VFB & Monitor PGOOD
1
Over Voltage Threshold
Over Voltage Hysteresis
MAX
UNIT
3.9
3.6
4.2
3.9
150
7.1
4.5
4.2
V
V
mV
mA
35
2
1
50
µA
µA
µA
3.3
3.5
V
8
mA
3.1
0.495
0.5
0.505
V
280
300
320
ns
8
-5
400
10
-2.4
12
0
ns
µA
mV
18
0.37
20
0.4
22
0.43
µA
V
1
FAULT PROTECTION
ISET Pin Output Current
Under Voltage Threshold
Under Voltage Hysteresis
TYP
RFF = 200K,
EN = HIGH, Fs = 300kHz
EN = LOW
EN = LOW
EN = LOW
VCC Shutdown Current
FF Shutdown Current
VIN Shutdown Current
INTERNAL LDO OUTPUT
LDO Output Voltage Range
Min Off Time
Soft-Start Current
Zero Current Threshold
MIN
Rising VFB
Rising VFB & Monitor PGOOD
1
Falling VFB
10
0.58
0.62
mV
0.66
V
10
mV
1
V
5
IR3871MPBF
Electrical Specifications (continued)
Unless otherwise specified, these specification apply over VIN = 12V, VCC = 5V, 0oC ≤ TJ ≤ 125oC.
PARAMETER
GATE DRIVE
Dead Time
NOTE
BOOTSTRAP PFET
Forward Voltage
UPPER MOSFET
Static Drain-to-Source OnResistance
LOWER MOSFET
Static Drain-to-Source OnResistance
LOGIC INPUT AND OUTPUT
EN High Logic Level
EN Low Logic Level
EN Input Current
PGOOD Pull Down
Resistance
1
TEST CONDITION
Monitor body diode conduction
on PHASE pin
MIN
TYP
5
MAX
UNIT
30
ns
I(BOOT) = 10mA
100
200
300
mV
VCC = 5V, ID = 7A, TJ = 25oC
14.5
20.8
25.5
mΩ
VCC = 5V, ID = 9A, TJ = 25oC
8
10
12.5
mΩ
2
-
11
25
0.6
V
V
µA
Ω
EN = 3.3V
50
Note 1: Guaranteed by design, not tested in production
Note 2: Upgrade to industrial/MSL2 level applies from date codes 1227 (marking explained on application
note AN1132 page 2). Products with prior date code of 1227 are qualified with MSL3 for Consumer
Market.
6
IR3871MPBF
TYPICAL OPERATING DATA
Tested with demoboard shown in Figure 7, VIN = 12V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25 oC, no airflow,
unless otherwise specified
100%
95%
EFF @ 8VIN
90%
75%
Efficiency
Efficiency
85%
EFF @ 19VIN
65%
55%
EFF @ 12VIN
45%
35%
0.01
80%
EFF @ 3.3VOUT, L = 3.3µH
70%
EFF @ 1.5VOUT, L = 2.2µH
60%
50%
0.1
1
EFF @ 1.05VOUT, L = 1.5µH
40%
0.01
10
0.1
Output Current (A)
Figure 1. Efficiency vs. Output Current for
VOUT = 1.05V
1200
300
5.0 Vout
4.0
3.0
2.0
1.0
1000
250
RFF (kOhm)
Frequency (kHz)
10
Figure 2. Efficiency vs. Output Current for
VIN = 12V
350
200
150
100
800
4.5
3.5
2.5
1.5
0.5
600
400
200
50
0
200
0
0
1
2
3
4
5
Output Current (A)
6
7
8
400
Figure 3. Switching Frequency vs. Output
Current
600
800
Frequency (kHz)
1000
Figure 4. Frequency vs. RFF
1.052
1.051
Output Voltage (V)
1.052
Output Voltage (V)
1
Output Current (A)
VOUT @ 12VIN
VOUT @ 19VIN
1.050
1.049
VOUT @ 8VIN
1.048
1.051
1.050
1.049
1.048
0
1
2
3
4
5
6
7
Output Current (A)
Figure 5. Output Voltage Regulation vs.
Output Current
8
6
8
10
12
14
16
18
20
Input Voltage (V)
Figure 6. Output Voltage Regulation vs.
Input Voltage at IOUT = 8A
7
IR3871MPBF
TYPICAL APPLICATION CIRCUIT
Demoboard Schematic: VOUT = 1.05V, Fs = 300kHz
+3 .3V
V CC
+V i ns
R1
op en
T P1
V INS
V IN
T P2
V IN
T P3
FCCM
C1
1u F
EN
+V i n1s
T P20
+V i n1s
R3
20 0K
-Vi ns
ISE T
1
13
V IN
1
C11
op en
C12
0.1uF
T P10
P GND
T P9
+V out1s
T P24
+V sws
5
4
4
C10
47 uF
T P12
V SWS
C15
op en
C16
op en
C17
op en
C18
op en
C19
op en
C26
op en
C27
op en
R12
op en
R8
2.55K
R9
0
R13
op en
10
+V out2s -Vo ut2 s
T P18
V OLT A GE S E NS E
5
-Vd d2s
-Vo ut1 s
9
4
+V i ns
1
+V i n1s
R11
op en
T P19
FB
+V out1s -Vo ut1 s
-Vd d1s
C24
op en
R7
2.80K
V out
C14
op en
+V dd2 s -Vd d2s
-Vd d1s
-Vi ns
C22
op en
C23
op en
+V dd2 s 3
T P25
-Vi n1s
+V dd1 s
8
T P15
-Vo ut1 s
-Vo ut1 s
C21
1u F
V CC
C9
33 0uF
7
-Vo ut1 s
+V dd2 s
C25
1u F
T P17
P GND
C8
op en
T P21
-Vs ws
T P26
A GND
T P16
V CC
C7
op en
3
14
15
FF
2
+3 .3V
-Vd d2s
T P22
+V sws
SS
NC1
C6
op en
C13
op en
12
FB
8
7
P HA SE
3
C20
0.1uF
IR3871
P GND
6
SS
P GOOD
GND1
T P7
V OUT
5 T P8
V OUTS
11
5
R6
op en
V CC
4
FB
T P14
+3 .3V
T P23
+V sws
V OUT
2
ISE T
NC2
3
P GOOD
V SW
U1
IR3 871
NC
10
2
B OOT
16
17
GND
1
R5
10 K
3V 3B P
C5
op en
T P13
SS
EN
+3 .3V
T P11
P GOOD
T P6
P GNDS
L1
1.5uH
9
1
2
V SW
T P5
P GND
C4
0.22u F
R4
10 .5K
6
4
3
FCCM
S W1
E N / FCCM
+ C3
68 uF
+V dd1 s -Vd d1s
T P4
EN
C2
22 uF
+V dd1 s 2
R2
10 K
R14
op en
R10
op en
Figure 7. Typical Application Circuit for VOUT = 1.05V, Fs = 300kHz
Demoboard Bill of Materials
Quantity
1
3
1
1
1
1
2
1
2
1
1
1
1
1
1
1
Reference
C4
C1, C21, C25
C2
C3
C9
C10
C12, C20
L1
R2, R5
R9
R3
R4
R7
R8
SW1
U1
Value
0.22uF
1uF
22uF
68uF
330uF
47uF
0.1uF
1.5uH
10K
0
200K
10.5K
2.8K
2.55K
SPST
IR3871
Description
CAP,CER,0.22uF,50V,10%,X7R,0603
CAP,CER,1.0uF,25V,X7R,0603
CAP,22uF,25V,CERAMIC,X5R,1210
CAP,68uF,25V,ELECT,FK,SMD
POSCAP, 330uF, 2.5V, SMD
CAP,CER,47uF,6.3V,X5R,0805
CAP,CER,0.1uF,50V,10%,X7R,0603
INDUCTOR, 1.5uH, 11A, 6.7mOhm,SMD
RES,10.0K,OHM,1/10W,1%,0603,SMD
RES,0.0,OHM,1/10W,1%,0603,SMD
RES,200K,OHM,1/10W,1%,0603,SMD
RES,10.5K,OHM,1/10W,1%,0603,SMD
RES,2.8K,OHM,1/10W,1%,0603,SMD
RES,2.55K,OHM,1/10W,1%,0603,SMD
SWITCH, DIP, SPST, SMT
5mm x 6mm QFN
Manufacturer
Murata Electronics
Murata Electronics
Panasonic
Panasonic
Sanyo
TDK
TDK
CYNTEC
Vishay/Dale
Vishay/Dale
Vishay/Dale
Vishay/Dale
Vishay/Dale
Vishay/Dale
C&K Components
IR
Part-Number
GRM188R71H224KA64D
GRM188R71E105KA12D
ECJ-4YB1E226M
EEV-FK1E680P
2R5TPE330M9
C2012X5R0J476M
C1608X7R1H104K
PCMB065T-1R5MS
CRCW060310K0FKEA
CRCW06030000Z0EAHP
CRCW0603200KFKEA
CRCW060310K5FKEA
CRCW06032K80FKEA
CRCW06032K55FKEA
SD02H0SK
IR3871MPBF
8
IR3871MPBF
TYPICAL OPERATING DATA
Tested with demoboard shown in Figure 7, VIN = 12V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, T A = 25oC, no airflow,
unless otherwise specified
EN
EN
PGOOD
PGOOD
SS
SS
VOUT
VOUT
5V/div 5V/div 1V/div 500mV/div
5ms/div
5V/div 5V/div 1V/div 500mV/div
200µs/div
Figure 9: Shutdown
Figure 8: Startup
VOUT
VOUT
PHASE
PHASE
iL
iL
20mV/div 5V/div 2A/div
10µs/div
Figure 10: DCM (IOUT = 0.1A)
20mV/div 5V/div 5A/div
2µs/div
Figure 11: CCM (IOUT = 6A)
PGOOD
PGOOD
SS
FB
VOUT
VOUT
IOUT
iL
5V/div 1V/div 1V/div 10A/div
1ms/div
Figure 12: Over Current Protection
(tested by shorting VOUT to PGND)
5V/div 1V/div 500mV/div 2A/div
50µs/div
Figure 13: Over Voltage Protection
(tested by shorting FB to VOUT)
9
IR3871MPBF
TYPICAL OPERATING DATA
Tested with demoboard shown in Figure 7, VIN = 12V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, T A = 25oC, no airflow,
unless otherwise specified
VOUT
VOUT
PHASE
PHASE
iL
iL
50mV/div 10V/div 2A/div
20µs/div
50mV/div 10V/div 2A/div
20µs/div
Figure 14: Load Transient 0-4A
Figure 15: Load Transient 4-8A
Figure 16: Thermal Image at VIN = 12V, IOUT =
8A (IR3871: 66oC, Inductor: 58oC, PCB: 40oC)
Figure 17: Thermal Image at VIN = 19V, IOUT =
8A (IR3871: 71oC, Inductor: 59oC, PCB: 44oC)
10
IR3871MPBF
CIRCUIT DESCRIPTION
PWM COMPARATOR
The PWM comparator initiates a SET signal
(PWM pulse) when the FB pin falls below the
reference (Vref) or the soft start (SS) voltage.
ON-TIME GENERATOR
The PWM on-time duration is programmed with
an external resistor (RFF) from the input supply
(VIN) to the FF pin. The simplified calculation for
RFF is shown in equation 1. The FF pin is held to
an internal reference after EN goes HIGH. A copy
of the current in RFF charges a timing capacitor,
which sets the on-time duration, as shown in
equation 2.
RFF
VOUT
(1)
1V 20 pF FSW
TON
RFF 1V 20 pF
(2)
VIN
CONTROL LOGIC
The control logic monitors input power sources,
sequences the converter through the soft-start
and protective modes, and initiates an internal
RUN signal when all conditions are met.
PGOOD
The PGOOD pin is open drain and it needs to be
externally pulled high. High state indicates that
output is in regulation. The PGOOD logic monitors
EN_DELAY, SS_DELAY, and under/over voltage
fault signals. PGOOD is released only when
EN_DELAY and SS_DELAY = HIGH and output
voltage is within the OV and UV thresholds.
PRE-BIAS STARTUP
IR3871 is able to start up into pre-charged output,
which prevents oscillation and disturbances of the
output voltage.
With constant on-time control, the output voltage
is compared with the soft start voltage (SS) or
Vref, depending on which one is lower, and will
not start switching unless the output voltage drops
below the reference. This scheme prevents
discharge of a pre-biased output voltage.
SHUTDOWN
The IR3871 will shutdown if VCC is below its
UVLO limit. The IR3871 can be shutdown by
pulling the EN pin below its lower threshold.
Alternatively, the output can be shutdown by
pulling the soft start pin below 0.3V.
VCC and 3VCBP pins are continuously monitored,
and the IR3871 will be disabled if the voltage of
either pin drops below the falling thresholds.
EN_DELAY will become HIGH when VCC and
3VCBP are in the normal operating range and the
EN pin = HIGH.
SOFT START
With EN = HIGH, an internal 10µA current source
charges the external capacitor (CSS) on the SS pin
to set the output voltage slew rate during the soft
start interval. The soft start time (tSS) can be
calculated from equation 3.
t SS
CSS 0.5V
(3)
10A
The feedback voltage tracks the SS pin until SS
reaches the 0.5V reference voltage (Vref), then
feedback is regulated to Vref. CSS will continue to
be charged, and when SS pin reaches VSS (see
Electrical Specification), SS_DELAY goes HIGH.
With EN_DELAY = LOW, the capacitor voltage
and SS pin is held to the FB pin voltage. A normal
startup sequence is shown in Figure 18.
Figure 18. Normal Startup
11
IR3871MPBF
CIRCUIT DESCRIPTION
UNDER/OVER VOLTAGE MONITOR
The IR3871 monitors the voltage at the FB node
through a 350ns filter. If the FB voltage is below
the under voltage threshold, UV# is set to LOW
holding PGOOD to be LOW. If the FB voltage is
above the over voltage threshold, OV# is set to
LOW, the shutdown signal (SD) is set to HIGH,
MOSFET gates are turned off, and PGOOD
signal is pulled low. Toggling VCC or EN will
allow the next start up. Figure 19 shows
PGOOD status change when UV/OV is
detected. The over voltage and under voltage
thresholds can be found in the Electrical
Specification section.
OVER CURRENT MONITOR
The over-current circuitry monitors the output
current during each switching cycle. The voltage
across the lower MOSFET, VPHASE, is
monitored for over current and zero crossing. The
OCP circuit evaluates VPHASE for an over
current condition typically 270ns after the lower
MOSFET is gated on. This delay functions to filter
out switching noise. The minimum lower gate
interval allows time to sample VPHASE.
The over current trip point is programmed with a
resistor from the ISET pin to PHASE pin, as
shown in equation 4, where Tj is the junction
temperature of Q2 at operation conditions, and 0.4
is the temperature coefficient (~4000 ppm/C) of
Q2 RDSON. When over current is detected, the
output gates are tri-state and SS voltage is pulled
to 0V. This initiates a new soft start cycle. If there
is a total of four OC events, the IR3871 will
disable switching, as shown in Figure 20. Toggling
VCC or EN will allow the next start up.
RSET
RDSON IOC
20 A
(1
Tj 25
100
0.4)
(4)
Figure 19(a). Under/Over Voltage Monitor
* typical filter delay
Figure 20. Over Current Protection
Figure 19(b). Over Voltage Protection
12
IR3871MPBF
CIRCUIT DESCRIPTION
GATE DRIVE LOGIC
The gate drive logic features adaptive dead
time, diode emulation, and a minimum lower
gate interval.
STABILITY CONSIDERATIONS
Constant-on-time control is a fast , ripple based
control scheme. Unstable operation can occur
if certain conditions are not met. The system
instability is usually caused by:
An adaptive dead time prevents the
simultaneous conduction of the upper and lower
MOSFETs. The lower gate voltage (LGATE)
must be below approximately 1V after PWM
goes HIGH before the upper MOSFET can be
gated on. Also, the upper gate voltage
(UGATE), the difference voltage between
UGATE and PHASE, must be below
approximately 1V after PWM goes LOW before
the lower MOSFET can be gated on.
• Switching noise coupled to FB input. This
causes the PWM comparator to trigger
prematurely after the 400ns minimum Q2 ontime. It will result in double or multiple pulses
every switching cycle instead of the expected
single pulse. Double pulsing can causes
higher output voltage ripple, but in most
application it will not affect operation. This can
usually be prevented by careful layout of the
ground plane and the FB sensing trace.
The control MOSFET is gated on after the
adaptive delay for PWM = HIGH and the
synchronous MOSFET is gated on after the
adaptive delay for PWM = LOW. The lower
MOSFET is driven ‘off’ when the signal
ZCROSS indicates that the inductor current has
reversed as detected by the PHASE voltage
crossing the zero current threshold. The
synchronous MOSFET stays ‘off’ until the next
PWM falling edge. When the lower peak of
inductor current is above zero, a forced
continuous current condition is selected. The
control MOSFET is gated on after the adaptive
delay for PWM = HIGH, and the synchronous
MOSFET is gated on after the adaptive delay for
PWM = LOW.
• Steady state ripple on FB pin being too small.
The PWM comparator in IR3871 requires
minimum 7mVp-p ripple voltage to operate
stably. Not enough ripple will result in similar
double pulsing issue described above. Solving
this may require using output capacitors with
higher ESR.
The synchronous MOSFET gate is driven on for
a minimum duration. This minimum duration
allows time to recharge the bootstrap capacitor
and allows the current monitor to sample the
PHASE voltage.
• For applications with all ceramic output
capacitors, the ESR is usually too small to
meet the stability criteria. In these
applications, external slope compensation is
necessary to make the loop stable. The ramp
injection circuit, composed of R6, C13, and
C14, shown in Figure 7 is required. The
inductor current ripple sensed by R6 and C13
is AC coupled to the FB pin through C14. C14
is usually chosen between 1 to 10nF, and C13
between 10 to 100nF. R6 should then be
chosen such that L/DCR = C13*R6.
• ESR loop instability. The stability criteria of
constant on-time is: ESR*Cout>Ton/2. If ESR
is too small that this criteria is violated then
sub-harmonic oscillation will occur. This is
similar to the instability problem of peakcurrent-mode control with D>0.5. Increasing
ESR is the most effective way to stabilize the
system, but the price paid is the larger output
voltage ripple.
13
IR3871MPBF
COMPONENT SELECTION
Selection of components for the converter is an
iterative process which involves meeting the
specifications
and
trade-offs
between
performance and cost. The following sections will
guide one through the process.
INDUCTOR SELECTION
Inductor selection involves meeting the steady
state output ripple requirement, minimizing the
switching loss of upper MOSFETs, meeting
transient response specifications and minimizing
the output capacitance. The output voltage
includes a DC voltage and a small AC ripple
component due to the low pass filter which has
incomplete
attenuation
of
the
switching
harmonics. Neglecting the inductance in series
with the output capacitor, the magnitude of the AC
voltage ripple is determined by the total inductor
ripple current flowing through the total equivalent
series resistance (ESR) of the output capacitor
bank.
INPUT CAPACITOR SELECTION
The main function of the input capacitor bank is
to provide the input ripple current and fast slew
rate current during the load current step up. The
input capacitor bank must have adequate ripple
current carrying capability to handle the total
RMS current. Figure 21 shows a typical input
current. Equation 6 shows the RMS input
current. The RMS input current contains the DC
load current and the inductor ripple current. As
shown in equation 5, the inductor ripple current
is unrelated to the load current. The maximum
RMS input current occurs at the maximum
output current. The maximum power dissipation
in the input capacitor equals the square of the
maximum RMS input current times the input
capacitor’s total ESR.
Input Current
IOUT
ΔI
ΔI
TON VIN VOUT
(5)
2L
TS
Figure 21. Typical Input Current Waveform.
One can use equation 5 to find the required
inductance. ΔI is defined as shown in Figure 21.
The main advantage of small inductance is
increased inductor current slew rate during a load
transient, which leads to a smaller output
capacitance requirement as discussed in the
Output Capacitor Selection section. The draw
back of using smaller inductances is increased
switching power loss in upper MOSFET, which
reduces the system efficiency and increases the
thermal dissipation.
Ts
IIN_RMS
1
f 2 t dt
Ts 0
1 ΔI
IOUT Ton Fs 1
3 IOUT
2
(6)
The voltage rating of the input capacitor needs
to be greater than the maximum input voltage
because of high frequency ringing at the phase
node. The typical percentage is 25%.
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IR3871MPBF
COMPONENT SELECTION
OUTPUT CAPACITOR SELECTION
Selection of the output capacitor requires meeting
voltage overshoot requirements during load
removal, and meeting steady state output ripple
voltage requirements. The output capacitor is the
most expensive converter component and
increases the overall system cost. The output
capacitor decoupling in the converter typically
includes the low frequency capacitor, such as
Specialty Polymer Aluminum, and mid frequency
ceramic capacitors.
The first purpose of output capacitors is to provide
current when the load demand exceeds the
inductor current, as shown in Figure 22. Equation
7 shows the charge requirement for a certain load.
The advantage provided by the IR3871 at a load
step is to reduce the delay compared to a fixed
frequency control method (in microseconds or (1D)*Ts). If the load increases right after the PWM
signal goes low, the longest delay will be equal to
the minimum lower gate on as shown in the
Electrical Specification table. The IR3871 also
reduces the inductor current slew time, the time it
takes for the inductor current to reach equality
with the output current, by increasing the
switching frequency up to 2.5MHz. The result
reduces the recovery time.
Load
Current
I STEP
Output
Charge
Inductor
Slew
Rate
Figure 22. Charge Requirement during Load Step
Q C V 0.5 Istep t
COUT
(7a)
1 L Istep2
VDROP 2 VIN VOUT
1
COUT
L ISTEP 2
VOS 2 VOUT 2
(8)
VOS
VOUT
VL
VDROP
IOUT
VESR
ISTEP
Figure 23. Typical Output Voltage Response
Waveform.
BOOT CAPACITOR SELECTION
The boot capacitor starts the cycle fully charged
to a voltage of VB(0). Cg equals 0.65nF in
IR3871. Choose a sufficiently small ΔV such
that VB(0)-ΔV exceeds the maximum gate
threshold voltage to turn on the high side
MOSFET.
V (0)
CBOOT Cg B 1 (9)
ΔV
t
Δt
The second purpose of the output capacitor is to
minimize the overshoot of the output voltage
when the load decreases as shown in Figure
23. By using the law of energy before and after
the load removal, equation 8 shows the output
capacitance requirement for a load step.
(7b)
The output voltage drop, VDROP, initially depends
on the characteristic of the output capacitor.
VDROP is the sum of the equivalent series
inductance (ESL) of the output capacitor times the
rate of change of the output current and the ESR
times the change of the output current. VESR is
usually much greater than VESL. The IR3871
requires a total ESR such that the ripple voltage at
the FB pin is greater than 7mV.
Choose a boot capacitor value larger than the
calculated CBOOT in equation 9. Equation 9 is
based on charge balance at CCM operation.
Usually the boot capacitor will be discharged to
a much lower voltage when the circuit is
operating in DCM mode at light load, due to
much longer Q2 off time and the bias current
drawn by the IC. Boot capacitance needs to be
increased if insufficient turn-on of Q1 is
observed at light load, typically larger than
0.1µF is needed. The voltage rating of this part
needs to be larger than VB(0) plus the desired
derating voltage. Its ESR and ESL needs to be
low in order to allow it to deliver the large
current and di/dt’s which drive MOSFETs most
efficiently. In support of these requirements a
ceramic capacitor should be chosen.
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IR3871MPBF
DESIGN EXAMPLE
Design Criteria:
Input Voltage, VIN, = 6V to 21V
Output Voltage, VOUT = 1.25V
Switching Frequency, FS = 400KHz
Inductor Ripple Current, 2ΔI = 3A
Maximum Output Current, IOUT = 6A
Over Current Trip, IOC = 9A
Overshoot Allowance, VOS = VOUT + 50mV
Undershoot Allowance, VDROP = 50mV
Find RFF :
RFF
2
Find RSET :
6.3 k
The RDSON of the lower MOSFET could be
expected to increase by a factor of 1.4 over
temperature.
Therefore, pick a 6.49kΩ, 1%
standard resistor.
Find a resistive voltage divider for VOUT = 1.25V:
VFB
R2 = 1.33kΩ, R1 = 1.96 kΩ, both 1% standard
resistors.
Choose the soft start capacitor:
Once the soft start time has chosen, such as
1000us to reach to the reference voltage, a 22nF
for CSS is used to meet 1000µs.
inductor
to
VOUT VIN VOUT
VIN 2ΔI Fs
1.25V 21V - 1.25V
21V 3 A 400kHz
1.0H
L
A
Panasonic
10µF
(ECJ3YB1E106M)
accommodates 6 Arms of ripple current at
300KHz. Due to the chemistry of multilayer
ceramic capacitors, the capacitance varies over
temperature and operating voltage, both AC and
DC. One 10µF capacitor is recommended. In a
practical solution, one 1µF capacitor is required
along with 10µF. The purpose of the 1µF
capacitor is to suppress the switching noise and
deliver high frequency current.
Choose an output capacitor:
R2
VOUT 0.5V
R2 R1
Choose an
specification:
1.25V 21V - 1.25V
1.75 A
2 21V 0.82H 400kHz
1.25V
1 1.75 A
IIN_RMS 6 A
1
1.5 A
21V
3 6A
Pick a standard value 158 kΩ, 1% resistor.
1.4 10m 9 A
20A
ΔI
Choose an input capacitor:
1.25V
156 k
1V 20 pF 400kHz
RSET
Choose an inductor with the lowest DCR and
AC power loss as possible to increase the
overall system efficiency. For instance, choose
an FDU0650-R82M manufactured by TOKO.
The inductance of this part is 820nH and has
4.2mΩ DCR. Ripple current needs to be
recalculated using the chosen inductor.
meet
the
design
To meet the undershoot specification, select a
set of output capacitors which has an equivalent
ESR of 10mΩ (50mV/5A). To meet the
overshoot specification, equation 8 will be used
to calculate the minimum output capacitance.
As a result, 160µF will be needed for 5A load
remover. Combine those two requirements, one
can choose a set of output capacitors from
manufactures such as SP-Cap (Specialty
Polymer Capacitor) from Panasonic or
POSCAP
from
Sanyo.
A
150µF
(EEFSL0D151R)
from
Panasonic
is
recommended. This capacitor has 9mΩ ESR
which leaves margin for the voltage drop of the
ESL during load step up. The typical ESL for
this capacitor is around 2nH.
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IR3871MPBF
LAYOUT RECOMMENDATION
Bypass Capacitor:
One 1µF high quality ceramic capacitor should be
placed as near VCC pin as possible. The other
end of capacitor can be connected to a via or
connected directly to GND plane. Use a GND
plane instead of a thin trace to the GND pin
because a thin trace have too much impedance.
Boot Circuit:
CBOOT should be placed near the BOOT and
PHASE pins to reduce the impedance when the
upper MOSFET turns on.
Power Stage:
Figure 24 shows the current paths and their
directions for the on and off periods. The on time
path has low average DC current and high AC
current. Therefore, it is recommended to place the
input ceramic capacitor, upper, and lower
MOSFET in a tight loop as shown in Figure 24.
VIN
ON
Q1
VOUT
IR3871
OFF
Q2
CIN
COUT
Figure 24. Current Path of Power Stage
The purpose of the tight loop from the input
ceramic capacitor is to suppress the high
frequency (10MHz range) switching noise and
reduce Electromagnetic Interference (EMI). If this
path has high inductance, the circuit will cause
voltage spikes and ringing, and increase the
switching loss. The off time path has low AC and
high average DC current. Therefore, it should be
laid out with a tight loop and wide trace at both
ends of the inductor. Lowering the loop resistance
reduces the power loss. The typical resistance
value of 1-ounce copper thickness is 0.5mΩ per
square inch.
17
IR3871MPBF
PCB Metal and Components Placement
Lead lands (the 13 IC pins) width should be equal to nominal part lead width. The minimum lead to
lead spacing should be ≥ 0.2mm to minimize shorting.
Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension. The
outboard extension ensures a large toe fillet that can be easily inspected.
Pad lands (the 4 big pads) length and width should be equal to maximum part pad length and width.
However, the minimum metal to metal spacing should be no less than; 0.17mm for 2 oz. Copper or
no less than 0.1mm for 1 oz. Copper or no less than 0.23mm for 3 oz. Copper.
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IR3871MPBF
Solder Resist
It is recommended that the lead lands are Non Solder Mask Defined (NSMD). The solder resist
should be pulled away from the metal lead lands by a minimum of 0.025mm to ensure NSMD pads.
The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist
onto the copper of 0.05mm to accommodate solder resist misalignment.
Ensure that the solder resist in between the lead lands and the pad land is ≥ 0.15mm due to the
high aspect ratio of the solder resist strip separating the lead lands from the pad land.
19
IR3871MPBF
Stencil Design
The Stencil apertures for the lead lands should be approximately 80% of the area of the lead lads.
Reducing the amount of solder deposited will minimize the occurrences of lead shorts. If too much
solder is deposited on the center pad the part will float and the lead lands will open.
The maximum length and width of the land pad stencil aperture should be equal to the solder resist
opening minus an annular 0.2mm pull back in order to decrease the risk of shorting the center land
to the lead lands when the part is pushed into the solder paste.
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IR3871MPBF
DIM
A
A1
b
b1
c
D
E
e
e1
e2
MILIMITERS
MIN
MAX
0.8
1
0
0.05
0.375
0.475
0.25
0.35
0.203 REF.
5.000 BASIC
6.000 BASIC
1.033 BASIC
0.650 BASIC
0.852 BASIC
INCHES
MIN
MAX
0.0315
0.0394
0
0.002
0.1477
0.1871
0.0098
0.1379
0.008 REF.
1.970 BASIC
2.364 BASIC
0.0407 BASIC
0.0256 BASIC
0.0259 BASIC
DIM
L
M
N
O
P
Q
R
S
t1, t2, t3
t4
t5
MILIMITERS
MIN
MAX
0.35
0.45
2.441
2.541
0.703
0.803
2.079
2.179
3.242
3.342
1.265
1.365
2.644
2.744
1.5
1.6
0.401 BASIC
1.153 BASIC
0.727 BASIC
INCHES
MIN
MAX
0.0138
0.0177
0.0962
0.1001
0.0277
0.0314
0.0819
0.0858
0.1276
0.1316
0.0498
0.05374
0.1042
0.1081
0.0591
0.063
0.016 BACIS
0.045 BASIC
0.0286 BASIC
IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105
TAC Fax: (310) 252-7903
This product has been designed and qualified for the Industrial Market (Note 2)
Visit us at www.irf.com for sales contact information
Data and specifications subject to change without notice. 03/12
21
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