IR3876MTRPBF

PD-97763 IR3876MPBF SupIRBuck TM 12A HIGHLY INTEGRATED WIDE-INPUT VOLTAGE, SYNCHRONOUS BUCK REGULATOR Features Description        The IR3876 SupIRBuckTM is an easy-to-use, fully integrated and highly efficient DC/DC voltage regulator. The onboard constant on time hysteretic controller and MOSFETs make IR3876 a space-efficient solution that delivers up to 12A of precisely controlled output voltage in o 60 C ambient temperature applications without airflow.      Input Voltage Range: 3V to 21V Output Voltage Range: 0.5V to 12V Continuous 12A Load Capability Constant On-Time control Excellent Efficiency at very low output current levels Compensation Loop not Required Programmable switching frequency, soft start, and over current protection Power Good Output Precision Voltage Reference (0.5V, +/-1%) Pre-bias Start Up Under/Over Voltage Fault Protection Ultra small, low profile 5mm x 6mm QFN Package Applications Programmable switching frequency, soft start, and over current protection allows for a very flexible solution suitable for many different applications and an ideal choice for battery powered applications.     Additional features include pre-bias startup, very precise 0.5V reference, over/under voltage shut down, power good output, and enable input with voltage monitoring capability. Notebook and desktop computers Game consoles Consumer electronics – STB, LCD, TV, printers General purpose POL DC-DC converters V5 VIN (3V-21V) (4.5V-7.5V) RFF VCC FF VIN BOOT 3VCBP C BOOT VOUT (0.5V-12V) L CLDO PHASE EN IR3876 C0 RISET PGND PGOOD SS CSS ISET R1 FB GND R2 1 IR3876MPBF ABSOLUTE MAXIMUM RATINGS (Voltages referenced to GND unless otherwise specified) • VIN. FF .………………………………………………. -0.3V to 25V • VCC, PGood, EN ………………….…....……..….… -0.3V to 8.0V • Boot ……………………………………..………….…. -0.3V to 33V • PHASE ……………………………………………....... -0.3V to 25V(DC), -5V(100ns) • Boot to PHASE …..…………………………….…….. -0.3V to 8V • ISET …..…………………………………………..……. -0.3V to 30V • PGND to GND ……………...……………………….... -0.3V to +0.3V • All other pins ……………...……………………….….. -0.3V to 3.9V • Storage Temperature Range .................................... -65°C To 150°C • Junction Temperature Range ................................... -10°C To 150°C • ESD Classification …………………………….……… JEDEC Class 1C • Moisture sensitivity level ..……………...…………….. JEDEC Level 2 @ 260°C (Note 2) Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications are not implied. PACKAGE INFORMATION 5mm x 6mm POWER QFN PHASE 12 VIN 13 11 PGND θJA = 35 o C / W θJ -PCB = 2 o C / W BOOT 14 17 GND FF 15 EN 16 10 VCC 9 NC 8 3VCBP 1 2 3 4 5 NC ISET PGOOD GND FB 6 SS 7 NC ORDERING INFORMATION PKG DESIG PACKAGE DESCRIPTION PIN COUNT PARTS PER REEL M IR3876MTRPbF 17 4000 M IR3876MTR1PbF 17 750 2 IR3876MPBF Block Diagram 3VCBP VCC VCC VBG FF VCC + SS + + - FB SET PWM ON-TIME + x0.8 GND VREF x1.2 - VIN Run PWM COMP GND GND BOOT 3VCBP LDO GATE DRIVE LOGIC UV# Run SOFT START FF 3VCBP VCC Run PHASE VCC SSDelay CONTROL LOGIC POR Zcross EN PGND DCM OC# PGOOD OVER CURRENT ISET 3 IR3876MPBF Pin Description NAME NUMBER NC 1 ISET 2 PGOOD 3 GND 4,17 I/O LEVEL ----- DESCRIPTION No connection Connecting resistor to PHASE pin sets over current trip point 5V Power good – pull up to 3.3V Reference Bias return and signal reference FB 5 3.3V Inverting input to PWM comparator, OVP / PGood sense SS 6 3.3V Set soft start slew-rate with a capacitor to GND NC 7 3VCBP 8 NC 9 VCC 10 5V Gate drive supply PGND 11 Reference Power return PHASE 12 VIN Phase node (or switching node) of MOSFET half bridge VIN 13 VIN Input voltage for the system. BOOT 14 VIN +VCC Bootstrapped gate drive supply – connect a capacitor to PHASE FF 15 VIN Input voltage feed forward – sets on-time with a resistor to VIN EN 16 5V Enable ----3.3V ----- No connection LDO output. A minimum of 1.0 µF ceramic capacitor is required No connection 4 IR3876MPBF Recommended Operating Conditions Min Max Unit VIN VCC Symbol Input Voltage Supply Voltage Definition 3 4.5 21* 7.5 V VOUT Output Voltage 0.5 12 IOUT Output Current 0 12 A Fs Switching Frequency N/A 1000 kHz TJ Junction Temperature 0 125 oC * Note: PHASE pin must not exceed 25V. Electrical Specifications Unless otherwise specified, these specification apply over VIN = 12V, VCC = 5V, 0oC ≤ TJ ≤ 125oC. PARAMETER BIAS SUPPLIES VCC Turn-on Threshold VCC Turn-off Threshold VCC Threshold Hysteresis VCC Operating Current NOTE TEST CONDITION CLDO = 1µF Output Current CONTROL LOOP Reference Accuracy, VREF VREF On-Time Accuracy RFF = 180K, TJ = 65oC 1 PGOOD Delay Threshold (VSS) EN = HIGH Measure at VPHASE Falling VFB & Monitor PGOOD 1 Over Voltage Threshold Over Voltage Hysteresis MAX UNIT 3.9 3.6 4.2 3.9 150 9.2 4.5 4.2 V V mV mA 35 2 1 50 µA µA µA 3.3 3.5 V 8 mA 3.1 0.495 0.5 0.505 V 280 300 320 ns 8 -5 400 10 -2.4 12 0 ns µA mV 18 0.37 20 0.4 22 0.43 µA V 1 FAULT PROTECTION ISET Pin Output Current Under Voltage Threshold Under Voltage Hysteresis TYP RFF = 200K, EN = HIGH, Fs = 300kHz EN = LOW EN = LOW EN = LOW VCC Shutdown Current FF Shutdown Current VIN Shutdown Current INTERNAL LDO OUTPUT LDO Output Voltage Range Min Off Time Soft-Start Current Zero Current Threshold MIN Rising VFB Rising VFB & Monitor PGOOD 1 Falling VFB 10 0.58 0.62 mV 0.66 V 10 mV 1 V 5 IR3876MPBF Electrical Specifications (continued) Unless otherwise specified, these specification apply over VIN = 12V, VCC = 5V, 0oC ≤ TJ ≤ 125oC. PARAMETER GATE DRIVE Dead Time NOTE BOOTSTRAP PFET Forward Voltage UPPER MOSFET Static Drain-to-Source OnResistance LOWER MOSFET Static Drain-to-Source OnResistance LOGIC INPUT AND OUTPUT EN High Logic Level EN Low Logic Level EN Input Current PGOOD Pull Down Resistance 1 TEST CONDITION Monitor body diode conduction on PHASE pin I(BOOT) = 10mA MIN TYP 5 MAX UNIT 30 ns 100 200 300 mV VCC = 5V, ID = 12A, TJ = 25oC 7 12 16 mΩ VCC = 5V, ID = 12A, TJ = 25oC 4 5.3 7 mΩ 2 - 11 25 0.6 V V µA Ω EN = 3.3V 50 Note 1: Guaranteed by design, not tested in production Note 2: Upgrade to industrial/MSL2 level applies from date codes 1227 (marking explained on application note AN1132 page 2). Products with prior date code of 1227 are qualified with MSL3 for Consumer Market. 6 IR3876MPBF TYPICAL OPERATING DATA 95% 100% 85% 90% 75% Efficiency Efficiency Tested with demoboard shown in Figure 7, VIN = 12.6V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25 oC, no airflow, unless otherwise specified 16VIN 65% 12.6VIN 55% 7VIN 45% 80% 1.05VOUT; L = 1.2uH, 2.9mΩ 70% 1.5VOUT; L = 1.2uH, 2.9mΩ 60% 3.3VOUT; L = 2.2uH, 4.2mΩ 50% 35% 0.01 0.1 1 10 40% 0.01 100 0.1 Output Current (A) 100 Figure 2. Efficiency vs. Output Current for VIN = 12.6V 350 1200 300 1000 250 RFF (kOhm) Frequency (kHz) 10 Output Current (A) Figure 1. Efficiency vs. Output Current for VOUT = 1.05V, L = 1.2µH (2.9mΩ) 200 150 100 50 5.0 Vout 4.0 3.0 2.0 1.0 800 4.5 3.5 2.5 1.5 0.5 600 400 200 0 0 2 4 6 8 10 0 200 12 Output Current (A) Figure 3. Switching Frequency vs. Output Current 400 600 800 Frequency (kHz) 1000 Figure 4. Frequency vs. RFF 1.058 1.057 VOUT @ 16VIN Output Voltage (V) 1.058 Output Voltage (V) 1 VOUT @ 12.6VIN 1.056 VOUT @ 7VIN 1.055 1.054 1.057 1.056 1.055 1.054 0 2 4 6 8 10 Output Current (A) Figure 5. Output Voltage Regulation vs. Output Current 12 6 7 8 9 10 11 12 13 14 15 16 17 Input Voltage (V) Figure 6. Output Voltage Regulation vs. Input Voltage at IOUT = 12A 7 IR3876MPBF TYPICAL APPLICATION CIRCUIT Demoboard Schematic: VOUT = 1.05V, Fs = 300kHz +3.3V VCC +Vins R1 open TP1 VINS VIN TP2 VIN TP3 FCCM C1 1uF EN TP4 EN 4 3 FCCM +Vin1s 1 2 TP5 PGND C4 0.1uF R4 7.87K VSW + C3 68uF TP20 +Vin1s R3 200K SW1 EN / FCCM C2 22uF -Vins TP6 PGNDS ISET TP23 +Vsws 13 14 15 16 17 L1 1.0uH U1 IR3876 VOUT VSW TP7 VOUT 1 VIN 1 2 C12 0.1uF TP10 PGND TP24 +Vsws TP12 VSWS C15 open C16 open C17 open C18 open C19 open C26 open C27 open R13 open 3 4 +Vdd2s Vout 10 +Vout2s -Vout2s +Vout1s -Vout1s -Vdd2s -Vout1s 9 2 TP18 VOLTAGE SENSE 5 -Vdd1s 1 +Vins R8 2.55K R9 0 +Vdd1s R12 open +Vin1s R11 open TP19 FB -Vdd2s C24 open R7 2.80K -Vdd1s C14 open +Vdd2s -Vdd1s -Vins C22 open C23 open +Vdd1s TP25 -Vin1s +Vdd1s 8 TP15 -Vout1s -Vout1s NC2 9 -Vout1s C21 1uF C25 1uF TP17 PGND C11 open TP21 -Vsws +Vdd2s VCC C10 47uF TP9 +Vout1s NC1 TP26 AGND TP16 VCC C9 330uF 5 SS +3.3V -Vdd2s TP22 +Vsws C8 open 3 BOOT FB 8 TP14 +3.3V 12 C7 open 7 7 C6 open C13 open 6 C20 0.1uF PHASE 4 6 SS IR3876 GND1 3V3BP TP13 SS PGOOD 3 5 PGND 4 FB R6 open ISET 5 TP8 VOUTS 11 3 PGOOD 2 NC VCC 2 10 1 R5 10K FF GND C5 open TP11 PGOOD EN +3.3V 4 R2 10K R14 open R10 open Figure 7. Typical Application Circuit for VOUT = 1.05V, Fs = 300kHz Bill of Materials Quantity 3 1 1 1 1 3 1 2 1 1 1 1 1 1 1 Reference C1, C21, C25 C2 C3 C9 C10 C4, C12, C20 L1 R2, R5 R9 R3 R4 R7 R8 SW1 U1 Value 1uF 22uF 68uF 330uF 47uF 0.1uF 1uH 10K 0 200K 7.87K 2.8K 2.55K SPST IR3876 Description CAP,CER,1.0uF,25V,X7R,0603 CAP,22uF,25V,CERAMIC,X5R,1210 CAP,68uF,25V,ELECT,FK,SMD SP-CAP, 330uF, 2V, 4.5mΩ, 20% CAP,CER,47uF,6.3V,X5R,0805 CAP,CER,0.1uF,50V,10%,X7R,0603 INDUCTOR, 1uH, 20A, 2.7mΩ,SMD RES,10.0kΩ,1/10W,1%,0603,SMD RES,0Ω,1/10W,1%,0603,SMD RES,200kΩ,1/10W,1%,0603,SMD RES,7.87kΩ,1/10W,1%,0603,SMD RES,2.8kΩ,1/10W,1%,0603,SMD RES,2.55kΩ,1/10W,1%,0603,SMD SWITCH, DIP, SPST, SMT 5mm x 6mm QFN Manufacturer Murata Electronics Panasonic Panasonic Panasonic TDK TDK CYNTEC Vishay/Dale Vishay/Dale Vishay/Dale Vishay/Dale Vishay/Dale Vishay/Dale C&K Components IR Part-Number GRM188R71E105KA12D ECJ-4YB1E226M EEV-FK1E680P EEF-SX0D331E4 C2012X5R0J476M C1608X7R1H104K PIMB103E-1R0MS-39 CRCW060310K0FKEA CRCW06030000Z0EAHP CRCW0603200KFKEA CRCW06037K87FKEA CRCW06032K80FKEA CRCW06032K55FKEA SD02H0SK IR3876MPBF 8 IR3876MPBF TYPICAL OPERATING DATA Tested with demoboard shown in Figure 7, VIN = 12.6V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25 oC, no airflow, unless otherwise specified EN EN PGOOD PGOOD SS SS VOUT VOUT 5V/div 5V/div 1V/div 500mV/div 5ms/div 5V/div 5V/div 1V/div 500mV/div 200µs/div Figure 9: Shutdown Figure 8: Startup VOUT VOUT PHASE PHASE iL iL 20mV/div 5V/div 2A/div 10µs/div Figure 10: DCM (IOUT = 0.1A) 20mV/div 5V/div 5A/div 2µs/div Figure 11: CCM (IOUT = 12A) PGOOD PGOOD SS FB VOUT VOUT IOUT iL 5V/div 1V/div 1V/div 10A/div 2ms/div Figure 12: Over Current Protection (tested by shorting VOUT to PGND) 5V/div 1V/div 500mV/div 2A/div 50µs/div Figure 13: Over Voltage Protection (tested by shorting FB to VOUT) 9 IR3876MPBF TYPICAL OPERATING DATA Tested with demoboard shown in Figure 7, VIN = 12.6V, VCC = 5V, Vout = 1.05V, Fs = 300kHz, TA = 25 oC, no airflow, unless otherwise specified VOUT VOUT PHASE PHASE iL iL 50mV/div 10V/div 5A/div 20µs/div Figure 14: Load Transient 0-8A 50mV/div 10V/div 5A/div 20µs/div Figure 15: Load Transient 4-12A Figure 16: Thermal Image at IOUT = 12A (IR3876: 81oC, Inductor: 58oC, PCB: 54oC) 10 IR3876MPBF CIRCUIT DESCRIPTION PWM COMPARATOR The PWM comparator initiates a SET signal (PWM pulse) when the FB pin falls below the reference (Vref) or the soft start (SS) voltage. ON-TIME GENERATOR The PWM on-time duration is programmed with an external resistor (RFF) from the input supply (VIN) to the FF pin. The simplified calculation for RFF is shown in equation 1. The FF pin is held to an internal reference after EN goes HIGH. A copy of the current in RFF charges a timing capacitor, which sets the on-time duration, as shown in equation 2. RFF  VOUT (1) 1V  20 pF  FSW TON  RFF 1V  20 pF (2) VIN CONTROL LOGIC The control logic monitors input power sources, sequences the converter through the soft-start and protective modes, and initiates an internal RUN signal when all conditions are met. VCC and 3VCBP pins are continuously monitored, and the IR3876 will be disabled if the voltage of either pin drops below the falling thresholds. EN_DELAY will become HIGH when VCC and 3VCBP are in the normal operating range and the EN pin = HIGH. PGOOD The PGOOD pin is open drain and it needs to be externally pulled high. High state indicates that output is in regulation. The PGOOD logic monitors EN_DELAY, SS_DELAY, and under/over voltage fault signals. PGOOD is released only when EN_DELAY and SS_DELAY = HIGH and output voltage is within the OV and UV thresholds. PRE-BIAS STARTUP IR3876 is able to start up into pre-charged output, which prevents oscillation and disturbances of the output voltage. With constant on-time control, the output voltage is compared with the soft start voltage (SS) or Vref, depending on which one is lower, and will not start switching unless the output voltage drops below the reference. This scheme prevents discharge of a pre-biased output voltage. SHUTDOWN The IR3876 will shutdown if VCC is below its UVLO limit. The IR3876 can be shutdown by pulling the EN pin below its lower threshold. Alternatively, the output can be shutdown by pulling the soft start pin below 0.3V. SOFT START With EN = HIGH, an internal 10µA current source charges the external capacitor (CSS) on the SS pin to set the output voltage slew rate during the soft start interval. The soft start time (tSS) can be calculated from equation 3. t SS  CSS  0.5V (3) 10A The feedback voltage tracks the SS pin until SS reaches the 0.5V reference voltage (Vref), then feedback is regulated to Vref. CSS will continue to be charged, and when SS pin reaches VSS (see Electrical Specification), SS_DELAY goes HIGH. With EN_DELAY = LOW, the capacitor voltage and SS pin is held to the FB pin voltage. A normal startup sequence is shown in Figure 17. Figure 17. Normal Startup 11 IR3876MPBF CIRCUIT DESCRIPTION UNDER/OVER VOLTAGE MONITOR The IR3876 monitors the voltage at the FB node through a 350ns filter. If the FB voltage is below the under voltage threshold, UV# is set to LOW holding PGOOD to be LOW. If the FB voltage is above the over voltage threshold, OV# is set to LOW, the shutdown signal (SD) is set to HIGH, MOSFET gates are turned off, and PGOOD signal is pulled low. Toggling VCC or EN will allow the next start up. Figure 18 shows PGOOD status change when UV/OV is detected. The over voltage and under voltage thresholds can be found in the Electrical Specification section. OVER CURRENT MONITOR The over-current circuitry monitors the output current during each switching cycle. The voltage across the lower MOSFET, VPHASE, is monitored for over current and zero crossing. The OCP circuit evaluates VPHASE for an over current condition typically 270ns after the lower MOSFET is gated on. This delay functions to filter out switching noise. The minimum lower gate interval allows time to sample VPHASE. The over current trip point is programmed with a resistor from the ISET pin to PHASE pin, as shown in equation 4, where Tj is the junction temperature of Q2 at operation conditions, and 0.4 is the temperature coefficient (~4000 ppm/C) of Q2 RDSON. When over current is detected, the output gates are tri-state and SS voltage is pulled to 0V. This initiates a new soft start cycle. If there is a total of four OC events, the IR3876 will disable switching, as shown in Figure 19. Toggling VCC or EN will allow the next start up. RSET  RDSON  IOC 20 A  (1  Tj  25 100  0.4) (4) Figure 18(a). Under/Over Voltage Monitor * typical filter delay Figure 19. Over Current Protection Figure 18(b). Over Voltage Protection 12 IR3876MPBF CIRCUIT DESCRIPTION GATE DRIVE LOGIC The gate drive logic features adaptive dead time, diode emulation, and a minimum lower gate interval. STABILITY CONSIDERATIONS Constant-on-time control is a fast , ripple based control scheme. Unstable operation can occur if certain conditions are not met. The system instability is usually caused by: An adaptive dead time prevents the simultaneous conduction of the upper and lower MOSFETs. The lower gate voltage (LGATE) must be below approximately 1V after PWM goes HIGH before the upper MOSFET can be gated on. Also, the upper gate voltage (UGATE), the difference voltage between UGATE and PHASE, must be below approximately 1V after PWM goes LOW before the lower MOSFET can be gated on. • Switching noise coupled to FB input. This causes the PWM comparator to trigger prematurely after the 400ns minimum Q2 ontime. It will result in double or multiple pulses every switching cycle instead of the expected single pulse. Double pulsing can causes higher output voltage ripple, but in most application it will not affect operation. This can usually be prevented by careful layout of the ground plane and the FB sensing trace. The control MOSFET is gated on after the adaptive delay for PWM = HIGH and the synchronous MOSFET is gated on after the adaptive delay for PWM = LOW. The lower MOSFET is driven ‘off’ when the signal ZCROSS indicates that the inductor current has reversed as detected by the PHASE voltage crossing the zero current threshold. The synchronous MOSFET stays ‘off’ until the next PWM falling edge. When the lower peak of inductor current is above zero, a forced continuous current condition is selected. The control MOSFET is gated on after the adaptive delay for PWM = HIGH, and the synchronous MOSFET is gated on after the adaptive delay for PWM = LOW. • Steady state ripple on FB pin being too small. The PWM comparator in IR3876 requires minimum 7mVp-p ripple voltage to operate stably. Not enough ripple will result in similar double pulsing issue described above. Solving this may require using output capacitors with higher ESR. The synchronous MOSFET gate is driven on for a minimum duration. This minimum duration allows time to recharge the bootstrap capacitor and allows the current monitor to sample the PHASE voltage. • For applications with all ceramic output capacitors, the ESR is usually too small to meet the stability criteria. In these applications, external slope compensation is necessary to make the loop stable. The ramp injection circuit, composed of R6, C13, and C14, shown in Figure 7 is required. The inductor current ripple sensed by R6 and C13 is AC coupled to the FB pin through C14. C14 is usually chosen between 1 to 10nF, and C13 between 10 to 100nF. R6 should then be chosen such that L/DCR = C13*R6. • ESR loop instability. The stability criteria of constant on-time is: ESR*Cout>Ton/2. If ESR is too small that this criteria is violated then sub-harmonic oscillation will occur. This is similar to the instability problem of peakcurrent-mode control with D>0.5. Increasing ESR is the most effective way to stabilize the system, but the price paid is the larger output voltage ripple. 13 IR3876MPBF COMPONENT SELECTION Selection of components for the converter is an iterative process which involves meeting the specifications and trade-offs between performance and cost. The following sections will guide one through the process. INDUCTOR SELECTION Inductor selection involves meeting the steady state output ripple requirement, minimizing the switching loss of upper MOSFETs, meeting transient response specifications and minimizing the output capacitance. The output voltage includes a DC voltage and a small AC ripple component due to the low pass filter which has incomplete attenuation of the switching harmonics. Neglecting the inductance in series with the output capacitor, the magnitude of the AC voltage ripple is determined by the total inductor ripple current flowing through the total equivalent series resistance (ESR) of the output capacitor bank. INPUT CAPACITOR SELECTION The main function of the input capacitor bank is to provide the input ripple current and fast slew rate current during the load current step up. The input capacitor bank must have adequate ripple current carrying capability to handle the total RMS current. Figure 20 shows a typical input current. Equation 6 shows the RMS input current. The RMS input current contains the DC load current and the inductor ripple current. As shown in equation 5, the inductor ripple current is unrelated to the load current. The maximum RMS input current occurs at the maximum output current. The maximum power dissipation in the input capacitor equals the square of the maximum RMS input current times the input capacitor’s total ESR. Input Current IOUT ΔI ΔI  TON  VIN  VOUT  (5) 2L TS Figure 20. Typical Input Current Waveform. One can use equation 5 to find the required inductance. ΔI is defined as shown in Figure 20. The main advantage of small inductance is increased inductor current slew rate during a load transient, which leads to a smaller output capacitance requirement as discussed in the Output Capacitor Selection section. The draw back of using smaller inductances is increased switching power loss in upper MOSFET, which reduces the system efficiency and increases the thermal dissipation. Ts IIN_RMS  1  f 2 t   dt Ts 0 1  ΔI   IOUT  Ton  Fs  1     3  IOUT  2 (6) The voltage rating of the input capacitor needs to be greater than the maximum input voltage because of high frequency ringing at the phase node. The typical percentage is 25%. 14 IR3876MPBF COMPONENT SELECTION OUTPUT CAPACITOR SELECTION Selection of the output capacitor requires meeting voltage overshoot requirements during load removal, and meeting steady state output ripple voltage requirements. The output capacitor is the most expensive converter component and increases the overall system cost. The output capacitor decoupling in the converter typically includes the low frequency capacitor, such as Specialty Polymer Aluminum, and mid frequency ceramic capacitors. The first purpose of output capacitors is to provide current when the load demand exceeds the inductor current, as shown in Figure 21. Equation 7 shows the charge requirement for a certain load. The advantage provided by the IR3876 at a load step is to reduce the delay compared to a fixed frequency control method (in microseconds or (1D)*Ts). If the load increases right after the PWM signal goes low, the longest delay will be equal to the minimum lower gate on as shown in the Electrical Specification table. The IR3876 also reduces the inductor current slew time, the time it takes for the inductor current to reach equality with the output current, by increasing the switching frequency up to 2.5MHz. The result reduces the recovery time. Load Current I STEP Output Charge Inductor Slew Rate Figure 21. Charge Requirement during Load Step Q  C  V  0.5  Istep  t COUT  (7a)  1 L  Istep2   VDROP  2 VIN  VOUT  1 COUT  L  ISTEP 2 VOS 2  VOUT 2 (8) VOS VOUT VL VDROP IOUT VESR ISTEP Figure 22. Typical Output Voltage Response Waveform. BOOT CAPACITOR SELECTION The boot capacitor starts the cycle fully charged to a voltage of VB(0). Cg equals 0.65nF in IR3876. Choose a sufficiently small ΔV such that VB(0)-ΔV exceeds the maximum gate threshold voltage to turn on the high side MOSFET.  V (0)  CBOOT  Cg   B  1 (9)  ΔV  t Δt The second purpose of the output capacitor is to minimize the overshoot of the output voltage when the load decreases as shown in Figure 22. By using the law of energy before and after the load removal, equation 8 shows the output capacitance requirement for a load step. (7b) The output voltage drop, VDROP, initially depends on the characteristic of the output capacitor. VDROP is the sum of the equivalent series inductance (ESL) of the output capacitor times the rate of change of the output current and the ESR times the change of the output current. VESR is usually much greater than VESL. The IR3876 requires a total ESR such that the ripple voltage at the FB pin is greater than 7mV. Choose a boot capacitor value larger than the calculated CBOOT in equation 9. Equation 9 is based on charge balance at CCM operation. Usually the boot capacitor will be discharged to a much lower voltage when the circuit is operating in DCM mode at light load, due to much longer Q2 off time and the bias current drawn by the IC. Boot capacitance needs to be increased if insufficient turn-on of Q1 is observed at light load, typically larger than 0.1µF is needed. The voltage rating of this part needs to be larger than VB(0) plus the desired derating voltage. Its ESR and ESL needs to be low in order to allow it to deliver the large current and di/dt’s which drive MOSFETs most efficiently. In support of these requirements a ceramic capacitor should be chosen. 15 IR3876MPBF DESIGN EXAMPLE Design Criteria: Input Voltage, VIN, = 7V to 16V Output Voltage, VOUT = 1.05V Switching Frequency, Fs = 300KHz Inductor Ripple Current, 2ΔI = 3A Maximum Output Current, IOUT = 12A Over Current Trip, IOC = 18A Overshoot Allowance, VOS = VOUT + 50mV Undershoot Allowance, VDROP = 50mV Find RFF : 1.05V RFF   175 k 1V  20 pF  300kHz Pick a standard value 178 kΩ, 1% resistor. Find RSET : RSET  1.4  5.2m  18 A 20A  6.55 k The RDSON of the lower MOSFET could be expected to increase by a factor of 1.4 over temperature. Therefore, pick a 6.65kΩ, 1% standard resistor. Choose an inductor with the lowest DCR and AC power loss as possible to increase the overall system efficiency. For instance, choose MPL1055-1R21R manufactured by Delta. The inductance of this part is 1.2µH and has 2.9mΩ DCR. Ripple current needs to be recalculated using the chosen inductor. ΔI  1.05V  16V - 1.05V   1.36 A 2 16V 1.2H  300kHz Choose an input capacitor: 1.05V 1  1.36 A  IIN_RMS  12 A   1   16V 3  12 A   3.1A 2 A Panasonic 10µF (ECJ3YB1E106M) accommodates 6 Arms of ripple current at 300KHz. Due to the chemistry of multilayer ceramic capacitors, the capacitance varies over temperature and operating voltage, both AC and DC. One 10µF capacitor is recommended. In a practical solution, one 1µF capacitor is required along with the 10µF. The purpose of the 1µF capacitor is to suppress the switching noise and deliver high frequency current. Find a resistive voltage divider for VOUT = 1.05V: VFB  R2  VOUT  0.5V R2  R1 R2 = 2.55kΩ, R1 = 2.80kΩ, both 1% standard resistors. Choose the soft start capacitor: Once the soft start time has chosen, such as 1000us to reach to the reference voltage, a 22nF for CSS is used to meet 1000µs. Choose an inductor to meet the design specification: VOUT  VIN  VOUT  VIN  2ΔI  Fs 1.05V  16V - 1.05V   16V  3 A  300kHz  1.1H L Choose an output capacitor: To meet the undershoot specification, select a set of output capacitors which has an equivalent ESR of 10mΩ (50mV/5A). To meet the overshoot specification, equation 7 will be used to calculate the minimum output capacitance. As a result, 300µF will be needed for 5A load removal. Combine those two requirements, one can choose a set of output capacitors from manufactures such as Sanyo or Rubycon. A 330µF (2SWZ330M R05) from Rubycon is recommended. This capacitor has 4.5mΩ ESR which leaves margin for the voltage drop of the ESL during load step up. 16 IR3876MPBF LAYOUT RECOMMENDATION Bypass Capacitor: One 1µF high quality ceramic capacitor should be placed as near VCC pin as possible. The other end of capacitor can be connected to a via or connected directly to GND plane. Use a GND plane instead of a thin trace to the GND pin because a thin trace have too much impedance. Boot Circuit: CBOOT should be placed near the BOOT and PHASE pins to reduce the impedance when the upper MOSFET turns on. Power Stage: Figure 23 shows the current paths and their directions for the on and off periods. The on time path has low average DC current and high AC current. Therefore, it is recommended to place the input ceramic capacitor, upper, and lower MOSFET in a tight loop as shown in Figure 23. VIN ON Q1 VOUT IR3876 Q2 OFF C IN C OUT Figure 23. Current Path of Power Stage The purpose of the tight loop from the input ceramic capacitor is to suppress the high frequency (10MHz range) switching noise and reduce Electromagnetic Interference (EMI). If this path has high inductance, the circuit will cause voltage spikes and ringing, and increase the switching loss. The off time path has low AC and high average DC current. Therefore, it should be laid out with a tight loop and wide trace at both ends of the inductor. Lowering the loop resistance reduces the power loss. The typical resistance value of 1-ounce copper thickness is 0.5mΩ per square inch. 17 IR3876MPBF PCB Metal and Components Placement Lead lands (the 13 IC pins) width should be equal to nominal part lead width. The minimum lead to lead spacing should be ≥ 0.2mm to minimize shorting. Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension. The outboard extension ensures a large toe fillet that can be easily inspected. Pad lands (the 4 big pads) length and width should be equal to maximum part pad length and width. However, the minimum metal to metal spacing should be no less than; 0.17mm for 2 oz. Copper or no less than 0.1mm for 1 oz. Copper or no less than 0.23mm for 3 oz. Copper. 18 IR3876MPBF Solder Resist It is recommended that the lead lands are Non Solder Mask Defined (NSMD). The solder resist should be pulled away from the metal lead lands by a minimum of 0.025mm to ensure NSMD pads. The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist onto the copper of 0.05mm to accommodate solder resist misalignment. Ensure that the solder resist in between the lead lands and the pad land is ≥ 0.15mm due to the high aspect ratio of the solder resist strip separating the lead lands from the pad land. 19 IR3876MPBF Stencil Design The Stencil apertures for the lead lands should be approximately 80% of the area of the lead lads. Reducing the amount of solder deposited will minimize the occurrences of lead shorts. If too much solder is deposited on the center pad the part will float and the lead lands will open. The maximum length and width of the land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back in order to decrease the risk of shorting the center land to the lead lands when the part is pushed into the solder paste. 20 IR3876MPBF DIM A A1 b b1 c D E e e1 e2 MILIMITERS MIN MAX 0.8 1 0 0.05 0.375 0.475 0.25 0.35 0.203 REF. 5.000 BASIC 6.000 BASIC 1.033 BASIC 0.650 BASIC 0.852 BASIC INCHES MIN MAX 0.0315 0.0394 0 0.002 0.1477 0.1871 0.0098 0.1379 0.008 REF. 1.970 BASIC 2.364 BASIC 0.0407 BASIC 0.0256 BASIC 0.0259 BASIC DIM L M N O P Q R S t1, t2, t3 t4 t5 MILIMITERS MIN MAX 0.35 0.45 2.441 2.541 0.703 0.803 2.079 2.179 3.242 3.342 1.265 1.365 2.644 2.744 1.5 1.6 0.401 BASIC 1.153 BASIC 0.727 BASIC INCHES MIN MAX 0.0138 0.0177 0.0962 0.1001 0.0277 0.0314 0.0819 0.0858 0.1276 0.1316 0.0498 0.05374 0.1042 0.1081 0.0591 0.063 0.016 BACIS 0.045 BASIC 0.0286 BASIC IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 This product has been designed and qualified for the Industrial Market (Note 2) Visit us at www.irf.com for sales contact information Data and specifications subject to change without notice. 5/2012 21