HI3304JIB

November 2002 DU CT CT E P RO RO DU SOLET BSTITUTE P ter at OB en LE S U por t C POSSIB chnical Sup rsil.com/tsc RA e e FO ww.int t o ur T contac TERSIL or w IN 1-888® HI3304 4-Bit, 25 MSPS, Flash A/D Converter Features • CMOS Low Power (Typ) . . . . . . . . . . . . . . . . . . . 35mW • Parallel Conversion Technique • Single Power Supply Voltage . . . . . . . . . . . . 3V to 7.5V • Sampling Rate at 5V Supply . . . . . . . . . . . . . . . . . 25MHz • 4-Bit Latched Three-State Output with Overflow and Data Change Outputs • Maximum Nonlinearity. . . . . . . . . . . . . . . . . . . . 1/8 LSB • Inherent Resistance to Latch-Up • Bipolar Input Range with Optional Second Supply • Input Bandwidth (Typ) . . . . . . . . . . . . . . . . . . . . . 40MHz • Linearity (INL, DNL): - HI3304JIP . . . . . . . . . . . . . . . . . . . . . . . . . . ±0.25 LSB - HI3304JIB . . . . . . . . . . . . . . . . . . . . . . . . . . ±0.25 LSB • Sampling Rate: - HI3304JIP . . . . . . . . . . . . . . . . . . . . . . . .25MHz (40ns) - HI3304JIB . . . . . . . . . . . . . . . . . . . . . . . .25MHz (40ns) Description The Intersil HI3304 is a CMOS parallel (FLASH) analog-todigital converter designed for applications demanding both low-power consumption and high speed digitization. Digitizing at 25MHz, for example, requires only about 35mW. The HI3304 operates over a wide, full-scale signal input voltage range of 0.5V up to the supply voltage. Power consumption is as low as 10mW, depending upon the clock frequency selected. Sixteen paralleled auto-balanced voltage comparators measure the input voltage with respect to a known reference to produce the parallel-bit outputs in the HI3304. Fifteen comparators are required to quantize all input voltage levels in this 4-bit converter, and the additional comparator is required for the overflow bit. A data change pin indicates when the present output differs from the previous, thus allowing compaction of data storage. Part Number Information PART NUMBER TEMP. RANGE (oC) -40 to 85 -40 to 85 PACKAGE 16 Ld PDIP 16 Ld SOIC PKG. NO. E16.3 M16.3 Applications • Video Digitizing • High Speed Data Acquisition • Digital Communication Systems • Radar Signal Processing HI3304JIP HI3304JIB Pinout HI3304 (PDIP, SOIC) TOP VIEW BIT 1 (LSB) 1 BIT 2 2 BIT 3 3 BIT 4 4 DATA CHANGE (DC) 5 OVERFLOW (OF) 6 CE2 7 VSS 8 16 VDD 15 CLK 14 VAA13 VREF 12 VREF + 11 VIN 10 VAA+ 9 CE1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2002. All Rights Reserved All other trademarks mentioned are the property of their respective owners. FN4137.2 1 HI3304 Functional Block Diagram φ2 VAA+ VDD 16 OUTPUT REGISTER DQ CLK THREE-STATE DRIVERS 5 DATA CHANGE φ2 φ1 VIN 11 1/ R 2 φ1 φ1 φ1 10 D COUNT 16 Q DQ CLK 6 OVERFLOW 12 VREF + R † CAB #16 LATCH 16 DQ CLK COUNT ENCODER 8 LOGIC Q D ARRAY LATCH 8 4 BIT 4 R † CAB #8 R DQ CLK 3 BIT 3 DQ CLK 2 BIT 2 R VREF - 1 13 50kΩ CLOCK 15 /2R D Q COUNT 1 †CAB COMPARATOR #1 φ1 (AUTO BALANCE) LATCH 0 14 VAA8 VSS DQ CLK 1 BIT 1 (LSB) 9 CE1 φ2 (SAMPLE UNKNOWN) 7 CE2 † Cascaded Auto Balance (CAB) NOTE: CE1 and CE2 inputs and data outputs have standard CMOS protection networks to VDD and VSS . Analog inputs and clock have standard CMOS protection networks to VAA+ and VAA-. 2 HI3304 Absolute Maximum Ratings DC Supply Voltage Range (VDD or VAA+) (Voltage Referenced to VSS or VAA- Terminal, Whichever is More Negative) . . . . . . . . . . . . . . . . . . -0.5V to +8V Input Voltage Range CE1, CE2 Inputs . . . . . . . . . . . . . . . . . . . VSS -0.5V to VDD +0.5V Clock, VREF+, VREF-, VIN Inputs . . . . . . VAA -0.5V to VAA +0.5V DC Input Current, Any Input . . . . . . . . . . . . . . . . . . . . . . . . . ± 20mA Thermal Information Thermal Resistance (Typical, Note 1) θJA (oC/W) PDIP Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90 SOIC Package. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100 Maximum Junction Temperature. . . . . . . . . . . . . . . . . . . . . . . 150oC Maximum Storage Temperature Range (TSTG) . . . .-65oC to 150oC Maximum Lead Temperature (Soldering 10s). . . . . . . . . . . . . 300oC (SOIC - Lead Tips Only) Operating Conditions Supply Voltage Range (VDD or VAA+). . . . . . . . . . . . . . . . . . 3V to 7.5V VAA+ Voltage Range . . . . . . . . . . . . . . . . . . VDD -1V to VDD +2.5V VAA- Voltage Range. . . . . . . . . . . . . . . . . . . . VSS -2.5V to VSS +1V Operating Temperature Range. . . . . . . . . . . . . . . . . . -40oC to 85oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTE: 1. θJA is measured with the component mounted on an evaluation PC board in free air. Electrical Specifications TA = 25oC, VREF+ = 2V, VDD = VAA+ = 5V, VAA- = VREF - = VSS = GND, fCLK = 25MHz Unless Otherwise Specified TEST CONDITIONS MIN TYP MAX UNITS PARAMETER SYSTEM PERFORMANCE Resolution Input Errors Integral Linearity Error Differential Linearity Error Offset Error (Unadjusted) Gain Error (Unadjusted) 4 - ±0.125 ±0.125 - ±0.25 ±0.25 ±1.0 ±1.0 Bits LSB LSB LSB LSB DYNAMIC CHARACTERISTICS Input Signal Level 0.5dB Below Full Scale Conversion Timing Aperture Delay fS = 25MHz, fIN = 100kHz fS = 25MHz, fIN = 5MHz fS = 25MHz, fIN = 100kHz fS = 25MHz, fIN = 5MHz fS = 25MHz, fIN = 100kHz fS = 25MHz, fIN = 5MHz Effective Number of Bits, ENOB fS = 25MHz, fIN = 100kHz fS = 25MHz, fIN = 5MHz ANALOG INPUTS Input Range Input Loading Full Scale Input Range Input Capacitance Input Current -3dB Input Bandwidth VIN = 2V (Note 2) (Notes 1, 4) 0.5 10 150 40 VAA 200 V pF µA MHz 3 23.7 23.6 ns dB dB Signal to Noise Ratio (SNR) RMS Signal = ---------------------------------RMS Noise Signal to Noise Ratio (SINAD) RMS Signal = --------------------------------------------------------------RMSNoise + Distortion Total Harmonic Distortion, THD - 23.4 22.8 - dB dB - -34.5 -31.0 3.67 3.57 - dBc dBc Bits Bits 3 HI3304 Electrical Specifications TA = 25oC, VREF+ = 2V, VDD = VAA+ = 5V, VAA- = VREF - = VSS = GND, fCLK = 25MHz Unless Otherwise Specified (Continued) TEST CONDITIONS MIN TYP MAX UNITS PARAMETER REFERENCE INPUTS Input Range VREF+ Range VREF- Range Input Loading DIGITAL INPUTS Digital Input Maximum VIN , Low CLOCK CE1, CE2 Minimum VIN , High CLOCK CE1, CE2 Input Leakage, Except CLK Input Leakage, CLK DIGITAL OUTPUTS Digital Outputs Output Low (Sink) Current Output High (Source) Current Three-State Leakage Current TIMING CHARACTERISTICS Conversion Timing Maximum Conversion Speed Auto-Balance Time (φ1) Sample Time (φ2) Output Timing Data Valid Delay Data Hold Time Output Enable Time Output Disable Time POWER SUPPLY CHARACTERISTICS Device Current, IAA Resistor Ladder Impedance (Note 4) (Note 4) VIN = 5V, CLK = Low VAA- +0.5 VAA640 - VAA+ VAA+ -0.5 960 V V Ω (Notes 3, 4) (Note 4) (Notes 3, 4) (Note 4) V = 0V, 5V (Note 3) 0.7 x VAA 0.7 x VDD - ±100 0.3 x VAA 0.3 x VDD ±1 ±150 V V V V µA µA VO = 0.4V VO = 4.6V VO = 0V, 5V 6 -3 - ±0.2 ±5 mA mA µA CLK = Square Wave 25 20 20 35 30 25 15 10 - MSPS ns ns ns ns ns ns 5000 40 - (Note 4) (Note 4) 15 - Continuous Clock Continuous φ2 Continuous φ1 - 5.5 0.4 2 1.5 5 10 mA mA mA mA mA Device Current, IDD VAA+ = 5V, VSS = CE1 = VAA- = CLK = GND VAA+ = 7V NOTES: Continuous Clock Continuous φ2 Continuous φ1 - 5 20 mA 1. Full scale input range, VREF + - VREF -, may be in the range of 0.5V to VAA+ -VAA- volts. Linearity errors increase at lower full scale ranges, however. 2. Input current is due to energy transferred to the input at the start of the sample period. The average value is dependent on input and VDD voltage. 3. The CLK input is a CMOS inverter with a 50kΩ feedback resistor. It operates from the VAA+ and VAA- supplies. It may be AC-coupled with a 1VP-P minimum source. 4. Parameter not tested, but guaranteed by design or characterization. 4 HI3304 Timing Diagrams DATA SHIFTED INTO OUTPUT REGISTERS COMPARATOR DATA LATCHED 1 CLOCK 0 1 B1 - B4, DC AND OF 0 φ1 AUTO BALANCE φ2 SAMPLE 1 AUTO BALANCE SAMPLE 2 AUTO BALANCE SAMPLE 3 DATA VALID 0 tHO tD DATA VALID 1 DATA VALID 2 FIGURE 1. TIMING DIAGRAM CE1 CE2 tDIS BITS 1-4 HIGH IMPEDANCE tEN tDIS tEN HIGH IMPEDANCE HIGH IMPEDANCE DC, OF FIGURE 2. OUTPUT ENABLE/DISABLE TIMING SAMPLE ENDS CLOCK SAMPLE ENDS φ2 OLD DATA φ1 tD φ2 NEW DATA CLOCK φ1 φ2 φ1 tD φ2 φ1 NEW DATA OUTPUT OUTPUT OLD DATA OLD DATA + 1 FIGURE 3A. With φ2 as standby state (fastest method, but standby limited to 5µs maximum) FIGURE 3B. With φ1 as standby state (indefinite standby, double pulse needed) SAMPLE ENDS CLOCK φ2 φ1 OLD DATA φ2 φ1 tD φ2 NEW DATA OUTPUT INVALID DATA FIGURE 3C. With φ2 as standby state (indefinite standby, lower power than 3B) FIGURE 3. PULSE-MODE TIMING DIAGRAMS 5 HI3304 Typical Performance Curves 40 8 38 7 34 IDD + IAA (MA) -25 0 25 50 75 100 36 tD (ns) 6 5 32 4 30 3 28 -50 2 TEMPERATURE (oC) 5 10 15 fS (MHz) 20 25 30 FIGURE 4. DATA DELAY vs TEMPERATURE 0.25 0.22 0.20 NON-LINEARITY (LSB) 0.17 0.15 0.12 0.10 0.07 0.05 0.02 DNL NON-LINEARITY (LSB) INL FIGURE 5. DEVICE CURRENT vs SAMPLE FREQUENCY 0.10 0.09 0.08 0.07 0.06 0.05 0.04 0.03 0.02 0.01 DNL INL 0.00 -40 -30 -20 -10 0 10 20 30 40 50 TEMPERATURE (oC) 60 70 80 90 0.00 1 2 3 REFERENCE VOLTAGE (V) 4 5 FIGURE 6. NON-LINEARITY vs TEMPERATURE FIGURE 7. NON-LINEARITY vs REFERENCE VOLTAGE 0.50 0.45 0.40 NON-LINEARITY (LSB) 0.35 ENOB (LSB) 35 0.30 0.25 0.20 0.15 0.10 0.05 0.00 15 20 25 fS (MHz) 30 DNL INL 4.00 3.80 3.60 3.40 3.20 3.00 2.80 2.60 2.40 2.20 2.00 -40 -30 -20 -10 0 10 20 30 40 50 TEMPERATURE (oC) 60 70 80 90 FIGURE 8. NON-LINEARITY vs SAMPLE FREQUENCY FIGURE 9. EFFECTIVE BITS vs TEMPERATURE 6 HI3304 Typical Performance Curves 4.00 3.80 3.60 3.40 ENOB (LSB) IDD (mA) 0 1 2 3 4 5 fI (MHz) 6 7 8 9 10 3.20 3.00 2.80 2.60 2.40 2.20 2.00 (Continued) 7.00 6.80 6.60 6.40 6.20 6.00 5.80 5.60 5.40 5.20 5.00 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90 TEMPERATURE (oC) FIGURE 10. EFFECTIVE BITS vs INPUT FREQUENCY FIGURE 11. DEVICE CURRENT vs TEMPERATURE Pin Descriptions PIN NUMBER 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 NAME Bit 1 Bit 2 Bit 3 Bit 4 DC OF CE2 VSS CE1 VAA+ VIN VREF+ VREFVAACLK VDD Bit 1 (LSB). Bit 2. Bit 3. Bit 4 (MSB). Data Change. Overflow. Three-State Output Enable Input, active low. See the Chip Enable Truth Table. Digital Ground. Three-State Output Enable Input, active high. See the Chip Enable Truth Table. Analog Power Supply, +5V. Analog Signal Input. Reference Voltage Positive Input. Reference Voltage Negative Input. Analog Ground. Clock Input. Digital Power Supply, +5V. Output Data Bits (High = True) DESCRIPTION CHIP ENABLE TRUTH TABLE CE1 0 1 X X = Don't Care CE2 1 1 0 BIT 1 - BIT 4 Valid Three-State Three-State DC, OF Valid Valid Three-State 7 HI3304 TABLE 1. OUTPUT CODE TABLE INPUT VOLTAGE (V) CODE DESCRIPTION Zero 1 LSB 2 LSB • • • • 1/ Full Scale -1 LSB 2 1/ Full Scale 2 1/ Full Scale +1 LSB 2 OUTPUT CODE 3.2V 0V 0 0.2 0.4 • • • • 1.4 1.6 1.8 • • • • 2.8 3.0 3.2 0.2 4.8V 0V 0 0.3 0.6 • • • • 2.1 2.4 2.7 • • • • 4.2 4.5 4.8 0.3 OF 0 0 0 • • • • 0 0 0 • • • • 0 0 1 B4 0 0 0 • • • • 0 1 1 • • • • 1 1 1 B3 0 0 0 • • • • 1 0 0 • • • • 1 1 1 B2 0 0 1 • • • • 1 0 0 • • • • 1 1 1 B1 0 1 0 • • • • 1 0 1 • • • • 0 1 1 DECIMAL COUNT 0 1 2 • • • • 7 8 9 • • • • 14 15 31 VREF + = 1V VREF - = -1V -1.000 -0.875 -0.750 • • • • -0.125 0 0.125 • • • • 0.750 0.875 1.000 0.125 1.6V 0V 0 0.1 0.2 • • • • 0.7 0.8 0.9 • • • • 1.4 1.5 1.6 0.1 2V 0V 0 0.125 0.250 • • • • 0.875 1.000 1.125 • • • • 1.750 1.875 2.000 0.125 • • • • Full Scale -1 LSB Full Scale Overflow Step Size NOTE: 1. The voltages listed are the ideal centers of each output code shown as a function of its associated reference voltage. See Ideal Transfer Curve Figure 6. The output code should exist for an input equal to the ideal center voltage ±1/2 of the step size. Description Device Operation A sequential parallel technique is used by the HI3304 converter to obtain its high speed operation. The sequence consists of the “Auto Balance” phase and the “Sample Unknown” phase (Refer to the circuit diagram). Each conversion takes one clock cycle (see Note). The “Auto Balance” (φ1) occurs during the Low period of the clock cycle, and the “Sample Unknown” (φ2) occurs during the High period of the clock cycle. NOTE: This device requires only a single-phase clock. The terminology of φ1 and φ2 refers to the High and Low periods of the same clock. charge to their associated tap voltages, priming the circuit for the next phase. In the “Sample Unknown” phase, all ladder tap switches are opened, the comparator amplifiers are no longer shorted, and VIN is switched to all 16 capacitors. Since the other end of the capacitor is now looking into an effectively open circuit, any voltage that differs from the previous tap voltage will appear as a voltage shift at the comparator amplifiers. All comparators whose tap voltages were lower than VIN will drive the comparator outputs to a “low” state. All comparators whose tap voltages were higher than VIN will drive the comparator outputs to a “high” state. A second, capacitorcoupled, auto-zeroed amplifier further amplifies the outputs. The status of all these comparator amplifiers are stored at the end of this phase (φ2), by a secondary latching amplifier stage. Once latched, the status of the 16 comparators is decoded by a 16 to 5 bit decode array and the results are clocked into a storage register at the rising edge of the next φ2. If the input is greater than 31/32 x VREF , the overflow output will go “high”. (The bit outputs will remain high). If the output differs from that of the previous conversion, the data change output will go “high”. A three-state buffer is used at the output of the 7 storage registers which are controlled by two chip-enable signals. CE1 will independently disable B1 through B4 when it is in a high state. CE2 will independently disable B1 through B4 and the OF and DC buffers when it is in the low state. During the “Auto Balance” phase, a transmission-gate switch is used to connect each of 16 commutating capacitors to their associated ladder reference tap. Those tap voltages will be as follows: VTAP(N) = [(VREF/16) x N] - [VREF/(2 x 16)] = VREF [(2N - 1)/32] Where: VTAP(N) = Reference ladder tap voltage at point N. VREF = Voltage across VREF - to VREF + N = Tap number (1 through 16) The other side of the capacitor is connected to a singlestage inverting amplifier whose output is shorted to its input by a switch. This biases the amplifier at its intrinsic trip point, which is approximately (VDD - VSS)/2. The capacitors now 8 HI3304 Continuous Clock Operation One complete conversion cycle can be traced through the HI3304 via the following steps. (Refer to timing diagram Figure 3). The rising edge of the clock input will start a “sample” phase. During this entire “High” state of the clock, the 16 comparators will track the input voltage and the 16 latches will track the comparator outputs. At the falling edge of the clock, all 16 comparator outputs are captured by the 16 latches. This ends the “sample” phase and starts the “auto balance” phase for the comparators. During this “Low” state of the clock the output of the latches propagates through the decode array and a 6-bit code appears at the D inputs of the output registers. On the next rising edge of the clock, this 6-bit code is shifted into the output registers and appears with time delay tD as valid data at the output of the three-state drivers. This also marks the start of a new “sample” phase, thereby repeating the conversion process for this next cycle. Pulse Mode Operation For sampling high speed nonrecurrent or transient data, the converter may be operated in a pulse mode in one of three ways. The fastest method is to keep the converter in the Sample Unknown phase, φ2, during the standby state. The device can now be pulsed through the Auto Balance phase with as little as 20ns. The analog value is captured on the leading edge of φ1 and is transferred into the output registers on the trailing edge of φ1. We are now back in the standby state, φ2, and another conversion can be started within 20ns, but not later than 5µs due to the eventual droop of the commutating capacitors. Another advantage of this method is that it has the potential of having the lowest power drain. The larger the time ratio between φ2 and φ1, the lower the power consumption. (See Timing Diagram Figure 3A). The second method uses the Auto Balance phase, φ1, as the standby state. In this state the converter can stay indefinitely waiting to start a conversion. A conversion is performed by strobing the clock input with two φ2 pulses. The first pulse starts a Sample Unknown phase and captures the analog value in the comparator latches on the trailing edge. A second φ2 pulse is needed to transfer the date into the output registers. This occurs on the leading edge of the second pulse. The conversion now takes place in 40ns, but the repetition rate may be as slow as desired. The disadvantage to this method is the slightly higher device dissipation due to the low ratio of φ2 to φ1. (See Timing Diagram Figure 3B). For applications requiring both indefinite standby and lowest power, standby can be in the φ2 (Sample Unknown) state with two φ1 pulses to generate valid data (see Figure 3C). The conversion process now takes 60ns. [Note that the above numbers do not include the tD (Output Delay) time.] Increased Accuracy In most case the accuracy of the HI3304 should be sufficient without any adjustments. In applications where accuracy is of utmost importance, two adjustments can be made to obtain better accuracy; i.e., offset trim and gain trim. Offset Trim In general offset correction can be done in the preamp circuitry by introducing a DC shift to VIN or by the offset trim of the op amp. When this is not possible the VREF - input can be adjusted to produce an offset trim. 1/ LSB. The equation is as follows: 2 The theoretical input voltage to produce the first transition is VIN (0 to 1 transition) = 1/2 LSB = 1/2(VREF/16) = VREF/32 Adjust offset by applying this input voltage and adjusting the VREF - voltage or input amplifier offset until an output code alternating between 0 and 1 occurs. Gain Trim In general the gain trim can also be done in the preamp circuitry by introducing a gain adjustment for the op amp. When this is not possible, then a gain adjustment circuit should be made to adjust the reference voltage. To perform this trim, VIN should be set to the 15 to overflow transition. That voltage is 1/2 LSB less than VREF + and is calculated as follows: VlN (15 to 16 transition) = VREF - VREF/32 = VREF (31/32) To perform the gain trim, first do the offset trim and then apply the required VIN for the 15 to overflow transition. Now adjust VREF+ until that transition occurs on the outputs. Layout, Input and Supply Considerations The HI3304 should be mounted on a ground-planed, printed-circuit board, with good high-frequency decoupling capacitors mounted as close as possible. If the supply is noisy, decouple VAA+ with a resistor as shown in Figure 12A. The HI3304 outputs current spikes to its input at the start of the auto-balance and sample clock phases. A low impedance source, such as a locally-terminated 50Ω coax cable, should be used to drive the input terminal. A fastsettling buffer such as the HA-5033, HA-5242, or CA3450 should be used if the source is high impedance. The VREF terminals also have current spikes, and should be well bypassed. Care should be taken to keep digital signals away from the analog input, and to keep digital ground currents away from the analog ground. If possible, the analog ground should be connected to digital ground only at the HI3304. Bipolar Operation The HI3304, with separate analog (VAA+, VAA-) and digital (VDD , VSS) supply pins, allows true bipolar or negative input operation. The VAA- pin may be returned to a negative supply (observing maximum voltage ratings to VAA+ or VDD and recommended rating to VSS), thus allowing the VREF potential also to be negative. Figure 12B shows operation with an input range of -1V to +1V. Similarly, VAA+ and VREF + could be maintained at a higher voltage than VDD , for an input range above the digital supply. 9 HI3304 Digital Input and Output Levels The clock input is a CMOS inverter operating from and with logic input levels determined by the VAA supplies. If VAA+ or VAA- are outside the range of the digital supplies, it may be necessary to level shift the clock input to meet the required 30% to 70% of VAA input swing. Figure 12B shows an example for a negative VAA-. An alternate way of driving the clock is to capacitively couple the pin from a source of at least 1VP-P . An internal 50kΩ feedback resistor will keep the DC level at the intrinsic trip point. Extremely non-symmetrical clock waveforms should be avoided, however. The remaining digital inputs and outputs are referenced to VDD and VSS . If TTL or other lower voltage sources are to drive the HI3304, either pull-up resistors or CD74HCT series “QMOS” buffers are recommended. 5-Bit Resolution To obtain 5-bit resolution, two HI3304s can be wired together. Necessary ingredients include an open-ended ladder network, an overflow indicator, three-state outputs, and chipenable controls - all of which are available on the HI3304. The first step for connecting a 5-bit circuit is to totem-pole the ladder networks, as illustrated in Figure 13. Since the absolute-resistance value of each ladder may vary, external trim of the mid-reference voltage may be required. The overflow output of the lower device now becomes the fifth bit. When it goes high, all counts must come from the upper device. When it goes low, all counts must come from the lower device. This is done simply by connecting the lower overflow signal to the CE1 control of the lower A/D converter and the CE2 control of the upper A/D converter. The three-state outputs of the two devices (bits 1 through 4) are now connected in parallel to complete the circuitry. 27Ω +5V SUPPLY HI3304 4.7µF TAN 2V REFERENCE 4.7µF TAN REMOTE 2V INTO 50Ω SOURCE 50Ω + 0.1µF CER 0.1µF CER VAA+ VDD CE2 VREF+ DC, OF, B1-B4 VIN CLK VREF- CE1 VAAVSS 0.1µF CER + 4.7µF TAN + OUTPUT DATA CMOS CLOCK SOURCE ANALOG GROUND DIGITAL GROUND FIGURE 12A. TYPICAL HI3304 UNIPOLAR CIRCUIT CONFIGURATION 27Ω +5V SUPPLY HI3304 4.7µF TAN + 0.1µF CER VAA+ VDD CE2 VREF + DC, OF, VIN B1 - B4 -1V REFERENCE 50Ω 0.1µF CER 0.1µF CER VREF - CLK CE1 VSS 10kΩ 0.1µF CER + 4.7µF TAN +1V REFERENCE REMOTE ±1V INTO 50Ω SOURCE 0.1µF CER OUTPUT DATA IN914 CMOS CLOCK SOURCE 0.001µF -5V SUPPLY VAA4.7µF TAN ANALOG GROUND DIGITAL GROUND FIGURE 12B. TYPICAL HI3304 BIPOLAR CIRCUIT CONFIGURATION 10 HI3304 +5V VAA+ VDD +FULL SCALE REF. VREF + VIN VREF VAAVSS CLK BUFFER INPUT 1kΩ ADJUST CENTER +5V DC OF B4 B3 B2 B1 CE1 CLOCK INPUT NC DC OF B4 B3 B2 B4 B3 B2 B1 +5V B5 MSB DECIMAL COUNT CE2 15 14 13 12 11 10 9 8 7 6 5 4 3 2 1 0 V0 V1 V2 V3 V4 V5 V6 V7 V8 V9 V10 V11 V12 V13 V14 V15 V16 INPUT VOLTAGE HI3304 NC NC OVERFLOW HI3304 CLK VAA+ VDD VREF + VIN VREF B1 VAACE1 VSS CE2 FIGURE 13. TYPICAL HI3304 5-BIT CONFIGURATION FIGURE 14. IDEAL TRANSFER CURVE Definitions Dynamic Performance Definitions Fast Fourier Transform (FFT) techniques are used to evaluate the dynamic performance of the HI3304. A low distortion sine wave is applied to the input, it is sampled, and the output is stored in RAM. The data is then transformed into the frequency domain with a 4096 point FFT and analyzed to evaluate the dynamic performance of the A/D. The sine wave input to the part is -0.5dB down from fullscale for all these tests. Signal-to-Noise (SNR) SNR is the measured RMS signal to RMS noise at a specified input and sampling frequency. The noise is the RMS sum of all of the spectral components except the fundamental and the first five harmonics. Signal-to-Noise + Distortion Ratio (SINAD) SINAD is the measured RMS signal to RMS sum of all other spectral components below the Nyquist frequency excluding DC. Effective Number of Bits (ENOB) The effective number of bits (ENOB) is derived from the SINAD data. ENOB is calculated from: ENOB = (SINAD - 1.76 + VCORR)/6.02, where: VCORR = 0.5dB. Total Harmonic Distortion (THD) THD is the ratio of the RMS sum of the first 5 harmonic components to the RMS value of the measured input signal. Operating and Handling Considerations HANDLING All inputs and outputs of CMOS devices have a network for electrostatic protection during handling. Recommended handling practices for CMOS devices are described in lCAN-6525. “Guide to Better Handling and Operation of CMOS Integrated Circuits.” OPERATING Operating Voltage During operation near the maximum supply voltage limit, care should be taken to avoid or suppress power supply turn-on and turn-off transients, power supply ripple, or ground noise; any of these conditions must not cause the power supply voltages to exceed the absolute maximum rating. Input Signals To prevent damage to the input protection circuit, input signals should never be greater than VDD or VAA+ nor less than VSS or VAA- (depending upon which supply the protection network is referenced. See Maximum Ratings). Input currents must not exceed 20mA even when the power supply is off. Unused Inputs A connection must be provided at every input terminal. All unused input terminals must be connected to either VDD or VSS , whichever is appropriate. Output Short Circuits Shorting of outputs to any supply potential may damage CMOS devices by exceeding the maximum device dissipation. 11 HI3304 All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation’s quality certifications can be viewed at www.intersil.com/design/quality Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries. For information regarding Intersil Corporation and its products, see www.intersil.com 12