IR3508MTRPBF

IR3508 DATA SHEET XPHASE3TM PHASE IC DESCRIPTION The IR3508 Phase IC combined with any IR XPhase3 Control IC provides a full featured and flexible way to implement a power solution for the latest high performance CPUs and ASICs. The “Control” IC provides overall system control and interfaces with any number of “Phase” ICs which each drive and monitor a single TM phase of a multiphase converter. The XPhase3 architecture results in a power supply that is smaller, less expensive, and easier to design while providing higher efficiency than conventional approaches. The IR3508 disables its current sense amplifiers when entering power savings mode. The recommended use for these Phase ICs is for applications without adaptive voltage positioning where two or more power stages will be operating in power savings mode. TM FEATURES IR3508 PHASE IC x x x x x x x x x x x x x x x x x Power State Indicator (PSI) interface provides the capability to maximize the efficiency at light loads. Anti-bias circuitry 7V/2A gate drivers (4A GATEL sink current) Support converter output voltage up to 5.1 V (Limited to VCCL-1.4V) Loss-less inductor current sensing Phase delay DFF bypassed during PSI assertion mode to improve output ripple performance Over-current protection during PSI assertion mode operation Feed-forward voltage mode control Integrated boot-strap synchronous PFET Only four external components per phase 3 wire analog bus connects Control and Phase ICs (VID, Error Amp, IOUT) 3 wire digital bus for accurate daisy-chain phase timing control without external components Debugging function isolates phase IC from the converter Self-calibration of PWM ramp, current sense amplifier, and current share amplifier Single-wire bidirectional average current sharing Small thermally enhanced 20L 4 X 4mm MLPQ package RoHS compliant APPLICATION CIRCUIT 12V EAIN 19 20 18 CSIN+ CSIN- EAIN 17 VCC NC 16 RCS SW 15 14 13 12 11 CBST L CCS IOUT PSI DACIN 1 2 3 4 5 IOUT PSI DACIN LGND PHSOUT GATEL PHSIN NC CLKIN PGND GATEH VOUT+ IR3508 BOOST VCCL NC COUT VOUT- 7 6 8 9 PHSIN PHSOUT CLKIN CVCCL VCCL Figure 1 Application Circuit Page 1 of 19 October 27, 2008 10 IR3508 ORDERING INFORMATION Part Number IR3508MTRPBF * IR3508MPBF * Samples only Package 20 Lead MLPQ (4 x 4 mm body) 20 Lead MLPQ (4 x 4 mm body) Order Quantity 3000 per reel 100 piece strips ABSOLUTE MAXIMUM RATINGS Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only and functional operation of the device, at these or any other conditions, beyond those indicated in the operational sections of the specifications are not implied. Operating Junction Temperature…………….. 0 to 150 C o o Storage Temperature Range………………….-65 C to 150 C MSL Rating………………………………………2 o Reflow Temperature…………………………….260 C PIN # 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 PIN NAME IOUT PSI DACIN LGND PHSIN NC PHSOUT CLKIN PGND GATEL NC VCCL BOOST GATEH SW VMAX 8V 8V 3.3V n/a 8V n/a 8V 8V 0.3V 8V n/a 8V 40V 40V 34V VMIN -0.3V -0.3V -0.3V n/a -0.3V n/a -0.3V -0.3V -0.3V -0.3V DC, -5V for 100ns n/a -0.3V -0.3V -0.3V DC, -5V for 100ns -0.3V DC, -5V for 100ns -0.3V -0.3V -0.3V -0.3V n/a ISOURCE 1mA 1mA 1mA n/a 1mA n/a 2mA 1mA 5A for 100ns, 200mA DC 5A for 100ns, 200mA DC n/a n/a 1A for 100ns, 100mA DC 3A for 100ns, 100mA DC 3A for 100ns, 100mA DC n/a 1mA 1mA 1mA n/a ISINK 1mA 1mA 1mA n/a 1mA n/a 2mA 1mA n/a 5A for 100ns, 200mA DC n/a 5A for 100ns, 200mA DC 3A for 100ns, 100mA DC 3A for 100ns, 100mA DC n/a 10mA 1mA 1mA 1mA n/a o 16 VCC 34V 17 CSIN+ 8V 18 CSIN8V 19 EAIN 8V 20 NC n/a Note: 1. Maximum GATEH – SW = 8V 2. Maximum BOOST – GATEH = 8V Page 2 of 19 October 27, 2008 IR3508 RECOMMENDED OPERATING CONDITIONS FOR RELIABLE OPERATION WITH MARGIN o o 8.0V ” VCC ” 28V, 4.75V ” VCCL ” 7.5V, 0 C ” TJ ” 125 C. 0.5V ” 9'$&,1 ” 9, 500kHz ” &/.,1 ” 0+], 250kHz ” 3+6,1 ”0+]. ELECTRICAL CHARACTERISTICS The electrical characteristics table lists the parametric range guaranteed to be within the recommended operating conditions. Typical values represent the median values, which are related to 25°C. CGATEH = 3.3nF, CGATEL = 6.8nF (unless otherwise specified) PARAMETER Gate Drivers GATEH Source Resistance GATEH Sink Resistance GATEL Source Resistance GATEL Sink Resistance GATEH Source Current GATEH Sink Current GATEL Source Current GATEL Sink Current GATEH Rise Time GATEH Fall Time GATEL Rise Time GATEL Fall Time GATEL low to GATEH high delay GATEH low to GATEL high delay Disable Pull-Down Resistance Clock CLKIN Threshold CLKIN Bias Current CLKIN Phase Delay PHSIN Threshold PHSOUT Propagation Delay PHSIN Pull-Down Resistance PHSOUT High Voltage PHSOUT Low Voltage Page 3 of 19 TEST CONDITION BOOST – SW = 7V. Note 1 BOOST – SW = 7V. Note 1 VCCL – PGND = 7V. Note 1 VCCL – PGND = 7V. Note 1 BOOST=7V, GATEH=2.5V, SW=0V. BOOST=7V, GATEH=2.5V, SW=0V. VCCL=7V, GATEL=2.5V, PGND=0V. VCCL=7V, GATEL=2.5V, PGND=0V. BOOST – SW = 7V, measure 1V to 4V transition time BOOST – SW = 7V, measure 4V to 1V transition time VCCL – PGND = 7V, Measure 1V to 4V transition time VCCL – PGND = 7V, Measure 4V to 1V transition time BOOST = VCCL = 7V, SW = PGND = 0V, measure time from GATEL falling to 1V to GATEH rising to 1V BOOST = VCCL = 7V, SW = PGND = 0V, measure time from GATEH falling to 1V to GATEL rising to 1V Note 1 MIN TYP 1.0 1.0 1.0 0.4 2.0 2.0 2.0 4.0 5 5 10 5 10 10 30 20 20 80 MAX 2.5 2.5 2.5 1.0 UNIT Ÿ Ÿ Ÿ Ÿ A A A A ns ns ns ns ns ns N 10 10 20 10 40 40 130 Compare to V(VCCL) CLKIN = V(VCCL) Measure time from CLKIN1V Compare to V(VCCL) Measure time from CLKIN > (VCCL * 50% ) to PHSOUT > (VCCL *50%). 10pF Load @ o 125 C 40 -0.5 40 35 4 30 45 0.0 75 50 15 100 0.6 0.4 57 0.5 125 55 35 170 % PA ns % ns N V I(PHSOUT) = -10mA, measure VCCL – PHSOUT I(PHSOUT) = 10mA 1 1 V October 27, 2008 IR3508 PARAMETER PWM Comparator PWM Ramp Slope EAIN Bias Current Minimum Pulse Width Current Sense Amplifier CSIN+/- Bias Current CSIN+/- Bias Current Mismatch Input Offset Voltage Gain Unity Gain Bandwidth Slew Rate Differential Input Range Differential Input Range Common Mode Input Range o Rout at TJ = 25 C o Rout at TJ = 125 C IOUT Source Current IOUT Sink Current Share Adjust Amplifier Input Offset Voltage Gain Unity Gain Bandwidth PWM Ramp Floor Voltage Maximum PWM Ramp Floor Voltage Minimum PWM Ramp Floor Voltage PSI Comparator Rising Threshold Voltage Falling Threshold Voltage Hysteresis Resistance Floating Voltage TEST CONDITION Vin=12V 0 ” EAIN ” 3V Note 1 MIN 42 -5 TYP 52.5 -0.3 55 0 0 0 32.5 6.8 6 -10 -5 0 2.3 3.6 0.5 0.5 -3 4 4 -116 120 -220 50 50 Note2 3.7 5.4 2.9 2.9 3 6 17 116 240 -100 MAX 57 5 70 200 50 1 35.0 8.8 UNIT mV/ %DC PA ns nA nA mV V/V MHz V/Ps mV mV V kŸ kŸ mA mA mV V/V kHz mV mV mV Note 1 CSIN+ = CSIN- = DACIN. Measure input referred offset from DACIN 0.5V ” 9'$&,1  9 C(IOUT)=10pF. Measure at IOUT. Note 1 0.8V ” 9'$&,1 ” 9 1RWH  0.5V ” 9'$&,1  9 1RWH  Note 1 Note 1 -200 -50 -1 30.0 4.8 3.0 4.7 1.6 1.4 0 5.0 8.5 0 180 -160 Note 1 CSIN+ = CSIN- = DACIN. Note 1 Note 1 IOUT Open, Measure relative to DACIN IOUT = DACIN – 200mV. Measure relative to floor voltage. IOUT = DACIN + 200mV. Measure relative to floor voltage. Note 1 Note 1 Note 1 520 400 50 200 800 620 550 70 500 700 650 120 850 1150 mV mV mV kŸ mV Page 4 of 19 October 27, 2008 IR3508 PARAMETER Body Brake Comparator Threshold Voltage with EAIN decreasing Threshold Voltage with EAIN increasing Hysteresis Propagation Delay OVP Comparator OVP Threshold TEST CONDITION Measure relative to Floor Voltage Measure relative to Floor Voltage MIN -300 -200 70 40 TYP -200 -100 105 65 MAX -110 -10 130 90 UNIT mV mV mV ns VCCL = 5V. Measure time from EAIN < V(DACIN) (200mV overdrive) to GATEL transition to < 4V. Step V(IOUT) up until GATEL drives high. Compare to V(VCCL) Propagation Delay V(VCCL)=5V, Step V(IOUT) up from V(DACIN) to V(VCCL). Measure time to V(GATEL)>4V. Synchronous Rectification Disable Comparator Threshold Voltage The ratio of V(CSIN-) / V(DACIN), below which V(GATEL) is always low. Negative Current Comparator Input Offset Voltage Note1 Propagation Delay Time Apply step voltage to V(CSIN+) – V(CSIN-). Measure time to V(GATEL)< 1V. Bootstrap Diode Forward Voltage I(BOOST) = 30mA, VCCL = 6.8V Debug Comparator Threshold Voltage Compare to V(VCCL) General VCC Supply Current 8V ” 9(VCC) < 10V VCC Supply Current 10V ” 9(VCC) ” 9 VCCL Supply Current BOOST Supply Current 4.75V ” 9(BOOST)-V(SW )” 8V DACIN Bias Current SW Floating Voltage Note 1: Guaranteed by design, but not tested in production Note 2: VCCL-0.5V or VCC – 2.5V, whichever is lower -1.0 15 -0.8 40 -0.4 70 V ns 66 75 86 % -16 100 0 200 16 400 mV ns 360 -250 1.1 1.1 3.1 0.5 -1.5 0.1 520 -150 4.0 2.0 8.0 1.5 -0.75 0.3 960 -50 6.1 4 12.1 3 1 0.4 mV mV mA mA mA mA PA V Page 5 of 19 October 27, 2008 IR3508 PIN DESCRIPTION PIN# 1 PIN SYMBOL IOUT PIN DESCRIPTION Output of the Current Sense Amplifier is connected to this pin through a 3kŸ resistor. Voltage on this pin is equal to V(DACIN) + 33 [V(CSIN+) – V(CSIN-)]. Connecting all IOUT pins together creates a share bus which provides an indication of the average current being supplied by all the phases. The signal is used by the Control IC for voltage positioning and over-current protection. OVP mode is initiated if the voltage on this pin rises above V(VCCL)- 0.8V. Logic low is an active low (i.e. low = low power state). Reference voltage input from the Control IC. The Current Sense signal and PWM ramp is referenced to the voltage on this pin. Ground for internal IC circuits. IC substrate is connected to this pin. Phase clock input. No connection. Phase clock output. Clock input. Return for low side driver and reference for GATEH non-overlap comparator. Low-side driver output and input to GATEH non-overlap comparator. No connection. Supply for low-side driver. Internal bootstrap synchronous PFET is connected from this pin to the BOOST pin. Supply for high-side driver. Internal bootstrap synchronous PFET is connected between this pin and the VCCL pin. High-side driver output and input to GATEL non-overlap comparator. Return for high-side driver and reference for GATEL non-overlap comparator. Supply for internal IC circuits. Non-Inverting input to the current sense amplifier, and input to debug comparator. Inverting input to the current sense amplifier, and input to synchronous rectification disable comparator. PWM comparator input from the error amplifier output of Control IC. Body Braking mode is initiated if the voltage on this pin is less than V(DACIN). No connection. 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 17 18 19 20 PSI DACIN LGND PHSIN NC PHSOUT CLKIN PGND GATEL NC VCCL BOOST GATEH SW VCC CSIN+ CSINEAIN NC Page 6 of 19 October 27, 2008 IR3508 SYSTEM THEORY OF OPERATION System Description The system consists of one control IC and a scalable array of phase converters, each requiring one phase IC. The control IC communicates with the phase ICs using three digital buses, i.e., CLOCK, PHSIN, PHSOUT and three analog buses, i.e., DAC, EA, and IOUT. The digital buses are responsible for switching frequency determination and accurate phase timing control without any external components. The analog buses are used for PWM control and current sharing between interleaved phases. The control IC incorporates all the system functions, i.e., VID, CLOCK signals, error amplifier, fault protections, current monitor, etc. The Phase IC implements the functions required by the converter of each phase, i.e., the gate drivers, PWM comparator and latch, over-voltage protection, phase disable circuit, current sensing and sharing, etc. PWM Control Method The PWM block diagram of the XPhase3 architecture is shown in Figure 1. Feed-forward voltage mode control with trailing edge modulation is used. A high-gain and wide-bandwidth voltage type error amplifier is implemented in the controller’s design to achieve a fast voltage control loop. Input voltage is sensed by the phase ICs to provide feedforward control. The feed-forward control compensates the ramp slope based on the change in input voltage. The input voltage can change due to variations in the silver box output voltage or due to the wire and PCB-trace voltage drop related to changes in load current. TM CONTROL IC CLOCK GENERATOR CLKOUT GATE DRIVE VOLTAGE PHSOUT CLKIN VIN PHASE IC CLK Q D VCC VCCH PHSOUT PHSIN PHSIN 1 2 RESET DOMINANT D Q R GATEH VID6 OFF SW CBST VOSNS+ VOUT COUT PWM COMPARATOR EAIN CLK Q + DFFRH VCCL GND GATEL VID6 OFF PGND VOSNS- PWM LATCH VID6 VO VDAC LGND SHARE ADJUST ERROR AMPLIFIER IOUT RCOMP RFB1 RFB RAMP DISCHARGE CLAMP + + FB RVSETPT CCOMP CFB DACIN PHSOUT RDRP RDRP1 IROSC IVSETPT VSETPT CDRP PHASE IC CLK Q D 1 2 RESET DOMINANT U248 D Q R IMON VDAC CLKIN PHSIN VDRP AMP Thermal Compensation + - VDRP VN RTHRM PWM COMPARATOR EAIN ENABLE CLK Q + VID6 OFF SW VCCL GATEL VID6 OFF + - DFFRH PWM LATCH VID6 + IIN RAMP DISCHARGE CLAMP BODY BRAKING COMPARATOR SHARE ADJUST ERROR AMPLIFIER ISHARE 3K VID6 VID6 + - CURRENT SENSE AMPLIFIER + VID6 VID6 + + DACIN Figure 1: PWM Block Diagram Page 7 of 19 October 27, 2008 - VID6 VID6 + + ERROR AMPLIFIER 3K - - - EAOUT + VDAC + - - - VID6 VID6 + + - + - + - REMOTE SENSE AMPLIFIER ENABLE BODY BRAKING COMPARATOR + - PSI PSI CURRENT SENSE AMPLIFIER CSIN+ CCS RCS CSIN- VCC VCCH GATEH CBST PGND PSI PSI CSIN+ CCS RCS CSIN- IR3508 Frequency and Phase Timing Control The oscillator is located in the Control IC and the system clock frequency is programmable from 250kHz to 9MHZ by an external resistor. The control IC system clock signal (CLKOUT) is connected to CLKIN of all the phase ICs. The phase timing of the phase ICs is controlled by the daisy chain loop, where the control IC phase clock output (PHSOUT) is connected to the phase clock input (PHSIN) of the first phase IC, and PHSOUT of the first phase IC is connected to PHSIN of the second phase IC, etc. The last phase IC is connected back to PHSIN of the control IC to complete the daisy chain loop. During power up, the control IC sends out clock signals from both CLKOUT and PHSOUT pins and detects the feedback at PHSIN pin to determine the phase number and monitor any fault in the daisy chain loop. When the PSI is asserted (active low), the phases are effectively removed from the daisy chain loop. Figure 2 shows the phase timing for a four phase converter. The switching frequency is set by the resistor ROSC. The clock frequency equals the number of phase times the switching frequency. Control IC CLKOUT (Phase IC CLKIN) Control IC PHSOUT (Phase IC1 PHSIN) P hase IC1 PWM Latch SET Phase IC 1 PHSOUT (Phase IC2 PHSIN) Phase IC 2 PHSOUT (Phase IC3 PHSIN) Phase IC 3 PHSOUT (Phase IC4 PHSIN) Phase IC4 PHSOUT (Control IC PHSIN) Figure 2: Four Phase Oscillator Waveforms PWM Operation The PWM comparator is located in the phase IC. Upon receiving the falling edge of a clock pulse, the PWM latch is set and the PWM ramp voltage begins to increase. In addition, the low side driver is turned off and the high side driver is turned on after the non-overlap time expires (GATEL < 1V). When the PWM ramp voltage exceeds the error amplifier’s output voltage, the PWM latch is reset and the internal ramp capacitor is quickly discharged to the output of the share adjust amplifier and remains discharged until the next clock pulse. This reset latch additionally turns off the high side driver and enables the low side driver after the non-overlap time concludes (Switch Node < 1V). The PWM latch is reset dominant allowing all phases to go to zero duty cycle within a few tens of nanoseconds in response to a load step decrease. Phases can overlap and go up to 100% duty cycle in response to a load step increase with turn-on gated by the clock pulses. An error amplifier output voltage greater than the common mode input range, of the PWM comparator, results in 100% duty cycle regardless of the voltage of the PWM ramp. This arrangement guarantees that the error amplifier is always in control and can demand 0 to 100% duty cycle as required. It also favors response to a load step decrease, which is appropriate, given that the low output to input voltage ratio of most systems. The inductor current will increase much more rapidly than decrease in response to load transients. This control method is designed to provide “single cycle transient response.” The inductor current will change in response to load transients within a single switching cycle maximizing the effectiveness of the power train and minimizing the output capacitor requirements. An additional advantage of the architecture is that differences in ground or input voltage, at the phases, have no effect on operation since the PWM ramps are referenced to VDAC. Figure 3 depicts PWM operating waveforms under various conditions. Page 8 of 19 October 27, 2008 IR3508 PHASE IC CLOCK PULSE EAIN PWMRMP VDAC GATEH GATEL STEADY-STATE OPERATION DUTY CYCLE INCREASE DUE TO LOAD INCREASE DUTY CYCLE DECREASE DUE TO VIN INCREASE (FEED-FORWARD) DUTY CYCLE DECREASE DUE TO LOAD DECREASE (BODY BRAKING) OR FAULT (VCCLUV, OCP, VID=11111X) STEADY-STATE OPERATION Figure 3: PWM Operating Waveforms Body Braking TM In a conventional synchronous buck converter, the minimum time required to reduce the current in the inductor in response to a load step decrease is; TSLEW L * ( I MAX  I MIN ) VO The slew rate of the inductor current can be significantly increased by turning off the synchronous rectifier in response to a load step decrease. The switch node voltage is then forced to decrease until conduction of the synchronous rectifier’s body diode occurs. This increases the voltage across the inductor from Vout to Vout + VBODYDIODE. The minimum time required to reduce the current in the inductor in response to a load transient decrease is now; TSLEW L * ( I MAX  I MIN ) VO  VBODYDIODE Since the voltage drop in the body diode is often comparable to the output voltage, the inductor current slew rate can be increased significantly. This patented technique is referred to as “body braking” and is accomplished through the “body braking comparator” located in the phase IC. If the error amplifier’s output voltage drops below the output voltage of the share adjust amplifier in the phase IC, this comparator turns off the low side gate driver. Lossless Average Inductor Current Sensing Inductor current can be sensed by connecting a series resistor and a capacitor network in parallel with the inductor and measuring the voltage across the capacitor, as shown in Figure 4. The equation of the sensing network is, vC ( s ) vL ( s ) 1 1  sRCS CCS iL ( s ) RL  sL 1  sRCS CCS Page 9 of 19 October 27, 2008 IR3508 Usually the resistor Rcs and capacitor Ccs are chosen so that the time constant of Rcs and Ccs equals the time constant of the inductor which is the inductance L over the inductor DCR (RL). If the two time constants match, the voltage across Ccs is proportional to the current through L, and the sense circuit can be treated as if only a sense resistor with the value of RL was used. The mismatch of the time constants does not affect the measurement of inductor DC current, but affects the AC component of the inductor current. vL iL L RCS Current Sense Amp RL CCS c vCS VO CO CSOUT Figure 4: Inductor Current Sensing and Current Sense Amplifier The advantage of sensing the inductor current versus high side or low side sensing is that actual output current being delivered to the load is obtained rather than peak or sampled information about the switch currents. The output voltage can be positioned to meet a load line based on real time information. Except for a sense resistor in series with the inductor, this is the only sense method that can support a single cycle transient response. Other methods provide no information during either load increase (low side sensing) or load decrease (high side sensing). An additional problem associated with peak or valley current mode control for voltage positioning is that they suffer from peak-to-average errors. These errors will show in many ways but one example is the effect of frequency variation. If the frequency of a particular unit is 10% low, the peak to peak inductor current will be 10% larger and the output impedance of the converter will drop by about 10%. Variations in inductance, current sense amplifier bandwidth, PWM prop delay, any added slope compensation, input voltage, and output voltage are all additional sources of peak-to-average errors. Current Sense Amplifier A high speed differential current sense amplifier is located in the phase IC, as shown in Figure 4. Its gain is nominally 32.5, and the 3850 ppm/ºC increase in inductor DCR should be compensated in the voltage loop feedback path. The current sense amplifier can accept positive differential input up to 50mV and negative up to -10mV before clipping. The output of the current sense amplifier is summed with the DAC voltage and sent to the control IC and other phases through an on-chip 3KŸ UHVLVWRU FRQQHFWHG WR WKH ,287 SLQ 7KH ,287 SLQV RI DOO WKH SKDVHV DUH WLHG WRJHWKHU DQG WKH voltage on the share bus represents the average current through all the inductors and is used by the control IC for voltage positioning and current limit protection. The input offset of this amplifier is calibrated to +/- 1mV in order to reduce the current sense error. The input offset voltage is the primary source of error for the current share loop. In order to achieve very small input offset error and superior current sharing performance, the current sense amplifier continuously calibrates itself. This calibration algorithm creates ripple on IOUT bus with a frequency of fsw/896 in a multiphase architecture. Average Current Share Loop Current sharing between phases of the converter is achieved by the average current share loop in each phase IC. The output of the current sense amplifier is compared with the average current at the share bus. If current in a phase is smaller than the average current, the share adjust amplifier of the phase will pull down the starting point of the PWM ramp thereby increasing its duty cycle and output current; if current in a phase is larger than the average current, the share adjust amplifier of the phase will pull up the starting point of the PWM ramp thereby decreasing its duty cycle and output current. The current share amplifier is internally compensated so that the crossover frequency of the current Page 10 of 19 October 27, 2008 IR3508 share loop is much slower than that of the voltage loop and the two loops do not interact. For proper current sharing the output of current sense amplifier should not exceed (VCCL-1.4V) under all operating condition. IR3508 THEORY OF OPERATION Block Diagram The Block diagram of the IR3508 is shown in Figure 5, and specific features are discussed in the following sections. PSI_SY NC CLKIN . . . CLK Q D PHSOUT PHSIN D CLK Q PHASE DELAY DFF EAIN EAIN PWM COMPARATOR + PWM LATCH D PWM_CLK U183 Q R PWMQ . . . . . GATEH DRIVER BOOST GATEH SW CLK Q RESET DOMINANT DFFRSH GATEH NONGATEH NONOVERLAP OVERLAP LATCH COMPARATOR Q S + SET R DOMINANT RMPOUT PHSIN VCCL PWM RESET VCC VCC CALIBRATION DACIN-SHARE_ADJ PWM RAMP GENERATOR ANTI-BIAS LATCH Q R D CLK GATEL NONOVERLAP LATCH Q S 1V GATEL NONOVERLAP COMPARATOR 1V + + EAIN DACIN + SHARE_ADJ - 100mV 200mV OVP COMPARATOR VCCL + + NEGATIVE CURRENT LATCH Q RESET DOMINANT R S PSI_SYNC 0.8V IOUT DEBUG OFF (LOW=OPEN) - CURRENT SENSE AMPLIFIER X32.5 + 3K CSAOUT DEBUG COMPARATOR + IROSC + DACIN + X 0.75 CALIBRATION DACIN LGND PSI_SY NC Q R D CLK VCCL CALIBRATION 1V 8CLK Q R D CLK VCCL IROSC (CLKIN for 1-PHASE) PHSIN PSI COMPARATOR + 610mV 510mV Figure 5: Block diagram Tri-State Gate Drivers The gate drivers are design to provide a 2A source and sink peak current (Bottom gate driver can sink 4A). An adaptive non-overlap circuit monitors the voltage on the GATEH and GATEL pins to prevent MOSFET shootthrough current and minimizing body diode conduction. The non-overlap latch is added to eliminate erroneous triggering caused by the switching noise. A fault condition is communicated to the phase IC via the control IC’s error amplifier without an additional dedicated signal line. The error amplifier’s output is driven low in response to any fault condition detected by the controller, such as VCCL under voltage or output overload, disabling the phase IC TM TM and activating Body Braking . The IR3508 Body Braking comparator detects the low signal at the EAIN and drives the bottom gate output low. This tri-state operation prevents negative inductor current and negative output voltage during power-down. A synchronous rectification disable comparator is used to detect the converter’s CSIN- pin voltage, which represents local converter output voltage. If the voltage is below 75% of VDAC and negative current is detected, GATEL is driven low, which disables synchronous rectification and eliminates negative current during power-up. Page 11 of 19 October 27, 2008 - SHARE ADJUST AMPLIFIER + - NEGATIVE CURRENT COMPARATOR - + SYNCHRONOUS RECTIFICATION DISABLE COMPARATOR - BODY BRAKING COMPARATOR PSI_SY NC SET R DOMINANT GATEL DRIVER VCCL GATEL PGND 0.15V CSINCSIN+ 500K PSI IR3508 The gate drivers are pulled low if the supply voltage falls below the normal operating range. An 80kŸ UHVLVWRU LV connected across the GATEH/GATEL and PGND pins to prevent the GATEH/GATEL voltage from rising due to leakage or other causes under these conditions. PWM Ramp Every time the phase IC is powered up, the PWM ramp magnitude is calibrated to generate a 52.5 mV/% ramp (VCC=12V). For example, a 15 % duty ratio will generate a ramp amplitude of 787.5 mV (15 x 52.5 mV) with 12V supply applied to VCC. Feed-forward control is achieved by varying the PWM ramp proportionally with VCC voltage after calibration. In response to a load step-up, the error amplifier can demand 100 % duty cycle. As shown in Figure 6, 100 % duty is detected by comparing the PWM latch output (PWMQ) and its input clock (PWM_CLK). If the PWMQ is high when the PWM_CLK is asserted, the top FET turnoff is initiated. The top FET is again turned on once the RMPOUT drops within 200 mV of the VDAC. NORMAL OPERATION 100 % DUTY OPERATION CLKIN PHIN (2 Phase Design) EAIN RMPOUT PWMQ Figure 6: PWM Operation during normal and 100 % duty mode. Power State Indicator (PSI) function From a system perspective, the PSI input is controlled by the system and is forced low when the load current is lower than a preset limit and forced high when load current is higher than the preset limit. IR3508 can accept an active low signal on its PSI input and force the drivers into tri-state, effectively, forcing the phase IC into an off state. A PSI-assert signal activates three features in the Phase IC. First, it disconnects the IOUT pin from the ISHARE bus (from a system perspective). ISHARE is used to report current and is used for over-current protection. By disconnecting the disabled phase from the ISHARE bus, proper current reporting and over-current protection level are ensured. Secondly, the D Flip-Flop (DFF) is disabled, bypassing the Phase IC from the daisy chain loop. By removing the DFF from the daisy chain, the system ensures that proper phase delay is activated among the active phases. Finally, the gate drivers are forced to tri-state, disabling the phase IC from the power stage. Figure 7 shows the impact of PSI-assert on the gate drivers. After an 8 cycle PHSIN delay followed by a CLK falling edge, the PSI_SYNC goes from 0 to 1. This disables the gate drives and the DFF. Page 12 of 19 October 27, 2008 IR3508 PSI 8 PHSIN Delay CLK P SI_SYNC D_PWM LATCH Figure 7: PSI assertion. Debugging Mode If the CSIN+ pin is pulled up to VCCL voltage, IR3508 enters into debugging mode. Both drivers are pulled low and IOUT output is disconnected from the current share bus, which isolates this phase IC from other phases. However, the phase timing from PHSIN to PHSOUT does not change. Emulated Bootstrap Diode IR3508 integrates a PFET to emulate the bootstrap diode. If two or more top MOSFETs are to be driven at higher switching frequency, an external bootstrap diode connected from VCCL pin to BOOST pin may be needed. OUTPUT VOLTAGE (VO) OVP THRESHOLD 130mV IOUT(ISHARE) VCCL-800 mV GATEH (PHASE IC) GATEL (PHASE IC) FAULT LATCH ERROR AMPLIFIER OUTPUT (EAOUT) VDAC NORMAL OPERATION OVP CONDITION AFTER OVP Figure 8: Over-voltage protection waveforms Page 13 of 19 October 27, 2008 IR3508 Over Voltage Protection (OVP) The IR3508 includes over-voltage protection that turns on the low side MOSFET to protect the load in the event of a shorted high-side MOSFET, converter out of regulation, or connection of the converter output to an excessive output voltage. As shown in Figure 8, if IOUT pin voltage is above V(VCCL) – 0.8V, which represents over-voltage condition detected by control IC, the over-voltage latch is set. GATEL drives high and GATEH drives low. The OVP circuit overrides the normal PWM operation and within approximately 150ns will fully turn-on the low side MOSFET, which remains in conduction until IOUT drops below V(VCCL) – 0.8V when over voltage ends. The over voltage fault is latched in control IC and can only be reset by cycling the power to control IC. The error amplifier output (EAIN) is pulled down by control IC and will remain low. The lower MOSFETs alone can not clamp the output voltage however a SCR or N-MOSFET could be triggered with the OVP output to prevent processor damage. Operation at Higher Output Voltage The proper operation of the phase IC is ensured for output voltage up to 5.1V. Similarly, the minimum VCC for proper operation of the phase IC is 8 V. Operating below this minimum voltage, the current sharing performance of the phase IC is affected. DESIGN PROCEDURES - IR3508 Inductor Current Sensing Capacitor CCS and Resistor RCS The DC resistance of the inductor is utilized to sense the inductor current. Usually the resistor RCS and capacitor CCS in parallel with the inductor are chosen to match the time constant of the inductor, and therefore the voltage across the capacitor CCS represents the inductor current. If the two time constants are not the same, the AC component of the capacitor voltage is different from that of the real inductor current. The time constant mismatch does not affect the average current sharing among the multiple phases, but does affect the current signal IOUT as well as the output voltage during a load current transient if adaptive voltage positioning is being implemented. Measure the inductance L and the inductor DC resistance RL. Pre-select the capacitor CCS and calculate RCS as follows. L RL RCS (1) C CS Bootstrap Capacitor CBST Depending on the duty cycle and gate drive current of the phase IC, a capacitor in the range of 0.1uF to 1uF is needed for the bootstrap circuit. Decoupling Capacitors for Phase IC A 0.1uF-1uF decoupling capacitor is required at the VCCL pin. CURRENT SHARE LOOP COMPENSATION The internal compensation of current share loop ensures that crossover frequency of the current share loop is at least one decade lower than that of the voltage loop so that the interaction between the two loops is eliminated. The crossover frequency of current share loop is approximately 8 kHz. Page 14 of 19 October 27, 2008 IR3508 LAYOUT GUIDELINES The following layout guidelines are recommended to reduce the parasitic inductance and resistance of the PCB layout; therefore, minimizing the noise coupled to the IC. x Dedicate at least one middle layer for a ground plane, which is then split into signal ground plane (LGND) and power ground plane (PGND). x Separate analog bus (EAIN, DACIN, and IOUT) from digital bus (CLKIN, PSI, PHSIN, and PHSOUT) to reduce the noise coupling. x Connect PGND to LGND pins of each phase IC to the ground tab, which is tied to LGND and PGND planes respectively through vias. x Place current sense resistors and capacitors (RCS and CCS) close to phase IC. Use Kelvin connection for the inductor current sense wires, but separate the two wires by ground polygon. The wire from the inductor terminal to CSIN- should not cross over the fast transition nodes, i.e., switching nodes, gate drive outputs, and bootstrap nodes. x Place the decoupling capacitors CVCC and CVCCL as close as possible to VCC and VCCL pins of the phase IC respectively. x Place the phase IC as close as possible to the MOSFETs to reduce the parasitic resistance and inductance of the gate drive paths. x Place the input ceramic capacitors close to the drain of top MOSFET and the source of bottom MOSFET. Use combination of different packages of ceramic capacitors. x There are two switching power loops. One loop includes the input capacitors, top MOSFET, inductor, output capacitors and the load; another loop consists of bottom MOSFET, inductor, output capacitors and the load. Route the switching power paths using wide and short traces or polygons; use multiple vias for connections between layers. To Digital Bus To Analog Bus LGND PLANE To VIN LGND DACIN PHSIN NC PHSOUT CLKIN PGND IOUT To Gate Drive Voltage PSI NC EAIN CSIN+ VCC Cvccl GATEL BOOST VCCL GATEH NC SW To Bottom MOSFET Dbst Cbst PGND PLANE Page 15 of 19 To Switching Node To Top MOSFET To LGND Plane Ground Polygon To Inductor Sense October 27, 2008 Rcs Ccs CSIN - IR3508 PCB Metal and Component Placement x Lead land width should be equal to nominal part lead width. The minimum lead to lead spacing should be • 0.2mm to minimize shorting. x Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension + 0.05mm inboard extension. The outboard extension ensures a large and inspectable toe fillet, and the inboard extension will accommodate any part misalignment and ensure a fillet. x Center pad land length and width should be equal to maximum part pad length and width. However, the minimum metal to metal spacing should be • PP IRU  R] &RSSHU • PP IRU  R] &RSSHU DQG • 0.23mm for 3 oz. Copper) x Four 0.3mm diameter vias shall be placed in the pad land spaced at 1.2mm, and connected to ground to minimize the noise effect on the IC and to transfer heat to the PCB. Page 16 of 19 October 27, 2008 IR3508 Solder Resist x The solder resist should be pulled away from the metal lead lands and center pad by a minimum of 0.06mm. The solder resist mis-alignment is a maximum of 0.05mm and it is recommended that the lead lands are all Non Solder Mask Defined (NSMD). Therefore, pulling the S/R 0.06mm will always ensure NSMD pads. x The minimum solder resist width is 0.13mm. At the inside corner of the solder resist where the lead land groups meet, it is recommended to provide a fillet so a solder resist width of • PP UHPDLQV x Ensure that the solder resist in-between the lead lands and the pad land is • PP GXH WR WKH KLJK DVSHFW ratio of the solder resist strip separating the lead lands from the pad land. x The 4 vias in the land pad should be tented with solder resist 0.4mm diameter, or 0.1mm larger than the diameter of the via. Page 17 of 19 October 27, 2008 IR3508 Stencil Design x The stencil apertures for the lead lands should be approximately 80% of the area of the lead lands. Reducing the amount of solder deposited will minimize the occurrence of lead shorts. Since for 0.5mm pitch devices the leads are only 0.25mm wide, the stencil apertures should not be made narrower; openings in stencils < 0.25mm wide are difficult to maintain repeatable solder release. x The stencil lead land apertures should therefore be shortened in length by 80% and centered on the lead land. x The land pad aperture should be striped with 0.25mm wide openings and spaces to deposit approximately 50% area of solder on the center pad. If too much solder is deposited on the center pad the part will float and the lead lands will be open. x The maximum length and width of the land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back to decrease the incidence of shorting the center land to the lead lands when the part is pushed into the solder paste. Page 18 of 19 October 27, 2008 IR3508 PACKAGE INFORMATION 20L MLPQ (4 x 4 mm Body) – JA = 32 &: o JC = 3 C/W o Data and specifications subject to change without notice. This product has been designed and qualified for the Consumer market. Qualification Standards can be found on IR’s Web site. IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information. Page 19 of 19 October 27, 2008