IRS2609DSPBF

IRS2609DSPbF June 1, 2011 IRS2609DSPbF HALF-BRIDGE DRIVER Features • Floating channel designed for bootstrap operation Fully operational to +600 V Tolerant to negative transient voltage – dV/dt immune Gate drive supply range from 10 V to 20 V Undervoltage lockout for both channels 3.3 V, 5 V and 15 V input logic compatible Cross-conduction prevention logic Matched propagation delay for both channels High side output in phase with IN input Internal 530 ns dead-time Lower di/dt gate driver for better noise immunity Shut down input turns off both channels Integrated bootstrap diode RoHS compliant Packages • • • • • • • • • • • 8-Lead SOIC Product Summary VOFFSET IO+/VOUT ton/off (typ.) Dead Time 600 V max. 120 mA / 250 mA 10 V – 20 V 750 ns & 200 ns 530 ns Description The IRS2609D is a high voltage, high speed power MOSFET and IGBT drivers with dependent high and low side referenced output channels. Proprietary HVIC and latch immune CMOS technologies enable ruggedized monolithic construction. The logic input is compatible with Standard CMOS or LSTTL output, down to 3.3 V logic. The output drivers feature a high pulse current buffer stage designed for minimum driver cross-conduction. The floating channel can be used to drive an N-channel power MOSFET or IGBT in the high side configuration which operates up to 600 V. Applications: *Air Conditioner *Micro/Mini Inverter Drives *General Purpose Inverters *Motor Control Typical Connection www.irf.com 1 IRS2609DSPbF Qualification Information † Qualification Level Industrial†† Comments: This IC has passed JEDEC’s Industrial qualification. IR’s Consumer qualification level is granted by extension of the higher Industrial level. MSL2, 260°C (per IPC/JEDEC J-STD-020) Class 2 (per JEDEC standard JESD22-A114) Class B (per EIA/JEDEC standard EIA/JESD22-A115) Class I, Level A (per JESD78) Yes Moisture Sensitivity Level Human Body Model ESD Machine Model IC Latch-Up Test RoHS Compliant † Qualification standards can be found at International Rectifier’s web site http://www.irf.com/ †† Higher qualification ratings may be available should the user have such requirements. Please contact your International Rectifier sales representative for further information. www.irf.com 2 IRS2609DSPbF Absolute Maximum Ratings Absolute Maximum Ratings indicate sustained limits beyond which damage to the device may occur. All voltage parameters are absolute voltages referenced to COM. The thermal resistance and power dissipation ratings are measured under board mounted and still air conditions. Symbol VB VS VHO VCC VLO VIN COM dVS/dt PD RthJA TJ TS TL Definition High side floating absolute voltage High side floating supply offset voltage High side floating output voltage Low side and logic fixed supply voltage Low side output voltage Logic input voltage (IN & SD) Logic ground Allowable offset supply voltage transient Package power dissipation @ TA ≤ +25 °C Thermal resistance, junction to ambient Junction temperature Storage temperature Lead temperature (soldering, 10 seconds) Min. -0.3 VB - 2 0 VS - 0.3 -0.3 -0.3 COM -0.3 VCC - 20 — — — — -50 — Max. 620 VB + 0.3 VB + 0.3 20 VCC + 0.3 VCC + 0.3 VCC + 0.3 50 0.625 200 150 150 300 Units V V/ns W °C/W °C Recommended Operating Conditions For proper operation the device should be used within the recommended conditions. The VS and COM offset rating are tested with all supplies biased at 15 V differential. Symbol VB VS VSt VHO VCC VLO VIN Definition High side floating supply absolute voltage Static High side floating supply offset voltage Transient High side floating supply offset voltage High side floating output voltage Low side and logic fixed supply voltage Low side output voltage Logic input voltage (IN & SD) Min. VS +10 COM- 8(Note 1) -50 (Note2) VS 10 0 Max. VS +20 600 600 VB 20 VCC Units V VSS VCC TA Ambient temperature -40 125 °C Note 1: Logic operational for VS of -8 V to +600 V. Logic state held for VS of -8 V to – VBS. Note 2: Operational for transient negative VS of COM - 50 V with a 50 ns pulse width. Guaranteed by design. Refer to the Application Information section of this datasheet for more details. www.irf.com 3 IRS2609DSPbF Dynamic Electrical Characteristics VBIAS (VCC, VBS) = 15 V, COM = VCC, CL = 1000 pF, TA = 25 °C, DT = VSS unless otherwise specified. Symbol ton toff tsd MT tr t Definition Turn-on propagation delay Turn-off propagation delay Shut-down propagation delay Delay matching, HS & LS turn-on/off Turn-on rise time Turn-off fall time Deadtime: LO turn-off to HO turn-on(DTLO-HO) HO turn-off to LO turn-on (DTHO-LO) Delay matching time (t ON , t OFF) Deadtime matching = DTLO-HO - DTHO-LO & Min Typ Max Units Test Conditions — — — — — — 350 — — 750 250 250 — 150 50 530 — — 1100 400 400 60 220 80 800 60 60 ns VS = 0 V VS = 0 V VS = 0 V or 600 V VS = 0 V or 600 V f DT MT MDT VIN = 0 V & 5 V Without external deadtime Static Electrical Characteristics VBIAS (VCC, VBS) = 15 V, VCC = COM, DT = VCC and TA = 25 °C unless otherwise specified. The VIL, VIH and IIN parameters are referenced to VCC/COM and are applicable to the respective input leads: IN and SD. The VO, IO and Ron parameters are referenced to COM and are applicable to the respective output leads: HO and LO. Symbol VIH VIL VOH VOL ILK IQBS IQCC IIN+ IINISD, TH+ ISD, THVCCUV+ VBSUV+ VCCUVVBSUVVCCUVH VBSUVH IO+ IORbs Definition logic “1” input voltage for HO & logic “0” for LO logic “0” input voltage for HO & logic “1” for LO High level output voltage, VBIAS - VO Low level output voltage, VO Offset supply leakage current Quiescent VBS supply current Quiescent VCC supply current Logic “1” input bias current Logic “0” input bias current SD input positive going threshold SD input negative going threshold VCC and VBS supply undervoltage positive going Threshold VCC and VBS supply undervoltage negative going Threshold Hysteresis Output high short circuit pulsed current Output low short circuit pulsed current Bootstrap resistance Min Typ Max Units Test Conditions 2.2 — — — — — — — 0.8 0.3 — 45 — 0.8 1.4 0.6 50 70 A V IO = 20 mA IO = 20 mA VB = VS = 600 V VIN = 0 V or 4 V VIN = 0 V or 4 V VIN = 4 V VIN = 0 V 1000 2000 3000 — — — — 8.0 7.4 — 120 250 — 5 — 15 10 8.9 8.2 0.7 200 350 200 20 2 30 20 9.8 9.0 — — V mA — — Ohm VO = 0 V, PW ≤ 10 us VO = 15 V, PW ≤ 10 us www.irf.com 4 IRS2609DSPbF Functional Block Diagrams Lead Definitions Symbol IN SD VB HO VS VCC LO COM Description Logic input for high and low side gate driver outputs (HO and LO), in phase Logic input for shutdown High side floating supply High side gate drive output High side floating supply return Low side and logic fixed supply Low side gate drive output Low side return Lead Assignments IRS2609DS www.irf.com 5 IRS2609DSPbF Application Information and Additional Details Informations regarding the following topics are included as subsections within this section of the datasheet. • • • • • • • • • • • • • IGBT/MOSFET Gate Drive Switching and Timing Relationships Deadtime Matched Propagation Delays Shut down Input Input Logic Compatibility Undervoltage Lockout Protection Shoot-Through Protection Integrated Bootstrap Functionality Negative VS Transient SOA PCB Layout Tips Integrated Bootstrap FET limitation Additional Documentation IGBT/MOSFET Gate Drive The IRS2609D HVICs are designed to drive MOSFET or IGBT power devices. Figures 1 and 2 illustrate several parameters associated with the gate drive functionality of the HVIC. The output current of the HVIC, used to drive the gate of the power switch, is defined as IO. The voltage that drives the gate of the external power switch is defined as VHO for the high-side power switch and VLO for the low-side power switch; this parameter is sometimes generically called VOUT and in this case does not differentiate between the high-side or low-side output voltage. VB (or VCC) VB (or VCC) IO+ HO (or LO) + HO (or LO) VHO (or VLO) VS (or COM) VS (or COM) IO- Figure 1: HVIC sourcing current Figure 2: HVIC sinking current www.irf.com 6 IRS2609DSPbF Switching and Timing Relationships The relationships between the input and output signals of the IRS2609D are illustrated below in Figures 3, 4. From these figures, we can see the definitions of several timing parameters (i.e. tON, tOFF, tR, and tF) associated with this device. Figure 3: Switching time waveforms Figure 4: Input/output timing diagram Deadtime This family of HVICs features integrated deadtime protection circuitry. The deadtime for these ICs is fixed; other ICs within IR’s HVIC portfolio feature programmable deadtime for greater design flexibility. The deadtime feature inserts a time period (a minimum deadtime) in which both the high- and low-side power switches are held off; this is done to ensure that the power switch being turned off has fully turned off before the second power switch is turned on. This minimum deadtime is automatically inserter whenever the external deadtime is shorter than DT; external deadtimes larger than DT are not modified by the gate driver. Figure 5 illustrates the deadtime period and the relationship between the output gate signals. The deadtime circuitry of the IRS2609D is matched with respect to the high- and low-side outputs. Figure 6 defines the two deadtime parameters (i.e., DTLO-HO and DTHO-LO); the deadtime matching parameter (MDT) associated with the IRS2609D specifies the maximum difference between DTLO-HO and DTHO-LO. Matched Propagation Delays The IRS2609D family of HVICs is designed with propagation delay matching circuitry. With this feature, the IC’s response at the output to a signal at the input requires approximately the same time duration (i.e., tON, tOFF) for both the low-side channels and the high-side channels; the maximum difference is specified by the delay matching parameter (MT). The propagation turn-on delay (tON) of the IRS2609D is matched to the propagation turn-on delay (tOFF). www.irf.com 7 IRS2609DSPbF Shut down Input The IRS2609D family of HVICs is equipped with a shut down (/SD) input pin that is used to shutdown or enable the HVIC. When the /SD pin is in the high state the HVIC is able to operate normally. When the /SD pin is in low state the HVIC is tristated. 50% 50% IN 90% HO DTLO-HO 10% LO 90% DTHO-LO 10% MDT = DTLO-HO - DTHO-LO Figure 5: Shut down Figure 6: Dead time Definition Figure 7: Delay Matching waveform Definition Input Logic Compatibility The inputs of this IC are compatible with standard CMOS and TTL outputs. The IRS2609D has been designed to be compatible with 3.3 V and 5 V logic-level signals. The IRS2609D features an integrated 5.2 V Zener clamp on the /SD. Figure 8 illustrates an input signal to the IRS2609D, its input threshold values, and the logic state of the IC as a result of the input signal. www.irf.com 8 IRS2609DSPbF Input Signal (IRS23364D) V IH VIL Input Logic Level High Low Low Figure 8: HIN & LIN input thresholds Undervoltage Lockout Protection This family of ICs provides undervoltage lockout protection on both the VCC (logic and low-side circuitry) power supply and the VBS (high-side circuitry) power supply. Figure 9 is used to illustrate this concept; VCC (or VBS) is plotted over time and as the waveform crosses the UVLO threshold (VCCUV+/- or VBSUV+/-) the undervoltage protection is enabled or disabled. Upon power-up, should the VCC voltage fail to reach the VCCUV+ threshold, the IC will not turn-on. Additionally, if the VCC voltage decreases below the VCCUV- threshold during operation, the undervoltage lockout circuitry will recognize a fault condition and shutdown the high- and low-side gate drive outputs, and the FAULT pin will transition to the low state to inform the controller of the fault condition. Upon power-up, should the VBS voltage fail to reach the VBSUV threshold, the IC will not turn-on. Additionally, if the VBS voltage decreases below the VBSUV threshold during operation, the undervoltage lockout circuitry will recognize a fault condition, and shutdown the high-side gate drive outputs of the IC. The UVLO protection ensures that the IC drives the external power devices only when the gate supply voltage is sufficient to fully enhance the power devices. Without this feature, the gates of the external power switch could be driven with a low voltage, resulting in the power switch conducting current while the channel impedance is high; this could result in very high conduction losses within the power device and could lead to power device failure. Figure 9: UVLO protection Shoot-Through Protection The IRS2609D high-voltage ICs is equipped with shoot-through protection circuitry (also known as cross-conduction prevention circuitry). www.irf.com 9 IRS2609DSPbF Integrated Bootstrap Functionality The IRS2609D embeds an integrated bootstrap FET that allows an alternative drive of the bootstrap supply for a wide range of applications. A bootstrap FET is connected between the floating supply VB and VCC (see Fig. 10). Vcc BootFet Vb Figure 10: Semplified BootFET connection The integrated bootstrap feature can be used either in parallel with the external bootstrap network (diode and resistor) or as a replacement of it. The use of the integrated bootstrap as a replacement of the external bootstrap network may have some limitations at very high PWM duty cycle, corresponding to very short LIN pulses, due to the bootstrap FET equivalent resistance RBS. The summary for the bootstrap state follows: • Bootstrap turns-off (immediately) or stays off when at least one of the following conditions are met: 1- /SD is low 2- /SD is high, IN is low and VB is high (> 1.1*VCC) 3- /SD is high, IN is high (DT period excluded) 4- /SD is high, IN is high and VB is high (> 1.1*VCC) (during DT period) • Bootstrap turns-on when: 1- /SD in high, IN is low and VB is low (< 1.1(VCC)) 2- /SD in high, IN is high and VB is low (< 1.1(VCC)) (during the DT period). Please refer to the BootFET timing diagram for more details. www.irf.com 10 IRS2609DSPbF IN DT HO DT LO /SD BootStrap Fet VB 1.1*Vcc + Figure 11: BootFET timing diagram www.irf.com 11 IRS2609DSPbF Negative VS Transient SOA A common problem in today’s high-power switching converters is the transient response of the switch node’s voltage as the power switches transition on and off quickly while carrying a large current. A typical 3-phase inverter circuit is shown in Figure 12; here we define the power switches and diodes of the inverter. If the high-side switch (e.g., the IGBT Q1 in Figures 13 and 14) switches off, while the U phase current is flowing to an inductive load, a current commutation occurs from high-side switch (Q1) to the diode (D2) in parallel with the low-side switch of the same inverter leg. At the same instance, the voltage node VS1, swings from the positive DC bus voltage to the negative DC bus voltage. Figure 12: Three phase inverter DC+ BUS Q1 ON IU VS1 D2 Q2 OFF DC- BUS Figure 13: Q1 conducting Figure 14: D2 conducting Also when the V phase current flows from the inductive load back to the inverter (see Figures 15 and 16), and Q4 IGBT switches on, the current commutation occurs from D3 to Q4. At the same instance, the voltage node, VS2, swings from the positive DC bus voltage to the negative DC bus voltage. Figure 15: D3 conducting Figure 16: Q4 conducting www.irf.com 12 IRS2609DSPbF However, in a real inverter circuit, the VS voltage swing does not stop at the level of the negative DC bus, rather it swings below the level of the negative DC bus. This undershoot voltage is called “negative VS transient”. The circuit shown in Figure 17 depicts one leg of the three phase inverter; Figures 18 and 19 show a simplified illustration of the commutation of the current between Q1 and D2. The parasitic inductances in the power circuit from the die bonding to the PCB tracks are lumped together in LC and LE for each IGBT. When the high-side switch is on, VS1 is below the DC+ voltage by the voltage drops associated with the power switch and the parasitic elements of the circuit. When the high-side power switch turns off, the load current momentarily flows in the low-side freewheeling diode due to the inductive load connected to VS1 (the load is not shown in these figures). This current flows from the DC- bus (which is connected to the COM pin of the HVIC) to the load and a negative voltage between VS1 and the DC- Bus is induced (i.e., the COM pin of the HVIC is at a higher potential than the VS pin). Figure 17: Parasitic Elements Figure 18: VS positive Figure 19: VS negative In a typical motor drive system, dV/dt is typically designed to be in the range of 3-5 V/ns. The negative VS transient voltage can exceed this range during some events such as short circuit and over-current shutdown, when di/dt is greater than in normal operation. International Rectifier’s HVICs have been designed for the robustness required in many of today’s demanding applications. An indication of the IRS2609D’s robustness can be seen in Figure 20, where there is represented the IRS2609D Safe Operating Area at VBS=15V based on repetitive negative VS spikes. A negative VS transient voltage falling in the grey area (outside SOA) may lead to IC permanent damage; viceversa unwanted functional anomalies or permanent damage to the IC do not appear if negative Vs transients fall inside SOA. At VBS=15V in case of -VS transients greater than -16.5 V for a period of time greater than 50 ns; the HVIC will hold by design the high-side outputs in the off state for 4.5 µs. www.irf.com 13 IRS2609DSPbF Figure 20: Negative VS transient SOA for IRS2608D @ VBS=15V Even though the IRS2609D has been shown able to handle these large negative VS transient conditions, it is highly recommended that the circuit designer always limit the negative VS transients as much as possible by careful PCB layout and component use. PCB Layout Tips Distance between high and low voltage components: It’s strongly recommended to place the components tied to the floating voltage pins (VB and VS) near the respective high voltage portions of the device. Please see the Case Outline information in this datasheet for the details. Ground Plane: In order to minimize noise coupling, the ground plane should not be placed under or near the high voltage floating side. Gate Drive Loops: Current loops behave like antennas and are able to receive and transmit EM noise (see Figure 21). In order to reduce the EM coupling and improve the power switch turn on/off performance, the gate drive loops must be reduced as much as possible. Moreover, current can be injected inside the gate drive loop via the IGBT collector-to-gate parasitic capacitance. The parasitic auto-inductance of the gate loop contributes to developing a voltage across the gate-emitter, thus increasing the possibility of a self turn-on effect. Figure 21: Antenna Loops www.irf.com 14 IRS2609DSPbF Supply Capacitor: It is recommended to place a bypass capacitor (CIN) between the VCC and COM pins. A ceramic 1 µF ceramic capacitor is suitable for most applications. This component should be placed as close as possible to the pins in order to reduce parasitic elements. Routing and Placement: Power stage PCB parasitic elements can contribute to large negative voltage transients at the switch node; it is recommended to limit the phase voltage negative transients. In order to avoid such conditions, it is recommended to 1) minimize the high-side emitter to low-side collector distance, and 2) minimize the low-side emitter to negative bus rail stray inductance. However, where negative VS spikes remain excessive, further steps may be taken to reduce the spike. This includes placing a resistor (5 or less) between the VS pin and the switch node (see Figure 22), and in some cases using a clamping diode between COM and VS (see Figure 23). See DT04-4 at www.irf.com for more detailed information. Figure 22: VS resistor Integrated Bootstrap FET limitation Figure 23: VS clamping diode The integrated Bootstrap FET functionality has an operational limitation under the following bias conditions applied to the HVIC: • • VCC pin voltage = 0V AND VS or VB pin voltage > 0 In the absence of a VCC bias, the integrated bootstrap FET voltage blocking capability is compromised and a current conduction path is created between VCC & VB pins, as illustrated in Fig.24 below, resulting in power loss and possible damage to the HVIC. Figure 24: Current conduction path between VCC and VB pin Relevant Application Situations: www.irf.com 15 IRS2609DSPbF The above mentioned bias condition may be encountered under the following situations: • In a motor control application, a permanent magnet motor naturally rotating while VCC power is OFF. In this condition, Back EMF is generated at a motor terminal which causes high voltage bias on VS nodes resulting unwanted current flow to VCC. • Potential situations in other applications where VS/VB node voltage potential increases before the VCC voltage is available (for example due to sequencing delays in SMPS supplying VCC bias) Application Workaround: Insertion of a standard p-n junction diode between VCC pin of IC and positive terminal of VCC capacitors (as illustrated in Fig.25) prevents current conduction “out-of” VCC pin of gate driver IC. It is important not to connect the VCC capacitor directly to pin of IC. Diode selection is based on 25V rating or above & current capability aligned to ICC consumption of IC - 100mA should cover most application situations. As an example, Part number # LL4154 from Diodes Inc (25V/150mA standard diode) can be used. VCC VCC VCC Capacitor VB VSS (or COM) Figure 25: Diode insertion between VCC pin and VCC capacitor Note that the forward voltage drop on the diode (VF) must be taken into account when biasing the VCC pin of the IC to meet UVLO requirements. VCC pin Bias = VCC Supply Voltage – VF of Diode. Additional Documentation Several technical documents related to the use of HVICs are available at www.irf.com; use the Site Search function and the document number to quickly locate them. Below is a short list of some of these documents. DT97-3: Managing Transients in Control IC Driven Power Stages AN-1123: Bootstrap Network Analysis: Focusing on the Integrated Bootstrap Functionality DT04-4: Using Monolithic High Voltage Gate Drivers AN-978: HV Floating MOS-Gate Driver ICs www.irf.com 16 IRS2609DSPbF Parameters trend in temperature Figures 26-49 provide information on the experimental performance of the IRS2609D(S) HVIC. The line plotted in each figure is generated from actual lab data. A large number of individual samples from multiple wafer lots were tested at three temperatures (-40 ºC, 25 ºC, and 125 ºC) in order to generate the experimental (Exp.) curve. The line labeled Exp. consist of three data points (one data point at each of the tested temperatures) that have been connected together to illustrate the understood trend. The individual data points on the curve were determined by calculating the averaged experimental value of the parameter (for a given temperature). Turn-On Propagation Delay (ns) Turn-Off Propagation Delay (ns) 1500 1200 900 600 300 0 -50 -25 0 25 50 75 100 125 Temperature (oC) Exp. 500 400 300 Exp. 200 100 0 -50 -25 0 25 50 75 100 125 Temperature (oC) Fig. 26. Turn-on Propagation Delay vs. Temperature Turn-On Rise Time (ns) Turn-Off fall Time (ns) Fig. 27. Turn-off Propagation Delay vs. Temperature 125 100 75 50 Exp. 250 200 150 100 Exp. 50 0 -50 -25 0 25 50 o 25 0 ` 75 100 125 -50 -25 0 25 50 o 75 100 125 Temperature ( C) Temperature ( C) Fig. 28. Turn-on Rise Time vs. Temperature Fig. 29. Turn-off Rise Time vs. Temperature www.irf.com 17 IRS2609DSPbF 4 4 VCCUV hysteresis (V) 3 2 VBSUV hysteresis (V) 3 2 1 0 1 0 -50 Exp. Exp. -25 0 25 50 75 100 125 -50 -25 0 25 50 75 100 125 Temperature (oC) Temperature (oC) Fig. 30. VCC Supply UV Hysteresis vs. Temperature 10 VCC Quiescent Current (mA) VBS Quiescent Current ( A) 100 80 60 Fig. 31. VBS Supply UV Hysteresis vs. Temperature 8 6 4 2 0 -50 -25 0 25 50 75 100 125 Temperature (oC) Exp. Exp. 40 20 0 -50 -25 0 25 50 o ` 75 100 125 Temperature ( C) Fig. 32. VCC Quiescent Supply Current vs. Temperature 12 Exp. Fig. 33. VBS Quiescent Supply Current vs. Temperature 12 VCCUV+ Threshold (V) VCCUV- Threshold (V) 9 6 9 6 Exp. 3 0 -50 -25 0 25 50 o 3 0 75 100 125 -50 -25 0 25 50 o 75 100 125 Temperature ( C) Temperature ( C) Fig. 35. VCCUV+ Threshold vs. Temperature Fig. 36. VCCUV- Threshold vs. Temperature www.irf.com 18 IRS2609DSPbF 12 Exp. 12 VBSUV+ Threshold (V) 6 VBSUV- Threshold (V) 9 9 6 Exp. 3 0 -50 -25 0 25 50 75 100 125 Temperature (oC) 3 0 -50 -25 0 25 50 o 75 100 125 Temperature ( C) Fig. 37. VBSUV+ Threshold vs. Temperature Fig. 38. VBSUV- Threshold vs. Temperature 400 High Level Output Voltage (mV) 400 Low Level Output Voltage (mV) 300 200 EXP. 300 200 Exp. 100 0 -50 -25 0 25 50 o 100 0 -50 -25 0 25 50 o 75 100 125 75 100 125 Temperature ( C) Temperature ( C) Fig. 38. Low Level Output Voltage vs. Temperature Fig. 39. High Level Output Voltage vs. Temperature 8 500 Bootstrap Resistance ( ) 400 300 200 Exp. 6 IN VTH+ (V) 4 Exp. 100 0 -50 -25 0 25 50 75 100 125 Temperature (oC) 2 0 -50 -25 0 25 50 75 100 125 Temperature (oC) Fig. 40. Bootstrap Resistance vs. Temperature Fig. 41. IN VTH+ vs. Temperature www.irf.com 19 IRS2609DSPbF 8 6 IN VTH- (V) 8 6 HIN VTH+ (V) Exp. 4 2 0 -50 -25 0 25 50 o 4 Exp. 2 0 75 100 125 -50 -25 0 25 50 o 75 100 125 Temperature ( C) Temperature ( C) Fig. 42. LIN VTH- vs. Temperature Fig. 43. HIN VTH+ vs. Temperature 50 40 Tbson_VccTYP(ns) 30 20 10 0 8 6 HIN VTH- (V) 4 Exp. 2 Exp. 0 -50 -25 0 25 50 75 100 125 Temperature (oC) -50 -25 0 25 50 o 75 100 125 Temperature ( C) Fig. 44. HIN VTH- vs. Temperature 500 Shut-down propagation delay (ns) Fig. 45. Tbson_VCCTYP vs. Temperature 1000 800 400 300 200 100 0 -50 -25 0 25 50 75 100 125 Temperature (oC) Exp. Deadtime (ns) 600 400 200 0 -50 Exp. -25 0 25 50 o 75 100 125 Temperature ( C) Fig. 46. Shut-down Propagation Delay vs. Temperature Fig. 47. Deadtime vs. Temperature www.irf.com 20 IRS2609DSPbF 50 40 MDT (ns) 30 MT (ns) 20 10 0 -50 -25 0 25 50 o Exp. 30 25 20 15 Exp. 10 5 0 75 100 125 -50 -25 0 25 50 75 100 125 Temperature ( C) Temperature (oC) Fig. 48. Delay Matching vs. Temperature Fig. 49. Deadtime Matching vs. Temperature www.irf.com 21 IRS2609DSPbF Case Outlines www.irf.com 22 IRS2609DSPbF Tape and Reel Details: 8L-SOIC LOADED TAPE FEED DIRECTION B A H D F C NOTE : CONTROLLING DIM ENSION IN M M E G CARRIER TAPE DIMENSION FOR Metric Code Min Max A 7.90 8.10 B 3.90 4.10 C 11.70 12.30 D 5.45 5.55 E 6.30 6.50 F 5.10 5.30 G 1.50 n/a H 1.50 1.60 8SOICN Imperial Min Max 0.311 0.318 0.153 0.161 0.46 0.484 0.214 0.218 0.248 0.255 0.200 0.208 0.059 n/a 0.059 0.062 F D C E B A G H REEL DIMENSIONS FOR 8SOICN Metric Code Min Max A 329.60 330.25 B 20.95 21.45 C 12.80 13.20 D 1.95 2.45 E 98.00 102.00 F n/a 18.40 G 14.50 17.10 H 12.40 14.40 Imperial Min Max 12.976 13.001 0.824 0.844 0.503 0.519 0.767 0.096 3.858 4.015 n/a 0.724 0.570 0.673 0.488 0.566 www.irf.com 23 IRS2609DSPbF ORDER INFORMATION 8-Lead SOIC IRS2609DSPbF 8-Lead SOIC Tape & Reel IRS2609DSTRPbF The information provided in this document is believed to be accurate and reliable. However, International Rectifier assumes no responsibility for the consequences of the use of this information. International Rectifier assumes no responsibility for any infringement of patents or of other rights of third parties which may result from the use of this information. No license is granted by implication or otherwise under any patent or patent rights of International Rectifier. The specifications mentioned in this document are subject to change without notice. This document supersedes and replaces all information previously supplied. For technical support, please contact IR’s Technical Assistance Center http://www.irf.com/technical-info/ WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245 Tel: (310) 252-7105 www.irf.com 24 IRS2609DSPbF Revision History Revision Date 1.5 03-17-08 1.6 03-17-08 1.6a 1.7 03-21-08 04-18-08 May 8, 08 06-18-08 08-18-2009 Comments/Changed items Added application note to include negative Vs curve Added Qualification Information on Page 2, Disclaimer information on Page 25, and updated information on Pages 21-23 Removed revision letter from JEDEC standards under Qualification Information table. Added “RoHS compliant” statement to front page, Changed latch up level to A, added MT parameter. Changed file name from using revision to using date, Page1: corrected IGBT, Page5: corrected p/n on lead assignment diagram to IRS2609DS Corrected internal dead time on front page to 530ns instead of 540ns. Removed reference to trapezoidal modulation in Integrated Bootstrap Functionality section www.irf.com 25